Drive control device, linear motor system, and drive control method

Abstract

The present invention is provided with: a coil current control means (5) that individually controls the current in each of a plurality of coils (3) for a non-three-phase linear motor (100); and a movable element position estimating means (6) that has a speed electromotive force calculation unit (61) for computing, from the current (Is) and a voltage command (V* s) which was obtained from a motor constant of the linear motor (100) and the coil current control means (5), the speed electromotive force (Es) generated in each among the plurality of coils (3), that has a position calculator (63) for calculating, from the speed electromotive force (Es), an estimated position (θ-hat) of a movable element (1) on a travel path, and that estimates at least one of the estimated position (θ-hat) and an estimated velocity (θ-dot-hat).

Description

[Technical Field]

[0001] This application relates to a drive control device, a linear motor system, and a drive control method. [Background technology]

[0002] Linear synchronous motors (LSMs) have long been used in industries such as railways and factory automation (FA). Recently, in the FA industry, LSMs with the primary stator windings (coils) located on the ground and the secondary magnets located on the movable part have been attracting attention.

[0003] To control the thrust of an LSM, it is necessary to know the magnetic pole position of the movable element, just as with rotating machines. However, with LSMs, there is a challenge in that the cost of position sensors increases as the distance the device can travel increases. In response to this, a position sensorless control method has been disclosed for LSMs that detects the position using the induced voltage generated in a coil, also called the speed electromotive force, similar to that used in rotating machines (see, for example, Patent Document 1). [Prior art documents] [Patent Documents]

[0004] [Patent Document 1] Japanese Patent Publication No. 2002-223587 (paragraph 0022, Figure 4, paragraphs 0033-0034, Figure 7) [Overview of the Initiative] [Problems that the invention aims to solve]

[0005] On the other hand, recent LSMs have seen miniaturization of the movable parts and narrowing of the pitch between movable parts, and the number of non-three-phase motors is increasing. However, non-three-phase LSMs differ from ordinary (three-phase) LSMs in that the phase difference of the induced voltage is not 120 degrees. In addition, with miniaturization, the influence of end effects has become relatively larger, and the narrowing of the pitch between movable parts has made it necessary to individually control the current of each coil, resulting in an increasing number of LSMs to which the dq transformation theory cannot be applied.

[0006] In particular, LSMs divide the stator to improve power supply efficiency or to arrange multiple movable elements on the stator. However, sensorless control becomes more difficult where a movable element spans two stators, and existing sensorless control methods are no longer applicable.

[0007] This application discloses technology to solve the above-mentioned problems, and aims to provide a linear motor system that efficiently drives a non-three-phase LSM without a position sensor. [Means for solving the problem]

[0008] The drive control device disclosed herein comprises a power supply device for individually controlling the current of each of the multiple coils in a non-three-phase linear synchronous motor comprising a movable element and a stator in which a plurality of coils are arranged to form a path for the movable element; an induced voltage calculation unit for calculating an induced voltage generated in each of the multiple coils from the motor constants of the linear synchronous motor and current and voltage information for each coil obtained from the power supply device; a position calculation unit for calculating the magnetic pole position of the movable element in the path from the induced voltage; a movable element position estimation means for estimating at least one of the magnetic pole position and the travel speed of the movable element; and a control calculation means for controlling the operation of the power supply device based on the estimation result output from the movable element position estimation means. The movable element position estimation means includes a multiphase two-phase converter that transforms the induced voltage into a two-phase sine wave using a coordinate transformation matrix that utilizes the phase difference of the induced voltages occurring in adjacent coils among the plurality of coils, and the position calculator calculates the magnetic pole position based on the two-phase sine wave. It is characterized by being equipped with [this feature].

[0009] The drive control method disclosed herein is characterized by comprising: an induced voltage calculation step for a non-three-phase linear synchronous motor comprising a movable element and a stator in which a plurality of coils are arranged to form a path for the movable element, calculating the induced voltage generated in each of the plurality of coils from current and voltage information for each coil and the motor constants of the linear synchronous motor; a multiphase two-phase conversion step for converting the induced voltage into a two-phase sinusoidal wave using a coordinate transformation matrix that utilizes the phase difference of the induced voltages generated in adjacent coils among the plurality of coils; a position calculation step for calculating the magnetic pole position of the movable element in the path from the two-phase sinusoidal wave; and a control step for individually controlling the current of each of the plurality of coils based on the magnetic pole position. [Effects of the Invention]

[0010] According to the drive control device or drive control method disclosed in this application, the position of the movable element can be estimated from the induced voltage of the coil for a non-three-phase LSM, thus enabling the creation of a linear motor system that is driven without a position sensor. [Brief explanation of the drawing]

[0011] [Figure 1] This is a block diagram illustrating the configuration of a drive control device and a linear motor system using the drive control device according to Embodiment 1. [Figure 2] This is a block diagram illustrating the configuration of the coil current control means in the drive control device according to Embodiment 1. [Figure 3] This is a waveform diagram showing the speed electromotive force generated in each coil in the linear motor system according to Embodiment 1. [Figure 4] This is a waveform diagram obtained by performing a static two-phase conversion on the speed electromotive force generated in each coil in the linear motor system according to Embodiment 1. [Figure 5] This waveform diagram shows the result of calculating the arctangent of the speed electromotive force after static two-phase conversion in the linear motor system according to Embodiment 1. [Figure 6]This is a block diagram illustrating the configuration of the movable element position estimation means in the drive control device according to Embodiment 1. [Figure 7] This is a block diagram illustrating the configuration of the position calculator for the movable element position estimation means in the drive control device according to Embodiment 1. [Figure 8] This is a flowchart illustrating the operation of the drive control device and the drive control method according to Embodiment 1. [Figure 9] This is a block diagram showing an example of the hardware configuration of the part of the drive control device according to Embodiment 1 that performs calculation processing. [Figure 10] This is a block diagram illustrating the configuration of the movable element position estimation means in the drive control device according to Embodiment 2. [Figure 11] This is a flowchart illustrating the operation of the drive control device and the drive control method according to Embodiment 2. [Figure 12] This is a block diagram illustrating the configuration of the movable element position estimation means in the drive control device according to Embodiment 3. [Figure 13] This waveform diagram illustrates the position estimation error when performing an arctangent calculation due to the influence of end effects in the linear motor system according to Embodiment 3. [Figure 14] This is a block diagram showing an example in which the velocity calculator of the movable element position estimation means is configured as a type 3 in the drive control device according to Embodiment 4. [Figure 15] This is a block diagram showing the configuration of the proportional-integral double integrator of the speed calculator in the drive control device according to Embodiment 4. [Modes for carrying out the invention]

[0012] Embodiment 1. Figures 1 to 8 illustrate the configuration and operation of a drive control device and a linear motor system using the drive control device according to Embodiment 1, as well as a drive control method. Figure 1 is a block diagram illustrating the configuration of the drive control device and the linear motor system including the linear motor that it controls, and Figure 2 is a block diagram illustrating the configuration of the coil current control means portion of the drive control device.

[0013] Figure 3 is a waveform diagram showing the speed electromotive force generated in each of the three adjacent coils of the stator when the movable element of the linear motor is driven at a constant speed in a linear motor system, with the speed electromotive force on the vertical axis and the movable element position on the horizontal axis. Figure 4 is a waveform diagram showing the speed electromotive force after static two-phase conversion processing of the speed electromotive force shown in Figure 3, and Figure 5 is a waveform diagram showing the result of arctangent calculation of the speed electromotive force after static two-phase conversion shown in Figure 4.

[0014] Furthermore, Figure 6 is a block diagram illustrating the configuration of the movable element position estimation means of the drive control device, Figure 7 is a block diagram illustrating the configuration of the position calculator of the movable element position estimation means, and Figure 8 is a flowchart illustrating the operation of the drive control device and the drive control method.

[0015] The drive control device, drive control method, and linear motor system according to the embodiment of the present application will be described in detail below with appropriate reference to the drawings. In each figure, the same reference numerals indicate the same or corresponding parts.

[0016] The linear motor system according to Embodiment 1, as shown in Figure 1, consists of a linear motor 100, which is a non-three-phase LSM, and a drive control device 8 that drives and controls the linear motor 100. It is characterized by estimating the position or speed of the movable element 1 of the linear motor 100 based on the induced voltage generated in the coil 3 of the stator 2. Before describing the characteristic features, the basic configuration and operation of the drive control of the non-three-phase LSM will be explained.

[0017] The linear motor 100 has three permanent magnets 11a to 11c (referred to as "permanent magnet 11" if not individually distinguished) arranged in a straight line on the movable element 1, and magnetically coupled by a back yoke 10. On the other hand, the stator 2 has multiple coils 3a to 3i (referred to as "coil 3" if not individually distinguished) arranged to form the track for the movable element 1. Due to space limitations, only nine coils 3 are shown in the diagram, but in reality, hundreds or more coils 3 may be arranged. These coils 3 are magnetically coupled by a core back 4. Also, although only one movable element 1 is depicted in the diagram, in an actual linear motor 100, multiple movable elements 1 may exist on the track.

[0018] In the linear motor 100, the movable element 1 can be moved by controlling the current flowing through each coil 3. To control the current flowing through each coil 3, the drive control device 8 is provided with coil current control means 5a to 5i (referred to as "coil current control means 5" when not individually distinguished) corresponding to each of the multiple coils 3. In order for the multiple movable elements 1 to move in close proximity, it is necessary to individually control the current of each of the multiple coils 3, so each of the multiple coil current control means 5 individually controls the current of each coil 3 according to the current command determined by the control calculation means 7.

[0019] Each of the multiple coil current control means 5 can be configured, for example, as shown in Figure 2. In the figure, coil 3 is represented by an inductor symbol. A variable voltage source 52 is connected to coil 3. The variable voltage source 52 can be any circuit as long as it can independently control the current of each coil 3 by outputting an arbitrary voltage. The variable voltage source 52 can be configured as, for example, a single-phase inverter circuit. The current detection means 51 detects the current flowing through coil 3. The current control means 53 controls the current so that the current command matches the current flowing through the coil, and determines the voltage command. As a calculation method for current control, for example, PID control (proportional-integral-derivative control) is well known. The variable voltage source 52 operates according to the voltage command determined by the current control means 53, and the desired current flows through coil 3.

[0020] As explained in the background technology section, recent LSMs are characterized by "miniaturization of the movable elements" and "narrowing of the pitch between movable elements." To achieve this, a non-three-phase motor structure that cannot be driven by 120-degree energization is used. By removing the premise of "driving with 120-degree energization," the degree of freedom of the magnetic structure increases, making it easier to increase the thrust density.

[0021] Although this is merely one example, the linear motor 100 shown in Figure 1 is a type of LSM that is driven by a 135° current flow. The ratio of the lateral length of one movable element to the lateral length of one coil is 4:1, and three permanent magnets 11a to 11c are attached to the movable element 1. We will refer to this structure as a "3-pole 4-slot structure". If the lateral length of one magnetic pole is 180 degrees, then the electrical angle θ per coil in the 3-pole 4-slot structure is c As shown in equation (1), this is 135 degrees.

[0022]

number

[0023] By changing the combination of the length of the movable element 1 and the arrangement of the permanent magnets 11, various structures can be considered, such as a "5-pole 6-slot structure" and a "6-pole 8-slot structure". The phase difference of the current in each coil 3 changes depending on the number of pole slots, but the electrical angle θ of one coil remains the same regardless of the structure. c This can be calculated using a similar formula. In general rotating machines, from the perspective of cost performance, θ c Three-phase motors with a 120-degree rotational frequency (=120 degrees) are often used. However, in recent LSMs, flexibility of operation and added value such as cableless movable parts are becoming more important than cost performance, so non-three-phase motor structures that cannot be driven with 120-degree rotational frequency are being adopted.

[0024] As shown in Figure 1, the drive control device 8 can individually control the current of each coil 3, so thrust can be generated regardless of the combination of pole slots. However, to control the thrust of the LSM, it is necessary to know the magnetic pole position of the movable element, just as in the case of a rotating machine. Therefore, the magnetic pole position of the movable element 1 is measured by the movable element position estimation means 6. At the same time, the moving speed of the movable element 1 may also be calculated.

[0025] Of course, any known means, such as optical encoders, magnetic encoders, or video cameras, may be used to detect the movable element position and velocity. However, as mentioned above, this increases costs, so the movable element position is estimated from the voltage command and current of each coil 3.

[0026] In rotating machinery, the principle of position estimation generally utilizes the induced voltage, also known as the speed electromotive force, generated during rotation, and this method has been applied to LSMs as well. However, it is unclear whether conventional position estimation methods can be applied to non-three-phase LSMs that cannot be driven with 120-degree energization. Furthermore, non-three-phase LSMs have fewer magnetic poles and larger edge distortion in the induced voltage waveform, which also makes position estimation difficult. In this application, "non-three-phase" refers to energization at a distance of 5 degrees or more from 120 degrees (115 degrees or less, or 125 degrees or more), and it is assumed that conventional position estimation methods cannot be applied within that range.

[0027] Therefore, the drive control device 8 of the present invention makes it possible to estimate the position of the movable element of a non-three-phase LSM that cannot be driven with 120 degrees of energization using the speed electromotive force without a sensor, as shown below. For example, when the movable element 1 is driven at a constant speed, the waveforms of the three adjacent coils 3d to 3f with coil 3e as the center are as shown in Figure 3. In the figure, the three waveforms V3d, V3e, and V3f are the speed electromotive forces generated in coils 3d, 3e, and 3f, respectively, and the point where the center of coil 3e and the center of the movable element 1 coincide is considered to be zero degrees.

[0028] The waveform V3e, shown by the solid line, is basically sinusoidal, although some distortion is observed within the ±180-degree range. On the other hand, the waveform distortion is due to the non-uniformity of the magnetic properties at the ends of the movable element, i.e., the end effect, which is not so unusual in LSMs with a small number of magnetic poles. Beyond the ±180-degree range, the waveform V3e shown by the solid line gradually attenuates and eventually becomes zero. The other two waveforms, V3d and V3f, have the same shape, but their phases are shifted by 135 degrees each. Due to space limitations, the waveforms of the other coils 3 (coils 3a-3c, 3g-3i) are not shown here, but the speed electromotive forces of these coils 3 also have waveforms that are similarly shifted by 135 degrees each.

[0029] In rotating machinery control, it is common practice to use coordinate transformation to convert such multiphase AC waveforms into two-phase AC waveforms. However, in the case of non-three-phase LSMs that cannot be driven with 120-degree energization, it is not even certain whether they can be converted into two-phase AC waveforms. To the best of the inventors' knowledge, sensorless control technology and coordinate transformation technology for non-three-phase LSMs that cannot be driven with 120-degree energization are not publicly known, and none have yet become clear to those skilled in the art.

[0030] Therefore, we decided to perform position estimation using the following approach. Here, the rate electromotive force for 9 coils of coil 3 is given by dot θK. m9 We will represent this using the vector shown. In this application, in order to convert this into a two-phase sinusoidal waveform with two orthogonal axes, we use the coordinate transformation matrix A9 shown in equation (3) and calculate it as shown in equation (2).

[0031]

number

[0032] If the motor structure is not a 3-pole 4-slot type, the electrical angle θ per coil cAlthough the value of c is different from 135 degrees, the electrical angle θ per coil according to its structure

[0033] In the case of a three-phase rotating machine, since it is common to multiply the three-phase to two-phase coordinate conversion matrix or its inverse conversion matrix by a coefficient of (2 / 3) 0.5 accordingly, it is also possible to multiply the above coordinate conversion matrix by some coefficient. However, in a non-three-phase LSM that cannot perform 120-degree conduction, there is almost no merit in multiplying by a coefficient as in the case of a three-phase rotating machine, so it is not necessary to multiply by a coefficient deliberately. Therefore, no special coefficient is multiplied in Equation (2).

[0034] The number of Coil 3 used in the calculation can be increased or decreased as appropriate. For example, when expressing the speed electromotive force for 7 coils from Coil 3b to 3h by a vector called dot θK m7 the calculation formula and coordinate conversion matrix for converting this into a two-phase sine wave waveform on the orthogonal two axes are as shown in Equations (4) and (5).

[0035]

Equation

[0036] The waveform shown in Fig. 3 is converted by the stationary two-phase conversion, that is, by Equations (4) and (5), and the calculated waveform is shown in Fig. 4. This is the calculation result when centered on Coil 3e. However, as long as the mover 1 is within the range of ±180 degrees, although some distortion can be seen, the two waveforms Va and Vb have almost the same amplitude, the phase difference is about 90 degrees, and a two-phase sine wave-like waveform is obtained.

[0037] Furthermore, it can be seen that when the distance between the central coil 3e and the movable element 1 exceeds ±180 degrees, the waveform of the recoil electromotive force after static two-phase conversion begins to attenuate. The waveform in this range cannot be said to be a two-phase sine wave. Increasing the number of coils 3 used in the calculation expands the range in which two-phase sine waves can be produced, but it also increases the amount of calculation required. In the case of this type of LSM, the number of coils can exceed several hundred, so it is best not to use too many coils 3 in the calculation.

[0038] In the drive control device 8 of this invention, the position of the movable element is estimated using the recoil electromotive force after the static two-phase conversion described above. Figure 5 shows the waveform Wr obtained by the four-quadrant arctangent (inverse tangent) calculation process for the recoil electromotive force after the static two-phase conversion shown in Figure 4. For reference, the waveform Wi obtained by the four-quadrant arctangent calculation process for an ideal two-phase sine wave with no end attenuation is also shown.

[0039] When the four-quadrant arctangent calculation is performed on an ideal two-phase sine wave, a sawtooth wave waveform (waveform Wi) is obtained. The waveform Wr obtained by the four-quadrant arctangent processing on the recoil electromotive force after static two-phase conversion also has a shape similar to a sawtooth wave, but as the position of the movable element 1 moves away from the origin, it deviates from the sawtooth wave waveform. Since the difference from the sawtooth wave waveform corresponds to the position estimation error, Figure 5 shows that position estimation by arctangent calculation is possible as long as the distance between the central coil 3e and the movable element 1 is within a range of approximately ±180 degrees.

[0040] It should be noted that this arctangent calculation only yields the direction of the recoil electromotive force vector. Even if the movable element position is the same, if the direction of movement of the movable element 1 is different, the direction of the recoil electromotive force vector will differ by 180 degrees. Therefore, if the movable element 1 can move in both directions, it is necessary to correct the movable element position according to the sign of the movable element velocity. However, since position sensorless control using recoil electromotive force is intended for operation in the medium to high speed range, the sign of the movable element velocity is known, so the correction is not particularly difficult.

[0041] Based on the above findings, the drive control device 8 according to Embodiment 1 uses a voltage command vector V * s and current vector I s To calculate the estimated position hat θ, the movable element position estimation means 6, configured as shown in Figure 6, is provided.

[0042] The movable element position estimation means 6 uses a voltage command vector V * s and current vector I s From the recoil electromotive force E s A speed electromotive force calculation unit 61 calculates the speed electromotive force E s The velocity electromotive force E in stationary Cartesian coordinates is obtained by multiphase two-phase conversion (stationary two-phase sine wave conversion). αβ It includes a multiphase two-phase converter 62 that calculates the velocity electromotive force E in stationary Cartesian coordinates. αβ The system includes a position calculator 63 that calculates an estimated position hat θ from the data. The position calculator 63 may also acquire additional information to determine the driving direction of the movable element 1. Here, the sign(dot θ) is the code of the speed command. * This is set to be input to the position calculator 63.

[0043] Furthermore, the movable element position estimation means 6 may include a velocity calculator 64 that calculates the estimated velocity dot hat θ from the estimated position hat θ. The velocity calculator 64 calculates the estimated velocity dot hat θ by performing a differential operation or a noise reduction operation on the estimated position hat θ.

[0044] Velocity electromotive force E obtained by velocity electromotive force calculation unit 61 s The method for calculating the induced voltage will be explained. The voltage equation for linear motor 100 is given by equation (6).

[0045]

number

[0046] Stator voltage vector V s Instead, the voltage command vector V * s If we decide to use this, the rate electromotive force vector E s This can be found using equation (7).

[0047]

number

[0048] Note that the rate electromotive force VectorE s The movable element magnetic flux vector φ m It is defined as the time derivative of the movable element flux vector φ. m Since is a function of the movable element position θ, and the movable element position θ is a function of time t, the recoil electromotive force vector E s This can also be written as equation (8).

[0049]

number

[0050] As mentioned above, when the linear motor 100 is driven (movable element 1 travels along the track provided by the stator 2), the recoil electromotive force of each coil 3 has a waveform as shown in Figure 3, and the recoil electromotive force can be roughly calculated using equation (7). Since the recoil electromotive force is a function of the movable element position θ, the movable element position θ can be estimated from this information. In this case, using the coordinate transformation equations shown in equations (2) to (5) above makes it easier to estimate the movable element position θ.

[0051] The position calculator 63 calculates the velocity electromotive force E in stationary Cartesian coordinates, for example, as shown in Figure 7. αβ and the speed command sign (dot θ) *The system includes an inverse tangent calculator 631 that estimates the movable part position by performing an inverse tangent calculation. It also includes an edge detection unit 632 that detects edges from the result of the inverse tangent calculation, an offset amount calculation unit 633 that calculates the offset amount, and an adder 634 that adds the offset amount calculated by the offset amount calculation unit 633 to the movable part position estimated by the inverse tangent calculator 631.

[0052] The inverse tangent unit 631 performs a four-quadrant inverse tangent calculation on the speed electromotive force, which has been coordinate-transformed to two-phase AC, to estimate the position of the movable element. When the movable element 1 moves in both forward and reverse directions, the calculation result needs to be corrected by 180 degrees according to the direction of movement of the movable element 1, so here the sign of the speed command sign(θ) * The sign (θ) of the velocity command is input to the arctangent unit 631, but if the movable element 1 moves in only one direction, the sign (θ) of the velocity command is input to the arctangent unit 631. * You do not need to enter ().

[0053] In linear motor systems requiring non-three-phase LSMs, hundreds or more coils 3 may be positioned on the ground side, resulting in a range of motion of the movable element 1 that can exceed tens of thousands of degrees in electrical angle terms. However, the output range of the inverse tangent calculator 631 is only ±180 degrees. If current were to be supplied to all coils 3, the phase of current supply could be determined solely by the output value of the inverse tangent calculator 631. However, supplying current to coils 3 located far from the movable element 1 is energy inefficient. Therefore, current is usually supplied only to coils 3 in the vicinity of the movable element 1. To achieve this, a mechanism is needed to determine the approximate position of the movable element 1.

[0054] The simplest way to roughly determine the position of the movable element 1 is to detect and count discontinuous changes (edges) in the result of the arctangent calculation. The edge detection unit 632 adds one count value when the result of the arctangent calculation changes abruptly from +180 degrees to -180 degrees. Conversely, it subtracts one count value when it changes abruptly from -180 degrees to +180 degrees. The offset amount calculation unit 633 outputs the approximate position (offset amount) of the movable element 1 by multiplying the value counted by the edge detection unit 632 by 360 degrees. This is added to the result of the arctangent calculation by the adder 634, and the resulting addition is used as the output (estimated position hat θ) of the position calculator 63.

[0055] There are various other methods for determining the offset amount. For example, the rate electromotive force vector E s One possible method for determining the offset amount is to square each element and examine their relative magnitudes. Another method involves determining the offset amount by obtaining a rough position using an inexpensive sensor such as a video camera, and then estimating the precise position using arctangent calculation.

[0056] The operation of the drive control device 8 according to Embodiment 1, that is, the flow of the drive control method, is summarized in the flowchart of Figure 8. First, the recoil electromotive force calculation unit 61 receives the voltage command vector V from each of the multiple coil current control means 5. * s and current vector I s From the recoil electromotive force E s The (induced voltage) is calculated (step S100). The multiphase two-phase converter 62 receives the speed electromotive force E output from the speed electromotive force calculation unit 61. s The velocity electromotive force E in stationary Cartesian coordinates is obtained by multiphase two-phase conversion (stationary two-phase sine wave conversion). αβ The calculation is performed (step S110).

[0057] The position calculator 63 receives the reciprocating electromotive force E in stationary Cartesian coordinates output from the multiphase two-phase converter 62. αβThe movable element position is estimated by the four-quadrant arctangent calculation (steps S120-S140). At this time, the arctangent calculator 631 calculates the rate electromotive force E αβ The four quadrants inverse tangent calculation is performed (step S120), and the result is output to the adder 634 and the edge detection unit 632.

[0058] The edge detection unit 632 detects edges from the result of the arctangent calculation and outputs a count value, increased or decreased according to the type of edge (positive or negative direction), to the offset amount calculation unit 633 (step S130). The offset amount calculation unit 633 outputs the value obtained by multiplying the count value by 360 degrees as the offset amount to the adder 634 (step S140). The adder 634 adds the offset amount to the result of the arctangent calculation and outputs the movable element position as the estimated position hat θ (step S150).

[0059] Then, based on the outputted estimated position hat θ, the control calculation means 7 controls the operation of the coil current control means 5. In step S150, the velocity calculator 64 may calculate the estimated velocity dot hat θ from the estimated position hat θ.

[0060] By performing this process, it is possible to estimate the position of the movable element in the linear motor 100, which is a non-three-phase LSM. The drive control device 8 of this invention can calculate the estimated position hat θ of the movable element 1 of the linear motor 100 without using a sensor and control the drive, thereby reducing costs.

[0061] Incidentally, in rotating machinery, various position estimation methods and position sensorless control methods have been devised in addition to the method of performing an arctangent calculation on the speed electromotive force. Such technologies may be combined with the form disclosed in this application.

[0062] Furthermore, the movable element position estimation means 6 or the control calculation means 7 that constitute the drive control device 8 may be configured by a single hardware 6H comprising a processor 6H1 and a storage device 6H2, as shown in Figure 9. Although not shown, the storage device 6H2 comprises a volatile storage device such as random access memory and a non-volatile auxiliary storage device such as flash memory. Alternatively, a hard disk may be provided as an auxiliary storage device instead of flash memory. The processor 6H1 executes the program input from the storage device 6H2. In this case, the program is input from the auxiliary storage device to the processor 6H1 via the volatile storage device. The processor 6H1 may also output data such as calculation results to the volatile storage device of the storage device 6H2, or it may save the data to the auxiliary storage device via the volatile storage device.

[0063] Embodiment 2. In Embodiment 2, an example is described in which the coils from which information is collected for position estimation calculations are limited according to the position of the movable element. Figure 10 is a block diagram corresponding to Figure 6 of Embodiment 1 for explaining the configuration of the movable element position estimation means in the drive control device according to Embodiment 2, and Figure 11 is a flowchart for explaining the operation of the drive control device and the drive control method. Except for the configuration and operation for limiting the coils from which information is collected, it is the same as in Embodiment 1, and the explanation of the same parts will be omitted, and Figures 1 to 5, 7 and 9 used in Embodiment 1 will be referenced.

[0064] The movable element position estimation means 6 according to Embodiment 2 differs from the movable element position estimation means 6 described in Figure 6 of Embodiment 1 in that a nearby coil selector 65 is added, as shown in Figure 10.

[0065] In linear motor systems that require non-three-phase LSMs, it is not uncommon to have a track with hundreds or more coils 3 arranged on the ground. Even in such cases, the coordinate transformation formula described in Embodiment 1 can be extended to estimate the position of the movable element using the information of the recoil electromotive force of all coils 3. However, this method has the problem that the amount of computation required for position estimation increases as the scale of the device increases. Furthermore, if there are multiple movable elements 1 on the track of the linear motor 100, the information of the recoil electromotive force of all coils 3 will include the recoil electromotive force information of multiple movable elements 1, but it is extremely difficult to estimate the position information of each movable element 1 from that information.

[0066] Therefore, in Embodiment 2, a proximity coil selector 65 is provided in the movable element position estimation means 6 so as to estimate the movable element position using only the voltage command and current of the coil 3 located near the movable element 1 among the multiple coils 3.

[0067] The proximity coil selector 65 selects several coils 3 located near a given movable element 1 based on the estimated position hat θ of the movable element. In the case of the 3-pole 4-slot linear motor 100 shown in Figure 1, it is sufficient to select about 5 to 7 coils 3. More specifically, it selects coils 3 directly below the movable element 1, and coils 3 within an electrical angle of less than one coil's length (135 degrees in this example) extending outward from both ends in the direction of travel. In Figure 1, coils 3c to 3f directly below the movable element 1 and coils 3b and 3g within a range of less than 135 degrees extending outward from both ends are selected. The selected coils 3 are referred to here as "proximity coils". The proximity coil selector 65 receives voltage command vectors V from each of the multiple coil current control means 5. * s and current vector I s The following is input, but from among them, the voltage command V for the nearby coil is selected. * sn and current I sn Only this information is transmitted to the recurrent electromotive force calculation unit 61.

[0068] The speed electromotive force calculation unit 61 calculates the voltage command V of the nearby coil. * sn and current Isn The rate electromotive force of the nearby coil is calculated using the following method. The multiphase two-phase converter 62 calculates the rate electromotive force of the nearby coil E sn The coordinates are transformed to values ​​on a stationary Cartesian coordinate system. The operation of the other blocks is the same as in Embodiment 1. That is, as shown in the flowchart of Figure 11, in the operation described in Figure 8 of Embodiment 1, a step S90 is added before step S100 to select a nearby coil using the nearby coil selector 65, and thereafter (steps S100~) the operation is the same as in Embodiment 1.

[0069] By providing the nearby coil selector 65 in this way, the amount of computation required for estimating the movable element position can be reduced by excluding voltage and current information from coils other than the nearby coil from the calculation. Furthermore, even when multiple movable elements 1 are present on the track, the position or velocity of each movable element 1 can be estimated individually.

[0070] Embodiment 3. Embodiment 3 describes an example in which errors due to the influence of end effects in position estimation are reduced. Figure 12 is a block diagram illustrating the configuration of the movable element position estimation means in the drive control device according to Embodiment 3, and Figure 13 is a waveform diagram illustrating the position estimation error when performing arctangent calculation due to the influence of end effects in a linear motor system.

[0071] Except for the configuration and operation to reduce errors due to edge effects, the details are the same as in Embodiments 1 and 2. Therefore, the explanation of the similar parts will be omitted, and Figures 1 to 5, 7, and 9 used in Embodiment 1, and Figure 11 used in Embodiment 2 will be referenced.

[0072] In the movable element position estimation means 6 described in Figure 10 of Embodiment 2, the velocity calculator 64 performed only velocity calculations. However, in the movable element position estimation means 6 according to Embodiment 3, as shown in Figure 12, the velocity calculator 64 of the movable element position estimation means 6 is equipped with a filter means to reduce errors, so that it performs not only velocity estimation but also position estimation with reduced errors.

[0073] In Embodiment 3, the signal output by the position calculator 63, which was referred to as "estimated position hat θ" in Embodiments 1 and 2, is renamed "first estimated position," and the velocity calculator 64 obtains a second estimated position hat θ by removing the periodic error component from the first estimated position. F The following is output. During this calculation, the estimated velocity dot-hat θ can be calculated, so the estimated velocity signal is output simultaneously. The velocity calculator 64 is configured as a type 2 controller consisting of an adder 641, a proportional-integral calculator 642, and an integrator 643. In other words, in step S150 explained in Figure 8 of Embodiment 1, in addition to calculating the first estimated position, the second estimated position dot-hat θ is obtained by removing the error component from the calculated first estimated position. F This means that the calculation has been modified to do so.

[0074] The reason for adopting this configuration in Embodiment 3 will be explained. In the LSM (linear motor 100) with the structure described in Figure 1, a movable element 1 with a small number of magnetic poles is used, so the so-called end effect tends to be large. The end effect refers to the performance degradation caused by magnetic asymmetry at the ends of the movable element. Due to the effect of the end effect, there is a large distortion in the speed electromotive force of this type of LSM. This causes an error in the estimation of the movable element position. Since this error is due to the motor structure, the inverse tangent calculation by the multiphase two-phase converter 62 results in a position estimation error of a periodic pulsating waveform as shown in Figure 13.

[0075] While the movable element velocity can be calculated by differentiating the movable element position, if such pulsating components are superimposed on the estimated position information, then large pulsations will also be superimposed on the estimated velocity. Controlling the linear motor 100 using movable element position or velocity superimposed with pulsations will not yield a very good control response. Therefore, it is necessary to remove these pulsating components from the movable element position and velocity information.

[0076] Furthermore, there are cases where the voltage command or current changes suddenly, or where noise is superimposed on the detected current value. In such cases, the output of the position calculator 63 fluctuates, but the actual movable element position may not have changed yet. Therefore, it is necessary to prevent the estimated movable element position from changing too sensitively.

[0077] The simplest method for removing the high-frequency pulsation component from the estimated movable element position is to use a low-pass filter. However, a simple low-pass filter increases the DC component of the position estimation error. For example, when movable element 1 is moving at a constant speed, its position changes in a ramp-like manner. If this movable element position is input to a simple first-order low-pass filter, a steady-state error occurs in the filter's input and output. Since a DC error in the estimation of the movable element position prevents efficient thrust generation, filtering must be performed in a way that eliminates DC errors in the steady state.

[0078] Applying a low-pass filter with a low cutoff frequency relative to the estimated velocity, even if intended to eliminate pulsations, is not ideal. This is because applying a low-pass filter with a low cutoff frequency prevents rapid control of the movable element.

[0079] Thus, it is not easy to eliminate the influence of end effects from the position estimation results or velocity estimation results of a linear motor 100, which has large end effects. Therefore, in Embodiment 3, the velocity calculator 64 was configured as described above as a method to eliminate the influence of end effects from the position estimation results and velocity estimation results with a simple configuration.

[0080] The mechanism of the velocity calculator 64 will now be explained. As mentioned above, the first estimated position hat θ output from the position calculator 63 has a pulsation component superimposed on it due to edge effects, and some kind of filter is needed to remove this. We consider performing this filtering by feedback calculation of the difference between input and output. At this time, it is important not only to remove the pulsation due to edge effects, but also to make the steady difference between the input and output of the filter zero.

[0081] Here, we consider how to make the steady-state difference between the input and output of the filter zero when the movable element 1 is moving at a constant speed. Since the movable element position changes in a ramp-like manner at this time, we should adopt a controller structure that makes the steady-state error with respect to the ramp response zero. It is generally known as the "internal model principle" in classical control theory that a necessary condition for making the steady-state error with respect to the ramp response zero is to use a controller of type 2 or higher.

[0082] A Type 2 controller is a controller that includes two integrators connected in series. The speed calculator 64 includes a proportional-integral calculator 642 and an integrator 643. Since the proportional-integral calculator 642 has an internal integrator, two integrators are connected in series in this system. Therefore, the speed calculator 64 is a Type 2 controller.

[0083] Furthermore, in cases where the movable element speed changes at a constant rate, if the steady-state difference between the filter's input and output is to be zero, a Type 3 controller can be used. As will be discussed later, this can be achieved simply by adding a double integrator path to the speed calculator 64 of this embodiment, so it is not particularly difficult to implement.

[0084] Regarding the ramp response, according to the internal model principle, the first estimated position and the second estimated position hat θ, which is the output of integrator 643, are given by the first estimated position and the second estimated position hat θ. F Since they consistently coincide, the input to integrator 643 can be considered as velocity information. From this, it can be said that this system can calculate velocity without performing differential operations.

[0085] The proportionality term in the proportional-integral operator 642 is necessary to manipulate the damping coefficient of the system. This will be explained in more detail. The transfer function of the proportional-integral operator 642 can be written as shown in equation (9).

[0086]

number

[0087] Then, in the velocity calculator 64, the second estimated position hat θ is calculated from the first estimated position hat θ. F The transfer function up to this point can be expressed as shown in equation (10).

[0088]

number

[0089] Furthermore, the transfer function from the first estimated position hat θ to the estimated velocity dot hat θ can be expressed as shown in equation (11).

[0090]

number

[0091] Furthermore, the following equation (Equation (12)) is well-known as the transfer function in quadratic normal form.

[0092]

number

[0093] By comparing the coefficients in equations (10) to (12), we can determine the damping coefficient ζ and natural angular frequency ω of the velocity calculator 64. n It can be seen that this is determined by the following formulas (Equations (13) and (14)).

[0094]

number

[0095] Proportional gain K of proportional-integral unit 642 PWhen ω is zero, the damping coefficient is zero. It is known that when the damping coefficient is zero, the output signal undergoes sustained oscillation. Therefore, it is necessary to design the proportional gain so that the damping coefficient becomes the desired value. On the other hand, the natural angular frequency ω of the system n The integral gain K I This is determined by the integral gain K. I By adjusting this setting, you can change the frequency range at which the effects of edge effects are removed.

[0096] As described above, by using the movable element position estimation means 6 according to Embodiment 3, the influence of end effects can be eliminated from the position estimation result or velocity estimation result with a simple configuration. Using this position estimation system, high-performance control of the linear motor 100 can be realized at low cost.

[0097] Embodiment 4 In Embodiment 3, the case where the speed is constant was discussed, so the speed calculator was a Type 2 controller. In Embodiment 4, an example of configuring the speed calculator to handle the case where the acceleration is constant will be described. Figure 14 is a block diagram showing an example in which the speed calculator of the movable element position estimation means is configured as a Type 3 in the drive control device according to Embodiment 4, and Figure 15 is a block diagram showing the configuration of the proportional-integral-double integrator of the speed calculator.

[0098] Except for the fact that the velocity calculator is configured as a Type 3, the details are the same as in Embodiments 1 to 3. Therefore, the explanation of the similar parts will be omitted, and Figures 1 to 5, 7, and 9 used in Embodiment 1, and Figure 11 used in Embodiment 2 will be referenced.

[0099] A constant acceleration is an example of a pattern in which velocity gradually increases or decreases. In this case, if we want to make the steady-state difference between the input and output of the filter zero, a Type 3 controller is required. Therefore, in this embodiment 4, as shown in Figure 14, the velocity calculator 64 is equipped with a proportional-integral double integral calculator 644 instead of the proportional-integral calculator 642 shown in embodiment 3.

[0100] This section explains the difference between the proportional-integral unit 642 and the proportional-integral double integrator 644. As shown in Figure 15, the proportional-integral double integrator 644 includes a proportional unit 6441, an integral unit 6442, a double integrator 6443, and an adder 6444. The proportional-integral double integrator 644 is equivalent to the proportional-integral unit 642 with the addition of the double integrator 6443. The double integrator 6443 outputs the result of integrating the input signal and then integrating that signal again. The output of the double integrator 6443 is multiplied by a coefficient as appropriate to stabilize the controller. The adder 6444 calculates the sum of the outputs of the proportional unit 6441, the integral unit 6442, and the double integrator 6443, and the result of this calculation is the output of the proportional-integral double integrator 644.

[0101] We will also explain the mathematical differences. The transfer function of the proportional-integral double integral operator 644 is expressed as shown in equation (15).

[0102]

number

[0103] Comparing equation (9) and equation (15), equation (15) contains the double integral term K. II / s 2 A has been added. In Embodiment 4, this double integral term and the integrator 643 are connected in series, so the number of integrators connected in series in the velocity calculator 64 is 3. Therefore, this is a type 3 controller. According to the internal model principle, it is known that a type 3 controller can make the steady-state error for a constant acceleration input zero. However, for this to be possible, the proportional gain K P , integral gain K I Double integral gain K II It is necessary to design it properly and ensure that the system is stable.

[0104] An example of a gain design method for the 644 proportional-integral double integral calculator is shown below. In the velocity calculator 64, the second estimated position hat θ is calculated from the first estimated position hat θ. F The transfer function up to this point can be expressed as shown in equation (16).

[0105]

number

[0106] In this system as well, the input to integrator 643 can be considered as the estimated velocity. Here, the transfer function from the first estimated position hat θ to the estimated velocity dot hat θ can be expressed as shown in equation (17).

[0107]

number

[0108] There are several possible polar configurations, but for example, the triple root polar configuration is commonly used in cubic systems, so it is good to use this. In the case of the triple root polar configuration, the denominator polynomial is as shown in equation (18).

[0109]

number

[0110] Here, -ω n ω is the pole of the system. To ensure the stability of the system, n Assume that > ​​0. This system has three poles, but all three poles coincide at a single point on the real axis. That point of coincidence is -ω n This type of polar configuration is called a "triple root polar configuration."

[0111] Once the poles of the system are determined, the values ​​of each gain can be found by comparing the coefficients in equation (18). The gains such that the poles of this system have triple roots are shown in equation (19).

[0112]

number

[0113] In the case of the triple root pole configuration described above, the system is stable because the real axis component of all poles is negative. If the system is stable, other pole configuration methods other than the triple root can be used. For example, if the gain design is performed using a third-order Butterworth pole configuration, the gain can be determined as shown in equation (20).

[0114]

number

[0115] In the above design, the pole is -ω n ,-ω n It is located at exp(±jπ / 3). In this case as well, the real axis component of all poles is negative, so the system is stable.

[0116] The gain can be designed with some degree of freedom, but to stabilize this system, at least K is required. P >0, K I It must be >0. Therefore, the proportional term or single integral term cannot be omitted from the proportional-integral double integral operator 644.

[0117] As described above, the velocity calculator 64, which has a properly designed proportional-integral double integral calculator 644, has the characteristic that no steady-state error occurs before and after the filter input / output, even when the acceleration is constant. Therefore, by using the movable element position estimation means 6 according to Embodiment 4, the influence of end effects can be eliminated from the position estimation result or velocity estimation result with a simple configuration. Using this position estimation system, high-performance control of the LSM (linear motor 100) can be realized at low cost.

[0118] In addition, by applying the method disclosed in the present application, it is possible to handle even cases where the change in the position or speed of the mover 1 is steeper. For example, when it is desired to make the steady-state deviation zero in a case where the acceleration changes at a constant rate, a type-4 controller may be employed, that is, a proportional-integral-double-integral-triple-integral calculator may be used. Similarly, controllers of type 5 or higher can also be considered.

[0119] Although various exemplary embodiments and examples are described in the present application, the various features, aspects, and functions described in one or more of the embodiments are not limited to the application to a specific embodiment, but are applicable to the embodiments alone or in various combinations. Therefore, countless variations not illustrated are assumed to be within the scope of the technology disclosed in the specification of the present application. For example, it includes cases where at least one component is modified, added, or omitted, and further cases where at least one component is extracted and combined with components of other embodiments.

[0120] As described above, according to the drive control device 8 of the present application, for a non-three-phase linear synchronous motor (linear motor 100) composed of a mover 1 and a stator 2 in which a plurality of coils 3 are arranged to form a travel path of the mover 1, a power supply device (coil current control means 5) that individually controls the current of each of the plurality of coils 3, a motor constant (armature resistance R s , inductance matrix L, etc.) of the linear synchronous motor (linear motor 100) and information on current and voltage for each coil 3 obtained from the power supply device (coil current control means 5) (voltage command vector V * s , current vector I s ), an induced voltage calculation unit (speed electromotive force calculation unit 61) that calculates the induced voltage (speed electromotive force E s ) generated in each of the plurality of coils 3, and the induced voltage (speed electromotive force E sThe LSM includes a position calculator 63 that calculates the magnetic pole position (estimated position hat θ) of the movable element 1 on the track from the position calculator 6, a movable element position estimation means 6 that estimates at least one of the magnetic pole position (estimated position hat θ) and the travel speed of the movable element 1 (estimated speed dot hat θ), and a control calculation means 7 that controls the operation of the power supply unit (coil current control means 5) based on the estimation results output from the movable element position estimation means 6. This makes it possible to estimate the position of the movable element 1 without a position sensor and drive it efficiently, even in a non-three-phase LSM where the movable element 1 spans multiple stators 2.

[0121] The movable element position estimation means 6 determines the induced voltage (speed electromotive force E) generated in adjacent coils 3 among the multiple coils 3. s Using the phase difference of ) a coordinate transformation matrix is ​​used to generate the induced voltage (rate electromotive force E s ) is a two-phase sine wave (velocity electromotive force E αβ The position calculator 63 has a multiphase two-phase converter 62 that performs coordinate transformation to a two-phase sine wave (speed electromotive force E αβ By configuring the system to calculate the magnetic pole position (estimated position hat θ) based on ), the position of the movable element 1 in a non-three-phase LSM can be reliably estimated.

[0122] The position calculator 63 generates a two-phase sine wave (velocity electromotive force E αβ The device has an inverse tangent calculation unit (inverse tangent calculator 631) that calculates the inverse tangent of the four quadrants, and by calculating the magnetic pole position (estimated position hat θ) from the inverse tangent of the four quadrants, the position of the movable element 1 can be estimated with high accuracy.

[0123] The position calculator 63 has an edge detection unit 632 that detects sawtooth wave edges obtained by calculating the four-quadrant inverse tangent. By calculating the magnetic pole position (estimated position hat θ) from the sawtooth wave edges, the approximate position of the movable element 1 can be grasped and the coil 3 to be energized can be carefully selected, thus enabling more efficient driving.

[0124] The mover position estimation means 6 selects a coil 3 that the mover 1 is approaching among the plurality of coils 3 (for example, a coil 3 within a range less than one electrical angle of the coil 3 outward from both ends in the traveling direction directly below the mover 1), and narrows down to the selected coil 3 to output information on the current and voltage for each coil 3 (voltage command V * sn , current I sn ) to the induced voltage calculation unit (speed electromotive force calculation unit 61). By having a proximity coil selector 65, even when the track becomes long, the position of the mover 1 can be estimated without a position sensor without excessive computational complexity.

[0125] The mover position estimation means 6 is configured to form a type-2 or higher controller using an integrator, and has a speed calculator 64 that calculates a second magnetic pole position (second estimated position hat θ F ) obtained by removing an error from the magnetic pole position (estimated position hat θ) output from the position calculator 63 and the traveling speed (estimated speed dot hat θ). By doing so, the position estimation error and speed estimation error due to the end effect can be removed, and a high-quality estimation signal can be obtained.

[0126] Further, according to the linear motor system of the present application, since it includes a linear synchronous motor (linear motor 100) in which the currents of each of the plurality of coils 3 are individually controlled by the above-described drive control device 8 and power supply device (coil current control means 5), the position of the mover 1 in a non-three-phase LSM can be estimated without a position sensor and driven efficiently.

[0127] Further, according to the drive control method of the present application, for a non-three-phase linear synchronous motor (linear motor 100) composed of a mover 1 and a stator 2 in which a plurality of coils 3 are arranged to form a track for the mover 1, information on the current and voltage for each coil 3 (voltage command vector V * s , current vector I s ) and the motor constants (armature resistance R s , inductance matrix L, etc.) of the linear synchronous motor (linear motor 100), the induced voltage (speed electromotive force E s) an induced voltage calculation step, calculate the induced voltage (speed electromotive force E) generated in adjacent coils among multiple coils 3 s Using a coordinate transformation matrix based on the phase difference of ), the induced voltage is converted to a two-phase sinusoidal (speed electromotive force E) αβ A multiphase two-phase conversion step (step S110) that transforms the coordinates to a two-phase sine wave (speed electromotive force E αβ The system includes a position calculation step (step S150) which calculates the magnetic pole position (estimated position hat θ) of the movable element 1 on the track from the calculated magnetic pole position (estimated position hat θ), and a control step which individually controls the current of each of the multiple coils based on the calculated magnetic pole position (estimated position hat θ). This makes it possible to estimate the position of the movable element 1 without a position sensor and drive it efficiently, even in a non-three-phase LSM where the movable element 1 spans multiple stators 2. [Explanation of Symbols]

[0128] 1: Movable element, 100: Linear motor (linear synchronous motor), 2: Stator, 3: Coil, 5: Coil current control means (power supply device), 6: Movable element position estimation means, 61: Speed ​​electromotive force calculation unit (induced voltage calculation unit), 62: Multiphase two-phase converter, 63: Position calculator, 631: Arcthine tangent calculator, 632: Edge detection unit, 633: Offset amount calculation unit, 64: Velocity calculation vessel, Es: Reaction electromotive force (induced voltage), Eαβ: Reaction electromotive force (two-phase sine wave), Is: Current vector, Isn: Current, V*s: Voltage command vector, V*sn: Voltage command, Hat θ: Estimated position (magnetic pole position), Hat θF: Second estimated position (second magnetic pole position), Dot hat θ: Estimated speed (traveling speed).

Claims

1. A power supply device for a non-three-phase linear synchronous motor, which consists of a movable element and a stator in which multiple coils are arranged to form a path for the movable element, that individually controls the current of each of the multiple coils. The system includes an induced voltage calculation unit that calculates the induced voltage generated in each of the plurality of coils from the motor constants of the linear synchronous motor and the current and voltage information for each coil obtained from the power supply unit, a position calculation unit that calculates the magnetic pole position of the movable element on the track from the induced voltage, and a movable element position estimation means that estimates at least one of the magnetic pole position and the travel speed of the movable element, and The system includes a control calculation means for controlling the operation of the power supply based on the estimation results output from the movable element position estimation means, The movable element position estimation means includes a multiphase two-phase converter that transforms the induced voltage into a two-phase sinusoidal wave using a coordinate transformation matrix that utilizes the phase difference of the induced voltages occurring in adjacent coils among the plurality of coils. The drive control device is characterized in that the position calculator calculates the magnetic pole position based on the two-phase sinusoidal wave.

2. The drive control device according to claim 1, wherein the position calculator has an inverse tangent calculation unit that calculates the four quadrants inverse tangent of the two-phase sinusoidal wave, and calculates the magnetic pole position from the four quadrants inverse tangent.

3. The drive control device according to claim 2, wherein the position calculator has an edge detection unit that detects sawtooth wave edges obtained by the calculation of the four quadrants inverse tangent, and calculates the magnetic pole position from the sawtooth wave edges.

4. The drive control device according to claim 1, characterized in that the movable element position estimation means has a proximity coil selector that selects a coil from among the plurality of coils to which the movable element is close, and outputs current and voltage information for each coil to the induced voltage calculation unit, focusing on the selected coil.

5. The movable element position estimation means is The drive control device according to claim 1, comprising a controller of type 2 or more using an integrator, and a speed calculator that calculates the second magnetic pole position obtained by removing errors from the magnetic pole position output from the position calculator and the travel speed.

6. A drive control device according to any one of claims 1 to 5, and The linear synchronous motor, in which the current of each of the multiple coils is individually controlled by the power supply, A linear motor system characterized by having the following features.

7. For a non-three-phase linear synchronous motor consisting of a movable element and a stator in which multiple coils are arranged to form a path for the movable element, an induced voltage calculation step is performed to calculate the induced voltage generated in each of the multiple coils from the current and voltage information of each coil and the motor constants of the linear synchronous motor. A multiphase two-phase transformation step in which the induced voltage is transformed into a two-phase sinusoidal wave using a coordinate transformation matrix that utilizes the phase difference of the induced voltages occurring in adjacent coils among the plurality of coils, A position calculation step of calculating the magnetic pole position of the movable element on the track from the two-phase sinusoidal wave, and A control step of individually controlling the current in each of the plurality of coils based on the calculated magnetic pole position, A drive control method characterized by including the following.

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