Vibration amplification circuit and power transmission system

Abstract

To provide a vibration amplification circuit enabling voltage enlargement of amplitude of vibration to power supply voltage or more, and a power transmission system capable of effectively transmitting power using double resonance even when the vibration amplification circuit is used in an electric supply side resonance circuit and a transmission distance is separated.SOLUTION: A vibration amplification circuit comprises a DC power supply 5, an exciting element Q1 of which one terminal is connected with a high potential side terminal of the DC power supply 5, a recirculation control element Q2 of which one terminal is connected with the other terminal of the exciting element Q1 and which performs switching operation in a complementary manner for the exciting element Q1, a transmission side capacitor C1 of which one terminal is connected with a connection node of the other terminal of the exciting element Q1 and the one terminal of the recirculation control element Q2, and a transmission side coil L1 of which one terminal is connected with the other terminal of the transmission side capacitor C1 and of which the other terminal is connected with a connection node of the other terminal of the recirculation control element Q2 and a low potential side terminal of the DC power supply 5.SELECTED DRAWING: Figure 2

Description

[Technical Field]

[0001] The present invention relates to a contactless power transmission system (wireless power transmission system) that does not use expensive AC power supply circuits such as switching power supplies in the power supply circuit, and more particularly to a vibration amplification circuit based on a novel idea of ​​inversely amplifying damped vibrations, and a power transmission system that uses this vibration amplification circuit as a power supply side resonant circuit and transmits electrical energy from the power supply side resonant circuit to the power receiving side resonant circuit by mutual resonance (double resonance) between the power supply side resonant circuit and the power receiving side resonant circuit. [Background technology]

[0002] The present inventors proposed a contactless power transmission device that does not use expensive AC power supply circuits such as switching power supplies in the power supply circuit, based on a non-AC theory that focuses on transient response and considering the double resonance between the power supply side resonant circuit and the power receiving side resonant circuit (see Patent Document 1). However, the invention described in Patent Document 1 has the problem that it is difficult to send power to a distant power receiving side resonant circuit because the power supply side resonant circuit can oscillate up to the power supply voltage but not beyond that. For example, in the invention described in Patent Document 1, there was a tendency for effective power transmission to become impossible when the distance (transmission distance) between the power sending side coil and the power receiving side coil was 40 mm or more. In addition, in the contactless power transmission using double resonance described in Patent Document 1, there was a problem that the vibration amplitude was attenuated by the parasitic resistance inherent in the resonant circuit. [Prior art documents] [Patent Documents]

[0003] [Patent Document 1] World Intellectual Property Organization International Bureau International Publication 2020 / 039594 Pamphlet [Overview of the project] [Problems that the invention aims to solve]

[0004] In view of the above problems, the present invention aims to provide a vibration amplification circuit that can amplify the vibration amplitude beyond the magnitude of the power supply voltage to an integer multiple of the power supply voltage, and a power transmission system that can effectively transmit power using double resonance even when the distance between the transmitting coil and the receiving coil is 40 mm or more, by using this vibration amplification circuit in the power supply side resonant circuit. [Means for solving the problem]

[0005] The gist of the present invention is an oscillatory amplification circuit comprising (a) a DC power supply, (b) an excitation element with one terminal connected to the high-potential side terminal of the DC power supply, (c) a freewheeling control element with one terminal connected to the other terminal of the excitation element and switching in a complementary manner to the excitation element, (d) a power-transmitting capacitor with one terminal connected to the connection node between the other terminal of the excitation element and one terminal of the freewheeling control element, and (e) a power-transmitting coil with one terminal connected to the other terminal of the power-transmitting capacitor, and the other terminal connected to the connection node between the other terminal of the freewheeling control element and the low-potential side terminal of the DC power supply.

[0006] A second aspect of the present invention is a power transmission system comprising: (a) a DC power supply; (b) an excitation element with one terminal connected to the high-potential terminal of the DC power supply; (c) a freewheeling control element with one terminal connected to the other terminal of the excitation element and switching in a complementary manner to the excitation element; (d) a transmitting capacitor with one terminal connected to the connection node between the other terminal of the excitation element and one terminal of the freewheeling control element; (e) a transmitting coil with one terminal connected to the other terminal of the transmitting capacitor, and the other terminal connected to the connection node between the other terminal of the freewheeling control element and the low-potential terminal of the DC power supply; (f) a receiving coil positioned spaced apart from the transmitting coil and receiving magnetic energy from the transmitting coil without contact; (g) a receiving capacitor connected in parallel to the receiving coil and storing the magnetic energy accumulated in the receiving coil as electrostatic energy; and (h) a load connected in parallel with the receiving coil. [Effects of the Invention]

[0007] According to the present invention, a vibration amplification circuit capable of expanding the vibration amplitude beyond the magnitude of the power supply voltage to an integer multiple of the power supply voltage, and a power transmission system that can effectively transmit power even when the distance between the transmitting coil and the receiving coil is exceptionally large, such as 600 mm or more, can be provided by using this vibration amplification circuit in the power supply side resonant circuit. [Brief explanation of the drawing]

[0008] [Figure 1A] This is a schematic diagram showing the general structure of an example of a power transmission system according to the first embodiment of the present invention. [Figure 1B] This is an equivalent circuit illustrating an example of the load for the circuit shown in Figure 2. [Figure 2] This is a schematic circuit diagram showing the primary and secondary circuits that constitute an example of a power transmission system according to the first embodiment. [Figure 3] Figure 2 shows waveform diagrams illustrating the changes in the terminal voltage of the transmitting side capacitor and the terminal voltage of the transmitting side coil in the primary circuit shown. [Figure 4A] This is a schematic diagram illustrating the initial state of the primary circuit of the power transmission system according to the first embodiment, where the excitation element is turned on (conducting state) and the freewheel control element is turned off (blocked state) at time t=t0=0. [Figure 4B] Following Figure 4A, this is a schematic diagram illustrating the transfer of electromagnetic energy in the circuit state set for the first excitation cycle, where the excitation element of the primary circuit is in the ON state and the freewheel control element is in the OFF state in the time domain t0 < t < t1. [Figure 4C] Following Figure 4B, this is a schematic diagram illustrating the complementary operation in which the excitation element of the primary circuit transitions to off and the freewheel control element transitions to on at time t1. [Figure 4D] Following Figure 4C, this is a schematic diagram illustrating the transfer of electromagnetic energy in the circuit state set for the first recirculation cycle, where the excitation element of the primary circuit is in the off state and the recirculation control element is in the on state in the time domain t1 < t < t2. [Figure 4E]Following Figure 4D, this is a schematic diagram illustrating a complementary transition in which, at time t2, the excitation element of the primary circuit turns on and the freewheel control element turns off, initiating the second excitation cycle. [Figure 5A] This figure shows the waveforms indicating the changes in the terminal voltage of the transmitting capacitor and the transmitting coil in the primary circuit shown in Figure 2, extended from the timing shown in Figure 3 to the timing t3, and also includes the waveform showing the change in the terminal voltage of the receiving coil in the secondary circuit. [Figure 5B] This waveform diagram shows the changes in the terminal voltages of the transmitting capacitor, the transmitting coil, and the receiving coil, as shown in Figure 5A, extended to the timing of the transmission mode. [Figure 6] This is a schematic bird's-eye view illustrating another example of a spacing control mechanism for adjusting the plane spacing (transmission spacing) between coils when the power transmission system according to the first embodiment is applied to charging the battery of an electric vehicle (EV). [Figure 7] This is a schematic circuit diagram showing the primary and secondary circuits that constitute an example of a power transmission system according to a first modification of the first embodiment of the present invention. [Figure 8] This is a schematic circuit diagram showing the primary and secondary circuits that constitute an example of a power transmission system according to a second modification of the first embodiment of the present invention. [Figure 9] This is a schematic circuit diagram showing the primary and secondary circuits that constitute an example of a power transmission system according to a third modified example of the first embodiment of the present invention. [Modes for carrying out the invention]

[0009] Next, a representative embodiment of the present invention will be described as the first embodiment with reference to the drawings. In the following drawings, identical or similar parts are denoted by the same or similar reference numerals. However, it should be noted that the drawings are schematic, and the relationship between thickness and planar dimensions, the ratio of the thicknesses of each component, etc., may differ from reality. Therefore, specific thicknesses and dimensions should be determined by referring to the following explanation. Furthermore, it goes without saying that there are parts where the relationships and ratios of dimensions differ between drawings.

[0010] Furthermore, the first embodiment shown below illustrates an apparatus and method for embodying the technical concept of the present invention, and the technical concept of the present invention does not limit the material, shape, structure, arrangement, etc. of the components to those described below. The technical concept of the present invention can be modified in various ways within the technical scope defined by the claims described in the patent claims. Moreover, the directions of "left and right" and "up and down" in the following description are merely definitions for the convenience of explanation and do not limit the technical concept of the present invention. Therefore, for example, if the paper is rotated 90 degrees, "left and right" and "up and down" are swapped when read, and of course, if the paper is rotated 180 degrees, "left" becomes "right" and "right" becomes "left". Similarly, the direction of the spiral helix as shown in Figure 6 is merely a selection for the convenience of explanation, and it is possible to select right-handed to left-handed or left-handed to right-handed depending on the actual design circumstances.

[0011] (First Embodiment) The power transmission system according to the first embodiment of the present invention is a contactless power transmission system that supplies wavelet-shaped electromagnetic energy to a vehicle 31a having a power receiving circuit 27a via double resonance from a power supply device 29a, as shown in Figure 1A. "Wavelet-shaped electromagnetic energy" refers to a packet of electromagnetic energy that exhibits time-localized vibration characteristics, rather than steady sinusoidal vibrations. The power receiving circuit 27a includes a load (storage battery) 6. The power transmission system according to the first embodiment includes a power supply device 29a that supplies wavelet-shaped electromagnetic energy to the power receiving circuit 27a without contact, and a primary-side operation unit 33 connected to the power supply device 29a that sends commands to the power supply device 29a.

[0012] Various structures and mechanisms can be employed in the primary side operation unit 33; for example, the primary side operation unit 33 may be equipped with an imaging device. In the embodiment equipped with an imaging device, a mechanism can be provided that automatically associates the vehicle height of the vehicle 31a with the image of the vehicle 31a captured by the imaging device using an AI function. Figure 1A illustrates a schematic diagram showing that the transmitting coil L1 on the power supply device 29a side and the receiving coil L2 on the vehicle 31a side face each other, and that wavelet-shaped electromagnetic energy is transmitted from the transmitting coil L1 to the receiving coil L2 without contact.

[0013] As shown in Figure 1A, the power supply device 29a mainly consists of a power supply panel 11 in which the power transmitting coil L1 is housed in a disc-shaped dielectric, a spacing control mechanism 32 mounted on the power supply panel 11 that controls the spacing between the power transmitting coil L1 and the power receiving coil L2, a drive control circuit 34a that controls the power supply current flowing to the power transmitting coil L1 and the spacing control mechanism 32, a transmission data storage device 342a and a program storage device 342b connected to the drive control circuit 34a, and the power supply panel 11 whose transmission current is controlled by the drive control circuit 34a. The drive control circuit 34a and the power receiving circuit 27a send and receive wavelet-shaped electromagnetic energy to each other via the power transmitting coil L1 and the power receiving coil L2, causing double resonance. In the embodiment shown in Figure 1A, the spacing control mechanism 32 is an up-and-down movement mechanism, and various well-known mechanisms can be employed, such as a hydraulic up-and-down mechanism, an electromagnet-based up-and-down mechanism, or a movement mechanism that rotates a ball spiral with a stepper motor. On the other hand, in the embodiment shown in Figure 6, the spacing control mechanism 32 is a horizontal movement mechanism, but similarly, various mechanisms such as a hydraulic horizontal movement mechanism, an electromagnet-based horizontal movement mechanism, and a ball-helical horizontal movement mechanism can be employed.

[0014] Figure 1A is an example, and it is also possible to omit the power supply panel 11 that houses the power transmitting coil L1 and use the power transmitting coil L1 in an exposed state. In Figure 1A, the power receiving coil L2 is also illustrated as being housed in a power receiving panel 12 made of a disc-shaped dielectric, but it is also possible to omit the power receiving panel 12 that houses the power receiving coil L2 and use the power receiving coil L2 in an exposed state. Power is supplied from the power transmitting coil L1 to the power receiving coil L2 by double resonance of the power supply side resonant circuit and the power receiving side resonant circuit.

[0015] The power supply panel 11 is installed or buried in the ground so that its upper surface is parallel to the lower surface of the power receiving panel 12. In the state before power supply work, the upper surface of the power supply panel 11 is parallel to the flat surface 30 on the ground, so that a vehicle 31a can drive on a uniform flat surface and enter. The power supply device 29a is installed, for example, in a parking space, and while the vehicle 31a is parked, it faces the power receiving panel 12 and supplies wavelet-shaped electromagnetic energy to the power receiving panel 12 mounted on the vehicle 31a by double resonance.

[0016] Load 6 is a battery represented by an equivalent circuit as shown in Figure 1B, and stores wavelet-shaped electromagnetic energy supplied from the power supply device 29a via the power receiving panel 12 by double resonance. The vehicle 31a is, for example, a hybrid electric vehicle (HEV), a plug-in electric vehicle (PEV), or an electric vehicle (EV), and runs on the electromagnetic energy stored in the battery as load 6. The primary side operation unit 33 outputs a power supply start signal indicating the start of power supply or a power supply stop signal indicating the stop of power supply to the power supply device 29a based on operation from outside the vehicle 31a. If the primary side operation unit 33 determines the vehicle height of the vehicle 31a using the AI ​​function, it also transmits the vehicle height data of the vehicle 31a to the drive control circuit 34a of the power supply device 29a.

[0017] The drive control circuit 34a controls the power supply panel 11 and performs various drive controls related to the movement of electromagnetic energy due to double resonance between the power supply side resonant circuit and the power receiving side resonant circuit. For example, when a power supply start signal is input from the primary side operation unit 33, the drive control circuit 34a controls the current of the power supply panel 11 to supply wavelet-shaped electromagnetic energy that oscillates at a set period. The drive control circuit 34a also includes an arithmetic logic circuit (ALU) that acquires the oscillation characteristics of the electromagnetic energy returned from the power receiving circuit 27a and calculates the conditions under which the transmission efficiency due to double resonance between the power supply panel 11 and the power receiving panel 12 is maximized. The transmission data storage device 342a and program storage device 342b shown in Figure 1A are connected to the ALU.

[0018] The drive control circuit 34a selects a drive timing to perform complementary switching of the excitation element Q1 and the freewheel control element Q2 shown in Figure 2, and outputs a command to operate the primary-side switching element drive circuit 340a shown in Figure 2 at the selected drive timing. The primary-side switching element drive circuit 340a sends control signals to the control terminals of the excitation element Q1 and the freewheel control element Q2 shown in Figure 2, and drives the excitation element Q1 and the freewheel control element Q2 to switch on and off complementaryly. If the excitation element Q1 is a thyristor such as a field-effect transistor (FET), electrostatic induction transistor (SIT), gate turn-off thyristor (GTO), or electrostatic induction thyristor (SI thyristor), the gate electrode of these power semiconductor elements corresponds to the "control terminal" of the excitation element Q1. If the excitation element Q1 is a bipolar transistor (BJT), the base electrode of the BJT becomes the control terminal of the excitation element Q1. Similarly, if the freewheel control element Q2 is an FET, SIT, GTO thyristor, SI thyristor, etc., the gate electrode of these power semiconductor elements corresponds to the "control terminal" of the freewheel control element Q2. If the freewheel control element Q2 is a BJT, the base electrode of the BJT becomes the control terminal of the freewheel control element Q2.

[0019] The ALU can be configured using a microprocessor (MPU) implemented as a microchip to form a computer system. Alternatively, the ALU in the computer system may be a digital signal processor (DSP) with enhanced arithmetic capabilities and specialized for signal processing, or a microcontroller (MCU) equipped with memory and peripheral circuits for embedded device control. Or, the main CPU of a current general-purpose computer may be used as the ALU. Furthermore, some or all of the ALU's configuration may be made up of a programmable logic device (PLD) such as a field-programmable gate array (FPGA).

[0020] The computer system is configured including the ALU included in the drive control circuit 34a shown in Figure 2. In the computer system including the drive control circuit 34a, the transmission data storage device 342a can be any combination appropriately selected from a group including multiple registers, multiple cache memories, main memory, and auxiliary storage. Furthermore, the cache memory may be a combination of primary and secondary cache memories, and may also have a hierarchy including a tertiary cache memory. If the PLD constitutes part or all of the ALU, the transmission data storage device 342a can be configured as a memory element such as a memory block included in part of the logical blocks that constitute the PLD. Furthermore, the ALU may have a structure in which a CPU core-like array and a PLD-like programmable core are mounted on the same chip. This CPU core-like array includes a hard macro CPU pre-mounted inside the PLD and a soft macro CPU configured using the logical blocks of the PLD. In other words, the PLD may have a configuration in which software processing and hardware processing are mixed inside.

[0021] The features of the power transmission system according to the first embodiment shown in Figure 1A can be represented by the power supply side resonant circuits (34a, 2a) and the power receiving side resonant circuits (27a, 3a), as shown in Figure 2. The power supply side resonant circuits (34a, 2a) are a circuit topology that forms part of the power supply device 29a shown in Figure 1A, and include a drive control circuit 34a and a second power supply side resonant circuit 2a. The power receiving side resonant circuits (27a, 3a) correspond to a structure including the power receiving circuit 27a and the power receiving panel 12 in Figure 1A, but are shown in Figure 2 as a circuit topology including the power receiving circuit 27a and the secondary side circuit 3a. As will be described later, in the power supply side resonant circuits (34a, 2a) shown in Figure 2, the "first power supply side resonant circuit" and the "second power supply side resonant circuit 2a" are formed transiently in a time-division manner in a complementary manner. Therefore, the power supply side resonant circuit (34a, 2a) in Figure 2 is represented as a combination of a second power supply side resonant circuit 2a, which is formed transiently in a time-division manner, and a drive control circuit 34a. The power supply side resonant circuit (34a, 2a) includes a DC power supply 5 that supplies a DC voltage of a constant voltage (power supply voltage) E0, an excitation element Q1 with one terminal connected to the high-potential side terminal (positive terminal) of the DC power supply 5, a freewheel control element Q2 with one terminal connected to the other terminal of the excitation element Q1 and which switches in a complementary manner to the excitation element Q1, and a power transmission side capacitor C1 with one terminal connected to the other terminal of the excitation element Q1. Since one terminal of the power transmission side capacitor C1 is also connected to one terminal of the freewheel control element Q2, one terminal of the power transmission side capacitor C1 is connected to the connection node between the other terminal of the excitation element Q1 and one terminal of the freewheel control element Q2. The other terminal of the transmitting capacitor C1 is connected to one terminal of the transmitting coil L1. The other terminal of the transmitting coil L1 is connected to the connection node between the other terminal of the freewheeling control element Q2 and the low-potential terminal (negative terminal) of the DC power supply 5.

[0022] The secondary circuit 3a of the receiving-side resonant circuit (27a,3a) has a receiving-side coil L2 that is spaced apart from and opposite to the transmitting-side coil L1 and receives magnetic energy from the transmitting-side coil L1 without contact, and a receiving-side capacitor C2 that is connected in parallel to the receiving-side coil L2 and stores the magnetic energy accumulated in the receiving-side coil L2 as electrostatic energy. As can be seen from Figure 2, the supply-side resonant circuit (34a,2a) is configured by adding the transmitting-side coil L1 to the drive control circuit 34a, so if we eliminate the overlapping parts in set theory notation, the supply-side resonant circuit (34a,2a) can be expressed as the supply-side resonant circuit (34a,L1). Similarly, the receiving-side resonant circuit (27a,3a) is configured by adding the receiving-side coil L2 to the receiving circuit 27a, so if we eliminate the overlapping parts notation, the receiving-side resonant circuit (27a,3a) can be expressed as the receiving-side resonant circuit (27a,L2). As shown in Figure 2, the receiving circuit 27a of the receiving-side resonant circuits (27a, 3a) has a configuration in which the series connection circuit of the load-side diode D2 and the load 6 is connected in parallel to the receiving-side capacitor C2 and the receiving-side coil L2, respectively.

[0023] In Figure 2, the equivalent internal resistance of the DC power supply 5 is shown as r1, and the parasitic resistance r of the power transmission capacitor C1 is shown. p1 It is assumed that such a thing exists. The other terminal of the transmitting coil L1 is connected to the other terminal of the excitation element Q1 and the low-potential terminal of the DC power supply 5. A detector 28, which acts as a voltmeter, is connected between one terminal of the transmitting coil L1 and the other terminal of the transmitting coil L1, and the transmitting coil voltage V is the terminal voltage of the transmitting coil L1. L1 The detector 28 can measure the voltage amplification effect of the power supply side resonant circuit (34a, 2a) as an oscillation amplification circuit, as well as the return voltage due to the electromagnetic energy that has returned from the receiving side resonant circuit (27a, 3a) to the power supply side resonant circuit (34a, 2a) in double resonance. The output of the detector 28 is fed back to the primary side switching element drive circuit 340a. However, the detector 28 is not an essential circuit element, and may be omitted depending on the design specifications.

[0024] In the implementation circuit shown in Figure 2, considering the freewheeling current from the power-transmitting coil L1, the first freewheeling diode FWD1 is connected in parallel between the source and drain of the MOSFET acting as the excitation element Q1, and the second freewheeling diode FWD2 is connected in parallel between the source and drain of the MOSFET acting as the freewheeling control element Q2, as protection elements. Similarly, to prevent the freewheeling current from the power-transmitting coil L1 from flowing back into the DC power supply 5, the power supply-side diode D1 is connected in series between the DC power supply 5 and the excitation element Q1.

[0025] As shown in Figure 2, the power supply side resonant circuit (34a, 2a) of the power transmission system according to the first embodiment is a ladder-type circuit in which the DC power supply 5, the freewheeling control element Q2, and the power transmission side coil L1 are arranged in parallel vertically like three steps of a ladder. The series circuit of the excitation element Q1 and the power transmission side capacitor C1 is configured to form one of the side frames of the horizontally laid ladder (the upper frame in the arrangement in Figure 2). One terminal of the freewheeling control element Q2, which is located corresponding to the central step of the three steps of the ladder, is connected to the connection node of the excitation element Q1 and the power transmission side capacitor C1, and the excitation element Q1, power transmission side capacitor C1, and freewheeling control element Q2 constitute a T-type circuit. The excitation element Q1 and the freewheeling control element Q2 perform complementary on / off operations so that the freewheeling current from the power transmission side coil L1 does not reach the DC power supply 5.

[0026] The transmitting capacitor C1 stores electrostatic energy supplied from the DC power supply 5 and magnetic energy supplied as a return current from the transmitting coil L1, and the primary side charge / discharge voltage V is the voltage across the transmitting capacitor C1. C1 The voltage is increased. The transmitting coil L1 stores the electrostatic energy sent from the transmitting capacitor C1 as magnetic energy, and at the same time returns this magnetic energy to the transmitting capacitor C1, it magnetically couples with the receiving coil L2 of the secondary circuit 3a, sending and receiving magnetic energy to and from the secondary circuit 3a.

[0027] Since the excitation element Q1 and the reflux control element Q2 perform complementary on / off operations, when the excitation element Q1 is in the conduction state, the reflux control element Q2 is in the cutoff state. When the excitation element Q1 is in the conduction state, a closed loop composed of the DC power supply 5, the excitation element Q1, the power transmission side capacitor C1, and the power transmission side coil L1 constitutes a transient "first power supply side resonance circuit" composed of an RLC series resonance circuit. The equivalent internal resistance r1 of the DC power supply 5 constituting the first power supply side resonance circuit, the parasitic resistance of the power transmission side capacitor C1, and the parasitic resistance of the power transmission side coil L1 are combined to form a parasitic resistance r p1 , and the on-resistance r on1 of the excitation element Q1 result in an equivalent floating resistance stray1 R1 = r on1 = r1 + r p1 ……(1) which is the resistance component of the RLC series resonance circuit.

[0028] Assuming the current transiently flowing through the first power supply side resonance circuit is i1, when the excitation element Q1 is in the conduction state, the transient phenomenon of the first power supply side resonance circuit can be

Equation

[0029] When the equivalent floating resistance R1 of the first power supply side resonance circuit shown in Equation (1) is sufficiently small, that is, when the attenuation constant α = R1 / (2L1) and the natural angular frequency ω0 = (L1C1) -1 / 2 are defined, α 2 < ω0 2 ……(3) When this holds, the solution to equation (2) is, as is well known in transient phenomena theory,

number

[0030] On the other hand, when the freewheel control element Q2 is conducting, the closed loop formed by the freewheel control element Q2, the transmitting capacitor C1, and the transmitting coil L1 constitutes a transient "second power supply side resonant circuit 2a" consisting of an RLC series resonant circuit. Therefore, as described above, in the power transmission system according to the first embodiment, the first power supply side resonant circuit and the second power supply side resonant circuit 2a are switched in a time-division manner by causing complementary switching operations of the excitation element Q1 and the freewheel control element Q2. The parasitic resistance r is the sum of the parasitic resistance of the transmitting capacitor C1 and the parasitic resistance of the transmitting coil L1. p1 , the on-resistance r of the freewheel control element Q2 on2 So, R²=r stray2 =r on2 +r p1 ...(5) However, this is the resistance component R2 of the RLC series resonant circuit that constitutes the second power supply side resonant circuit 2a.

[0031] If i2 is the transient current flowing through the second power supply side resonant circuit 2a, then the transient phenomenon of the second power supply side resonant circuit 2a when the freewheel control element Q2 is in the conducting state is:

number

[0032] If we differentiate both sides of equation (2) with respect to t and show the result in equation (7a), and differentiate both sides of equation (6) with respect to t and show the result in equation (7b), we can see that both result in differential equations of the same form.

number

number

[0033] The difference between equation (2) and equation (6) lies in the boundary conditions, and the coefficients A1 and A2 are determined by the boundary conditions. Therefore, in the time-division feeding side resonant circuits (34a, 2a), both the current i1 flowing through the first feeding side resonant circuit and the current i2 flowing through the time-division second feeding side resonant circuit 2a have a frequency (ω0 2 -α 2 ) 1 / 2This shows the waveform of under-damping (damped oscillation) which oscillates in a sinusoidal manner and decays exponentially. In the power supply side resonant circuits (34a, 2a) of the power transmission system according to the first embodiment, the excitation element Q1 and the freewheeling control element Q2 perform complementary on / off operations, and the initial vibrations of the time-divided damped oscillation of the first power supply side resonant circuit and the initial vibrations of the time-divided damped oscillation of the second power supply side resonant circuit 2a are alternately combined to create a "compound oscillation" which amplifies the vibration peak value in the power supply side resonant circuits (34a, 2a), and the power supply side resonant circuits (34a, 2a) function as a "vibration amplification circuit". In reality, since only the initial vibrations of the damped oscillation are isolated and alternately combined, the compound oscillation does not become a damped oscillation but rather an "amplified oscillation" in which the voltage is amplified.

[0034] The excitation element Q1 and the freewheeling control element Q2 control the process of cutting out the initial vibrations of a damped oscillation, as exemplified by equation (4), and alternately connecting them in the first power supply side resonant circuit and the second power supply side resonant circuit 2a to generate a composite oscillation. That is, the excitation element Q1 and the freewheeling control element Q2 alternately repeat the operation of the excitation cycle in the first power supply side resonant circuit and the freewheeling cycle in the second power supply side resonant circuit 2a, generating a composite oscillation in the power supply side resonant circuits (34a, 2a) to amplify the voltage. When the oscillation peak value of the composite oscillation reaches the desired value, as shown in Figure 5B, they switch to the transmission mode and transmit the electrical energy of the high-voltage composite oscillation generated in the power supply side resonant circuits (34a, 2a) to the receiving side resonant circuits (27a, 3a). In "transmission mode," the excitation element Q1 is always in an interrupted state, and the freewheeling control element Q2 is always in a conductive state, resulting in complementary operation.

[0035] Thus, in the "amplification mode," the complementary repetitive operation of the excitation element Q1 and the freewheel control element Q2 is performed a specific number of times to amplify the signal to a desired amplitude value. In the "transmission mode," the primary-side switching element drive circuit 340a drives and controls the operation to transmit the electrical energy of the power supply side resonant circuit (34a, 2a), which has been amplified to a desired amplitude value in the amplification mode, to the power receiving side resonant circuit (27a, 3a) using double resonance. Specifically, the primary-side switching element drive circuit 340a uses the power supply side resonant circuit (34a, 2a) as an "oscillation amplification circuit" to generate wavelet-shaped high-voltage electromagnetic energy, and further drives and controls the operation to cause double resonance between the resonance in the power supply side resonant circuit (34a, 2a) and the resonance in the power receiving side resonant circuit (27a, 3a). In the amplification mode, when the excitation element Q1 is conducting, it is an "excitation cycle," and when the freewheel control element Q2 is conducting, it is a "freewheel cycle." In amplification mode, excitation cycles and freewheeling cycles alternate periodically. The repeated operation of excitation and freewheeling cycles in amplification mode amplifies the power supply voltage E0 to a voltage greater than the power supply voltage E0, resulting in large-amplitude oscillations. The energy of these large-amplitude electrical oscillations in the power supply side resonant circuit (34a, 2a) is transmitted to the secondary side circuit 3a in transmission mode.

[0036] However, even in amplification mode, there is transmission due to leakage of electrical energy from the power supply side resonant circuit (34a, 2a) to the secondary side circuit 3a. Here, "power supply voltage E0" is the terminal voltage E0 of the DC power supply 5 shown in Figure 2. In amplification mode, the repeated operation of excitation cycle and freewheel cycle causes the damped oscillation of the RLC series resonant circuit to be amplified so that the oscillation peak is gradually enlarged. That is, the ALU constituting the drive control circuit 34a has a natural angular frequency ω0=(L1C1) so that the primary side switching element drive circuit 340a can repeatedly operate the excitation cycle and freewheel cycle at the desired timing. -1 / 2The ALU can perform logical operations to determine the following, using the results of preliminary experiments as necessary data. The ALU can further perform logical operations to determine the number of excitation cycles and freewheeling cycles in amplification mode, using the results of preliminary experiments as data. The ALU can further perform logical operations to determine the timing for switching to transmission mode simultaneously with stopping amplification mode, the duration of transmission mode, and the timing for returning from transmission mode to amplification mode, using the results of preliminary experiments as data. The ALU then transmits the results of the logical operations as commands to the primary-side switching element drive circuit 340a. Furthermore, the ALU constituting the drive control circuit 34a controls the drive control circuit 34a by transmitting necessary commands to the drive control circuit 34a so that the power supply side resonant circuits (34a, 2a) and the power receiving side resonant circuits (27a, 3a) can achieve double resonance via the power transmitting side coil L1 and the power receiving side coil L2.

[0037] In the power transmission system according to the first embodiment, the power supply side resonant circuits (34a, 2a) cut off the transient damping oscillations inherent in the RLC series resonant circuit at their initial oscillations before they attenuate, through the complementary switching operation of the excitation element Q1 and the freewheeling control element Q2. In the power supply side resonant circuits (34a, 2a), the damping oscillations can be converted into amplified oscillations that are gradually amplified to a specific amplitude value larger than the power supply voltage E0, thereby realizing large-amplitude wavelet-shaped electromagnetic energy oscillations. Therefore, according to the power transmission system according to the first embodiment, as illustrated in Figure 6, effective power transmission is possible even when the distance (transmission distance) d between the transmitting coil and the receiving coil is 600 mm or more. The DC power supply 5 that supplies the power supply voltage E0 may be a pseudo constant voltage source, and may have a simple structure that is merely rectified, and may be a power supply with a waveform containing a large ripple component. In other words, the DC power supply 5 does not require precise and complex circuits, such as complex smoothing circuits, to minimize ripple components. The control circuit and peripheral circuits of the DC power supply 5 are simple, durable, and inexpensive. Furthermore, since the power supply side resonant circuits (34a, 2a) and the double resonance phenomenon do not rely on AC theory, high-frequency inverters and switching power supplies that generate highly accurate AC components are not required. Thus, because complex and expensive circuits such as expensive constant voltage sources, high-frequency inverters, and switching power supplies are not used, the control circuit and peripheral circuits of the power supply side resonant circuits (34a, 2a) are simple and durable. Therefore, the power supply side resonant circuits (34a, 2a) that can employ the inexpensive DC power supply 5 are easy to design and inexpensive.

[0038] The load 6 constituting the power receiving circuit 27a of the power receiving side resonant circuit (27a, 3a) can be a rechargeable battery such as the on-board lithium (Li) ion battery of the vehicle 31a exemplified in Figure 1A. In Figure 1B, the equivalent circuit of a lithium-ion battery is schematically shown as a series-parallel circuit of resistors and capacitors. A lithium-ion battery includes resistance of the current collector and electrolytic fluid, and capacitors and resistors of the electrical double layer formed at the interface within the battery. As shown in Figure 2, the load-side diode D2 is connected so that the anode faces the secondary side circuit 3a and the cathode faces the load 6, and the charging current I CThe direction of current flow is restricted to one direction. In Figure 2, the stray resistance including the on-resistance of load 6 is shown as r2, and the parasitic resistance r of the receiving capacitor C2 is shown. p2 It is assumed that such a thing exists. Therefore, the receiving side resonant circuit (27a, 3a) consists of a receiving side coil L2, a receiving side capacitor C2, a stray resistance r2 and a parasitic resistance r p2 This is an RLC resonant circuit equipped with [a specific feature / feature].

[0039] [A: Amplification Mode] In the following, the operation of the amplification mode, which expands the oscillation amplitude to an integer multiple of the power supply voltage E0 by dividing it into periods (2π / ω0) that are half the switching period (2π / ω0) determined by the resonant frequencies of the first power supply side resonant circuit and the second power supply side resonant circuit 2a, will be explained in chronological order. - (a-1) First excitation cycle: t0 ≤ t < t1- The primary side charge / discharge voltage V obtained by actual measurement when the excitation element Q1 and the freewheel control element Q2 are driven on and off in a complementary manner. C1 and the transmitting coil voltage V L1 The transient response waveform is shown in Figure 3. The transient response waveform shown in Figure 3 explains the transient phenomenon at a transmission distance d = 600 mm, as defined in the schematic diagram in Figure 6. As the transmission distance d increases, the equivalent coupling coefficient K decreases, and the transient response waveform shown in Figure 3 corresponds to the case where the equivalent coupling coefficient K = 0.05. When the equivalent coupling coefficient K decreases, the effect of double resonance between the power supply side resonant circuit (34a, 2a) and the power receiving side resonant circuit (27a, 3a) weakens.

[0040] Furthermore, the power transmission system according to the first embodiment is a transient phenomena theory based on non-AC theory and must be distinguished from AC theory. However, the coupling coefficient K derived from AC theory that defines the steady state... ACThe equivalent pseudo-coupling coefficient in the transient state defined during transient response is defined as the "equivalent coupling coefficient k" and is used to correspond with AC theory. Unlike AC theory, the equivalent coupling coefficient k is strictly speaking a time-dependent parameter k=k(t), similar to the dynamic mutual inductance M=M(t). Even in non-AC theory that considers transient phenomena, the coupling coefficient K from AC theory is used for transient phenomena (transient response mutual induction) as an interaction between the time constant inherent in the circuit characteristics of the supply side resonant circuit (34a,2a) and the time constant inherent in the circuit characteristics of the receiving side resonant circuit (27a,3a). AC It can be evaluated using the equivalent coupling coefficient k, which is a similar "magnetic coupling degree". The equivalent coupling coefficient K = 0.05 in the transient response waveform shown in Figure 3 is the coupling coefficient K in the AC theory that defines the steady state. AC It can be approximated as a value that is almost equivalent to it.

[0041] In the first embodiment, the charging voltage VC of the load 6 is S We assume that the initial value of is a high value, close to the charging completion voltage (close to full charge). When the freewheel control element Q2 is kept in the off state (blocked state) and the excitation element Q1 is turned on (conducted state) at t=t0=0 as shown in Figure 4A, the power supply voltage E0 from the DC power supply 5 is stepped into the first power supply side resonant circuit, but the relationship in equation (2) must be satisfied. The equivalent internal resistance r of the second term on the left side of equation (2) stray1 If the voltage loss due to R1 is sufficiently small and negligible, then the left side of equation (2) should be the transmitting coil voltage of the first term V L1 and the primary side charge / discharge voltage V of item 3 C1 That remains, V C1 +V L1 =E0……(9) As shown above, approximate simplified expressions are possible.

[0042] The DC power supply 5 supplies the power supply voltage E0 to the transmitting capacitor C1 of the first power supply side resonant circuit such that equation (9) is satisfied when the excitation element Q1 is in the conduction state. At time t=t0=0, the transmitting capacitor C1 is not charged and the primary side charge / discharge voltage V C1Since = 0, from equation (9), E0=V L1 (t=t0=0)=ΔV L10 ...(10) This is necessary, and as shown by the solid line in Figure 3, the transmitting coil voltage V L1 ΔV L10 = Performs an impulse ascent of E0.

[0043] From the third term on the left side of equation (2), the primary side charge / discharge voltage V C1 This can be expressed as in equation (11). In equation (11), at t=t0=0, V C1 = 0V. In the first half-period t0 ≤ t < t1 after the step input at t=t0, the primary side charge / discharge voltage V is as shown in Figure 3. C1 The current begins to rise. The charging current to the transmitting capacitor C1 begins to rise according to equation (11), but in the time domain of the first half-period t0 < t < t1, the oscillation waveform shown by equation (11) remains as a half-period oscillation waveform with only the initial oscillation with the maximum amplitude cut off:

number

[0044] On the other hand, from the first term on the left side of equation (2) and equation (9), the transmitting coil voltage V L1 This can be expressed as shown in equation (12).

number

[0045] In equation (11), ω0 2 ≫ α 2 If we consider it this way, then equation (11) becomes, V C1 ≒E0(1-e -αt cosω0t) ……(13) It can be expressed as follows. Similarly, in equation (12), ω0 2 ≫ α 2 If we consider it this way, then equation (12) becomes, V L1 ≒E0e -αt cosω0t ……(14) It can be expressed as follows: At time t=t1=π / ω0, when charging of the transmitting capacitor C1 is complete and cosω0t1=-1, Formula (13 ) from the primary side charge / discharge voltage V C1 The maximum value is shown by the dashed line in Figure 3.

number

[0046] After the step input at t=t0, when t0 < t < t1, the transmitting coil L1 discharges the charging current to the transmitting capacitor C1 as a return current. Therefore, the transmitting coil voltage V, which is the voltage across the terminals of the transmitting coil L1, is lost. L1The primary side charge / discharge voltage V satisfies equations (9) and (13). C1 As increases, it decreases as shown by the solid line in Figure 3. At the timing (t1-t0) / 2=t1 / 2=π / (2ω0), cosω0t=0. After that, the transmitting coil voltage V L1 Furthermore, in the region π / 2 < ω0t < π, the primary side charge / discharge voltage V satisfies equations (9) and (13). C1 In opposite phase to the increase, it decreases further while maintaining a negative value. As shown by the dashed line in Figure 3, the primary side charge / discharge voltage V is immediately before time t=t1=π / ω0. C1 When approaching ≈ 2E0, in order to satisfy equation (9), the transmitting coil voltage V L1 It needs to be approximately -E0.

[0047] - (a-2) First reflux cycle: t1 ≤ t < t2- Primary charge / discharge voltage V C1 At time t=t1=π / ω0 when the voltage reaches its maximum value, a complementary switching transition is performed, as shown in Figure 4C, where the excitation element Q1 is turned off (blocked state) and the freewheel control element Q2 is turned on (conducted state). When the excitation element Q1 is turned off and the freewheel control element Q2 is turned on at t=t1, the freewheel cycle operation of the second feed-side resonant circuit 2a, which consists of an RLC series resonant circuit formed by the freewheel control element Q2, the transmitting-side capacitor C1, and the transmitting-side coil L1, begins. Therefore, at time t=t1, the transmitting-side coil voltage V L1 Equation (13) shows the change and the transmitting coil voltage V L1 Equation (14), which shows the change, is cut off in the first oscillation and does not become a damped oscillation. In the operation of the recirculation cycle by the second power supply side resonant circuit 2a, the relationship in equation (6) must be satisfied. The equivalent internal resistance r of the second term on the left side of equation (6) stray2 If the voltage loss due to R2 is sufficiently small and negligible, the left side of equation (6) should be the transmitting coil voltage of the first term V L1 and the primary side charge / discharge voltage V of item 3 C1 That remains, V C1 +V L1 =0 ...(16) Approximate simplified expressions are possible, as follows.

[0048] As can be seen from FIG. 3, at the timing immediately before t=t1, the primary side charge / discharge voltage V C1 (t→t1)≈2E0, and the power transmission side coil voltage V L1 (t→t1)≈-E0. However, if V C1 (t=t1)≈2E0, then from Equation (16), V L1 (t=t1)=V L1 (t→t1)+ΔV L11 ≈-2E0……(17) Therefore, as shown by the solid line in FIG. 3, at time t=t1, the power transmission side coil voltage V L1 undergoes an impulse-like sudden voltage transition of ΔV L11 =-E0. Comparing Equation (17) with Equation (14), in the range t1≤t<t2 of the second half cycle, the power transmission side coil voltage V​​​​​​​​​​​​​​​​​​​​​​​​​​​​​​​Equation (14), which shows the change of, can be replaced by equation (19). Similar to equation (18), the number 1 in the expression of t1 written in the exponential notation on the right side of equation (19) is a subscript. At the timing t1 ≤ t < t2, e -α(t-t1) is e -α(t-t1) It can be approximated as ≈ 1. That is, the damped oscillation of the LCR series resonant circuit, at timings t1 ≤ t < t2, shows a transient response waveform of a large amplitude oscillation of 2E0 as a new initial oscillation, different from the damped oscillation of equations (13) and (14). Comparing equations (13) and (19), it is equivalent to changing the initial state to a state where a DC voltage 2E0 is supplied from the DC power supply 5 to the transmitting side capacitor C1 at the boundary condition at time t=t1, which is shifted in the time axis direction by the equivalent of t1. In this way, by complementary on / off control of the excitation element Q1 and the freewheel control element Q2, and by transitioning from the excitation cycle in the first power supply side resonant circuit to the freewheel cycle in the second power supply side resonant circuit 2a, the power supply voltage E0 supplied by the DC power supply 5 can be doubled.

[0050] That is, as shown in Figure 4C, when a complementary transition occurs at t=t1, where the excitation element Q1 is turned off and the freewheel control element Q2 is turned on, the initial value of the virtual equivalent power supply voltage 2E0 from the DC power supply 5 in the second power supply side resonant circuit 2a is changed to a circuit state equivalent to the state in which the initial conditions were changed. However, even in this case, it is necessary to satisfy the voltage relationship in equation (16). In the time domain t1 ≤ t < t2, the primary side charge / discharge voltage V C1 The equivalent expansion of the power supply voltage is as shown in equation (19). At t=t1 in equation (19), the initial condition V C1 = 2E0. After changing the initial condition t=t1, for t1≦t < t2, as shown in Figure 4D, the discharge current from the transmitting capacitor C1 flows to the transmitting coil L1 via the recirculation control element Q2, and the electrostatic energy stored in the transmitting capacitor C1 decreases, so according to equation (19), the primary side charge / discharge voltage V C1 It begins to decline.

[0051] On the other hand, the transmission side coil voltage V L1As shown in equation (18), the power supply voltage is equivalently expanded. At t=t1 in equation (19), V C1 Since = 2E0, from the requirement of equation (16), the value of equation (18) at t=t1 is V L = -2E0. After changing the initial condition t=t1, for t1≦t < t2, as can be seen from Figure 3, the discharge current from the transmitting capacitor C1 flows into the transmitting coil L1 as magnetic energy, and according to equation (18), the transmitting coil voltage V L1 It will begin to rise.

[0052] That is, at time t=t2=2π / ω0, when the discharge from the transmitting capacitor C1 is complete and cosω0t2=1, the primary side charge / discharge voltage V is as shown in equation (19). C1 The minimum value is -2E0, as shown by the dashed line in Figure 3. On the other hand, after changing the initial condition t=t1, for t1≦t < t2, the transmitting coil L1 stores the discharge current from the transmitting capacitor C1 as magnetic energy. Therefore, the transmitting coil voltage V, which is the terminal voltage of the transmitting coil L2, is obtained. L1 The primary side charge / discharge voltage V satisfies equations (16) and (18). C1 As decreases, it increases as shown by the solid line in Figure 3. At the timing t=3π / (2ω0), cosω0t=0, so the transmitting coil voltage V L1 The primary side charge / discharge voltage V becomes 0. C1 However, as shown by the dashed line in Figure 3, at the timing t = 3π / (2ω0), cosω0t = 0, so the primary side charge / discharge voltage V C1 = 0.

[0053] Transmitting coil voltage V L1 Furthermore, in the region 3π / 2 < ω0t < 2π, the primary side charge / discharge voltage V satisfies equations (16) and (18). C1 It decreases in the opposite phase and increases further in the positive value region. Primary side charge / discharge voltage V C1 Furthermore, in the region 3π / 2 < ω0t < 2π, the transmitting coil voltage V satisfies equations (16) and (19). L1In the opposite phase to the increase, it further decreases in the negative region. However, at time t=t2, the transmitting coil voltage V L1 Equation (18) shows the change and the primary side charge / discharge voltage V C1 Equation (19), which shows the change, is cut off at the first oscillation, and does not become a damped oscillation of a continuous RLC series resonant circuit. As shown by the dashed line in Figure 3, the primary side charge / discharge voltage V is just before time t=t2=2π / ω0. C1 When approaching ≈-2E0, in order to satisfy equation (16), the transmitting coil voltage V L1 It needs to be approximately equal to 2E0.

[0054] - (a-3) Second excitation cycle: t2 ≤ t < t3- Primary charge / discharge voltage V C1 At the time t=t2=2π / ω0 when the value is at its minimum, a complementary switching transition is performed to turn off the freewheel control element Q2 and turn on the excitation element Q1, as shown in Figure 4E. When the freewheel control element Q2 is turned off and the excitation element Q1 is turned on at t=t2, the operation of the excitation cycle by the first power supply side resonant circuit, which consists of an RLC series resonant circuit composed of the DC power supply 5, the excitation element Q1, the power transmission side capacitor C1, and the power transmission side coil L1, begins. The operation of the excitation cycle by the first power supply side resonant circuit must satisfy the relationship in equation (9).

[0055] As can be seen from Figure 3, at the timing just before t=t2, the primary side charge / discharge voltage V C1 (t→t2)≈-2E0, and the transmitting coil voltage V L1 (t→t2)≈2E0. However, V C1 (t=t2)≒ -2E If it is 0, then from equation (9), V L1 (t=t2=V) L1 (t→t2)+ΔV L12 ≈3E0……(20) Therefore, as shown by the solid line in Figure 3, at time t=t2, the transmitting coil voltage V L1 ΔV L12The voltage transition is an impulsive, sharp rise of =E0.

[0056] Comparing equation (20) with equations (14) and (18), in the third half-period, t2 ≤ t < t3, the transmitting coil voltage V L1 This is shifted in the time axis direction by the equivalent of t2, V L1 ≈3E0e -α(t-t2) cosω0(t-t2) ……(21) It is clear that it needs to be expressed as follows. The number 2 in the expression of t2 written in the exponential notation on the right side of equation (21) is a subscript. Considering equation (9), from equation (21) the primary side charge / discharge voltage V C1 too, V C1 ≒E0{1-3e -α(t-t2) cosω0(t-t2)} ……(22) It is clear that it needs to be expressed in this way. Similar to equation (21), the number 2 in the expression of t2 written in the exponential notation on the right side of equation (22) is a subscript. At the timing t2 ≤ t < t3, e -α(t-t2) is e -α(t-t2) It can be approximated as approximately 1.

[0057] That is, at time t=t2, the transmitting coil voltage V L1 Equation (18), which shows the change in V, is replaced by equation (21), and the primary side charge / discharge voltage V C1Equation (19), which shows the change, is replaced by equation (22). As a result, the changes in equations (18) and (19) remain as a waveform equivalent to half a period of the initial oscillation with maximum amplitude in the time domain t1 ≤ t < t2 of the second half-period. At the timing t2 ≤ t < t3 of the third half-period, the waveform of the damped oscillation of the LCR series resonant circuit shows a transient response waveform of a large amplitude oscillation of 3E0 as a new initial oscillation shown by equations (21) and (22). Compared with equations (13) and (19), equation (22) has a technical effect equivalent to changing the initial state to a state where a DC voltage 3E0 is supplied from the DC power supply 5 at the boundary condition at t=t2, which is shifted in the time axis direction by the equivalent of t2. In this way, by complementary on / off control of the excitation element Q1 and the freewheel control element Q2, and by repeatedly switching between the excitation cycle in the first power supply side resonant circuit and the freewheel cycle in the second power supply side resonant circuit 2a, the power supply voltage E0 supplied by the DC power supply 5 can be equivalently boosted to three times its original value.

[0058] That is, as shown in Figure 4E, when the excitation element Q1 is turned ON and the freewheel control element Q2 is turned OFF at t=t2, the system switches to the new oscillation waveform shown by equations (21) and (22), and the initial value is changed to a new circuit state in which a virtual equivalent power supply voltage 3E0 is supplied from the DC power supply 5. Even when the equivalent power supply voltage 3E0 is supplied from the DC power supply 5 to the first power supply side resonant circuit and the system switches to the new oscillation waveform shown by equations (21) and (22), the voltage relationship in equation (9) must be satisfied. In the time domain t2 ≤ t < t3, the primary side charge / discharge voltage V C1 The equivalent expansion of the power supply voltage is as shown in equation (22). In equation (22), at t=t2=2π / ω0, the initial condition V C1 = -2E0. After changing the initial condition t=t2, for t2≦t < t3, as shown in Figure 4B, the charging current to the transmitting capacitor C1 flows from the DC power supply 5 through the excitation element Q1, and electrostatic energy is accumulated in the transmitting capacitor C1. Therefore, according to equation (22), the primary side charge / discharge voltage V is shown by the dashed line in Figure 5A. C1 It begins to rise.

[0059] On the other hand, the transmission side coil voltage VL1 As shown in equation (21), the power supply voltage is seemingly expressed as an equivalent expansion. In equation (22), at t=t2, V C1 Since = -2E0, from the requirement of equation (9), the value of equation (21) at t=t2 is V L =3E0. After changing the initial condition t=t2, for t2≦t < t3, as can be seen from Figure 5A, the magnetic energy stored in the transmitting coil L1 flows into the transmitting capacitor C1 as a charging current, and according to equation (21), the transmitting coil voltage V L1 The voltage begins to decline. Figure 5A is an extended waveform diagram of Figure 3, and therefore, similar to Figure 3, it represents the transient response waveform when the equivalent coupling coefficient K = 0.05 at a transmission distance d = 600 mm.

[0060] At time t=t3=3π / ω0, when charging from the transmitting capacitor C1 is complete and cosω0t3=-1, the primary side charge / discharge voltage V is as shown in equation (22). C1 The maximum value is 4E0, as shown by the dashed line in Figure 5A. After changing the initial condition t=t2, for t2≦t < t3, the transmitting coil L1 releases magnetic energy as a charging current to the transmitting capacitor C1. Therefore, the transmitting coil voltage V, which is the terminal voltage of the transmitting coil L2, is released. L1 The primary side charge / discharge voltage V satisfies equations (9) and (21). C1 As increases, it decreases as shown by the solid line in Figure 5A. At the timing t=5π / (2ω0), cosω0t=0, so the transmitting coil voltage V L1 The primary side charge / discharge voltage V becomes 0. C1 However, as shown by the dashed line in Figure 5A, at the timing t = 5π / (2ω0), cosω0t = 0, so the primary side charge / discharge voltage V C1 = 0.

[0061] Transmitting coil voltage V L1 Furthermore, in the region 5π / 2 < ω0t < 3π, the primary side charge / discharge voltage V satisfies equations (9) and (21). C1 In the negative region, it decreases further in the opposite phase to the increase of the primary side charge / discharge voltage V. C1Furthermore, in the region 5π / 2 < ω0t < 2π, the transmitting coil voltage V satisfies equations (9) and (22). L1 In the region of positive values, it increases further in the opposite phase to the decrease. As shown by the dashed line in Figure 5A, the primary side charge / discharge voltage V at time t=t3=3π / ω0 C1 When approaching ≈ 4E0, in order to satisfy equation (9), the transmitting coil voltage V L1 It needs to be approximately -3E0.

[0062] - (a-4) Second reflux cycle: t3 ≤ t < t4 - Primary charge / discharge voltage V C1 At the time t=t3=3π / ω0 when the value reaches its maximum, a complementary switching transition is performed to turn off the excitation element Q1 and turn on the freewheel control element Q2. When the excitation element Q1 is turned off and the freewheel control element Q2 is turned on at t=t3, the freewheel cycle operation by the second feed-side resonant circuit 2a, which consists of an RLC series resonant circuit formed by the freewheel control element Q2, the transmitting capacitor C1, and the transmitting coil L1, begins. The operation of the freewheel cycle by the second feed-side resonant circuit 2a must satisfy the relationship in equation (16).

[0063] As can be seen from Figure 5A, at the timing just before t=t3, the primary side charge / discharge voltage V C1 (t→t3)≈4E0, and the transmitting coil voltage V L1 (t→t3)≈-3E0. However, V C1 If (t=t3)≈4E0, then from equation (16), V L1 (t=t3=V) L1 (t→t3)+ΔV L13 ≒-4E0……(23) Therefore, as shown by the solid line in Figure 5A, at time t=t3, the transmitting coil voltage V L1 ΔV L13 The voltage transition is an impulsive, sharp drop of -E0.

[0064] Comparing equation (23) with equations (14), (18), and (21), in the fourth half-period, t3 ≤ t < t4, the transmitting coil voltage V L1 This is shifted in the time axis direction by the equivalent of t3, V L1 ≈4E0e -α(t-t3) cosω0(t-t3) ……(24) It is clear that it needs to be expressed as follows. The number 3 in the expression of t3 written in the exponential notation on the right side of equation (24) is a subscript. Considering equation (16), from equation (24) the primary side charge / discharge voltage V C1 too, V C1 ≈-4E0e -α(t-t3) cosω0(t-t3) ……(25) It is clear that it needs to be expressed in this way. Similar to equation (24), the number 3 in the expression of t3 written in the exponential notation on the right side of equation (25) is a subscript. At the timing t3 ≤ t < t4, e -α(t-t3) is e -α(t-t3) It can be approximated as approximately 1.

[0065] That is, at time t=t3, the transmitting coil voltage V L1 Equation (21), which shows the change in V, is replaced by equation (25), and the primary side charge / discharge voltage V C1Equation (22), which shows the change, is replaced by equation (24). As a result, at time t=t2, the waveform change shown by equation (21), which is smoothly continuous with equation (18), remains in the time domain t2≦t < t3 of the third half-period as a waveform corresponding to half a period of the initial oscillation with maximum amplitude. Also, at time t=t2, the waveform change shown by equation (22), which is continuous with equation (19) as a distorted waveform, remains in the time domain t2≦t < t3 of the third half-period as a waveform corresponding to half a period of the initial oscillation with maximum amplitude. The damped oscillation of the LCR series resonant circuit shows a transient response waveform of a large amplitude oscillation of 4E0 as the initial oscillation at timing t3≦t < t4 of the fourth half-period. In comparison with equations (13), (19), and (22), equation (25) represents a technical effect equivalent to changing the initial state to a state where a constant DC voltage 4E0 is supplied from the DC power supply 5 to the transmitting capacitor C1 at the boundary condition at t=t3, which is shifted in the time axis direction by the equivalent of t3. In this way, by complementary on / off control of the excitation element Q1 and the freewheel control element Q2, and by time-division alternating the excitation cycle in the first power supply side resonant circuit and the freewheel cycle in the second power supply side resonant circuit 2a, the power supply voltage E0 supplied by the DC power supply 5 can be equivalently boosted by four times.

[0066] That is, as shown in Figure 4C, if a complementary transition occurs at t=t3, where the excitation element Q1 is turned off and the freewheel control element Q2 is turned on, the circuit state is changed to one equivalent to the state in which the virtual equivalent power supply voltage 4E0 of the DC power supply 5 is applied to the second power supply side resonant circuit 2a as an initial condition. However, in this case as well, the voltage relationship in equation (16) must be satisfied. In the time domain t3 ≤ t < t4, the primary side charge / discharge voltage V C1 The power supply voltage is equivalently expanded as shown in equation (25). At t=t3 in equation (25), the initial condition V is as shown by the dashed line in Figure 5B. C1=4E0. Figure 5B is an extended waveform diagram of Figure 5A, and, similar to Figures 3 and 5A, it is the transient response waveform when the equivalent coupling coefficient K = 0.05 at a transmission distance d = 600 mm. After the initial condition change at t = t3, for t3 ≤ t < t4, as shown in Figure 4D, the discharge current from the transmitting capacitor C1 flows to the transmitting coil L1 via the recirculation control element Q2, and the electrostatic energy stored in the transmitting capacitor C1 decreases, so the primary side charge / discharge voltage V is calculated according to equation (25). C1 It begins to decline.

[0067] On the other hand, the transmission side coil voltage V L1 As shown in equation (24), the power supply voltage is equivalently expanded. At t=t3 in equation (25), V C1 Since =4E0, from the requirement of equation (16), the value of equation (24) at t=t3 is V L = -4E0. After changing the initial condition t=t3, for t3≦t < t4, as can be seen from Figure 5B, the discharge current from the transmitting capacitor C1 flows into the transmitting coil L1 as magnetic energy, and according to equation (24), the transmitting coil voltage V L1 It will begin to rise.

[0068] At time t=t4=4π / ω0, when the discharge from the transmitting capacitor C1 is complete and cosω0t4=1, the primary side charge / discharge voltage V is as shown in equation (25). C1 The minimum value is -4E0, as shown by the dashed line in Figure 5B. After changing the initial condition t=t3, for t3≦t < t4, the transmitting coil L1 stores the discharge current from the transmitting capacitor C1 as magnetic energy. Therefore, the transmitting coil voltage V, which is the terminal voltage of the transmitting coil L2, is obtained. L1 The primary side charge / discharge voltage V satisfies equations (16) and (24). C1 As decreases, it increases as shown by the solid line in Figure 5B. At the timing t = 7π / (2ω0), cosω0t = 0, so the transmitting coil voltage V L1 The primary side charge / discharge voltage V becomes 0. C1 However, as shown by the dashed line in Figure 5B, at the timing t = 7π / (2ω0), cosω0t = 0, so the primary side charge / discharge voltage VC1 = 0.

[0069] Transmitting coil voltage V L1 Furthermore, in the region 7π / 2 < ω0t < 4π, the primary side charge / discharge voltage V satisfies equations (16) and (24). C1 It decreases in the opposite phase and increases further in the positive value region. Primary side charge / discharge voltage V C1 Furthermore, in the region 7π / 2 < ω0t < 4π, the transmitting coil voltage V satisfies equations (16) and (25). L1 In the negative region, it decreases further in the opposite phase to the increase. As shown by the dashed line in Figure 5B, the primary side charge / discharge voltage V at time t=t4=4π / ω0 C1 When approaching ≈-4E0, in order to satisfy equation (16), the transmitting coil voltage V L1 It needs to be approximately 4E0.

[0070] - (a-5) Third excitation cycle: t4 ≤ t < t5- Primary charge / discharge voltage V C1 At the time t=t4=4π / ω0 when the value is at its minimum, a complementary switching transition is performed, similar to the state shown in Figure 4E, where the freewheeling control element Q2 is turned off and the excitation element Q1 is turned on. When the freewheeling control element Q2 is turned off and the excitation element Q1 is turned on at t=t4, the operation of the excitation cycle by the first power supply side resonant circuit, which consists of an RLC series resonant circuit composed of the DC power supply 5, the excitation element Q1, the power transmission side capacitor C1, and the power transmission side coil L1, begins. The operation of the excitation cycle by the first power supply side resonant circuit must satisfy the relationship in equation (9).

[0071] As can be seen from Figure 5B, at the timing just before t=t4, the primary side charge / discharge voltage V C1 (t→t4)≈-4E0, and the transmitting coil voltage V L1 (t→t4)≈4E0. However, V C1 If (t=t4)≈4E0, then from equation (9), V L1 (t=t4=V) L1 (t→t4)+ΔV L14 ≒5E0……(26) Therefore, as shown by the solid line in Figure 5B, at time t=t4, the transmitting coil voltage V L1 ΔV L14 The voltage transition is an impulsive, sharp rise of =E0.

[0072] Comparing equation (20) with equations (14), (24), and (26), in the fifth half-period, t4 ≤ t < t5, the transmitting coil voltage V L1 This is shifted in the time axis direction by the equivalent of t4. V L1 ≈5E0e -α(t-t4) cosω0(t-t4) ……(27) It can be seen that it needs to be expressed as follows. The number 4 in the expression of t4 written in the exponential notation on the right side of equation (27) is a subscript. Considering equation (9), from equation (27) the primary side charge / discharge voltage V C1 too, V C1 ≒E0{1-5e -α(t-t4) cosω0(t-t4)} ……(28) It is clear that it needs to be expressed in this way. Similar to equation (27), the number 4 in the expression of t4 written in the exponential notation on the right side of equation (28) is a subscript. At the timing t4 ≤ t < t5, e -α(t-t4) is e -α(t-t4) It can be approximated as approximately 1.

[0073] That is, at time t=t4, the transmitting coil voltage V L1 Equation (24), which shows the change in V, is replaced by equation (27), and the primary side charge / discharge voltage V C1Equation (25), which shows the change, is replaced by equation (28). Then, at time t=t3, the waveform change shown by equation (24), which is continuous with equation (22) as a distorted waveform, remains as a waveform equivalent to half a period of the initial oscillation with maximum amplitude in the time domain t3≦t < t4 of the fourth half-period. On the other hand, at time t=t3, the waveform change shown by equation (25), which is smoothly continuous with equation (21), remains as a waveform equivalent to half a period of the initial oscillation with maximum amplitude in the time domain t3≦t < t4 of the fourth half-period. The damped oscillation of the LCR series resonant circuit shows a transient response waveform of a large amplitude oscillation of 5E0 as an initial oscillation at the timing t4≦t < t5 of the fifth half-period. Comparing equations (13) and (25) with equation (28), equation (28) represents a technical effect equivalent to changing the initial state to a state where a constant DC voltage 5E0 is supplied from the DC power supply 5 to the transmitting capacitor C1 at the boundary condition at t=t4, which is shifted in the time axis direction by the equivalent of t4. In this way, by complementary on / off control of the excitation element Q1 and the freewheel control element Q2, and by alternately repeating the excitation cycle in the first power supply side resonant circuit and the freewheel cycle in the second power supply side resonant circuit 2a in a time-division manner, the power supply voltage E0 supplied by the DC power supply 5 can be equivalently boosted by five times.

[0074] That is, similar to the state shown in Figure 4E, if the excitation element Q1 is turned ON and the freewheel control element Q2 is turned OFF at t=t4, the circuit state of the first power supply side resonant circuit is changed to one equivalent to the state in which the virtual equivalent power supply voltage 5E0 of the DC power supply 5 has been changed from its initial condition. However, in this case as well, the voltage relationship in equation (9) must be satisfied. In the time domain t4 ≤ t < t5, the primary side charge / discharge voltage V C1 The equivalent expansion of the power supply voltage is as shown in equation (28). In equation (28), at t=t4=4π / ω0, the initial condition V C1 = -4E0. After changing the initial condition t=t4, for t4≦t < t5, as shown in Figure 4B, the charging current to the transmitting capacitor C1 flows from the DC power supply 5 through the excitation element Q1, and electrostatic energy is accumulated in the transmitting capacitor C1. Therefore, according to equation (28), the primary side charge / discharge voltage V is shown by the dashed line in Figure 5A. C1 It begins to rise.

[0075] On the one hand, the power transmission side coil voltage V L1 is seemingly expressed as an equivalent expansion of the power supply voltage as shown in Equation (27). At t = t4 in Equation (28), since V C1 = -4E0, from the requirement of Equation (9), the value at t = t4 in Equation (27) is V L = 5E0. In the region of t4 ≤ t < t5 after the initial condition change at t = t4, as can be seen from Figure 5A, the magnetic energy stored in the power transmission side coil L1 flows into the power transmission side capacitor C1 as a charging current, and according to Equation (27), the power transmission side coil voltage V L1 starts to decline.

[0076] At the time t = t5 = 9π / ω0, when the charging from the power transmission side capacitor C1 is completed and cosω0t5 = -1, as shown in Equation (28), the primary side charge-discharge voltage V C1 becomes the maximum value of 6E shown by the dashed line in Figure 5A. In the region of t4 ≤ t < t5 after the initial condition change at t = t4, the power transmission side coil L1 releases magnetic energy as a charging current to the power transmission side capacitor C1. Therefore, the power transmission side coil voltage V L1 which is the voltage between the terminals of the power transmission side coil L4, decreases as shown by the solid line in Figure 5B as the primary side charge-discharge voltage V C1 increases. At the timing of t = 9π / (2ω0), since cosω0t = 0, the power transmission side coil voltage V L1 = 0. The primary side charge-discharge voltage V C1 also becomes 0 at the timing of t = 9π / (2ω0) as shown by the dashed line in Figure 5B because cosω0t = 0, so the primary side charge-discharge voltage V C1 = 0.

[0077] The power transmission side coil voltage V L1 further decreases in the region of 9π / 2 < ω0t < 5π in a reverse phase to the increase of the primary side charge-discharge voltage V C1 and satisfies Equations (9) and (27) in the negative value region. The primary side charge-discharge voltage V C1In the region of 9π / 2 < ω0t < 5π, in order to further satisfy equations (9) and (28), the voltage V of the power transmission side coil L1 decreases in the opposite phase and further increases in the positive value region. As shown by the dashed line in Fig. 5B, at time t = t5 = 5π / ω0, when the primary side charge-discharge voltage V C1 ≒ 6E0, in order to satisfy equation (9), the voltage V of the power transmission side coil shown by the solid line in Fig. 5B L1 ≒ -5E0 is required.

[0078] When the primary side charge-discharge voltage V C1 reaches its maximum value at time t = t5 = 5π / ω0, a complementary switching transition is performed to turn off the excitation element Q1 and turn on the reflux control element Q2. When the excitation element Q1 is turned off and the reflux control element Q2 is turned on at t = t5, the operation of the reflux cycle by the second power supply side resonance circuit 2a composed of the reflux control element Q2, the power transmission side capacitor C1, and the power transmission side coil L1 starts. In the operation of the reflux cycle by the second power supply side resonance circuit 2a, it is necessary to satisfy the relationship of equation (16).

[0079] As can be seen from Fig. 5B, at the timing immediately before t = t5, the primary side charge-discharge voltage V C1 (t → t5) ≒ 6E0, and the voltage V of the power transmission side coil L1 (t → t5) ≒ -5E0. However, if V C1 (t = t5) ≒ 6E0, then from equation (16), V L1 (t = t5) = V L1 (t → t5) + ΔV L15 ≒ -6E0……(29) Therefore, as shown by the solid line in Fig. 5B, at time t = t5, the voltage V of the power transmission side coil L1 undergoes an impulse-like steep voltage transition with ΔV L15 = -E0.

[0080] Comparing equation (23) with equations (14), (24), (27), and (29), when t5 ≤ t < t4, the voltage V of the power transmission side coil L1This is shifted in the time axis direction by the equivalent of t5. V L1 ≒6E0e -α(t-t5) cosω0(t-t5) ……(30) It is clear that it needs to be expressed as follows. The number 5 in the expression of t5 written in the exponential notation on the right side of equation (30) is a subscript. Considering equation (16), from equation (30) the primary side charge / discharge voltage V C1 too, V C1 ≈-6E0e -α(t-t5) cosω0(t-t5) ……(31) It is clear that it needs to be expressed in this way. Similar to equation (30), the number 5 in the expression of t5 written in the exponential notation on the right side of equation (31) is a subscript. At the timing t5 ≤ t < t6, e -α(t-t5) is e -α(t-t5) It can be approximated as approximately 1.

[0081] That is, at time t=t5, the transmitting coil voltage V L1 Equation (27), which shows the change, is replaced by equation (30), and the primary side charge / discharge voltage V C1Equation (28), which shows the change, is replaced by equation (31). The changes in equations (27) and (28) remain as a half-period vibration waveform in the time domain t4 ≤ t < t5 of the fifth half-period, with only the initial vibration of the maximum amplitude cut off. At time t=t4, the waveform change shown by equation (27), which is continuous with equation (24) as a distorted waveform, remains as a half-period vibration waveform in the time domain t4 ≤ t < t5 of the fifth half-period, with only the initial vibration of the maximum amplitude cut off. On the other hand, at time t=t4, the waveform change shown by equation (28), which is smoothly continuous with equation (25), remains as a half-period vibration waveform in the time domain t4 ≤ t < t5 of the fifth half-period, with only the initial vibration of the maximum amplitude cut off. At the timing t5 ≤ t < t6 of the sixth half-period, the damped oscillation of the LCR series resonant circuit shows a transient response waveform of oscillation with a large initial value of 6E0. Comparing equations (13), (25), (28) with equation (31), equation (31) represents a technical effect equivalent to changing the initial state to a state where a DC voltage 6E0 is supplied from the DC power supply 5 at the boundary condition at t=t5, which is shifted in the time axis direction by the equivalent of t5. That is, by complementary on / off control of the excitation element Q1 and the freewheel control element Q2, and by time-division alternating the excitation cycle in the first power supply side resonant circuit and the freewheel cycle in the second power supply side resonant circuit 2a, the power supply voltage E0 supplied by the DC power supply 5 can be equivalently boosted by six times.

[0082] [B: Transmit mode; t≧t5] As described above, when a complementary transition occurs at t=t5, turning the excitation element Q1 to the off state and the freewheel control element Q2 to the on state, the virtual equivalent power supply voltage 6E0 of the DC power supply 5 becomes equivalent to the circuit state applied as an initial condition to the second power supply side resonant circuit 2a. In the power transmission system according to the first embodiment, it is determined that the desired oscillation amplitude of the power supply side resonant circuits (34a, 2a) has been obtained when the power supply voltage E0 supplied by the DC power supply 5 is equivalently boosted by six times. This determination in the power transmission system according to the first embodiment is just one example, and it is not necessary to limit it to a six-fold increase in amplitude value. By repeating the excitation cycle p times and the freewheel cycle (p-1) times... n=2p ……(32a) or n = 2p - 1 ……(32b) The power supply voltage E0 supplied by the DC power supply 5 can be amplified by a factor of two. p is a positive integer greater than or equal to 1, and n is a positive integer greater than or equal to 2. In the power transmission system according to the first embodiment, the case where n=6 is selected is shown only as an example, and the value of n can be arbitrarily selected.

[0083] For timings t0 ≤ t < t5, the operation of the amplification mode, in which the excitation element Q1 and the freewheel control element Q2 are complementary on / off control over a half-period π, and the excitation cycle and freewheel cycle are switched over over a half-period π, was described. In the conditions exemplified by the power transmission system according to the first embodiment, as shown in Figure 5B, instead of the amplification mode operation that switches over over a half-period π, the system switches to a "transmission mode" operation in which a transmission mode period of about 5 to 10π, which is longer than a half-period π, is selected. During the transmission mode period, the excitation element Q1 is kept in the off state at all times, and the freewheel control element Q2 is kept in the on state at all times, thereby realizing the waveforms of the damped oscillations shown in equations (30) and (31) as shown in Figure 5B.

[0084] Equation (30) is given by the transmitting coil voltage V, as can be seen from Figure 5B. L1 As it oscillates as a cosine wave of frequency ω0, the envelope of the oscillation peak (amplitude) moves from the maximum value at time t5 to e ―α(t-t5) This shows the waveform of underdamping (damped oscillation) which decays exponentially. On the other hand, as can be seen from Figure 5B, equation (31) is given by the primary side charge / discharge voltage V C1 However, the transmission side coil voltage V L1 The waveform showing the change is in opposite phase to the cosine wave of frequency ω0, and the envelope of the vibration peak (amplitude) changes from the maximum value at time t5 to e ―α(t-t5) The waveform shows an under-damping (damped oscillation) that decays exponentially. In the transmission mode, the damped oscillation of the power supply side resonant circuit (34a, 2a) shown in equations (30) and (31) is transmitted to the power receiving side resonant circuit (27a, 3a) via contactless power transmission (wireless power transmission) using double resonance.

[0085] Since the excitation element Q1 is kept in the off state and the freewheel control element Q2 is kept in the on state, and the operation of the second power supply side resonant circuit 2a is fixed, in the transmission mode, the voltage relationship in equation (16) must be satisfied. In the time domain of the sixth half-period t5 ≤ t < t6 shown in Figure 5B, the primary side charge / discharge voltage V C1 The power supply voltage is equivalently expanded as shown in equation (31). At t=t5 in equation (31), the initial condition V is as shown by the dashed line in Figure 5B. C1 = 6E0. After switching to the transmission mode at t=t5, for t5≦t < t6, as shown in Figure 4D, the discharge current from the transmitting capacitor C1 flows to the transmitting coil L1 via the recirculation control element Q2, and the electrostatic energy stored in the transmitting capacitor C1 decreases, so the primary side charge / discharge voltage V is defined in equation (31). C1 It begins to decline.

[0086] On the other hand, the transmission side coil voltage V L1 As shown in equation (30), the power supply voltage is equivalently expanded. At t=t5 in equation (31), V C1 Since =6E0, from the requirement of equation (16), the value of equation (30) at t=t5 is V L = -6E0. In the sixth half-period t5 ≤ t < t6 after switching to the transmission mode at t=t5, as can be seen from Figure 5B, the discharge current from the transmission capacitor C1 flows into the transmission coil L1 as magnetic energy, and as defined by equation (30), the transmission coil voltage V L1 It will begin to rise.

[0087] At time t=t6=6π / ω0, when the discharge from the transmitting capacitor C1 is complete and cosω0t6=1, the primary side charge / discharge voltage V is as shown in equation (31). C1 The minimum value is -6E0, as shown by the dashed line in Figure 5B. At t=t5, the primary side charge / discharge voltage V C1 Since the maximum value was 6E0, the primary side charge / discharge voltage V was in the time domain t5 ≤ t < t6 of the 6th half-cycle when in transmit mode. C1The voltage is not amplified. In the sixth half-cycle t5 ≤ t < t6 after switching to the transmission mode at t=t5, the transmitting coil L1 stores the discharge current from the transmitting capacitor C1 as magnetic energy. Therefore, the transmitting coil voltage V, which is the terminal voltage of the transmitting coil L2, is stored. L1 The primary side charge / discharge voltage V satisfies equations (16) and (30). C1 As decreases, it increases as shown by the solid line in Figure 5B. At the timing t = 11π / (2ω0), cosω0t = 0, so the transmitting coil voltage V L1 The primary side charge / discharge voltage V becomes 0. C1 However, as shown by the dashed line in Figure 5B, at the timing t = 11π / (2ω0), cosω0t = 0, so the primary-secondary charge / discharge voltage V C1 = 0.

[0088] Transmitting coil voltage V L1 Furthermore, in the region 11π / 2 < ω0t < 6π, the primary side charge / discharge voltage V satisfies equations (16) and (30). C1 The decrease is in opposite phase, and it further increases in the positive value region. Primary-secondary charge / discharge voltage V C1 Furthermore, in the region 11π / 2 < ω0t < 6π, the transmitting coil voltage V satisfies equations (16) and (31). L1 In the opposite phase to the increase, it further decreases in the negative region. As shown by the dashed line in Figure 5B, the primary side charge / discharge voltage V at time t=t6=6π / ω0 C1 When the voltage becomes approximately -6E0, the transmitting coil voltage V shown by the solid line in Figure 5B is... L1 Since it is approximately 6E0, the requirement of equation (16) is satisfied.

[0089] At t=t5, the transmission coil voltage V L1 Since the minimum value was -6E0, the transmitting coil voltage V was in the time domain t5 ≤ t < t6 of the 6th half-cycle when in transmit mode. L1 This process does not amplify the voltage, but instead shapes the distorted waveform into a cosine wave. Primary side charge / discharge voltage V C1In the time domain t5 ≤ t < t6 of the sixth half-period, the voltage is not amplified and the waveform is maintained. Therefore, the time domain t5 ≤ t < t6, which is the first half-period in transmit mode, is defined as the "waveform shaping cycle". The waveform shaping cycle corresponds to the third freewheel cycle from the definition of the amplification mode. If the sum of the half-period (=π) excitation cycle and the half-period (=π) freewheel cycle is considered to be one period of the switching period (=2π), then the end of the third freewheel cycle corresponds to the end of the third period. The waveform shown in Figure 5B means that the voltage has been amplified sixfold by three periods of complementary switching. This is equivalent to repeating the freewheel cycle defined in equation (32b) p times. From equations (32a) and (32b), including the waveform shaping cycle, which is the first half-period of transmit mode, the voltage value is amplified by n=2p times through the repetition of p periods of complementary switching.

[0090] At the end of the waveform shaping cycle, the primary side charge / discharge voltage V C1 Even at the point t=t6=6π / ω0 when the value is at its minimum, the freewheeling control element Q2 remains ON and the excitation element Q1 remains OFF in transmission mode. Since the freewheeling control element Q2 is ON and the excitation element Q1 is OFF at t=t6, the operation of the second feed-side resonant circuit 2a, which is an RLC series resonant circuit formed by the freewheeling control element Q2, the transmitting capacitor C1, and the transmitting coil L1, continues. The constraint condition in equation (16) is required for the operation of the transmission mode by the second feed-side resonant circuit 2a.

[0091] As can be seen from Figure 5B, at the timing just before t=t6, the primary side charge / discharge voltage V C1 (t→t6)≈-6E0, maintaining the maximum value of the vibration amplitude, and the transmitting coil voltage V L1 The maximum value of the oscillation amplitude is maintained at (t→t6)≈6E0, and the relationship in equation (16) is satisfied for both. Equation (30) is given by the transmitting coil voltage V for t6≦t. L1 This is shifted in the time axis direction by the equivalent of t6, V L1 ≒6E0e -α(t-t6) cosω0(t-t6) ……(33) It can be expressed as follows. The number 6 in the expression of t6 described in the exponent display part on the right side of Equation (33) is a subscript. At time t = t5, the change in the waveform shown by Equation (30) continuous as the distortion waveform in Equation (27) remains as a half-cycle vibration waveform in the time region of t5 ≦ t < t6 in the sixth half-cycle. However, as shown in FIG. 5B, at t = t6, Equation (30) and Equation (33) are smoothly connected and have a cosine wave waveform. That is, in the waveform shaping cycle defined in the time region of t5 ≦ t < t6 in the sixth half-cycle, the voltage is not enlarged, but the waveform of the transmission-side coil voltage V L1 has the effect of shaping the waveform into a normal cosine wave. At t = t6, e -α(t-t6) = 1. That is, as can be seen from FIG. 5B at the timing of t6 ≦ t, the transmission-side coil voltage V L1 vibrates with a cosine wave of frequency ω0, and the envelope of the vibration peak (amplitude) exponentially decays from the maximum value of 6E0 to e ―α(t-t6) , showing the waveform of typical underdamped (damped vibration) of the LCR series resonance circuit.

[0092] Considering the constraint of Equation (16), from Equation (33), the primary-side charge / discharge voltage V C1 also V C1 ≈ -6E0e -α(t-t6) cosω0(t - t6) ……(34) is expressed as follows. Similar to Equation (33), the number 6 in the expression of t6 described in the exponent display part on the right side of Equation (34) is a subscript. At time t = t5, the change in the waveform shown by Equation (31) smoothly continuous with Equation (28) remains as a half-cycle vibration waveform in the time region of t5 ≦ t < t6 in the sixth half-cycle. However, as shown in FIG. 5B, at t = t6, Equation (31) and Equation (34) are smoothly connected. In the waveform shaping cycle defined in the time region of t5 ≦ t < t6 in the sixth half-cycle, the voltage is not enlarged, but similar to the case of the transmission-side coil voltage V L1 , the waveform of the primary-side charge / discharge voltage V C1 is shaped into a cosine wave. At t = t6, e -α(t-t6)= 1. That is, at timing t6 ≤ t, the transmitting coil voltage V is as shown in Figure 5B. L1 The waveform showing the change is in opposite phase to the primary side charge / discharge voltage V. C1 As it oscillates as a cosine wave of frequency ω0, the envelope of the oscillation peak (amplitude) moves from the minimum value of -6E0 at time t6 to e ―α(t-t6) This shows a waveform of under-damping (damped oscillation), typical of an LCR series resonant circuit, which decays exponentially.

[0093] For t6 ≤ t, as can be seen from Figure 5B, the magnetic energy stored in the transmitting coil L1 flows into the transmitting capacitor C1 as a charging current, and the transmitting coil voltage V is determined by equation (33). L1 The voltage begins to decrease. At time t=t7=7π / ω0, when charging from the transmitting capacitor C1 is complete and cosω0t7=-1, the primary side charge / discharge voltage V is as shown in equation (34). C1 The peak value of the vibration is e, as shown by the dashed line in Figure 5B. ―α(t-t6) The energy decays exponentially. For t6 ≤ t, the magnetic energy stored in the transmitting coil L1 is released as a charging current to the transmitting capacitor C1. Therefore, the transmitting coil voltage V, which is the terminal voltage of the transmitting coil L7, is released. L1 The primary side charge / discharge voltage V satisfies equations (16) and (33). C1 As increases, it decreases as shown by the solid line in Figure 5B. At the timing t = 13π / (2ω0), cosω0t = 0, so the transmitting coil voltage V L1 The primary side charge / discharge voltage V becomes 0. C1 However, as shown by the dashed line in Figure 5B, at the timing t = 13π / (2ω0), cosω0t = 0, so the primary-secondary charge / discharge voltage V C1 = 0.

[0094] Transmitting coil voltage V L1 Furthermore, in the region 13π / 2 < ω0t < 7π, the primary side charge / discharge voltage V satisfies equations (16) and (33). C1 In the opposite phase to the increase, it further decreases in the negative value region. Primary-secondary charge / discharge voltage V C1Furthermore, in the region 13π / 2 < ω0t < 7π, the transmitting coil voltage V satisfies equations (16) and (34). L1 In the opposite phase to the decrease, it further increases in the positive region. As shown in Figure 5B, the primary side charge / discharge voltage V at time t=t7=7π / ω0 C1 The oscillation peak (maximum value) is shown, and the transmitting coil voltage V satisfies equation (16). L1 This indicates the oscillation peak (minimum value). That is, the primary side charge / discharge voltage V C1 and the transmitting coil voltage V L1 The envelope of the oscillation peak (amplitude) is obtained from the maximum amplitude at time t6 to e ―α(t-t6) This shows a waveform of under-damping (damped oscillation), typical of an LCR series resonant circuit, which decays exponentially.

[0095] In transmission mode, the primary side charge / discharge voltage V C1 Even at the point when the voltage reaches its maximum value, t=t7=7π / ω0, the excitation element Q1 remains off and the freewheel control element Q2 remains on. If the excitation element Q1 remains off and the freewheel control element Q2 remains on at t=t7, the transient phenomenon of the second feed-side resonant circuit 2a, which is an RLC series resonant circuit formed by the freewheel control element Q2, the transmitting capacitor C1, and the transmitting coil L1, continues. Therefore, in the time domain of the transmission mode t7≦t, the transmitting coil voltage V L1 The vibration peak value is defined in equation (33) and e ―α(t-t6) The voltage decays exponentially, and the primary side charge / discharge voltage V C1 The vibration peak value is also defined in equation (34) and e ―α(t-t6) It decays exponentially.

[0096] In the transmission mode, an electromotive force is generated in the power receiving coil L2 coupled by the dynamic mutual inductance M = M(t) due to the magnetic field generated around the power transmission coil L1 by the current flowing through the power transmission coil L1, and a current flows. At this time, if the characteristics of the power supply side resonance circuit (34a, 2a) and the power receiving side resonance circuit (27a, 3a) are harmonized to enable double resonance, the current flowing through the power receiving coil L2 is efficiently charged to the power receiving capacitor C2 by this double resonance. That is, power is efficiently transmitted from the power supply side resonance circuit (34a, 2a) to the power receiving side resonance circuit (27a, 3a) by double resonance.

[0097] In the power transmission system according to the first embodiment, in the amplification mode, the excitation element Q1 and the reflux control element Q2 are complementarily turned on and off to repeat the excitation cycle and the reflux cycle, thereby amplifying the vibration peak of the composite vibration as shown in FIG. 5B. Then, in the transmission mode, the excitation element Q1 is turned off and the reflux control element Q2 is turned on, and the power is efficiently transmitted from the power supply side resonance circuit (34a, 2a) to the power receiving side resonance circuit (27a, 3a) by double resonance. However, as shown by the dashed-dotted line in FIGS. 5A and 5B, in the power transmission system according to the first embodiment, actually, even in the time region of t0 ≦ t < t5 in the amplification mode, the power receiving coil voltage V due to the current flowing through the power receiving coil L2 L2 is generated.

[0098] As shown by the dashed-dotted line in FIGS. 5A and 5B, the power receiving coil voltage V, which is the voltage between the terminals of the power receiving coil L2 L2 depends on the magnitude and characteristics of the dynamic mutual inductance M = M(t), which is the effect of the magnetic field generated around the power transmission coil L1 by the current flowing through the power transmission coil L1, and the transient response characteristics of the secondary circuit 3a. For example, in the time region of t0 ≦ t < t1 in the amplification mode, the power receiving coil voltage V L2 is, as V L2 ≒-(E0 / 2)sinω0t ……(35) A waveform that crosses zero at \(t = t_0=0\) can be exemplified as an approximate expression.

[0099] That is, although it is in the amplification mode, power transmission is already being carried out from the power supply side resonance circuit to the power reception side resonance circuit. According to the exemplified equation (35), the voltage \(V\) of the power reception side coil L2 decreases along the waveform of a sine wave, and at the timing of \(t_{1 / 2}=\frac{\pi}{2\omega_0}\), since \(\sin\omega_0t = 1\), the voltage \(V\) of the power reception side coil L2 reaches the minimum value. After that, the voltage \(V\) of the power reception side coil L2 increases along the waveform of a sine wave from the minimum value while maintaining a negative value according to equation (35) in the region of \(\frac{\pi}{2}<\omega_0t<\pi\). As shown by the dashed line in FIGS. 5A and 5B, \(\sin\omega_0t = 0\) at the time \(t = t_1=\frac{\pi}{\omega_0}\).

[0100] As shown by the dashed line in FIGS. 5A and 5B, in the time region of \(t_1\leq t < t_2\) in the amplification mode, the voltage \(V\) of the power reception side coil L2 is shifted in the time axis direction by the amount corresponding to \(t_1\), and \(V\) L2 \(\approx E_0\sin\omega_0(t - t_1)\cdots(36)\) and a waveform that crosses zero at \(t = t_1\) can be exemplified as an approximate expression. The amplitude and phase of the exemplified equation (36) also depend on the magnitude and characteristics of the dynamic mutual inductance \(M = M(t)\) between the power transmission side coil \(L_1\) and the power transmission side coil \(L_1\), and further on the transient response characteristics of the secondary side circuit 3a, etc.

[0101] What is important is that at the timing of \(t_1\leq t < t_2\), reflecting that the voltage \(V\) of the power transmission side coil L1 has an amplitude of \(2E_0\), which is twice the power supply voltage \(E_0\) supplied from the DC power supply 5 as shown in equation (18), and the phase is delayed, but the voltage \(V\) of the power reception side coil shown in equation (36) L2 and the voltage \(V\) of the power transmission side coil shown in equation (35) L1It has been amplified to twice that amount, enabling high-efficiency wireless transmission. It should also be noted that, while the time is within t1 ≦ t < t2 in the amplification mode, high-efficiency power transmission has already occurred from the power supply-side resonance circuit to the power reception-side resonance circuit. The power reception-side coil voltage V L2 increases along the waveform of a sine wave, and at the timing of t = 3π / (2ω0), sinω0t = 1, so the power reception-side coil voltage V L2 = reaches its maximum value. The power reception-side coil voltage V L2 decreases sinusoidally from the maximum value in accordance with Equation (36) in the region of 3π / 2 < ω0t < 2π. As shown by the dashed-dotted line in FIGS. 5A and 5B, sinω0t = 0 at the time t = t2 = 2π / ω0.

[0102] In the time region of t2 ≦ t < t3 in the amplification mode, the power reception-side coil voltage V L2 is shifted in the time axis direction by the amount corresponding to t2, V L2 ≒ -(3 / 2)E0sinω0(t - t2) ……(37) and can be exemplified as an approximate expression of a waveform that crosses zero at t = t2. The amplitude and phase of the exemplified Equation (37) also depend on the magnitude and characteristics of the dynamic mutual inductance M = M(t) between the power transmission-side coil L1 and the power transmission-side coil L1, and further on the transient response characteristics of the secondary-side circuit 3a. What should be noted here is that, at the timing of t2 ≦ t < 3, reflecting that the power transmission-side coil voltage V L1 has an amplitude of 3E0, which is three times the power supply voltage E0 supplied from the DC power supply 5 as shown in Equation (21), and the phase is delayed, but the power reception-side coil voltage V L2 shown by Equation (37) is amplified to three times the power transmission-side coil voltage V L1 shown by Equation (35), enabling high-efficiency wireless transmission.

[0103] Moreover, as can be seen from the comparison with Equations (35) and (36), as time elapses in the amplification mode, the power transmission-side coil voltage V L1The voltage increases, and although there is a phase delay, the receiving coil voltage V is wirelessly transmitted by double resonance. L2 A characteristic of the power transmission system according to the first embodiment is that the voltage V is also gradually increasing. L2 The voltage decreases along the sinusoidal waveform, and at the timing t=5π / (2ω0), sinω0t=1, so the receiving coil voltage V L2 The voltage V of the receiving coil is the minimum value. L2 In the region 5π / 2 < ω0t < 3π, sinω0t increases from its minimum value in the negative region along the sinusoidal waveform, as follows according to equation (37). As shown by the dashed lines in Figures 5A and 5B, sinω0t = 0 at time t = t3 = 3π / ω0.

[0104] In the time domain of amplification mode, t3 ≤ t < 4, the receiving coil voltage V L2 This is shifted in the time axis direction by the equivalent of t3, V L2 ≈2E0sinω0(t-t3) ……(38) Thus, a waveform that crosses zero at t=t3 can be illustrated as an approximate representation. The amplitude and phase of the illustrated equation (38) depend on the magnitude and characteristics of the dynamic mutual inductance M=M(t), as well as the transient response characteristics of the secondary circuit 3a. What should be considered here is the transmitting coil voltage V at the timing t3≦t<4. L1 As shown in equation (24), the amplitude of the voltage 4E0 is four times the power supply voltage E0 supplied from the DC power supply 5, and the phase is lagging, but the receiving coil voltage V shown in equation (38) L2 However, the transmitting coil voltage V shown in equation (35) L1 This allows for amplification up to four times, enabling highly efficient wireless transmission.

[0105] Moreover, the transmitting coil voltage V increases with the time elapsed in the amplification mode. L1 As a result of the gradual increase, the receiving coil voltage V transmitted wirelessly by double resonance L2However, as can be seen from the comparison with equations (35) to (37), it gradually increases along the sinusoidal waveform. As shown by the dashed line in Figure 5B, at the timing t = 7π / (2ω0), sinω0t = 1, so the receiving coil voltage V L2 This is the maximum value. Receiving coil voltage V L2 The receiving coil voltage V L2 The voltage changes along the sine wave waveform, and in the region 7π / 2 < ω0t < 4π, it begins to decrease from its maximum value in the positive value region, as follows (38). As shown by the dashed line in Figure 5B, sinω0t = 0 at time t = t4 = 4π / ω0, so the receiving coil voltage V L2 This equals 0V.

[0106] In the time domain of amplification mode, t4 ≤ t < 5, the receiving coil voltage V L2 This is shifted in the time axis direction by the equivalent of t4. V L2 ≒-(5 / 2)E0sinω0(t-t4) ……(39) Thus, a waveform that crosses zero at t=t4 can be used as an approximate representation. The amplitude and phase of equation (39) depend on the magnitude and characteristics of the dynamic mutual inductance M=M(t), as well as the transient response characteristics of the secondary circuit 3a. What should be considered here is the transmission coil voltage V at the timing t4≦t <5. L1 As shown in equation (27), the amplitude of the voltage 5E0 has become five times the power supply voltage E0 supplied from the DC power supply 5, and the phase is lagging, but the receiving coil voltage V shown in equation (39) L2 However, the transmitting coil voltage V shown in equation (35) L1 This allows for amplification up to five times, enabling highly efficient wireless transmission.

[0107] Moreover, the transmitting coil voltage V increases with the time elapsed in the amplification mode. L1 As a result of the gradual increase, the receiving coil voltage V transmitted wirelessly by double resonance L2However, as can be seen from the comparison with equations (35) to (38), it is gradually increasing. As shown by the dashed line in Figure 5B, the receiving coil voltage V L2 The voltage changes along the sine wave waveform, and at the timing t=9π / (2ω0), sinω0t=1, so the receiving coil voltage V L2 The voltage V is minimized along the sinusoidal waveform. L2 In the region 9π / 2 < ω0t < 5π, sinω0t starts increasing from its minimum value in the negative region, following the sinusoidal waveform. As shown by the dashed line in Figure 5B, at time t=t5=5π / ω0, sinω0t=1, so the receiving coil voltage V L2 This equals 0V.

[0108] In the time domain t5 ≤ t < t6 of the transmission mode, the receiving coil voltage V L2 This is shifted in the time axis direction by the equivalent of t5. V L2 ≈3E0sinω0(t-t5) ……(40) Thus, a waveform that crosses zero at t=t5 can be used as an approximate representation. The amplitude and phase of equation (40) depend on the magnitude and characteristics of the dynamic mutual inductance M=M(t), as well as the transient response characteristics of the secondary circuit 3a. What is important in the power transmission system according to the first embodiment is that at the timing of time t5≦t < t6, the transmitting coil voltage V L1 As shown in equation (30), the amplitude of the voltage 6E0 is six times the power supply voltage E0 supplied from the DC power supply 5, and the phase is lagging, but the receiving coil voltage V shown in equation (40) L2 However, the transmitting coil voltage V shown in equation (35) L1 This allows for amplification up to six times, enabling highly efficient wireless transmission.

[0109] Furthermore, the transmitting coil voltage V increases with the transition from amplification mode to transmission mode. L1 As a result of the gradual increase, the receiving coil voltage V transmitted wirelessly by double resonance L2However, as can be seen from the comparison with equations (35) to (39), it is increasing. Receiving coil voltage V L2 The voltage changes along the sine wave waveform, and at the timing t=11π / (2ω0), sinω0t=1, so the receiving coil voltage V L2 This is the maximum value. Receiving coil voltage V L2 In the region 11π / 2 < ω0t < 6π, the voltage begins to decrease in the positive value region along the sinusoidal waveform from the maximum value. As shown by the dashed line in Figure 5B, sinω0t = 0 at time t = t6 = 6π / ω0, so the receiving coil voltage V L2 This equals 0V.

[0110] In the time domain t6 ≤ t of the transmission mode, the receiving coil voltage V L2 This is shifted in the time axis direction by the equivalent of t6, V L2 ≒-4E0sinω0(t-t6) ……(41) Thus, a waveform that crosses zero at t=t6 can be used as an approximate representation. The amplitude and phase of equation (41) depend on the magnitude and characteristics of the dynamic mutual inductance M=M(t), as well as the transient response characteristics of the secondary circuit 3a. What is important in the power transmission system according to the first embodiment is that at the timing of time t6≦t, the transmitting coil voltage V L1 As shown in equation (33), the amplitude of the voltage 6E0 is six times the power supply voltage E0 supplied from the DC power supply 5, and the phase is lagging, but the receiving coil voltage V shown in equation (41) L2 However, the transmitting coil voltage V shown in equation (35) L1 This allows for amplification up to eight times, enabling highly efficient wireless transmission.

[0111] Furthermore, the transmitting coil voltage V increases with the passage of time in the transmission mode. L1 As a result of the gradual increase, the receiving coil voltage V transmitted wirelessly by double resonance L2 However, as can be seen from the comparison with equation (40), it is increasing. At the timing t = 13π / (2ω0), sinω0t = 1, so the receiving coil voltage VL2 The voltage V is minimized along the sinusoidal waveform. L2 In the region 13π / 2 < ω0t < 7π, it starts increasing from a minimum value in the negative region, following a sinusoidal waveform, as in (41). At time t=t7=7π / ω0, sinω0t=0, so the receiving coil voltage V L2 This equals 0V.

[0112] As the transmission time elapses, the receiving coil voltage V becomes as shown in Figure 5B. L2 In the time domains t7 ≤ t < t8 and t8 ≤ t < t9, the receiving coil voltage V in the time domain t0 ≤ t < t1 is... L2 8 times, time t9 ≤ t < t 10 In the time domain, the receiving coil voltage V in the time domain t0 ≤ t < t1. L2 It has become 9.8 times. Furthermore, as the transmission time elapses, the receiving coil voltage V L2 This is time t in Figure 5B. 10 In the time domain ≤ t, the receiving coil voltage V in the time domain t0 ≤ t < t1 is... L2 An amplitude more than 10 times greater has been achieved.

[0113] The amplification constant β is dependent on time t. β = β(t) ……(42) If defined as such, the receiving coil voltage V L2 teeth, V L2 ≒-(E0 / 2)e βt sinω0t ……(43) The function changes in the form of . Compared with Figure 5B, in the time domain t0 ≤ t < t7, β(t) ≈ α, and the amplification constant β is approximately the same value as the damping constant α. Figure 5B shows that in the time domain t7 ≤ t, β(t) ≈ 0. Although not shown in Figure 5B, as time progresses further, β(t) < 0.

[0114] On the other hand, due to double resonance, electrical energy is wirelessly transmitted from the power supply side resonant circuit (34a, 2a) to the power receiving side resonant circuit (27a, 3a), resulting in a power receiving side coil voltage V L2 As the voltage increased, the transmission coil voltage V L1 Figure 5B shows that the voltage is attenuated exponentially by the attenuation constant α. As shown in equation (33), the transmitting coil voltage V L1 It vibrates as a cosine wave with frequency ω0, and the envelope of the vibration peak (amplitude) is e ―α(t-t6) In this form, the change is an exponentially damped oscillation (underdamping). Note that, as shown in Figure 5B, at time t 11 In the time domain ≤ t, the transmitting coil voltage V L1 The waveform is distorted from a sine wave of damped oscillation. Time t 11 In the time domain ≤ t, the effect of reverse transmission due to the return of electrical energy from the secondary circuit 3a to the power supply side resonant circuit (34a,2a) becomes significant, and the power supply side coil voltage V L1 This is likely because it does not become small enough.

[0115] The coupling coefficient K in AC theory AC When the relationship between the transmitting coil L1 and the receiving coil L2 results in an equivalent coupling coefficient K that can be approximated to approximately 0.6, it is suitable for contactless power transmission using double resonance between the supplying resonant circuit (34a, 2a) and the receiving resonant circuit (27a, 3a). Conductor cross-sectional area: 16 mm² 2 In the case of a spiral planar coil with 9 turns of wiring cable, to achieve an equivalent coupling coefficient K=0.6, the transmission distance d needs to be approximately 0mm to 20mm. On the other hand, the coupling coefficient K in AC theory... AC In order to achieve a relationship between the transmitting coil L1 and the receiving coil L2 that results in an equivalent coupling coefficient K that can be approximated to 0.1, the transmission distance d is approximately 100 mm.

[0116] Figures 3, 5A, and 5B show the transient response waveforms when the transmission distance d = 600 mm and the equivalent coupling coefficient K = 0.05. According to the vibration amplification circuit of the first embodiment, despite the simplified circuit configuration, the vibration amplitude can be amplified to an integer multiple of the power supply voltage E0 supplied by the DC power supply 5, as shown in equations (32a) and (32b). Therefore, by using the vibration amplification circuit of the first embodiment as the power supply side resonant circuit (34a, 2a), effective double resonance between the power supply side resonant circuit (34a, 2a) and the power receiving side resonant circuit (27a, 3a) becomes possible even when the transmission distance d is large and the equivalent coupling coefficient K is small.

[0117] Figure 6 illustrates a contactless power transmission system with the transmitting coil L1 and receiving coil L2 schematically shown in an exaggerated (enlarged) manner. According to the power transmission system of the first embodiment, by using the vibration amplification circuit of the first embodiment in the power supply side resonant circuit (34a, 2a), even when the transmission distance d = 600 mm, as defined as the distance between the transmitting coil L1 and the receiving coil L2, efficient contactless power transmission (wireless power transmission) can be achieved as shown in Figures 3, 5A, and 5B. Furthermore, in order to charge the load 6, which is an on-board rechargeable battery of the vehicle 31b, using the power transmission system of the first embodiment, the rear wheel stopper 11 is used as a magnetic coupling degree control mechanism to control the transmission distance d between the transmitting coil L1 and the receiving coil L2 to approximately 600 mm, thereby enabling efficient contactless power supply.

[0118] As described above, the power transmission system according to the first embodiment solves the problem of conventional contactless power transmission systems, which suffer from attenuation due to parasitic resistance and can only amplify up to the power supply voltage E0, enabling effective contactless power transmission (wireless power transmission) even over long distances of d ≤ 600 mm. Moreover, the contactless power transmission system using the vibration amplification circuit according to the first embodiment as the power supply side resonant circuit (34a, 2a) does not require a high-frequency inverter or switching power supply, thus simplifying the circuit configuration, preventing damage to circuit elements, and being inexpensive.

[0119] (First embodiment, first modified example) As shown in Figure 7, the power transmission system according to the first modified example of the first embodiment of the present invention is a transmission system that performs amplitude amplification by voltage expansion in the power supply side resonant circuits (34b, 2b), and supplies wavelet-shaped electromagnetic energy from the power supply side resonant circuits (34b, 2b) to the power receiving side resonant circuits (27a, 3a) without contact, thereby creating double resonance between the power supply side resonant circuits (34b, 2b) and the power receiving side resonant circuits (27a, 3a). As shown in Figure 7, the power supply side resonant circuits (34b, 2b) include a drive control circuit 34b and a second power supply side resonant circuit 2b, and the power receiving side resonant circuits (27a, 3a) include a power receiving circuit 27a and a secondary side circuit 3a. The second power supply side resonant circuit 2b is a circuit that is transiently formed in a time-division manner inside the power supply side resonant circuits (34a, 2b) in a complementary manner to the first power supply side resonant circuit.

[0120] The power transmission system according to the first modification of the first embodiment has a configuration in which an adjustment element Q3 is added to the circuit configuration of the power transmission system according to the first embodiment shown in Figure 1A. As shown in Figure 7, the control electrodes (gate electrodes, etc.) of the excitation element Q1, the freewheeling control element Q2, and the adjustment element Q3 are each driven and controlled by the primary-side switching element drive circuit 340b. Unlike the excitation element Q1 and the freewheeling control element Q2, which switch in a complementary manner, the adjustment element Q3 is a circuit element that is always on (conducting state) and is a variable resistor element and variable capacitor element intended to adjust the impedance of the RLC series resonant circuit. The adjustment element Q3 is added with the aim of improving the efficiency of contactless power transmission by double resonance between the power supply side resonant circuit (34a, 2a) and the power receiving side resonant circuit (27a, 3a) by adjusting the impedance of the RLC series resonant circuit. In other words, in the power transmission system according to the first modified example of the first embodiment, the excitation element Q1 and the freewheeling control element Q2 are switched in a complementary manner, and the first power supply side resonant circuit and the second power supply side resonant circuit 2a are switched alternately in a time-division manner, which is the same as the circuit configuration described in Figure 2. However, by adding an adjustment element Q3 which is a variable resistor element and a variable capacitance element, the efficiency of contactless power transmission by double resonance is increased.

[0121] As shown in Figure 7, the excitation element Q1, power transmission control element Q2, and adjustment element Q3 can be power semiconductor elements including FETs, SITs, BJTs, and thyristors such as GTO thyristors and SI thyristors, similar to the circuit configuration shown in Figure 2. However, voltage-driven switching elements such as MOSFETs, MISFETs, MISSITs, IGBTs, and MOS-controlled SI thyristors are preferred for the excitation element Q1 and power transmission control element Q2, but since the adjustment element Q3 is used as a variable resistor and variable capacitor, junction FETs, junction SITs, BJTs, GTO thyristors, and junction SI thyristors are preferred. If the adjustment element Q3 is a BJT, the control electrode of the adjustment element Q3 is the base electrode, and the base electrode of the BJT is current-controlled by the base current supplied from the primary-side switching element drive circuit 340b. As the adjustment element Q3, a wide channel structure with a widened gate spacing, particularly for using a normally-on junction SIT as a variable resistor element, or a structure that uses changes in junction capacitance such as a BJT or thyristor can be employed. Furthermore, a super-stepped junction can be used as a variable capacitance element that efficiently changes capacitance.

[0122] In the implementation circuit shown in Figure 7, considering the freewheeling current from the transmitting coil L1, the first freewheeling diode FWD1 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the excitation element Q1, the second freewheeling diode FWD1 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the freewheeling control element Q2, and the third freewheeling diode FWD3 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the adjustment element Q3. However, since the adjustment element Q3 is not a switching element, the third freewheeling diode FWD3 can be omitted. Similar to the circuit shown in Figure 2, to prevent the freewheeling current from the transmitting coil L1 from flowing back to the DC power supply 5, the power supply diode D1 is connected in series between the DC power supply 5 and the excitation element Q1. In the implementation circuit shown in Figure 7, the equivalent impedance X of the load 6 is also Leq Charging capacity C s It is expressed as an approximation.

[0123] Similar to the first embodiment, the charging voltage VC SAssume that the initial value is a sufficiently high voltage, close to fully charged. First, at time t0, similar to that shown in Figure 4A, with the freewheel control element Q2 in the off state and the excitation element Q1 and adjustment element Q3 in the on state, when the power supply voltage E0 from the DC power supply 5 is stepped in, at time t=t0=0, the transmitting side capacitor C1 is not charged and the primary side charge / discharge voltage V C1 Since = 0, by the requirement of equation (9), the transmitting coil voltage V L1 ΔV L10 = Performs an impulse ascent of E0.

[0124] When t0 < t < t1, the transmitting coil L1 discharges the charging current to the transmitting capacitor C1 as a return current. Therefore, the transmitting coil voltage V L1 The primary side charge / discharge voltage V satisfies equations (9) and (13). C1 As V increases, the transmitting coil voltage V increases, as shown by the solid line in Figure 3. L1 The primary side charge / discharge voltage V decreases. C1 The voltage increases as shown by the dashed line in Figure 3. The primary side charge / discharge voltage V is immediately before time t=t1=π / ω0. C1 When it approaches ≈ 2E0, the transmitting coil voltage V can be determined from the requirement of equation (9). L1 It becomes approximately -E0.

[0125] At time t=t1=π / ω0, if the excitation element Q1 is turned off while the adjustment element Q3 remains ON, and the freewheel control element Q2 is turned ON, then, according to the requirements of equation (16), the primary side charge / discharge voltage V C1 As shown in equation (19), the equivalent expansion of the power supply voltage occurs, and the transmitting coil voltage V L1 As shown in equation (18), the power supply voltage is equivalently expanded. At time t=t2=2π / ω0, a complementary switching transition is performed in which the recirculation control element Q2 is turned off and the excitation element Q1 is turned on while the adjustment element Q3 remains in the ON state. When the recirculation control element Q2 is turned off and the excitation element Q1 is turned on at t=t2, by the requirement of equation (9), the primary side charge / discharge voltage V C1 As shown in equation (20), the equivalent expansion of the power supply voltage occurs, and the transmitting coil voltage V L1As shown in equation (21), the power supply voltage is equivalently expanded. Thereafter, a similar operation of equivalent expansion of the power supply voltage is continued by complementary switching between the excitation element Q1 and the power transmission control element Q2, while the adjustment element Q3 remains in the ON state. Although the adjustment element Q3 does not function effectively as a switching element, as a variable resistor and variable capacitor element, it can adjust the impedance of the RLC series resonant circuit, thereby increasing the efficiency of contactless power transmission by double resonance between the power supply side resonant circuit (34b, 2b) and the power receiving side resonant circuit (27a, 3a).

[0126] As described above, the power transmission system according to the first modification of the first embodiment can achieve the same remarkable effect as the power transmission system according to the first embodiment, even over long distances of d ≤ 600 mm, by amplifying the vibration amplitude to an integer multiple of the power supply voltage E0 in the power supply side resonant circuit (34b, 2b), enabling efficient contactless power transmission (wireless power transmission). Furthermore, by adding an adjustment element Q3 equipped with the functions of a variable resistor and a variable capacitor to the power supply side resonant circuit (34b, 2b), the efficiency of contactless power transmission by double resonance is increased, enabling a more effective contactless power transmission system even when the transmission distance d is long.

[0127] (Second modified example of the first embodiment) A power transmission system according to a second modification of the first embodiment of the present invention is a system that utilizes the voltage amplification effect in the power supply side resonant circuit (34b, 2b) as a vibration amplification circuit as shown in Figure 8. It is characterized by supplying wavelet-shaped electromagnetic energy, which has been amplified by voltage from the power supply side resonant circuit (34b, 2b), to the power receiving side resonant circuit (27b, 3b), thereby enabling contactless power transmission between the power supply side resonant circuit (34b, 2b) and the power receiving side resonant circuit (27b, 3b) through double resonance without contact. As shown in Figure 8, the power supply side resonant circuit (34b, 2b) comprises a drive control circuit 34b and a second power supply side resonant circuit 2b, and the power receiving side resonant circuit (27b, 3b) comprises a power receiving circuit 27b and a secondary side circuit 3b. The second power supply side resonant circuit 2b is a circuit that is transiently formed in a time-division manner inside the power supply side resonant circuit (34a, 2b) in a complementary manner to the first power supply side resonant circuit.

[0128] The power receiving circuit 27b of the power transmission system according to the second modification of the first embodiment has a configuration in which a power receiving control element Q4 is added to the power transmission system shown in Figure 7. The control electrode (gate electrode) of the newly added power receiving control element Q4 is driven and controlled by a secondary-side switching element drive circuit 270a provided on the vehicle side. On the other hand, as shown in Figure 8, the control electrodes (gate electrode or base electrode) of the excitation element Q1, the freewheeling control element Q2, and the adjustment element Q3 are driven and controlled by a primary-side switching element drive circuit 340b. Similar to the power transmission systems shown in Figures 2 and 7, in the power supply side resonant circuits (34b, 2b), the excitation element Q1 and the freewheeling control element Q2 are switched complementaryly, and the first power supply side resonant circuit and the second power supply side resonant circuit 2a are switched alternately in a time-division manner. Power semiconductor elements such as FETs, similar to those in the power transmission systems shown in Figures 2 and 7, are used as the excitation element Q1, freewheeling control element Q2, adjustment element Q3, and power receiving control element Q4 shown in Figure 8. Due to the requirement for low internal resistance and market availability, it is considered industrially advantageous to use MOSFETs as the excitation element Q1, freewheeling control element Q2, and power receiving control element Q4 in the implementation circuit shown in Figure 8. On the other hand, as already mentioned, a junction-type structure such as a junction-type SIT or BJT is preferred for the adjustment element Q3 used as a variable resistor or variable capacitor.

[0129] As explained in the power transmission system shown in Figure 2, Joule heat generation is significant in high-power contactless power transmission systems. In the power transmission system according to the second modification, only four power semiconductor elements are needed, used as the excitation element Q1, freewheeling control element Q2, adjustment element Q3, and power receiving control element Q4. Therefore, a cooling structure to prevent element damage due to heat generation can be easily designed, and the generation of stray resistance, stray capacitance, and stray inductance can also be minimized. However, as explained in the first modification, the adjustment element Q3, used as a variable resistor or variable capacitance element, is assumed to be used in a constantly ON state, and the adjustment element Q3 can be omitted. Therefore, the number of power semiconductor elements can be reduced to three. In the end, since only simple control of the three switching elements, the excitation element Q1, freewheeling control element Q2, and power receiving control element Q4, is required, it is also easy to design a system that suppresses Joule heat generation by increasing the voltage of the second power supply side resonant circuit 2b.

[0130] In the implementation circuit shown in Figure 8, the first freewheeling diode FWD1 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the excitation element Q1, the second freewheeling diode FWD1 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the freewheeling control element Q2, the third freewheeling diode FWD3 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the adjustment element Q3, and the fourth freewheeling diode FWD4 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the power receiving control element Q4. However, since the adjustment element Q3 is not a switching element, the third freewheeling diode FWD3 can be omitted. As shown in Figure 8, the fourth freewheeling diode FWD4 is provided in the direction that allows the freewheeling current from the power receiving coil L2 to flow, so it is provided in the opposite direction to the third freewheeling diode FWD3. Similar to the circuit shown in Figure 2, etc., in order to prevent the freewheeling current from the power transmitting coil L1 from flowing back to the DC power supply 5, the power supply side diode D1 is connected in series between the DC power supply 5 and the excitation element Q1. In the implementation circuit shown in Figure 8, the equivalent impedance X of the load 6 is also Leq Charging capacity C s It is expressed as an approximation.

[0131] Similar to the first embodiment, the charging voltage VC S The power transmission system according to the second modification of the first embodiment will be described assuming that the initial value is a sufficiently high voltage close to full charge, and that the adjustment element Q3 used as a variable resistor or variable capacitor is always in the ON state. First, at time t0, similar to that shown in Figure 4A, the freewheeling control element Q2 and the power receiving control element Q4 are turned OFF, and the excitation element Q1 is turned ON. When the excitation element Q1 is turned ON and the power supply voltage E0 from the DC power supply 5 is stepped in, at time t=t0=0, the transmitting capacitor C1 is not charged and the primary side charge / discharge voltage V C1 Since = 0, from the requirement of equation (9), the transmitting coil voltage V L1 ΔV L10 An impulse rise of =E0 occurs. Primary charge / discharge voltage V of the transmitting capacitor C1. C1 It is charged to a constant voltage while ringing.

[0132] When t0 < t < t1, the transmitting coil L1 discharges the charging current to the transmitting capacitor C1 as a return current. Therefore, the transmitting coil voltage V L1 The primary side charge / discharge voltage V satisfies equations (9) and (13). C1 As V increases, the transmitting coil voltage V increases, as shown by the solid line in Figure 3. L1 The value decreases along the cosine function, and the primary side charge / discharge voltage V C1 As shown by the dashed line in Figure 3, it increases according to the cosine function. The primary side charge / discharge voltage V is immediately before time t=t1=π / ω0. C1 When it approaches ≈ 2E0, the transmitting coil voltage V can be determined from the requirement of equation (9). L1 It becomes approximately -E0.

[0133] At time t=t1=π / ω0, similar to that shown in Figure 4C, if the excitation element Q1 is turned off and the freewheel control element Q2 is turned on while the adjustment element Q3 remains in the ON state, then, according to the requirements of equation (16), the primary side charge / discharge voltage V C1 As shown in equation (19), the equivalent expansion of the power supply voltage occurs, and the transmitting coil voltage V L1After an impulsive instantaneous change, the waveform changes with an equivalent expansion of the power supply voltage as shown in equation (18). Since the adjustment element Q3 is always on, the electromagnetic energy stored in the transmitting capacitor C1 is stored in the transmitting coil L1, and furthermore, a double resonance occurs between the supply side resonant circuit (34b, 2b) and the receiving side resonant circuit (27b, 3b). When the electromagnetic energy stored in the transmitting capacitor C1 begins to move to the transmitting coil L1, the primary side charge / discharge voltage V C1 It decreases according to the cosine function shown in equation (19).

[0134] At time t=t1, the excitation element Q1 is turned off, and after a certain period of time, the power receiving control element Q4 is turned on. The electromagnetic energy transmitted to the power receiving coil L2 by the double resonance from the power supply side resonant circuit (34b,2b) to the power receiving side resonant circuit (27b,3b) charges the power receiving side capacitor C2 if the power receiving control element Q4 is turned on. When charging of the power receiving side capacitor C2 begins, the secondary side charge / discharge voltage V of the power receiving side capacitor C2 is reached. C2 The secondary charge / discharge voltage V starts increasing from a negative maximum and becomes a positive value. C2 While the charging current I is taking a negative value CS No current flows, but the secondary side charge / discharge voltage V C2 When it becomes a positive value, the charging current I CS It begins to stand up.

[0135] Charging current I CS At the moment the power starts to rise, the power receiving control element Q4 is turned off. The turn-off of the power receiving control element Q4 is due to the double resonance between the power supply side resonant circuit (34b, 2b) and the power receiving side resonant circuit (27b, 3b), which controls the secondary side charge / discharge voltage V C2 When it reaches its maximum, the primary side charge / discharge voltage V C1 This is the point at which the voltage becomes 0V, which is 3π / (2ω0). In the case of the power transmission system shown in Figure 7, the secondary side charge / discharge voltage V C2 When it reaches its maximum, a very small current flows through the receiving coil L2, but in the power transmission system according to the second modified example, the receiving control element Q4 is in the off state, so the charging current I is more effectively utilized. CS The load begins to move to load 6. Charging current I CSThe value continues to increase even after the power receiving control element Q4 turns off, reaching a peak value, then decreases to zero.

[0136] After a certain period of time, the secondary charge / discharge voltage V C2 The maximum value of decreases along the cosine function, but the charging voltage VC S If the charging current I is high, C Secondary side charge / discharge voltage V C2 The decrease in the maximum value is small, and the effect on the double resonance between the power supply side resonant circuit (34b,2b) and the power receiving side resonant circuit (27b,3b) is minimal. Charging current I C After the current becomes 0A, turning the power receiving control element Q4 back on starts charging the power receiving capacitor C2. The electromagnetic coupling between the second power supply side resonant circuit 2b and the secondary side circuit 3b causes the primary side charge / discharge voltage V of the power transmitting side capacitor C1. C1 The voltage begins to decrease from OV to negative. Discharge of the receiving side capacitor C2 begins, and freewheeling occurs due to double resonance between the supply side resonant circuit (34b,2b) and the receiving side resonant circuit (27b,3b). The freewheeling current flowing through the transmitting side coil L1 charges the transmitting side capacitor C1, and the primary side charge / discharge voltage V C1 However, it starts to increase from a negative maximum value to a positive value. Secondary charge / discharge voltage V C2 The voltage begins to decrease along the cosine function, reaches a negative maximum, and then becomes 0V.

[0137] As described above, the power transmission system according to the second modified example of the first embodiment solves the problem of conventional contactless power transmission systems, which suffer from attenuation due to parasitic resistance and can only amplify up to the power supply voltage E0. Even when the transmission distance d is long, the vibration amplitude is amplified to an integer multiple of the power supply voltage E0, enabling efficient contactless power transmission (wireless power transmission).

[0138] (Third modified example of the first embodiment) A third modified example of the first embodiment of the present invention is a power transmission system that utilizes the voltage amplification effect in a power supply side resonant circuit (34b, 2b) as a vibration amplification circuit, as shown in Figure 9. The system is characterized by supplying wavelet-shaped electromagnetic energy, which has been amplified by voltage from the power supply side resonant circuit (34b, 2b), to the power receiving side resonant circuit (27c, 3b), thereby enabling contactless power transmission between the power supply side resonant circuit (34b, 2b) and the power receiving side resonant circuit (27c, 3b) through double resonance. As shown in Figure 9, the power supply side resonant circuit (34b, 2b) comprises a drive control circuit 34b and a second power supply side resonant circuit 2b, and the power receiving side resonant circuit (27c, 3b) comprises a power receiving circuit 27c and a secondary side circuit 3b. The second power supply side resonant circuit 2b is a circuit that is transiently formed in a time-division manner inside the power supply side resonant circuit (34a, 2b) in a complementary manner to the first power supply side resonant circuit.

[0139] The power receiving circuit 27c, which constitutes the power receiving side resonant circuits (27c, 3b) of the power transmission system according to the third modified example, has a configuration in which a load transfer control element Q5 is added to the power receiving circuit 27b of the power transmission system according to the second modified example of the first embodiment. The power supply side resonant circuits (34b, 2b) of the power transmission system according to the third modified example have complementary switching operations of the excitation element Q1 and the freewheeling control element Q2, and the first power supply side resonant circuit and the second power supply side resonant circuit 2a are alternately switched in a time-division manner, which is a technical feature common to the circuit configuration described in Figure 2. The excitation element Q1, freewheeling control element Q2, adjustment element Q3, power receiving control element Q4 and load transfer control element Q5 shown in Figure 9 can be power semiconductor elements such as FETs, similar to the circuits shown in Figures 2, 7 and 8. The control electrodes (gate electrodes) of the excitation element Q1, the freewheel control element Q2, and the regulating element Q3 are driven and controlled by the primary-side switching element drive circuit 340b, while the control electrodes (gate electrodes or base electrodes) of the power receiving control element Q4 and the load transfer control element Q5 are driven and controlled by the secondary-side switching element drive circuit 270b provided on the vehicle side. Considering the requirement for low internal resistance, and given current market availability, it is preferable to use MOSFETs as the excitation element Q1, freewheel control element Q2, power receiving control element Q4, and load transfer control element Q5 in the implementation circuit shown in Figure 9. A junction-type structure such as a junction-type SIT or BJT is preferred for the regulating element Q3.

[0140] In the power transmission system according to the third modified example, only five power semiconductor elements are needed, used as the excitation element Q1, freewheel control element Q2, adjustment element Q3, power receiving control element Q4, and load transfer control element Q5. Therefore, a cooling structure to prevent Joule heat generation can be easily designed, and the generation of stray resistance, stray capacitance, and stray inductance can also be minimized. However, as explained in the first modified example, the adjustment element Q3 is assumed to be used in a constantly ON state, and therefore the adjustment element Q3 can be omitted. Consequently, it is also possible to use only four power semiconductor elements. In the end, since only simple control of the four switching elements—excitation element Q1, freewheel control element Q2, power receiving control element Q4, and load transfer control element Q5—is required, it is also easy to design a system that suppresses Joule heat generation by increasing the voltage of the second power supply side resonant circuit 2b.

[0141] In the implementation circuit shown in Figure 9, the first freewheeling diode FWD1 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the excitation element Q1, the second freewheeling diode FWD1 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the freewheeling control element Q2, the third freewheeling diode FWD3 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the adjustment element Q3, the fourth freewheeling diode FWD4 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the power receiving control element Q4, and the fifth freewheeling diode FWD5 is connected in parallel as a protection element between the source and drain of the MOSFET acting as the fourth semiconductor element Q4. However, since the adjustment element Q3 is not a switching element, the third freewheeling diode FWD3 can be omitted. As shown in Figure 9, the fourth freewheeling diode FWD4 is provided in the direction that allows the freewheeling current from the power receiving coil L2 to flow, so it is provided in the opposite direction to the third freewheeling diode FWD3, as in Figure 8. Similar to the circuits shown in Figures 2, 7, and 8, a power supply diode D1 is connected in series between the DC power supply 5 and the excitation element Q1 to prevent the return current from the transmitting coil L1 from returning to the DC power supply 5. In the implemented circuit shown in Figure 9, the equivalent impedance X of the load 6 is also Leq Charging capacity C s It is expressed as an approximation.

[0142] As shown in Figure 9, even when the power transmission system according to the second modified example is configured with the addition of a load transfer control element Q5, the essence of the voltage amplification effect of the power supply side resonant circuits (34b, 2b) that function as vibration amplification circuits remains unchanged. The basic operation of the vibration amplification circuit and the primary side charge / discharge voltage V in the double resonance between the power supply side resonant circuits (34b, 2b) and the power receiving side resonant circuits (27c, 3c) remain unchanged. C1 or secondary charge / discharge voltage V C2 The waveforms showing temporal changes (transient responses) and the characteristics of the double resonance are almost identical to those of the circuits shown in Figures 2, 7, and 8.

[0143] In the power transmission system according to the third modified example, the adjustment element Q3, which is used as a variable resistor or variable capacitance element, is assumed to be always in the ON state. First, at time t0, similar to that shown in Figure 4A, the power receiving control element Q4 and the load transfer control element Q5 are turned OFF, and the excitation element Q1 is turned ON. When the excitation element Q1 is turned ON and the power supply voltage E0 from the DC power supply 5 is stepped in, at time t=t0=0, the power transmitting capacitor C1 is not charged and the primary side charge / discharge voltage V C1 Since = 0, from the requirement of equation (9), the transmitting coil voltage V L1 ΔV L10 An impulse rise of =E0 occurs. Primary charge / discharge voltage V of the transmitting capacitor C1. C1 It is charged to a constant voltage while ringing.

[0144] When t0 < t < t1, the transmitting coil L1 discharges the charging current to the transmitting capacitor C1 as a return current. Therefore, the transmitting coil voltage V L1 The primary side charge / discharge voltage V satisfies equations (9) and (13). C1 As V increases, the transmitting coil voltage V increases, as shown by the solid line in Figure 3. L1 The value decreases along the cosine function, and the primary side charge / discharge voltage V C1 As shown by the dashed line in Figure 3, it increases according to the cosine function. The primary side charge / discharge voltage V is immediately before time t=t1=π / ω0. C1 When it approaches ≈ 2E0, the transmitting coil voltage V can be determined from the requirement of equation (9). L1 It becomes approximately -E0.

[0145] At time t=t1=π / ω0, similar to that shown in Figure 4C, if the excitation element Q1, power receiving control element Q4, and load transfer control element Q5 are turned off while the adjustment element Q3 remains ON, and the freewheeling control element Q2 is turned ON, then, according to the requirements of equation (16), the primary side charge / discharge voltage V C1 As shown in equation (19), the equivalent expansion of the power supply voltage occurs, and the transmitting coil voltage V L1 After an impulsive instantaneous change, the waveform changes with an equivalent expansion of the power supply voltage as shown in equation (18). Since the adjustment element Q3 is always on, the electromagnetic energy stored in the transmitting capacitor C1 is stored in the transmitting coil L1, and furthermore, a double resonance occurs between the second power supply resonant circuit 2b and the secondary circuit 3b. When the electromagnetic energy stored in the transmitting capacitor C1 begins to move to the transmitting coil L1, the primary charge / discharge voltage V C1 It decreases according to the cosine function shown in equation (19).

[0146] After time t1, after a certain period of time, the excitation element Q1 and the load transfer control element Q5 are kept in the off state, and the power receiving control element Q4 is turned on. Since the adjustment element Q3 is always on, the electromagnetic energy stored in the transmitting capacitor C1 is stored in the transmitting coil L1, and furthermore, a double resonance occurs between the supply side resonant circuit (34b, 2b) and the receiving side resonant circuit (27c, 3c). When the electromagnetic energy stored in the transmitting capacitor C1 begins to move to the transmitting coil L1, the primary side charge / discharge voltage V C1 The voltage decreases according to the cosine function shown in equation (19), and becomes 0V at 3π / (2ω0). As shown in Figures 5A and 5B, the electromagnetic energy transmitted to the receiving coil L2 by the double resonance between the supply side resonant circuit (34b, 2b) and the receiving side resonant circuit (27c, 3c) charges the receiving side capacitor C2 because the receiving control element Q4 is in the ON state.

[0147] In the power transmission system according to the second modification of the first embodiment, at time t1, while the receiving capacitor C2 was being charged, a very small current was flowing to the load 6 side. However, in the power transmission system according to the third modification of the first embodiment, since the load transfer control element Q5 is in the off state, no current flows to the load 6 side, and charge is more effectively stored in the receiving capacitor C2. When charging of the receiving capacitor C2 begins, the secondary charge / discharge voltage V of the receiving capacitor C C2 The secondary charge / discharge voltage V starts increasing from a negative maximum and becomes a positive value. C2 When the value becomes positive, the load transfer control element Q5 is turned on, and the charging current I CS The load begins to flow to load 6.

[0148] At the same time that the load transfer control element Q5 is turned on, the power receiving control element Q4 is turned off. The off state of the power receiving control element Q4 is due to the double resonance between the power supply side resonant circuit (34b, 2b) and the power receiving side resonant circuit (27c, 3c), which controls the secondary side charge / discharge voltage V C2 When it reaches its maximum, the primary side charge / discharge voltage V C1 This is the point at which the voltage becomes 0V. Similar to the power transmission system according to the second modification of the first embodiment, the power receiving control element Q4 is in the off state, so the charging current I is more effectively charged. CS The load begins to move to load 6. Charging current I CS The value increases even after the power receiving control element Q4 is turned off, reaches a peak value, then decreases to zero.

[0149] As described above, the power transmission system according to the third modified example of the first embodiment solves the problems of conventional contactless power transmission systems, and even when the transmission distance d is long, the vibration amplitude can be amplified to an integer multiple of the power supply voltage E0, thus enabling efficient contactless power transmission (wireless power transmission).

[0150] (Other embodiments) As described above, the present invention has been described by first embodiments and modifications thereof, but the descriptions and drawings that constitute part of this disclosure should not be understood as limiting the present invention. Various alternative embodiments, examples and operational techniques will become apparent to those skilled in the art from this disclosure. For example, in the description of the first embodiment, a case in which a large vibration amplitude is obtained by applying a vibration amplification circuit to a power supply side resonant circuit (34a, 2a) and contactless power transmission (wireless power transmission) is performed was described as an example, but the vibration amplification circuit according to the first embodiment is not limited to contactless power transmission systems.

[0151] The remarkable effect of the vibration amplification circuit, as described in the first embodiment, which can amplify the vibration amplitude to an integer multiple of the power supply voltage E0 supplied by the DC power supply 5, as shown in equations (32a) and (32b), and the technical concept therefor, can be applied to various technical fields where it is desirable to achieve a vibration amplitude greater than or equal to the power supply voltage E0 supplied by the DC power supply 5. As described above, the present invention includes various embodiments, modifications, and operational techniques not described herein and in the drawings, and the technical scope of the present invention is defined solely by the inventive features relating to the claims that are reasonable from the above description. [Explanation of symbols]

[0152] 2a, 2b... Second power supply side resonant circuit, 3a, 3b... Secondary side circuit, 5... DC power supply, 6... Load, 11... Power supply panel, 12... Power receiving panel, 27a, 27b, 27c... Power receiving circuit, 28... Detector, 29a... Power supply device, 31a, 31b... Vehicle, 32... Interval control mechanism, 33... Primary side operation unit, 340a... Primary side switching element drive circuit, 270a, 270b... Secondary side switching element drive circuit, 342a... Transmission data storage device, 342b... Program storage device, 34a, 34b... Drive control circuit

Claims

1. A vibration amplification circuit comprising two partially superimposed resonant circuits having a common resonant frequency, wherein the two resonant circuits are time-divided into half-period operations of the resonant frequency, and a process is switched every half-period to boost the maximum amplitude of the vibration waveform of each of the two resonant circuits using transient voltage amplification and impulsive voltage transitions, DC power supply and An excitation element having one terminal connected to the high-potential side terminal of the DC power supply, One terminal of the excitation element is connected to the other terminal of the excitation element, and a freewheel control element operates in a switching manner complementary to the excitation element. A power transmission capacitor with one terminal connected to the connection node between the other terminal of the excitation element and one terminal of the freewheel control element, One terminal is connected to the other terminal of the power transmission capacitor, and the other terminal is connected to the power transmission coil which is connected to the connection node between the other terminal of the freewheel control element and the low-potential side terminal of the DC power supply. The two resonant circuits are comprised of a first power supply side resonant circuit formed by a closed loop of the DC power supply, the excitation element, the power transmission side capacitor, and the power transmission side coil when the excitation element is conducting, and a second power supply side resonant circuit formed by a closed loop of the freewheel control element, the power transmission side capacitor, and the power transmission side coil when the freewheel control element is conducting. The vibration amplification circuit is characterized by repeating the voltage boosting process until the voltage of the power-transmitting coil reaches a desired value that is an integer multiple of the power supply voltage generated by the DC power supply.

2. A power transmission system comprising: an amplification mode in which two partially superimposed resonant circuits have a common resonant frequency, the two resonant circuits are time-divided into half-period operations of the resonant frequency, and a process is switched every half-period to increase the maximum amplitude of the vibration waveform of each of the two resonant circuits using transient voltage amplification and impulsive voltage transitions; and a transmission mode in which, when a desired value is reached by the amplification mode, electromagnetic energy of the voltage of the desired value is transmitted, DC power supply and An excitation element having one terminal connected to the high-potential side terminal of the DC power supply, A recirculation control element is provided, which connects one terminal to the other terminal of the excitation element and performs switching operations complementary to the excitation element. A power transmission capacitor with one terminal connected to the connection node between the other terminal of the excitation element and one terminal of the freewheel control element, A power-transmitting coil is connected to one terminal of the other terminal of the power-transmitting capacitor, and the other terminal is connected to the connection node between the other terminal of the freewheeling control element and the low-potential side terminal of the DC power supply. A receiving coil is positioned at a distance from the transmitting coil and faces it, and receives magnetic energy from the transmitting coil without contact; A receiving-side capacitor connected in parallel to the receiving-side coil and storing the magnetic energy accumulated in the receiving-side coil as electrostatic energy, A load that constitutes a circuit further connected in parallel to the power receiving coil and the parallel circuit of the power receiving coil, The two resonant circuits are comprised of a first power supply side resonant circuit, which is formed by a closed loop of the DC power supply, the excitation element, the power transmission side capacitor, and the power transmission side coil when the excitation element is in a conductive state, and a second power supply side resonant circuit, which is formed by a closed loop of the freewheel control element, the power transmission side capacitor, and the power transmission side coil when the freewheel control element is in a conductive state. In the amplification mode, the voltage of the transmitting coil is increased to a desired value that is an integer multiple of the power supply voltage of the DC power supply. A power transmission system characterized in that, when the desired value is reached, the excitation element is kept in the off state and the freewheel control element is kept in the on state in the transmission mode.

3. The power transmission system according to claim 2, characterized in that the resistance component of the RLC series resonant circuit formed by the first power supply side resonant circuit includes the equivalent internal resistance of the DC power supply, the parasitic resistance of the power transmission side capacitor, the parasitic resistance of the power transmission side coil, and the on-resistance of the excitation element.

4. The power transmission system according to claim 3, characterized in that the resistance component of the RLC series resonant circuit formed by the second power supply side resonant circuit includes the parasitic resistance of the power transmission side capacitor, the parasitic resistance of the power transmission side coil, and the on-resistance of the freewheel control element.

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