Drive control device, linear motor system, and drive control method

Abstract

This drive control device comprises a mover position estimation means (6) and a control calculation means (7). The mover position estimation means (6) has: a speed electromotive force calculation unit (61) for calculating a speed electromotive force (Es) from a linear synchronous motor current (Is) and a voltage command (V* s) to be applied; an interference voltage calculation unit (69) for calculating an interference voltage (E-hatDisB, E-hatDisA) generated when the distance between movers (1) becomes close; and a position calculator (63) for calculating an estimated position (θ-hat) of each of a plurality of the movers 1 on a track on the basis of waveform data obtained by subtracting an interference voltage (E-hatDisB, E-hatDisA) from the speed electromotive force (Es). The control calculation means (7) controls the operation of a coil current control means (5) on the basis of the estimation result output from the mover position estimation means (6).

Description

Drive control device, linear motor system, and drive control method

[0001] The present application relates to a drive control device, a linear motor system, and a drive control method.

[0002] Linear synchronous motors (LSMs) have long been used in the railway industry, factory automation (FA), etc. Recently, the FA industry has been focusing on LSMs with the primary stator winding (coil) located on the ground and the secondary magnet located on the mover.

[0003] To control the thrust of an LSM, it is necessary to know the magnetic pole position of the mover, just as in the case of a rotating machine. However, with an LSM, the cost of the position sensor increases as the movable distance of the device increases. To address this issue, a position sensorless control method has been disclosed for an LSM, which detects the position using the induced voltage, also known as speed electromotive force, generated in a coil, just like in a rotating machine (see, for example, Patent Document 1).

[0004] JP 2002-223587 A (paragraph 0022, FIG. 4, paragraphs 0033-0034, FIG. 7)

[0005] However, in recent LSMs, it has become common to move multiple movers independently while keeping them close to each other, and the distance between the movers can become narrower than the width of the coil. When the distance between movers becomes extremely narrow, magnetic interference occurs between the adjacent movers, but the effects of magnetic interference have not been considered until now. However, the inventors of this application have discovered a phenomenon in which this magnetic interference causes errors in position or speed estimation without a position sensor.

[0006] The present application discloses technology for solving the above-mentioned problems, and aims to provide a linear motor system that takes into account magnetic interference that occurs between adjacent movers, accurately estimates position or speed without a position sensor, and efficiently drives an LSM.

[0007] The drive control device disclosed in the present application is characterized in that it comprises a power supply unit for individually controlling the current of each of the multiple coils of a linear synchronous motor composed of multiple movers and a stator on which multiple coils are arranged to form paths for the multiple movers, an induced voltage calculation unit for calculating an induced voltage generated in each of the multiple coils from the motor constants of the linear synchronous motor and information on the current and voltage for each coil obtained from the power supply unit, an interference voltage calculation unit for calculating an interference voltage generated when two of the multiple movers are close to each other, and a position calculator for calculating the magnetic pole position of each of the multiple movers on the path based on waveform data obtained by subtracting the interference voltage from the induced voltage, mover position estimation means for estimating at least one of the magnetic pole position and the traveling speed of the movers, and control calculation means for controlling the operation of the power supply unit based on the estimation result output from the mover position estimation means.

[0008] The drive control method disclosed in the present application is characterized by including, for a linear synchronous motor composed of a plurality of movers and a stator on which a plurality of coils are arranged to form paths for the plurality of movers, an induced voltage calculation step of calculating an induced voltage generated in each of the plurality of coils from information on the current and voltage for each coil and a motor constant of the linear synchronous motor, an interference voltage calculation step of calculating an interference voltage generated when two of the plurality of movers are close to each other, a position calculation step of calculating a magnetic pole position of each of the plurality of movers in the path based on waveform data obtained by subtracting the interference voltage from the induced voltage, and a control step of individually controlling the current of each of the plurality of coils based on the calculated magnetic pole position.

[0009] According to the drive control device or drive control method disclosed in the present application, accurate position estimation can be performed by taking into account magnetic interference between adjacent movers, so it is possible to obtain a linear motor system that can drive an LSM efficiently by accurately estimating the position or speed without a position sensor.

[0010] FIG. 4B is a block diagram illustrating the configuration of a drive control device and a linear motor system using the drive control device according to a first embodiment. FIG. 4C is a block diagram illustrating the configuration of a coil current control means in the drive control device according to the first embodiment. FIG. 4D is a waveform diagram illustrating a speed electromotive force generated in each coil when one mover travels in the linear motor system according to the first embodiment. FIG. 4A and FIG. 4B are waveform diagrams illustrating the speed electromotive forces generated in the coils by each of the two movers and the combined electromotive force of the two movers when the two movers travel close to each other in the linear motor system according to the first embodiment. FIG. 5A and FIG. 5B are waveform diagrams illustrating the α-axis speed electromotive force and the β-axis speed electromotive force, respectively, when the speed electromotive forces generated in the coils when one mover travels and the speed electromotive forces generated in the coils when two movers travel are subjected to stationary two-phase conversion in the linear motor system according to the first embodiment. FIG. 5D is a waveform diagram illustrating the result of arctangent calculation of the speed electromotive forces after stationary two-phase conversion in the linear motor system according to the first embodiment. FIG. 5E is a waveform diagram illustrating a position estimation error caused by moving the movers close to each other in the linear motor system according to the first embodiment. 15A and 15B are diagrams illustrating the configuration of a mover position estimation means in a drive control device according to a first embodiment. FIG. 15B is a block diagram illustrating the configuration of an interference voltage calculation unit of the mover position estimation means in a drive control device according to a first embodiment. FIG. 15C is a block diagram illustrating an example of a hardware configuration of a portion that executes calculation processing in a drive control device according to a first embodiment. FIG. 15D is a block diagram illustrating the configuration of a mover position estimation means in a drive control device according to a second embodiment. FIG. 15E is a block diagram illustrating the configuration of a position calculation unit of the mover position estimation means in a drive control device according to a second embodiment. FIG. 15F is a flowchart illustrating the operation of a drive control device and a drive control method according to a second embodiment. FIG. 15C is a block diagram illustrating the configuration of a speed calculation unit of the mover position estimation means in a drive control device according to a third embodiment.Fig. 10 is a block diagram for explaining the configuration of a speed calculator of a mover position estimation means in a drive control device according to a third embodiment. Fig. 11 is a block diagram for explaining an example in which the speed calculator of the mover position estimation means is configured as a type 3 speed calculator as a modified example in a drive control device according to a third embodiment. Fig. 12 is a block diagram for explaining the configuration of a mover position estimation means in a drive control device according to a fourth embodiment.

[0011] 1 to 10 are diagrams for explaining the configuration and operation of a drive control device and a linear motor system using the drive control device, as well as a drive control method, according to a first embodiment. Fig. 1 is a block diagram for explaining the configuration of a linear motor system including a drive control device and a linear motor that is the object of control by the drive control device, and Fig. 2 is a block diagram for explaining the configuration of the coil current control means portion of the drive control device.

[0012] Fig. 3 is a waveform diagram showing the speed electromotive forces generated in each of three adjacent coils of a stator when a mover of a linear motor is driven at a constant speed, with the speed electromotive forces on the vertical axis and the mover position on the horizontal axis. Fig. 4A is a waveform diagram showing the speed electromotive forces generated in the coils by two movers when the two movers are traveling close to each other, with the speed electromotive forces on the vertical axis and the mover position on the horizontal axis. Fig. 4B is a waveform diagram corresponding to Fig. 4A, showing the composite electromotive force generated in the coils by the two movers when the two movers are traveling close to each other.

[0013] Fig. 5A is a waveform diagram showing the α-axis speed electromotive force when a stationary two-phase conversion is performed on the speed electromotive force generated in the coil when one mover is traveling and the speed electromotive force generated in the coil when two movers are traveling, with the vertical axis representing the speed electromotive force and the horizontal axis representing the mover position, Fig. 5B is a waveform diagram corresponding to Fig. 5A for the β-axis speed electromotive force, Fig. 6 is a waveform diagram showing the result of a four-quadrant arctangent calculation of the speed electromotive force after stationary two-phase conversion, and Fig. 7 is a waveform diagram for explaining the position estimation error caused by moving movers traveling close to each other.

[0014] FIG. 8 is a block diagram for explaining the configuration of the mover position estimating means, and FIG. 9 is a block diagram for explaining the configuration of the interference voltage calculation section of the mover position estimating means.

[0015] Hereinafter, a drive control device, a drive control method, and a motor system according to embodiments of the present application will be described in detail with reference to the drawings. Note that the same reference numerals in the various drawings indicate the same or corresponding parts.

[0016] As shown in Fig. 1, the motor system according to the first embodiment is a linear motor system comprising a linear motor 100, which is an LSM, and a drive control device 8 that controls the drive of the linear motor 100. The linear motor 100 is characterized in that it estimates the position or speed of a plurality of movers 1A, 1B (which will be referred to as "movers 1" when not distinguished from one another) based on the induced voltage generated in the coil 3 of the stator 2, while taking into consideration the influence of magnetic interference caused by adjacent movers 1. However, before explaining the characteristic features, the basic configuration and operation related to the drive control of the LSM will be explained.

[0017] In the linear motor 100, a stator 2 has a plurality of coils 3a to 3i (referred to as "coils 3" when not distinguished from one another) arranged to form paths for a plurality of movers 1. Due to space limitations, only nine coils 3 are shown in the figure, but in reality, several hundred or more coils 3 may be arranged. These coils 3 are magnetically coupled by a core back 4.

[0018] The movers 1 are linear synchronous motors in which three permanent magnets 11Aa to 11Ac and 11Ba to 11Bc (referred to as "permanent magnets 11" when not distinguishing between them) are arranged in a straight line in each of the movers 1A and 1B, and are magnetically coupled by back yokes 10A and 10B. Due to space limitations, only two movers 1 are drawn, but in an actual product, more movers 1 may be present on the track.

[0019] In the linear motor 100, the mover 1 can be caused to move by controlling the current flowing through each coil 3. In order to control the current flowing through each coil 3, the drive control device 8 is provided with coil current control means 5a to 5i (referred to as "coil current control means 5" when not distinguishing between them) corresponding to each of the multiple coils 3. Since it is necessary to individually control the current of each of the multiple coils 3 so that the multiple movers 1 can move in close proximity to each other, each of the multiple coil current control means 5 individually controls the current of each coil 3 in accordance with the current command determined by the control calculation means 7.

[0020] Each of the multiple coil current control means 5 can be configured, for example, as shown in FIG. 2. In the figure, the coils 3 are represented by inductor symbols. A variable voltage source 52 is connected to the coils 3. The variable voltage source 52 can be any circuit as long as it can output an arbitrary voltage and independently control the current of each coil 3. The variable voltage source 52 may be configured, for example, as a single-phase inverter circuit. The current detection means 51 detects the current flowing through the coils 3. The current control means 53 controls the current flowing through the coils so that it matches the current command and determines the voltage command. A well-known current control method is, for example, PID control (proportional-integral-derivative control). The variable voltage source 52 operates according to the voltage command determined by the current control means 53, causing the desired current to flow through the coils 3.

[0021] As explained in the background art, recent LSMs feature "miniaturization of the mover" and "narrower pitch between the movers." To achieve this, a non-three-phase motor structure that cannot be driven by 120-degree energization is used. This is because removing the premise of "driving by 120-degree energization" increases the degree of freedom in the magnetic structure, making it easier to increase thrust density.

[0022] Although it is merely an example, the linear motor 100 shown in Fig. 1 is an LSM type that is driven by 135° current application. The ratio of the horizontal length of one mover to the horizontal length of one coil is 4:1, and three permanent magnets 11a to 11c are attached to the mover 1. This type of structure is referred to here as a "3-pole, 4-slot structure." If the horizontal length of one magnetic pole is 180 degrees, the electrical angle θ per coil in the 3-pole, 4-slot structure is c is 135 degrees as shown in equation (1).

[0023] For this reason, in order to drive the linear motor 100 shown in FIG. 1, the phases of the currents in the coils 3 must be shifted by 135 degrees.

[0024] By changing the combination of the length of the mover 1 and the arrangement of the permanent magnets 11, various structures can be considered, such as a "5-pole, 6-slot structure" or a "6-pole, 8-slot structure." The phase difference of the current in each coil 3 changes depending on the number of pole slots, but even if the structure changes, the electrical angle θ for one coil c can be calculated using the same formula. In general rotating machines, from the viewpoint of cost performance, θ c = 120 degrees. However, in recent LSMs, added value such as freedom of operation or cable-less movers is becoming more important than cost performance, so non-three-phase motor structures that cannot be driven by 120-degree current conduction are being adopted.

[0025] As shown in Figure 1, the drive control device 8 can individually control the current of each coil 3, so thrust can be generated regardless of the combination of pole slot numbers. However, to control the thrust of an LSM, it is necessary to know the magnetic pole position of the mover, just as in the case of a rotating machine. Therefore, the magnetic pole position of the mover 1 is measured by the mover position estimation means 6. At this time, the moving speed of the mover 1 may also be calculated.

[0026] Of course, any known means may be used to detect the mover position and mover speed, such as an optical encoder, a magnetic encoder, a video camera, etc. However, since this increases costs as described above, the mover position is estimated from the voltage command and current of each coil 3.

[0027] The principle of position estimation in rotating machines is generally a method that utilizes induced voltage, also known as speed electromotive force, that occurs during rotation. As mentioned in the background art, this method has also been applied to LSMs, so we will consider whether it can be applied to the linear motor 100 described in Figure 1.

[0028] For example, when one mover 1 is driven at a constant speed without being placed close to other movers 1, the waveforms of the three coils 3d to 3f adjacent to coil 3e at the center are as shown in Figure 3. In the figure, the three waveforms V3d, V3e, and V3f are the speed electromotive forces generated in coils 3d, 3e, and 3f, respectively, and the point where the center of coil 3e and the center of mover 1 coincide is considered to be zero degrees.

[0029] The waveform V3e shown by the solid line is sinusoidal within the range of ±180 degrees. However, due to non-uniformity in the magnetic properties of the mover end, distortion of the waveform is observed even within the range of ±180 degrees. This phenomenon caused by non-uniformity in the magnetic properties of the mover end is generally known as the "end effect." As shown in Figure 1, an LSM with a small number of magnetic poles has a larger end effect than an LSM with a large number of magnetic poles (e.g., the number of magnetic poles is greater than the number of movers 1).

[0030] Beyond the range of ±180 degrees, waveform V3e, shown by the solid line, gradually attenuates and finally becomes zero. The other two waveforms (waveforms V3d and V3f) have the same shape, but their phases are shifted by 135 degrees. Due to space limitations, the waveforms of the other coils 3 (coils 3a to 3c, 3g to 3i) are not shown here, but the speed electromotive forces of these coils 3 also have waveforms shifted by 135 degrees.

[0031] In rotating machine control, it is common to convert such multi-phase AC waveforms into two-phase AC waveforms using coordinate transformation. However, in the case of non-three-phase LSMs that cannot be driven by 120-degree energization, it is not even clear whether conversion to a two-phase AC waveform is possible. To the inventors' knowledge, sensorless control techniques and coordinate transformation techniques for non-three-phase LSMs that cannot be driven by 120-degree energization are not publicly known, and none have yet become clear to those skilled in the art.

[0032] Therefore, the position estimation was carried out based on the following idea. Here, the speed electromotive force of the nine coils of the coil 3 is expressed as dot θK m9 In the present application, in order to convert this into a two-phase sinusoidal waveform on two orthogonal axes, a coordinate transformation matrix A9 shown in equation (3) is used to perform calculation as shown in equation (2).

[0033] where θ is the motor position, and the dot θ is the motor speed. The accent dot indicates time differentiation. As mentioned above, the electrical angle θ per coil in a 3-pole, 4-slot LSM is c is 135 degrees.

[0034] If the motor structure is not a 3-pole, 4-slot type, the electrical angle θ per coil c The value of is different from 135 degrees, but the electrical angle θ per coil according to the structure c If a similar coordinate transformation equation is formulated, a two-phase sinusoidal waveform on two orthogonal axes can be obtained.

[0035] In the case of a three-phase rotating machine, the three-phase to two-phase coordinate transformation matrix or its inverse transformation matrix is ​​(2 / 3) 0.5 Since it is common to multiply by a coefficient, the above coordinate transformation matrix can be multiplied by some other coefficient in line with this. However, in a non-three-phase LSM that cannot perform 120-degree conduction, unlike a three-phase rotating machine, there is almost no benefit to multiplying by a coefficient, so it is not necessary to multiply by a coefficient. For this reason, no special coefficient is used in equation (2).

[0036] The number of coils 3 used in the calculation may be increased or decreased as appropriate. For example, the speed electromotive force of seven coils, coils 3b to 3h, is calculated as dot θKm7 When this is expressed as a vector, the calculation formula and coordinate transformation matrix for converting this into a two-phase sinusoidal waveform on two orthogonal axes are as shown in Equation (4) and Equation (5).

[0037]

[0038] Similarly, if the speed electromotive force of the five coils, coils 3c to 3g, is expressed as a vector dot θKm5, the calculation formula and coordinate transformation matrix for converting this into a two-phase sinusoidal waveform on two orthogonal axes are as shown in Equation (6) and Equation (7).

[0039]

[0040] When multiple movers 1 are running close to each other, it is better not to consider the induced voltage of too many coils, due to the problem of interference voltage, which will be described later. Therefore, in the following explanation, stationary two-phase conversion will be performed using the speed electromotive force of five coils. Reducing the number of coils used for stationary two-phase conversion is also important in terms of reducing the amount of calculation.

[0041] When two movers 1 are traveling close to each other, the speed electromotive force generated in each coil 3 is the sum of the speed electromotive force VAm generated by mover 1A and the speed electromotive force VBm generated by mover 1B, as shown in Figure 4A. If the distance between the two movers is greater than the coil width, the two waveforms do not overlap, but if the distance between the movers is shorter, the ends of the two waveforms partially overlap. Specifically, when the ends of the two movers 1 are directly above one coil 3 at the same time, the speed electromotive force takes on this waveform.

[0042] Since only the sum of the speed electromotive forces of the respective movers 1 can be obtained from the coil 3 side, what can actually be obtained is a waveform (speed electromotive force VABm) in which the ends of two waveforms overlap, as shown in Fig. 4B. The waveform of the speed electromotive force VABm is sinusoidal in some sections, but is a very complex waveform overall. To the inventors' knowledge, no other technology has been proposed for estimating the positions or speeds of two movers 1 from such speed electromotive force information.

[0043] The waveforms Vα1 (dotted line) and Vα2 (solid line) of the α-axis speed electromotive force obtained by performing a stationary two-phase conversion on the speed electromotive force when one mover 1 is separated from the other movers 1 and when two movers 1 are closely spaced, respectively, while the motor is running at a constant speed, are shown in Figure 5A. The waveforms Vβ1 (dotted line) and Vβ2 (solid line) of the β-axis speed electromotive force obtained by performing a stationary two-phase conversion on the speed electromotive force when one mover 1 is separated from the other movers 1 and when two movers 1 are closely spaced, respectively, while the motor is running at a constant speed, are shown in Figure 5B.

[0044] Waveforms Vα1 and Vβ1 were calculated by converting waveform V3e in Figure 3, and waveforms Vα2 and Vβ2 were calculated by converting waveform VABm in Figure 4B using equations (6) and (7), respectively. This is the calculation result when coil 3e is the center. When one mover is traveling, the waveform Vα1 of the stationary α-axis speed electromotive force and the waveform Vβ1 of the stationary β-axis speed electromotive force are roughly two-phase sinusoidal waveforms obtained within the mover position range of ±180 degrees. In other words, within this range, the two waveforms Vα1 and Vβ1 have almost the same amplitude and a phase difference of about 90 degrees. By utilizing this characteristic, it is possible to easily estimate the position of mover 1.

[0045] On the other hand, with the stationary α-axis speed electromotive force, there is only a slight difference between the waveform Vα2 when two movers 1 are traveling close together and the waveform Vα1 when one mover 1 is traveling, but with the stationary β-axis speed electromotive force, there is a large difference between the waveforms Vβ2 and Vβ1. The range in which the speed electromotive force can be considered to be a two-phase sinusoidal wave when the movers 1 are traveling close together is much narrower than the range in which it can be considered to be a two-phase sinusoidal wave when the movers 1 are traveling independently. Therefore, it is very difficult to estimate the position or speed of two movers 1 from such speed electromotive force information using the technologies that have been considered so far.

[0046] In contrast, the drive control device 8 constituting the linear motor system of the present application is provided with mover position estimation means 6 that suppresses the influence of interference by evaluating (quantifying) it and estimates the mover position using the speed electromotive force after stationary two-phase conversion. Here, the influence of magnetic interference, which is the target of suppression, will be explained using the results (FIG. 6) of a four-quadrant arc tangent calculation performed on the speed electromotive force after stationary two-phase conversion shown in FIG. 5A and FIG. 5B, and the waveform of the position estimation error that occurs when vehicles travel close to each other (FIG. 7).

[0047] The estimated mover position calculated from the speed electromotive force when one mover 1 is traveling independently and away from the other movers 1 is called x1, and the estimated mover position calculated from the speed electromotive force when two movers 1 are traveling close to each other is called x2. As shown in Figure 6, when the four-quadrant arctangent is calculated from the speed electromotive force when one mover 1 is traveling independently, the estimated mover position x1 (dashed line) becomes a sawtooth waveform. Even when traveling independently, errors occur in the position estimation as the position of the mover 1 moves away from the origin, but if the distance between the central coil 3 and the mover 1 is within a range of approximately ±180 degrees, fairly good position estimation is possible.

[0048] On the other hand, the estimated mover position x2 (solid line) calculated from the speed electromotive force when two movers are traveling close to each other is significantly distorted in the negative mover position range and is not a sawtooth waveform. The difference in the waveforms between the estimated mover position x1 and the estimated mover position x2 almost directly results in a position estimation error. Therefore, as shown in Figure 7, it can be seen that the difference Δx (= x1 - x2) between the estimated mover position x1 and the estimated mover position x2, which is the position estimation error, increases when the mover position is in the negative range. In this way, it can be said that it is not a good idea to perform position estimation when information on the speed electromotive forces caused by multiple movers 1 is mixed.

[0049] Although the case where two movers 1 run close to each other has been described here, the same problem occurs when three or more movers 1 run close to each other. This type of problem is likely to become apparent when the end magnetic poles of adjacent movers 1 have the same polarity. In FIG. 1, the polarity of the permanent magnet 11Ac of mover 1A and the polarity of the permanent magnet 11Ba of mover 1B are both north poles, which corresponds to a case where the problem is likely to occur. This type of problem is less likely to occur when the north pole and the south pole are adjacent. However, regardless of the magnetic pole of the adjacent permanent magnets, the same problem essentially occurs, so position sensorless control when movers 1 run close to each other is difficult in either case.

[0050] Therefore, the linear motor system according to the first embodiment is provided with mover position estimating means 6 configured as shown in Fig. 8. For convenience, mover 1A will be referred to as mover A, and mover 1B will be referred to as mover B. The mover position estimating means 6 has a position estimating section 60A for mover A, a position estimating section 60B for mover B, and an interference voltage calculating section 69.

[0051] The position estimator 60A for the mover A and the position estimator 60B for the mover B respectively estimate the voltage command vector V * s and the current vector I s From this, the estimated position θ of the mover A is A and the estimated position of mover B, hat θ B Since the time differential value of the position signal is the velocity, the position estimation units 60A and 60B calculate the estimated velocity dot hat θ of the mover A. A and the estimated velocity of mover B, dot hat θ B is also configured to output.

[0052] The position estimation unit 60A and the position estimation unit 60B respectively estimate the voltage command vector V * s and the current vector I s The speed electromotive force of mover A and the speed electromotive force of mover B are calculated from the above. Then, each speed electromotive force is converted into a two-phase AC coordinate system, and further a four-quadrant arctangent calculation is performed to calculate the estimated position θ of mover A. A and the estimated position of mover B, hat θ BAlso, the estimated position hat θ A and the estimated position of mover B, hat θ B By differentiating each of these, the estimated velocity of the mover A is calculated as follows: A and the estimated velocity of mover B, dot hat θ B We also try to calculate it.

[0053] At this time, when the mover 1 moves in both forward and reverse directions, the calculation result needs to be corrected by 180 degrees depending on the moving direction of the mover 1. Therefore, here, the sign of the speed command (dot θ * A ), sign(dot θ * B ) is input, but if the mover 1 moves in only one direction, it is not necessary to input the sign of the speed command.

[0054] At this time, the interference voltage calculation unit 69 calculates the estimated positions hat θ of the two movers 1 as shown in FIG. A , hat θ B and the estimated speeds of the two movers 1, dot hat θ A , dot hat θ B Based on this, the speed electromotive force of each of the two movers 1 is calculated. When viewed from mover B, the speed electromotive force of mover A is a voltage interfering with the operation of mover B. At the same time, when viewed from mover A, the speed electromotive force of mover B is a voltage interfering with the operation of mover A.

[0055] These voltages are referred to as interference voltages E DisA , Hat E DisB The interference voltage calculation unit 69 calculates the speed electromotive force of the mover A (interference voltage E DisA ) for the mover A, and a speed electromotive force estimating unit 690A for the mover B that estimates the speed electromotive force (interference voltage E DisB ) for the mover B.

[0056] The speed electromotive force estimating units 690A and 690B check the position of each mover 1 against the speed electromotive force data map stored therein, and multiply the checked data by the estimated speed of each mover 1 and output the result.DisA , Hat E DisB Then, the interference voltage hat E DisA , Hat E DisB By using this, the influence of both movers 1 is subtracted from the sum of the speed electromotive forces of the two movers 1, thereby enabling good position estimation with the influence of magnetic interference suppressed.

[0057] The mover position estimation means 6 or the control calculation means 7 constituting the drive control device 8 may be configured as a single piece of hardware 6H including a processor 6H1 and a storage device 6H2, as shown in FIG. 10 . Although not shown, the storage device 6H2 includes a volatile storage device such as a random access memory and a non-volatile auxiliary storage device such as a flash memory. Alternatively, a hard disk auxiliary storage device may be used instead of the flash memory. The processor 6H1 executes a program input from the storage device 6H2. In this case, the program is input from the auxiliary storage device to the processor 6H1 via the volatile storage device. The processor 6H1 may output data such as calculation results to the volatile storage device of the storage device 6H2, or may store the data in the auxiliary storage device via the volatile storage device.

[0058] Embodiment 2. In the first embodiment, the description was given focusing on the configuration of the position estimation means for removing magnetic interference. In the second embodiment, the configuration of the position estimation means will be described in more detail. Fig. 11 is a block diagram for explaining the configuration of the mover position estimation means in the drive control device according to the second embodiment, Fig. 12 is a block diagram for explaining the configuration of the position calculator of the mover position estimation means, and Fig. 13 is a flowchart for explaining the operation of the drive control device and the drive control method.

[0059] The basic configuration of the linear motor system and the basic configuration for calculating the mover position are the same as those in the first embodiment, and therefore, a description of the similar parts will be omitted and Figures 1 to 10 used in the first embodiment will be used.

[0060] As described with reference to FIG. 8 in the first embodiment, the mover position estimation means 6 includes a position estimation unit 60A for mover A, a position estimation unit 60B for mover B, and an interference voltage calculation unit 69. As shown in FIG. 11 , the position estimation unit 60A includes a nearby coil selector 65A that selects, from among the multiple coils 3, the coil 3 near which the mover A is located as a nearby coil. Specifically, the coils 3 located directly below the mover 1 and within a range of less than the electrical angle of one coil 3 (135 degrees in this example) extending outward from both ends in the moving direction are selected. In FIG. 1 , for the mover 1A, the coils 3a to 3e located directly below and the unillustrated coils located to the left of coil 3a within a range of less than 135 degrees extending outward from both ends are selected. Similarly, for the mover 1B, the coils 3f to 3i located directly below, coil 3e within a range of less than 135 degrees extending outward from both ends, and the unillustrated coils located to the right of coil 3i are selected. The selected coils 3 are referred to as "neighboring coils" here. The neighboring coil selector 65A selects the voltage command vector V * s and the current vector I s From this, the voltage command V of the coil in the vicinity of the mover A is * snA and current I snA Only output.

[0061] Then, the voltage command V of the nearby coil output from the nearby coil selector 65A is * snA and current I snA From the velocity electromotive force E of the mover A snA The speed electromotive force calculation unit 61A calculates the speed electromotive force of the mover A, snA From the interference voltage hat E DisB Furthermore, the speed electromotive force (=E snA -E DisB ) is converted into a multiphase to two-phase form to obtain the velocity electromotive force E αβA The multi-phase to two-phase converter 62A is provided.

[0062] And the velocity electromotive force E in the stationary Cartesian coordinate system αβAFrom the estimated position θ of the mover A A The position calculator 63A may acquire information for determining the driving direction of the mover A as additional information. Here, the sign (dot θ * A ) is input to the position calculator 63A.

[0063] Furthermore, the position estimation unit 60A estimates the position θ A Estimated velocity from dot hat θ A The speed calculator 64A may be provided to calculate the estimated position hat θ A The estimated velocity of the mover A is calculated by performing a differential operation or noise elimination operation on A Calculate the following.

[0064] The reason for limiting the data output to the speed electromotive force calculation unit 61A is mainly to deal with the case where a plurality of movers 1 are present on the track and also because it helps to reduce the amount of calculation, but it is not always necessary to limit the data, and it is also possible to omit the nearby coil selector 65A. In that case, the speed electromotive force calculation unit 61A receives the voltage command vector V from each of the plurality of coil current control means 5. * s and the current vector I s will be output as is.

[0065] The position estimating section 60B for the mover B has the same configuration as the position estimating section 60A and operates in the same manner, and estimates the estimated position θ of the mover B. B and the estimated velocity dot hat θ B Calculate the following.

[0066] Speed ​​electromotive force E in the speed electromotive force calculation units 61A and 61B s The method for calculating the induced voltage will now be described. The voltage equation for the linear motor 100 is given by equation (8).

[0067] However, V s is the stator voltage vector, I s is the stator current vector, E sA and E sBare the velocity electromotive force vectors of mover A and mover B, R a is the armature resistance which is a motor constant, L is the inductance matrix, and p is the differential operator.

[0068] Stator voltage vector V s Instead of the voltage command vector V * s When using the sum of the speed electromotive force vectors E sA +E sB is calculated as in equation (9).

[0069]

[0070] The velocity electromotive force vectors E of the movers A and B are sA and E sB is the magnetic flux vector φ for each mover mA , φ mB are defined as the time derivatives of the mover magnetic flux vector φ m is a function of the mover position θ, and the mover position θ is a function of time t, so the speed electromotive force vector E sA and E sB can also be written as equations (10) and (11), respectively.

[0071] However, K mA (θ A ) and K mB (θ B ) is the velocity electromotive force coefficient vector shown in equation (11a), and the positions θ of the mover A and the mover B A , θ B is a function of

[0072] As mentioned above, when the linear motor 100 is driven (the mover 1 is traveling on the path defined by the stator 2), the speed electromotive force of each coil 3 has a waveform as shown in Figures 3, 4A, and 4B. If the nominal data of this waveform is known and the position and speed of the other mover 1 are also known, the speed electromotive force of mover A can be calculated from equation (12A).

[0073]

[0074] Speed ​​electromotive force E of mover A sAis the position θ of the mover A A Since the position θA of the mover A can be estimated from this information, the coordinate transformation formulas shown in the above formulas (2) to (7) can be used to estimate the position θA of the mover A. A This makes it easier to estimate.

[0075] The speed electromotive force of the mover B can also be estimated by equation (12B). B can also be estimated.

[0076] As described above, the position estimation units 60A and 60B have the same configuration and operate in the same way. Therefore, the position calculators 63A and 63B will be described as the position calculator 63 without distinguishing between those for the mover A and the mover B. The position calculator 63 calculates the velocity electromotive force E in the stationary orthogonal coordinates as shown in FIG. 12, for example. αβ and the sign of the speed command (dot θ * The apparatus also includes an edge detection unit 632 that detects edges from the results of the arctangent calculation, an offset amount calculation unit 633 that calculates an offset amount, and an adder 634 that adds the offset amount calculated by the offset amount calculation unit 633 to the mover position estimated by the arctangent calculation unit 631.

[0077] The arctangent calculator 631 performs a four-quadrant arctangent calculation on the speed electromotive force that has been coordinate-converted into two-phase AC, and estimates the mover position. When the mover 1 moves in both forward and reverse directions, the calculation result must be corrected by 180 degrees depending on the moving direction of the mover 1. For this reason, here, the sign (dot θ) of the speed command is used. * ) is input to the arctangent calculator 631. However, when the mover 1 moves only in one direction as described above, the sign (dot θ * ) is not required.

[0078] In a linear motor system in which multiple movers 1 travel on a track, several hundred coils 3 may be placed on the ground, meaning that the movable range of the mover 1 may be tens of thousands of degrees or more in electrical angle terms. However, the output range of the arctangent calculator 631 is only ±180 degrees. If current were to flow through all coils 3, the current conduction phase could be determined solely from the output value of the arctangent calculator 631, but flowing current through coils 3 located far away from the mover 1 is energy inefficient. Therefore, current is usually only passed through coils 3 in the vicinity of the mover 1. To do this, a mechanism is needed to grasp the approximate position of the mover 1.

[0079] The simplest mechanism for determining the rough position of the mover 1 is to detect discontinuous changes (edges) in the result of the arctangent calculation and count them. The edge detection unit 632 adds one to the count value when the result of the arctangent calculation suddenly changes from +180 degrees to -180 degrees. Conversely, it subtracts one from the count value when there is a sudden change from -180 degrees to +180 degrees. The offset amount calculation unit 633 multiplies the value counted by the edge detection unit 632 by 360 degrees and outputs the result as the rough position (offset amount) of the mover 1. The adder 634 adds this to the result of the arctangent calculation, and the result of this addition is used as the output (estimated position θ) of the position calculator 63.

[0080] There are various other methods for determining the offset amount. For example, s One possible method is to determine the offset by squaring each element and examining the magnitude relationship. Another possible method is to determine the offset by acquiring a rough position using an inexpensive sensor such as a video camera, and then estimate the precise position using an arctangent calculation.

[0081] The operation of the drive control device 8 according to the first and second embodiments, i.e., the flow of the drive control method, can be summarized as shown in the flowchart of Fig. 13. Here, the operation of the position estimation unit 60A will be mainly described, but in the parts where it is not necessary to distinguish between movers A and B, the symbols "A" and "B" used to distinguish between the movers will be omitted.

[0082] First, the nearby coil selector 65 selects the nearby coil (step S100), and the voltage command vector V * s and the current vector I s Among these, the voltage command V of the nearby coil * sn and current I sn Only the speed electromotive force is output to the speed electromotive force calculation unit 61.

[0083] The speed electromotive force calculation unit 61 calculates the voltage command V * sn and current I sn From the speed electromotive force E sn At this time, the interference voltage calculation unit 69 receives the estimated position hat θ output from the position estimation unit 60A and the position estimation unit 60B in the subsequent steps described later. A , estimated position θ B , estimated velocity dot hat θ A , estimated velocity dot hat θ B Then, the speed electromotive force estimating units 690A and 690B check the input position of each mover 1 against the data map stored inside each unit, multiply the checked data by the estimated speed of each mover 1, and calculate the interference voltage hat E DisA , Hat E DisB are calculated (step S120).

[0084] The subtractor 66 receives the speed electromotive force E of the mover A input from the speed electromotive force calculation unit 61. snA , the interference voltage E due to the mover B input from the interference voltage calculation unit 69 DisB Therefore, the speed electromotive force E snA From the interference voltage hat E DisB The multi-phase to two-phase converter 62 converts the speed electromotive force output from the subtractor 66 into a speed electromotive force E αβ is calculated (step S130).

[0085] The position calculator 63 calculates the speed electromotive force E αβ The estimated position hat θ is calculated by a four-quadrant arctangent calculation (step S140), and output to the interference voltage calculation unit 69 and the speed calculation unit 64. The speed calculation unit 64 calculates an estimated speed dot hat θ from the estimated position hat θ (step S150), and outputs it.

[0086] In step S140, the edge detection unit 632 detects an edge from the result of the arctangent calculation, and outputs a count value increased or decreased depending on the type of edge (positive or negative direction) to the offset amount calculation unit 633. The offset amount calculation unit 633 also outputs a value obtained by multiplying the count value by 360 degrees to the adder 634 as the offset amount, and the adder 634 outputs the mover position obtained by adding the offset amount to the result of the arctangent calculation as the estimated position hat θ.

[0087] As described above, by constructing a position estimation system such as that of embodiment 2 of the present application, it is possible to make the linear motor 100, which can operate with multiple movers 1 in close proximity, position sensorless, thereby reducing the cost of the system.

[0088] Third Embodiment In this third embodiment, a method for reducing the influence of the "end effect" caused by magnetic non-uniformity near the end of the mover will be described. Fig. 14 is a block diagram for explaining the configuration of a speed calculator of the position estimation means of the drive control device according to the third embodiment, and Figs. 15A and 15B are diagrams showing waveforms with time on the horizontal axis and mover position on the vertical axis of the estimated position output when a position estimate calculated with an error due to distortion of the induced voltage is filtered through a first-order low-pass filter as a comparative example when the mover is traveling at a constant speed, and waveforms with the same axis configuration of the estimated position output when filtered through a filter such that the steady-state error for the ramp response of the present application becomes zero.

[0089] Note that the configuration other than the elimination of the influence of the end effect is the same as in the first and second embodiments, and therefore a description of the similar parts will be omitted, and Figures 1 to 13 used in the first and second embodiments will be used. Furthermore, the influence of the end effect is a matter common to all movers, and therefore the reference symbols (A, B) that distinguish between movers will be omitted in the drawings and explanation.

[0090] By performing the processing described in the first and second embodiments, the estimated position θ of each mover 1 can be obtained. However, as shown in FIG. 3 , the speed electromotive force attenuates as the distance between the mover 1 and the coil 3 increases. Due to such magnetic nonuniformity at the ends (end effect), the speed electromotive force converted to stationary two-phase coordinates is somewhat distorted even in the case where one mover 1 is driven independently. This distortion of the speed electromotive force causes a pulsating error in the position estimation. If such a waveform is differentiated to obtain the mover speed, naturally, the pulsation will be superimposed on the estimated speed. Position estimation errors or speed estimation errors degrade the performance of position sensorless control, so it is necessary to reduce the position estimation errors and speed estimation errors as much as possible.

[0091] In the third embodiment, a configuration in which the velocity calculator has a pulsation removal function will be described as a method for accurately reducing the influence of the end effect from the position estimation result and the velocity estimation result with a simple configuration. In the third embodiment, the velocity calculator 64 of the position estimation unit 60 is configured as shown in FIG. 14 . The rest is the same as in the first and second embodiments. However, in the third embodiment, the estimated position θ output from the position calculator 63 is replaced with the first estimated position θ. Then, instead of the first estimated position θ output from the position calculator 63, the second estimated position θ output from the velocity calculator 64 is F is used as an estimate.

[0092] The speed calculator 64 calculates a second estimated position θ by removing the periodic error component from the first estimated position θ. FDuring this calculation, an estimated speed dot hat θ can be calculated, so that the speed calculator 64 also functions as a speed calculator, similar to the first and second embodiments. The speed calculator 64 is composed of an adder 641, a proportional integral calculator 642, and an integrator 643.

[0093] The reason why the velocity calculator 64 has such a configuration will be explained below. As described above, pulsation due to the end effect is superimposed on the first estimated position θ, and some kind of filter is required to remove this. Here, filtering is performed by feedback computing the difference between input and output. In this case, it is important not only to remove the pulsation due to the end effect, but also to reduce the steady-state difference between the input and output of the filter to zero.

[0094] Here, we consider the case where the mover 1 is traveling at a constant speed and the steady-state difference between the input and output of the filter is made zero. Since the mover position at this time changes in a ramp pattern, it is sufficient to adopt a controller structure that makes the steady-state error for the ramp response zero. The necessary condition for making the steady-state error for the ramp response zero is to use a type 2 or higher controller, which is generally known as the "internal model principle" in classical control theory.

[0095] A type 2 controller is a controller that includes two integrators connected in series. The pulsation eliminator 64 is provided with a proportional integral calculator 642 and an integrator 643. Since the proportional integral calculator 642 has an integrator inside, this system has two integrators connected in series. Therefore, the pulsation eliminator 64 is type 2.

[0096] The difference between removing pulsation due to the end effect when using a general low-pass filter as a comparative example and when using the pulsation remover 64 of the present invention will be described below. The mover position (first estimated position θ) input to each filter increases in a ramp shape as shown in Figures 15A and 15B.

[0097] On the other hand, the waveform hat θ when a first-order low-pass filter is applied LPFAs shown in Fig. 15A, the error Δθ between the two waveforms (input and output) in the steady state increases in a ramp shape with a slight delay from the first estimated position θ. DC Then, in the first-order low-pass filter, no matter how much time passes, Δθ DC never becomes zero. Δθ DC increases as the gradient of the first estimated position θ, i.e., the estimated velocity θ, increases. DC Since thrust cannot be generated efficiently in the presence of

[0098] On the other hand, in the filter using the velocity calculator 64 according to the third embodiment, the first estimated position hat θ and the corresponding filter output, the second estimated position hat θ F are guaranteed to eventually match, so after a certain time has passed, Δθ DC Therefore, if the speed calculator 64 of the third embodiment is used, the problem of deterioration in efficiency in the high-speed region, which occurs when a first-order low-pass filter is used, does not occur.

[0099] In addition, in cases where the mover speed changes at a constant rate, if it is desired to make the steady-state difference between the input and output of the filter zero, a type 3 controller can be used. This will be described in a modified example, but it is not particularly difficult to realize since all that is required is to add a double integrator path to the speed calculator 64 of this embodiment.

[0100] Regarding the ramp response, the internal model principle is used to calculate the first estimated position θ and the second estimated position θ, which is the output of the integrator 643. F Since the values ​​of the integrator 643 and the integrator 644 are constantly the same, the input to the integrator 643 can be regarded as velocity information. From this, it can be said that this system can apparently calculate velocity without performing a differential operation.

[0101] The proportional term of the proportional-plus-integral calculator 642 is necessary in order to manipulate the damping coefficient of the system. This will be explained in more detail below. The transfer function of the proportional-plus-integral calculator 642 can be written as in equation (13).

[0102] However, K P is the proportional gain, K I is the integral gain, and s is the Laplace transform operator.

[0103] Then, the speed calculator 64 calculates the second estimated position hat θ from the first estimated position hat θ. F The transfer function up to can be expressed as in equation (14).

[0104]

[0105] Further, the transfer function from the first estimated position hat θ to the estimated velocity dot hat θ can be expressed as in equation (15).

[0106] The denominator polynomials of equations (14) and (15) are the same.

[0107] The following equation (Equation (16)) is well known as a quadratic normal form transfer function.

[0108] where ζ is the damping coefficient of the system, ω n is the natural angular frequency of the system.

[0109] By comparing the coefficients of equations (14) to (16), the damping coefficient ζ and the natural angular frequency ω of the speed calculator 64 are obtained. n It can be seen that is determined by the following formulas (formulas (17) and (18)).

[0110]

[0111] Proportional gain K of the proportional integral calculator 642 P When is zero, the damping coefficient is zero. It is known that if the damping coefficient is zero, the output signal will oscillate continuously. Therefore, it is necessary to design the proportional gain so that the damping coefficient becomes the desired value. On the other hand, the natural angular frequency ω of the system n is the integral gain K I Therefore, the speed calculator 64 determines the integral gain K I By adjusting the frequency, it is possible to change the frequency up to which the influence of the end effect is eliminated.

[0112] A secondary effect of adopting such a configuration is that it becomes more resistant to sudden changes in the voltage command or current. It also becomes more resistant to current detection noise. If the voltage command or current changes suddenly, or if noise is introduced into the current detection, the first estimated position hat θ, which is the output of the position calculator 63, will fluctuate, but it is possible that the actual mover position has not yet changed. For this reason, it is necessary to prevent the estimated mover position from changing too sensitively. The second estimated position hat θ F Since the change in the first estimated position θ is more gradual than that in the first estimated position θ, it is less susceptible to the influence of such disturbance factors.

[0113] As described above, by using the drive control device 8 or drive control method according to the third embodiment, it is possible to eliminate the influence of the end effect from the position estimation result or the speed estimation result with a simple configuration. By using this position estimation system, high-performance control of the LSM can be achieved at low cost.

[0114] In the third embodiment, the case where the speed is constant was discussed, and therefore the speed calculator was a type 2 controller. In this modification, an example will be described in which the speed calculator is configured to handle the case where the acceleration is constant. Fig. 16 is a block diagram showing an example in which the speed calculator is configured as a type 3 controller in a drive control device according to the modification, and Fig. 17 is a block diagram showing the configuration of the proportional integral double integral calculator of the speed calculator.

[0115] A constant acceleration is an example of a pattern in which the speed gradually increases or decreases. In this case, if it is desired to reduce the steady-state difference between the input and output of the filter to zero, a Type 3 controller is required. Therefore, as shown in FIG. 16, the speed calculator 64 according to this modification is provided with a proportional-integral-double-integral calculator 644 instead of the proportional-integral calculator 642 described in FIG. 14.

[0116] The difference between the proportional integral calculator 642 and the proportional integral double integral calculator 644 will now be described. As shown in FIG. 17 , the proportional integral double integral calculator 644 includes a proportional calculator 6441, an integral calculator 6442, a double integral calculator 6443, and an adder 6444. The proportional integral double integral calculator 644 corresponds to the proportional integral calculator 642 with the double integral calculator 6443 added. The double integral calculator 6443 outputs a signal obtained by integrating an input signal and then further integrating it. The output of the double integral calculator 6443 is appropriately multiplied by a coefficient to stabilize the controller. The adder 6444 calculates the sum of the outputs of the proportional integral calculator 6441, the integral calculator 6442, and the double integral calculator 6443, and outputs the calculation result as the output of the proportional integral double integral calculator 6444.

[0117] The mathematical difference will also be explained. The transfer function of the proportional integral / double integral calculator 644 is expressed as in equation (19).

[0118] However, K P is the proportional gain, K I is the integral gain, K II is the double integral gain, and s is the Laplace transform operator.

[0119] Comparing equation (13) and equation (19), equation (19) contains the double integral term K II / s 2 is added. In this modified example, this double integral term and the integrator 643 are connected in series, so the number of integrators connected in series in the speed calculator 64 is three. Therefore, this is a type 3 controller. It is known that, based on the internal model principle, a type 3 controller can make the steady-state error for a constant acceleration input zero. However, to do this, the proportional gain K P , integral gain K I , double integral gain K II It is necessary to design appropriately so that the system is stable.

[0120] An example of a gain design method for the proportional-plus-double-integral calculator 644 is shown below. In the speed calculator 64, the first estimated position hat θ is converted into the second estimated position hat θ. FThe transfer function up to can be expressed as in equation (20).

[0121]

[0122] In this system as well, the input to the integrator 643 can be regarded as an estimated velocity. Here, the transfer function from the first estimated position hat θ to the estimated velocity dot hat θ can be expressed as in equation (21).

[0123] The denominator polynomials of Equation (20) and Equation (21) are the same. Once the transfer functions have been formulated, all that remains is to determine the poles of these transfer functions so that the system is stable.

[0124] There are several possible patterns for pole placement, but for example, triple-root pole placement is a pole placement that is often used in third-order systems, so it is a good idea to use this. The denominator polynomial in the triple-root pole placement case is shown in equation (22).

[0125]

[0126] Here, -ω n are the poles of the system. To ensure the stability of the system, ω n >0. This system has three poles, but all three poles overlap at one point on the real axis. The overlapping point is -ω n This type of pole placement is called "triple root pole placement."

[0127] Once the poles of the system are determined, the numerical values ​​of each gain can be obtained by comparing the coefficients of equation (22). The gains that make the poles of this system triple roots are shown in equation (23).

[0128]

[0129] In the case of the triple-root pole placement described above, the real axis components of all poles are negative, so the system is stable. Other pole placement methods besides the triple-root method may be used as long as the system is stable. For example, when gain design is performed using third-order Butterworth pole placement, the gain can be determined as shown in equation (24).

[0130]

[0131] In the above design, the pole is -ω n , -ω n The poles are located at exp(±jπ / 3). In this case too, the real axis components of all poles are negative, so the system is stable.

[0132] The gain can be designed freely to some extent, but to stabilize this system, at least K P >0, K I Therefore, the proportional term or the single integral term cannot be omitted from the proportional-integral-double integral calculator 644.

[0133] As described above, the speed calculator 64 having a properly designed proportional integral / double integral calculator 644 has the characteristic that no steady-state deviation occurs before or after the filter input / output even when the acceleration is constant. Therefore, by using the mover position estimating means 6 according to the modified example, it is possible to eliminate the influence of the end effect from the position estimation result or the speed estimation result with a simple configuration. Using this position estimation system, high-performance control of the LSM (linear motor 100) can be achieved at low cost.

[0134] It should be noted that by applying the method disclosed in this application, it is possible to deal with cases where the position or speed of the mover 1 changes even more abruptly. For example, if it is desired to reduce the steady-state error to zero when the acceleration changes at a constant rate, a type 4 controller can be adopted, and a proportional integral, double integral, triple integral calculator can be used. Similarly, type 5 or higher controllers can also be considered.

[0135] Fourth Embodiment In the above first to third embodiments, examples of estimating the positions of two adjacent movers have been described, but the drive control device or drive control method of the present application is applicable regardless of the number of adjacent movers. In this fourth embodiment, an example of estimating the positions of three adjacent movers will be described. Figure 18 is a block diagram for explaining the configuration of mover position estimation means in the drive control device according to the fourth embodiment.

[0136] Although the number of position estimation units and the number of interference voltage calculation units provided in the mover position estimation means are different, the configurations of each of the plurality of position estimation units and each of the plurality of interference voltage calculation units are basically the same as those explained in Embodiments 1 and 2. Therefore, explanations of similar parts will be omitted, and Figures 1 to 7 and Figures 12 to 17 used in Embodiments 1 to 3 will be used.

[0137] 18, the mover position estimating means 6 provided in the drive control device 8 according to the fourth embodiment simultaneously estimates the positions and velocities of three movers A, B, and C. The mover position estimating means 6 includes position estimating units 60A, 60B, and 60C corresponding to each mover 1, a first interference voltage calculating unit 69A that calculates the interference voltage between mover A and mover B, and a second interference voltage calculating unit 69B that calculates the interference voltage between mover B and mover C.

[0138] In the position estimation unit 60A for the mover A, the interference voltage E due to the mover B is DisBA Considering the estimated position θ A and the estimated velocity dot hat θ A The position estimation unit 60C for the mover C calculates the interference voltage E due to the mover B. DisBC Considering the estimated position θ C and the estimated velocity dot hat θ C On the other hand, the position estimation unit 60B for the mover B calculates the interference voltage E DisAB and the interference voltage E due to the mover C DisCB Considering the estimated position θ B and the estimated velocity dot hat θ B The following calculation is performed.

[0139] The first interference voltage calculation unit 69A calculates the estimated position θ of the mover A. A and the estimated velocity dot hat θ A , and the estimated position θ of the mover B B and the estimated velocity dot hat θ B Therefore, the interference voltage E from mover A to mover B DisAB and interference voltage E from mover B to mover A DisBA The second interference voltage calculation unit 69B calculates the estimated position θ of the mover B.B and the estimated velocity dot hat θ B , and the estimated position θ of the mover C C and the estimated velocity dot hat θ C Therefore, the interference voltage E from mover B to mover C DisBC and interference voltage E from mover C to mover B DisCB Calculate the following.

[0140] That is, position estimation units 60A, 60B, and 60C corresponding to movers A, B, and C, respectively, perform position estimation and velocity estimation taking into account interference voltage, as described in the first to third embodiments. The methodology is the same as that described in the first to third embodiments.

[0141] It is assumed that the movers A, B, and C are arranged in this order on the path. In other words, because mover B is present between mover A and C, mover A and C are far enough apart that no interference occurs between them. However, if the path of linear motor 100 is configured in a ring shape, the movers may be arranged in the order of mover C, mover A, and mover B depending on the operation pattern. In such cases, an interference voltage will occur between mover A and mover C, so it is advisable to provide an additional block that calculates the interference voltage between mover A and mover C. Furthermore, if there is a branch on the path, the positions of mover B and mover C may be swapped, and the movers may be arranged in the order of mover A, mover C, and mover B. Even in such cases, it is advisable to provide an additional block that calculates the interference voltage between mover A and mover C.

[0142] Although the fourth embodiment has been described with reference to a case in which there are three movers 1, the position sensorless control system can be configured in the same way regardless of the number of movers 1. What is important is that when movers 1 are close to each other, position estimation is performed taking into account voltage interference from other movers 1.

[0143] As described above, by using the mover position estimation means 6 of the present invention, it is possible to accurately estimate the mover position of the linear motor 100 that can drive a plurality of movers 1 in close proximity to each other. By using this position estimation system, it is possible to realize a highly efficient linear motor system at low cost.

[0144] Incidentally, in addition to the method of calculating the arctangent of the speed electromotive force, various other position estimation methods or position sensorless control methods have been devised for rotating machines. Such techniques may be combined with the present invention.

[0145] Although various exemplary embodiments and examples are described in this application, the various features, aspects, and functions described in one or more embodiments are not limited to the application of a particular embodiment, but may be applied to the embodiments alone or in various combinations. Therefore, countless modifications not illustrated are contemplated within the scope of the technology disclosed in this specification. For example, this includes cases where at least one component is modified, added, or omitted, or where at least one component is extracted and combined with components of another embodiment.

[0146] As described above, according to the drive control device 8 of the present application, for a linear synchronous motor (linear motor 100) that is configured with a plurality of movers 1 and a stator 2 on which a plurality of coils 3 are arranged and that form paths for the plurality of movers 1, a power supply device (coil current control means 5) that individually controls the current of each of the plurality of coils 3, a motor constant (armature resistance R s , inductance matrix L, etc.) and information on the current and voltage for each coil 3 obtained from the power supply device (coil current control means 5) (voltage command vector V * s , current vector I s ) generated in each of the coils 3 (speed electromotive force E s ) and an induced voltage calculation unit (speed electromotive force calculation unit 61) that calculates an interference voltage hat E that occurs when two of the multiple movers 1 are close to each other. DisB , Hat E DisA an interference voltage calculation unit 69 that calculates the induced voltage (speed electromotive force E s ) to the interference voltage E DisB , Hat E DisAand a position calculator 63 for calculating the magnetic pole position (estimated position hat θ) of each of the plurality of movers 1 on the track based on the waveform data obtained by subtracting the magnetic pole position (estimated position hat θ, second estimated position hat θ). F and a control calculation means 7 that controls the operation of the power supply device (coil current control means 5) based on the estimation result output from the mover position estimation means 6. As a result, even when the mover 1 comes into close proximity with the LSM, the influence of interference voltage can be suppressed and the position of the mover 1 can be accurately estimated without a position sensor, thereby enabling the linear motor 100 to be driven efficiently.

[0147] The mover position estimation means 6 estimates both the magnetic pole position and the running speed of each of the two movers (the estimated position hat θ or the second estimated position hat θ F and estimated velocity (dot hat θ), and the interference voltage calculation unit 69 calculates the interference voltage (dot hat θ) based on a data map of induced voltages stored in advance and the estimated values ​​calculated for each of the two movers. DisB , Hat E DisA If the above equation is calculated, the interference voltage can be calculated accurately by feedback control.

[0148] The mover position estimation means 6 selects, from the plurality of coils 3, a coil 3 to which any of the plurality of movers 1 is close (for example, a coil 3 directly below the mover 1 or within a range of less than the electrical angle of one coil 3 from both ends in the traveling direction toward the outside), and narrows down the coils 3 to the selected coils 3 to obtain information on the current and voltage for each coil 3 (voltage command V * sn , current I sn ) to the induced voltage calculation unit (speed electromotive force calculation unit 61), the position of the mover 1 can be estimated without a position sensor, even if the track is long, without excessively increasing the amount of calculation.

[0149] The mover position estimation means 6 is configured as a type 2 or more controller using an integrator, and calculates a second magnetic pole position (second estimated position hat θ) by removing an error from the magnetic pole position (first estimated position hat θ) output from the position calculator 63. F If the system is configured to include a speed calculator 64 that calculates the estimated speed (estimated speed dot hat θ) and the travel speed (estimated speed dot hat θ), it is possible to eliminate position estimation errors and speed estimation errors due to the end effect and obtain a high-quality estimated signal.

[0150] Furthermore, the linear motor system of the present application is equipped with the linear synchronous motor (linear motor 100) in which the current of each of the multiple coils 3 is individually controlled by the drive control device 8 and the power supply device (coil current control means 5), so even if the movers 1 move close to each other, the position of the movers 1 can be accurately estimated without a position sensor and the movers can be driven efficiently.

[0151] Furthermore, according to the drive control method of the present application, for a linear synchronous motor (linear motor 100) that is configured with a plurality of movers 1 and a stator 2 in which a plurality of coils 3 are arranged and that form paths for the plurality of movers 1, information on the current and voltage for each coil 3 (voltage command vector V * s , current vector I s ) and the motor constant (armature resistance R s , inductance matrix L, etc.), the induced voltage (speed electromotive force E s an induced voltage calculation step (step S110) for calculating an interference voltage E DisB , Hat E DisA an interference voltage calculation step (step S120) for calculating the induced voltage (speed electromotive force E s ) to the interference voltage E DisB , Hat E DisA Based on the waveform data from which the above-mentioned difference is subtracted, the magnetic pole positions of the plurality of movers 1 on the track (estimated position θ, second estimated position θ) are calculated. F ) and the calculated magnetic pole position (estimated position hat θ, second estimated position hat θ)F ) and a control step of individually controlling the current of each of the plurality of coils 3 based on the detected voltage. As a result, even when the movers 1 are in close proximity to each other in the LSM, the influence of the interference voltage can be suppressed, and the position of the mover 1 can be accurately estimated without a position sensor, thereby enabling the linear motor 100 to be driven efficiently.

[0152] 1: mover, 100: linear motor (linear synchronous motor), 2: stator, 3: coil, 5: coil current control means (power supply device), 6: mover position estimation means (position estimation means), 60: position estimation unit, 61: speed electromotive force calculation unit (induced voltage calculation unit), 62: polyphase to two-phase converter, 63: position calculation unit, 631: arctangent calculation unit, 632: edge detection unit, 633: offset amount calculation unit, 64: speed calculation unit, 69: interference voltage calculation unit, E s : Speed ​​electromotive force (induced voltage), E αβ : Speed ​​electromotive force (two-phase sine wave), I s : current vector, I sn : Current, V * s : voltage command vector, V * sn : Voltage command, hat E DisA , Hat E DisB : interference voltage, hat θ: estimated position, first estimated position, hat θ F : second estimated position, dot hat θ: estimated speed (traveling speed).

Claims

1. A power supply device for a linear synchronous motor including a plurality of movers and a stator having a plurality of coils arranged thereon to form paths for the plurality of movers, the power supply device controlling currents of the respective coils individually; a mover position estimating means for estimating at least one of the magnetic pole position and the traveling speed of the mover, the mover position estimating means comprising: an induced voltage calculating unit for calculating an induced voltage generated in each of the plurality of coils from a motor constant of the linear synchronous motor and information on the current and voltage for each coil obtained from the power supply device; an interference voltage calculating unit for calculating an interference voltage generated when two of the plurality of movers are close to each other; and a position calculator for calculating a magnetic pole position of each of the plurality of movers on the traveling path based on waveform data obtained by subtracting the interference voltage from the induced voltage; a control and calculation means for controlling the operation of the power supply device based on the estimation result output from the mover position estimation means; A drive control device comprising:

2. the mover position estimating means calculates estimated values ​​of both the magnetic pole position and the traveling speed of each of the two movers, 2. The drive control device according to claim 1, wherein the interference voltage calculation unit calculates the interference voltage based on a prestored data map of the induced voltage and the estimated value calculated for each of the two movers.

3. 2. The drive control device according to claim 1, wherein the mover position estimation means has a nearby coil selector that selects a coil to which any of the plurality of movers is close among the plurality of coils, and outputs information on the current and voltage of each of the selected coils to the induced voltage calculation unit.

4. The mover position estimation means 2. The drive control device according to claim 1, further comprising a speed calculator which calculates a second magnetic pole position obtained by removing an error from the magnetic pole position output from the position calculator and the traveling speed, the second magnetic pole position being configured as a type 2 or more controller using an integrator.

5. A drive control device according to any one of claims 1 to 4, and the linear synchronous motor in which the current of each of the plurality of coils is individually controlled by the power supply device; A linear motor system comprising:

6. an induced voltage calculation step of calculating an induced voltage generated in each of a plurality of coils of a linear synchronous motor, the induced voltage being generated in each of the plurality of coils from information on current and voltage for each coil and a motor constant of the linear synchronous motor, the linear synchronous motor being configured with a plurality of movers and a stator on which a plurality of coils are arranged to form paths for the plurality of movers; an interference voltage calculation step of calculating an interference voltage generated when two of the plurality of movers are close to each other; a position calculation step of calculating a magnetic pole position of each of the plurality of movers on the path based on waveform data obtained by subtracting the interference voltage from the induced voltage; and a control step of individually controlling the currents of the coils based on the calculated magnetic pole position; A drive control method comprising:

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