Motor control device and control method

Abstract

This motor control device for driving a motor by an inverter comprises: a transient component calculation unit for calculating a q-axis magnetic flux differential value on the basis of the difference between a q-axis magnetic flux command value and a q-axis magnetic flux detection value or the difference between a q-axis current command value and a q-axis current detection value; a d-axis magnetic flux estimation unit for calculating a d-axis magnetic flux estimation value on the basis of a q-axis voltage value, a q-axis current, an electrical angular velocity, and the q-axis magnetic flux differential value; and a magnet temperature calculation unit for calculating a magnet temperature estimation value on the basis of the d-axis magnetic flux estimation value. Thus, when the magnet temperature or the magnet magnetic flux is estimated, a transient component can be accurately calculated, and even in a transient state where the d-axis current or the q-axis current fluctuates, the estimation accuracy can be improved.

Description

Motor control device and control method

[0001] The present invention relates to a motor control device and a control method.

[0002] Prior art related to motor control is known, for example, from Patent Document 1 and Patent Document 2. Patent Document 1 discloses a drive device for a permanent magnet synchronous motor that estimates the magnetic flux amount or magnet temperature of the magnet of the permanent magnet synchronous motor based on a voltage command value and rotational speed information of the permanent magnet synchronous motor. Patent Document 2 also discloses an estimation method that performs coordinate transformation of the armature current to a current on a γ-δ axis, which is made up of the γ axis and the δ axis perpendicular to the γ axis, based on the rotor position and the γ axis, which is the direction in which the estimated total armature flux linkage is generated, and estimates the temperature or magnetic flux of the permanent magnet based on a control command for the synchronous machine and the current on the γ-δ axis.

[0003] Japanese Patent No. 6978602 Japanese Patent Application Laid-Open No. 2013-198340

[0004] In the conventional technology described in Patent Document 1, a magnet flux estimation calculation unit estimates the magnet flux by subtracting from the q-axis voltage the product of the resistance value and the q-axis current, the product of the q-axis inductance and the q-axis current derivative, and the product of the electrical angular velocity, the d-axis inductance, and the d-axis current, and dividing the result by the electrical angular velocity. Meanwhile, in order to estimate the magnet flux with high accuracy even in a transient state in which the d-axis current or the q-axis current is fluctuating, it is necessary to accurately calculate the product of the q-axis inductance and the q-axis current derivative, which is a component that fluctuates during a transient state (transient component) in the estimation calculation. Because noise present in the current detection value causes large noise to appear in the derivative value, it is difficult to accurately calculate the transient component. However, Patent Document 1 does not disclose anything about calculating the transient component, and errors are expected to occur in the magnet flux estimation value in a transient state, so there is room for improvement.

[0005] In the prior art described in Patent Document 2, the d-axis magnetic flux, which is the sum of the magnetic flux and the product of the d-axis inductance and the d-axis current, is estimated by subtracting the product of the resistance and the q-axis current and the product of the q-axis inductance and the q-axis current derivative from the q-axis voltage, and then dividing the result by the electrical angular velocity. Furthermore, the γ-axis, which is the direction in which the total armature flux linkage is generated, is estimated based on the calculated d-axis magnetic flux, and the armature current is coordinate-transformed to a current on the γ-δ axis, which is made up of the γ-axis and the δ-axis, which is perpendicular to the γ-axis. The magnet temperature or magnetic flux is calculated based on the control command for the synchronous machine and the current on the γ-δ axis. However, Patent Document 2 also does not disclose any calculation of the q-axis inductance and the q-axis current derivative, which are transient components. In other words, Patent Document 2 predicts that errors will occur in the estimated value of the d-axis magnetic flux during a transient state, and therefore there is room for improvement.

[0006] The present invention has been made in view of the above, and has an object to provide a motor control device and control method that can improve estimation accuracy even in a transient state in which the d-axis current or q-axis current is fluctuating by accurately calculating transient components when estimating magnet temperature or magnet magnetic flux.

[0007] The present application includes multiple means for solving the above-described problems. One example is a motor control device that drives a motor using an inverter, and includes: a transient component calculation unit that calculates a q-axis magnetic flux derivative based on the difference between a q-axis magnetic flux command value and a q-axis magnetic flux detection value, or the difference between a q-axis current command value and a q-axis current detection value; a d-axis magnetic flux estimation unit that calculates a d-axis magnetic flux estimate based on the q-axis voltage value, the q-axis current, the electrical angular velocity, and the q-axis magnetic flux derivative; and a magnet temperature calculation unit that calculates a magnet temperature estimate based on the d-axis magnetic flux estimate value.

[0008] According to the present invention, by accurately calculating the transient component when estimating the magnet temperature or magnet magnetic flux, it is possible to improve estimation accuracy even in a transient state in which the d-axis current or q-axis current is fluctuating.

[0009] 1 is a diagram showing an outline of the configuration of a motor control device; FIG. 1 is a diagram showing a schematic structure of a PM motor; FIG. 2 is a diagram showing the relationship between a rotor position and the phase of each stator winding; FIG. 3 is a diagram showing the configuration of an inverter and a current detection unit; FIG. 4 is a diagram showing the relationship between a PWM carrier signal and a voltage command value; FIG. 5 is a diagram showing an example configuration of a voltage command value calculation unit; FIG. 6 is a diagram showing another example configuration of the voltage command value calculation unit; FIG. 7 is a functional block diagram showing the processing content of a magnet temperature estimator; FIG. 8 is a functional block diagram showing an example of the processing content of a voltage detection unit; FIG. 9 is a diagram showing an example configuration of a transient component calculation unit; FIG. 10 is a diagram showing an example waveform of a magnet temperature estimated value when transient components are ignored; FIG. 11 is a diagram showing an example waveform of a magnet temperature estimated value when a q-axis magnetic flux differential value, which is a transient component, is calculated from a q-axis current detection value based on a difference method; Fig. 10 is a functional block diagram showing the processing contents of the transient component calculation unit. Fig. 11 is a functional block diagram showing another configuration example of the transient component calculation unit. Fig. 12 is a functional block diagram showing another configuration example of the magnet temperature estimation unit. Fig. 13 is a functional block diagram showing the processing contents of the transient component calculation unit in another configuration example of the magnet temperature estimation unit. Fig. 14 is a diagram showing an example waveform of the magnet temperature estimation value when the transient component is ignored. Fig. 15 is a diagram showing an example waveform of the magnet temperature estimation value when the influence of magnetic saturation is not taken into account in calculating the transient component. Fig. 16 is a diagram showing an example waveform of the magnet temperature estimation value according to the third embodiment.

[0010] Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. In the embodiments of the present invention shown below, various components do not necessarily have to exist independently of each other, and it is acceptable for multiple components to be formed as a single member, for one component to be formed by multiple members, for one component to be a part of another component, or for part of one component to overlap with part of another component.

[0011] First Embodiment A first embodiment of the present invention will be described with reference to FIGS.

[0012] (Motor Control Device 100) FIG. 1 is a diagram showing a schematic configuration of a motor control device.

[0013] In FIG. 1, a motor control device 100 includes a control unit 10, an electric motor (PM motor) 20, an inverter 30, a current detection unit 50, and an angle sensor 60 such as a resolver or an encoder.

[0014] Although not shown, the PM motor 20 is connected to a mechanism that mechanically or magnetically transmits mechanical output, and the motor control device 100 controls the rotation speed or torque of the PM motor 20 to a desired value.

[0015] Although this embodiment does not limit the operating principle of the electric motor 20, the following description will exemplify a case in which a permanent magnet synchronous motor (PMSM) having a permanent magnet in the rotor is used as the electric motor 20. Various types of electric motors are applicable as the electric motor 20, and the subject of this disclosure is an electric motor having a characteristic in which the magnetic flux of the permanent magnet changes depending on the temperature of the rotor.

[0016] The inverter 30 includes a DC voltage source 120, such as a battery, and a bridge circuit consisting of switching elements. The inverter 30 outputs a voltage by switching the switching elements in response to an input drive signal. Assuming that the switching elements ideally switch without delay, the voltage output from the inverter 30 is a pulsed voltage. Using the well-known concept of pulse width modulation, the pulsed voltage can be considered an AC voltage. In pulse width modulation, the on / off cycle of the switching elements, i.e., the switching frequency, is typically set sufficiently high relative to the frequency of the equivalent AC voltage (corresponding to the rotational frequency of a single-phase motor). However, it is also possible to consider the fundamental component of the pulsed voltage. For example, when the switching frequency is close to the rotational frequency of the PM motor, such as approximately 1 to 3 times, the fundamental component of the pulsed voltage can be considered to be applied to the PM motor 20. Therefore, in this embodiment, the output voltage of the inverter 30 is considered to have an AC waveform.

[0017] Instead of a DC voltage source, an AC power source may be converted into DC power by rectification, etc. Alternatively, a DC / DC converter may be used that controls a DC voltage source to a different voltage.

[0018] The control unit 10 includes a current command value calculation unit 11 , a voltage command value calculation unit 12 , a PWM signal generation unit 14 , a coordinate conversion unit 13 , an angular velocity calculation unit 15 , and a magnet temperature estimation unit 40 .

[0019] The current command value calculation unit 11 calculates and outputs a d-axis current command value and a q-axis current command value based on the rotation speed command value or the torque command value.

[0020] The coordinate conversion unit 13 receives the current detection value, which is information about the current flowing through the PM motor obtained by the current detection unit 50, and the motor electrical angle detection value obtained by the angle sensor 60, converts the three-phase UVW axes into dq axes, the definitions of which will be described later, and outputs the d-axis current detection value and the q-axis current detection value.

[0021] The angular velocity calculation unit 15 receives the motor electrical angle detection value and outputs the electrical angular velocity.

[0022] The voltage command value calculation unit 12 generates a d-axis voltage command value and a q-axis voltage command value based on the d-axis current command value and the q-axis current command value output by the current command value calculation unit 11, the d-axis current detection value and the q-axis current detection value output by the coordinate conversion unit 13, and the electrical angular velocity output by the angular velocity calculation unit 15.

[0023] The PWM signal generating unit 14 generates and outputs a PWM signal that controls the on / off of the switching elements that make up the inverter 30 based on the d-axis voltage command value and the q-axis voltage command value output by the voltage command value calculating unit 12 .

[0024] In FIG. 1, the rotation speed command value and torque command value are shown in the control unit 10, but they may be configured to be obtained from, for example, a higher-level control system or another control system not shown.

[0025] Furthermore, the magnet temperature estimator 40 receives as input the d-axis voltage command value, the q-axis voltage command value, the d-axis current command value, the q-axis current command value, the d-axis current detection value, the q-axis current detection value, and the electrical angular velocity, but may be configured to calculate the magnet temperature estimate using only some of the values ​​instead of inputting all of the values.

[0026] (PM Motor 20) FIG. 2 is a diagram showing a schematic structure of the PM motor 20. As shown in FIG.

[0027] In FIG. 2 , the PM motor 20 is composed of a stator 21 and a rotor 22. The stator 21 has multiple stator poles 28, each of which has a winding (coil) 25 wound around a stator core (stator iron core) 26. The rotor 22 has a permanent magnet 27. Note that FIG. 2 illustrates an example in which the number of stator poles (also referred to as the number of slots) is six and the number of rotor poles is two. However, this is not limiting. The numbers of magnetic poles of the stator 21 and rotor 22 can be freely selected, and the number of magnetic poles of the stator and rotor may not be equal. Furthermore, the multiple windings may be connected in parallel or in series. In this embodiment, the windings 25 of opposing stator poles 28 are connected in series to form one phase, and the PM motor 20 is described as having three-phase windings.

[0028] When a current flows through the windings 25, the stator magnetic poles 28 generate magnetic poles in the manner of electromagnets, and the polarity (north pole, south pole) can be changed depending on the direction of the current through the windings 25. In this embodiment, the rotation angle (rotational angular position) of the rotor 22 when a positive DC current flows through the windings 25 and the stator magnetic poles 28 become south poles, attracting the north poles of the permanent magnets 27 of the rotor 22, is defined as zero degrees. In the following description, the rotational angular position of the rotor is referred to as rotor position θd. If there are multiple windings 28, one of them is defined as the reference position. In this embodiment, the winding 25 located on the right side in FIG. 2 (i.e., the winding closest to the "U-phase") is defined as the reference position. Furthermore, the counterclockwise rotation of the rotor 22 is defined as positive rotation.

[0029] The processing within the motor control device 100 utilizes information about the rotor position θd of the rotor 22 of the PM motor 20, and the following description will be given assuming that the position information is detected by a resolver, encoder, or the like as the angle sensor 60. Naturally, it is also possible to use a configuration in which the position information is obtained by position sensorless control using a position estimation means that outputs an estimated rotational angle position of the PM motor 20 from the current flowing through the PM motor 20 and the voltage applied to the PM motor 20.

[0030] (Description of Coordinate Axes) Here, the definition of the coordinate axes will be explained. Fig. 3 is a diagram showing the relationship between the rotor position θd and the phase of each winding 25 of the stator 21.

[0031] The three-phase windings of UVW are arranged with a difference of 120 degrees in electrical angle. The d-axis is defined as the direction of the main magnetic flux of the permanent magnet provided in the rotor 22, and the dq-axis is defined as the q-axis, which is electrically advanced 90 degrees (90 degrees electrical angle) from the d-axis in the direction of rotation. The dq-axis is a rotating coordinate system. The definition of the d-axis can also be rephrased as the rotational angle position at which the magnetic flux of the permanent magnet 27 linking with the reference winding 25 is maximized.

[0032] (Inverter 30) FIG. 4 is a diagram showing the configuration of the inverter 30 and the current detection unit 50. As shown in FIG.

[0033] As shown in FIG. 4, the inverter 30 includes an inverter module 131, a DC voltage source 120, and a gate driver circuit 123. Here, the output of the DC voltage source 120 is assumed to be a DC voltage Edc. The inverter module 131 includes switching elements 32a to 32f (e.g., semiconductor switching elements such as IGBTs and MOS-FETs) and freewheeling diodes connected in parallel to the switching elements 32a to 32f. The switching elements 32a to 32f are collectively referred to as "switching elements 32." A shunt resistor 25 is connected in series to the DC voltage source 120. This resistor protects the switching elements 32 from excessive current flow.

[0034] Two pairs of these switching elements 32 are connected in series to form upper and lower arms for each phase. In FIG. 4 , the upper and lower arms for the U phase are formed by switching elements 32a and 32b, the V phase by switching elements 32c and 32d, and the W phase by switching elements 32e and 32f. The junctions of the upper and lower arms for each phase are connected to the motor 20. The gate driver circuit 123 receives a pulse-shaped drive signal (details of which will be described later) output by the PWM signal generator 14 shown in FIG. 1 and outputs drive signals 34a to 34f based on the pulse-shaped drive signal. While known techniques can be used to generate the drive signals, for example, as shown in FIG. 5 , the drive signals can be generated based on the magnitude relationship between a triangular-wave carrier signal and a voltage command value for each phase. The inverter module 131 controls the switching of each switching element 32 based on these drive signals 34a to 34f.

[0035] Depending on the switching state of the upper and lower arms, the voltage of each phase of the inverter 30 becomes either a DC voltage Edc or zero voltage. In the inverter 30, switching is performed at a frequency that is sufficiently higher than the frequency of the AC voltage appearing at the motor 20, so the output voltage of each phase of the inverter 30 can be freely adjusted by changing the switching ratio (switching duty) of the upper and lower arms. In other words, a three-phase AC voltage of any frequency can be applied to the electric motor 20, thereby enabling variable speed drive and torque control of the electric motor 20.

[0036] (Current Detector 50) The current detector 50 detects the current flowing in the U-phase and W-phase of the three-phase AC current flowing from the inverter 30 to the motor 20, and outputs the results as AC current detection values ​​Iu and Iw. Of course, it is possible to detect the AC current of all phases, but according to Kirchhoff's first law, if two of the three phases can be detected, the current of the remaining phase can be calculated from the detected two phases. The current detector 50, such as a CT (current transformer), can be provided on the lower arm of the inverter module 131.

[0037] The current detection unit 50 may employ, for example, a phase shunt current method in which a shunt resistor is added to the lower arm of the inverter module 131 instead of a CT, and the current of each phase flowing through the inverter 30 is detected from the current flowing through the shunt resistor. Instead of or in addition to the current detection unit 50, a single shunt current detection method may be employed in which the current on the AC side of the inverter 30 is detected from the DC current flowing through a shunt resistor 135 added to the DC side of the inverter 30. The single shunt current detection method utilizes the fact that the current flowing through the shunt resistor 135 changes over time depending on the conduction state of the switching elements 32 that constitute the inverter 30.

[0038] (Method of Driving PM Motor 20) When driving a motor having permanent magnets 27 in rotor 22, maximum torque can be obtained with minimum current by generating magnetic flux from winding 25 at a position 90 electrical degrees ahead in the direction of rotation of the magnetic flux from permanent magnet 27. Driving in this manner not only makes it possible to reduce the size and weight of the motor, but also has the effect of enabling the inverter 30 to be made smaller.

[0039] The aforementioned d-axis and q-axis can also be described as the magnet flux axis and the winding flux axis, respectively, and it is particularly important to appropriately control the current on the q-axis. Separating the current flowing through the motor into a field component and a torque component on the rotating coordinate axis and controlling the voltage phase and magnitude to control the motor's rotational speed or torque is generally called vector control. While PM motors are driven by control on the d- and q-axes, well-known coordinate transformation techniques can be used to convert from the three-phase UVW axes to the d- and q-axes.

[0040] (Configuration Example of Voltage Command Value Calculation Unit 12: Voltage Command Value Calculation Unit 12a) FIG. 6 is a diagram showing a configuration example of the voltage command value calculation unit.

[0041] Various conventional technologies (for example, the configurations and means described in Japanese Patent Application Laid-Open No. 2005-39912) can be applied to the functions of the voltage command value calculation unit 12, and an example configuration using these is shown in FIG. 6 as a voltage command value calculation unit 12a.

[0042] In FIG. 6 , a voltage command value calculation unit 12 a receives the d-axis and q-axis current command values ​​(Id* and Iq*) calculated by the current command value calculation unit 11 and the motor electrical angular velocity ωe as input, and performs vector calculation according to the following (Equation 1) and (Equation 2) to obtain a d-axis voltage command value Vd* and a q-axis voltage command value Vq*.

[0043] Vd*=R×Id**-ωe×Lq×Iqf** (Formula 1) Vq*=R×Iq**+ωe×Ld×Idf**+ωe×Φd0 (Formula 2) In the above formulas (1) and (2), R is the winding resistance value per phase of the motor 6, Ld is the d-axis inductance, Lq is the q-axis inductance, and Φd0 is the magnet magnetic flux. Φd0 is also called the induced voltage constant.

[0044] In Fig. 6, a d-axis current controller 114a and a q-axis current controller 114b are configured to ensure that the d-axis and q-axis currents flow as commanded. In the circuit configuration shown in Fig. 6, reference numeral 91 denotes a subtractor that calculates the difference between the current command values ​​(Id* and Iq*) and their detected values ​​(Idc and Iqc), 92 denotes a proportional circuit that adds a predetermined gain to the difference, 94 denotes an integrator, and 90 denotes an adder that adds the proportional and integral components to calculate a proportional-integral value. The outputs are the d-axis and q-axis current command values ​​(Id** and Iq**).

[0045] Id** and Iq** are multiplied by the winding resistance value R per phase of the motor 6 in multipliers 92g and 92i to obtain the first terms on the right-hand sides of the above (Equation 1) and (Equation 2).

[0046] In the second terms on the right-hand sides of the above (Equation 1) and (Equation 2), the d-axis and q-axis current command values ​​(Idf** and Iqf**) are values ​​obtained by filtering the q-axis and d-axis current command values ​​(Iq** and Id**) using filter circuits 98a and 98b in FIG. 6 . Multipliers 92h and 92j multiply Idf** and Iqf** by the q-axis inductance Lq and the d-axis inductance Ld, respectively, and also multiply them by the motor electrical angular velocity ωe to obtain the second terms on the right-hand sides of the above (Equation 1) and (Equation 2). Furthermore, multiplier 92k multiplies the motor electrical angular velocity ωe by the d-axis inductance Ld to obtain the third term on the right-hand side of the above (Equation 2).

[0047] A subtractor 91f subtracts the output of the multiplier 92h from the output of the multiplier 92g to obtain the d-axis voltage command value Vd* of the above (Equation 1). An adder 90d obtains the sum of the outputs of the multipliers 92i, 92j, and 92k to obtain the q-axis voltage command value Vq* of the above (Equation 2).

[0048] The circuit configuration example shown in Figure 6 is characterized by the fact that the voltage command value calculation unit 12a includes a d-axis current controller 114a and a q-axis current controller 114b connected in series to the voltage calculation, and first-order lag filters (low-pass filters) 98a and 98b with cutoff frequencies equivalent to the electrical time constant of the motor. These establish an inverse model of the motor, which has the effect of realizing ideal vector control even when the calculation period of the control unit 10 is limited. The voltage command value calculation unit 12a causes the detected current value to follow the current command value with a delay of the response time constant Tc. The response time constant Tc can be adjusted by the control parameters of the d-axis current controller 114a and the q-axis current controller 114b.

[0049] (Another Configuration Example of the Voltage Command Value Calculation Unit 12: Voltage Command Value Calculation Unit 12b) FIG. 7 is a diagram showing another configuration example of the voltage command value calculation unit.

[0050] Other conventional technologies (for example, the configurations and means described in Japanese Patent Application Laid-Open No. 2021-151003) can be applied to the functions of the voltage command value calculation unit 12, and an example configuration using this is shown in FIG. 7 as a voltage command value calculation unit 12b.

[0051] In this configuration example, the dq axis magnetic flux command value and magnetic flux detection value are calculated from the dq axis current command value and current detection value, and the magnetic flux detection value is controlled to be the magnetic flux command value.

[0052] To accurately control a motor with a high magnetic flux density for miniaturization, the effects of magnetic saturation must be considered. The d-axis and q-axis inductances fluctuate depending on the operating state due to the effects of magnetic saturation. Ignoring this phenomenon and maintaining constant d-axis and q-axis inductances for control results in poor controllability. One solution to this problem is to vary the d-axis and q-axis inductances used for control depending on the operating state. However, this approach complicates the control system, increasing the number of constants used in inductance-related calculations, making parameter identification difficult and increasing the load on the control device. The circuit configuration shown in Figure 7 calculates the magnetic flux command value and detected magnetic flux value while taking the effects of magnetic saturation into account, and controls the detected magnetic flux value to match the magnetic flux command value, thereby enabling accurate motor control without complicating the control system.

[0053] A d-axis magnetic flux command value calculation unit 80 calculates a d-axis magnetic flux command value Φd* from the d-axis current command value Id* and the q-axis current command value Iq*. Similarly, a q-axis magnetic flux command value calculation unit 81 calculates a q-axis magnetic flux command value Φq* from the d-axis current command value Id* and the q-axis current command value Iq*. A d-axis magnetic flux detected value calculation unit 82 calculates a d-axis magnetic flux detected value Φdc from the d-axis current detected value Idc and the q-axis current detected value Iqc. Similarly, a q-axis magnetic flux detected value calculation unit 83 calculates a q-axis magnetic flux detected value Φqc from the d-axis current detected value Idc and the q-axis current detected value Iqc.

[0054] The voltage command value calculation unit 84 calculates a d-axis voltage command value Vd* and a q-axis voltage command value Vq* so that the d-axis magnetic flux detection value Φdc is equal to the d-axis magnetic flux command value Φd*, and the q-axis magnetic flux detection value Φqc is equal to the q-axis magnetic flux command value Φq*. The voltage command value calculation unit 12b causes the magnetic flux detection value to follow the magnetic flux command value with a delay of a response time constant Tc. The response time constant Tc can be adjusted by the control parameters of the voltage command value calculation unit 84.

[0055] (Challenges in Improving Torque Accuracy) The motor control device 100 controls the rotation speed or torque of the PM motor 20 to a desired value, but when driving a PM motor, if the temperature of the magnet changes, the magnetic flux changes in response to the temperature, so there is a challenge in that torque accuracy will deteriorate unless some kind of response is taken.

[0056] If the magnet temperature can be measured, it is possible to improve torque accuracy by adjusting the current command value based on the temperature. However, directly connecting a temperature sensor to the magnet inside the rotor is difficult due to issues with durability and productivity. Therefore, estimating the magnet temperature is an effective solution.

[0057] Furthermore, in motor control devices for driving electric vehicles such as electric cars, the torque command value fluctuates depending on the driving conditions, frequently causing transient states in which the d-axis current or q-axis current fluctuates. Furthermore, with the aim of suppressing motor vibration and noise, preventing slippage, and driving electric vehicles stably, development of control that constantly fluctuates the torque command value is progressing, and the frequency of transient states is increasing. Therefore, it is necessary to accurately estimate magnet temperature not only in steady states, but also in transient states.

[0058] Therefore, in this embodiment, the transient component calculation unit calculates a q-axis magnetic flux differential value based on the difference between the q-axis magnetic flux command value and the q-axis magnetic flux detection value or the difference between the q-axis current command value and the q-axis current detection value, calculates a d-axis magnetic flux estimate value based on the q-axis voltage value, q-axis current value, electrical angular velocity, and q-axis magnetic flux differential value, and calculates a magnet temperature estimate value based on the d-axis magnetic flux estimate value. This makes it possible to accurately calculate the transient component when estimating the magnet temperature or magnet flux, and improves estimation accuracy even in a transient state where the d-axis current or q-axis current is fluctuating. This point will be explained in more detail below.

[0059] (Configuration and Operation of Magnet Temperature Estimation Unit) FIG. 8 is a functional block diagram showing the processing details of the magnet temperature estimation unit 40.

[0060] Here, we will explain the operation of estimating the magnet temperature while the motor is running. The relationship between the q-axis voltage value Vq, motor resistance value R, q-axis current value Iq, d-axis magnetic flux Φd, q-axis magnetic flux Φq, and electrical angular velocity ωe can be expressed by the following (Equation 3).

[0061] Vq = R x Iq + dΦq / dt + ωe x Φd (Equation 3) Here, the d-axis magnetic flux Φd and the q-axis magnetic flux Φq can be expressed by the following (Equation 4) and (Equation 5) using the d-axis inductance Ld, the d-axis current value Id, the magnet magnetic flux Φd0, the q-axis inductance Lq, and the q-axis current value Iq.

[0062] Φd=Ld×Id+Φd0 (Formula 4) Φq=Lq×Iq (Formula 5) Here, the value of the magnet magnetic flux Φd0 changes depending on the magnet temperature, and therefore the value of the d-axis magnetic flux Φd also changes depending on the magnet temperature.

[0063] Therefore, the d-axis magnetic flux Φd is estimated while the motor is running, and the magnet temperature can be estimated from the relationship between the d-axis magnetic flux and the magnet temperature. The d-axis magnetic flux estimator 41a calculates an estimated value Φd_est of the d-axis magnetic flux Φd using the following (Equation 6) based on the above (Equation 3).

[0064] Φd_est = (Vq - R × Iq ​​- dΦq / dt) / ωe (Equation 6) When calculating the d-axis magnetic flux estimate Φd_est using Equation 6, the q-axis voltage value Vq in Equation 6 is replaced with the q-axis voltage command value Vq*. Alternatively, as shown in FIG. 9 , the inverter output voltage is measured by a voltage sensor 70, the detected voltage value is input to the control unit 10, and the coordinate conversion unit 13a converts the detected voltage value from the three-phase UVW axes to the dq axes to obtain the d-axis voltage detected value Vdc and the q-axis voltage detected value Vqc. The d-axis current value Iq in Equation 6 may be replaced with the q-axis current detected value Iqc or the q-axis current command value Iq*. Calculation of the q-axis magnetic flux differential value dΦq / dt, which is a transient component, will be described later.

[0065] The d-axis magnetic flux estimated value Φd_est is input to the magnet magnetic flux calculation unit 43a, which calculates the magnet temperature estimated value. The magnet magnetic flux Φd0 estimated value Φd0_est is calculated by the following (Equation 7) based on the above (Equation 4).

[0066] Φd0_est=Φd_est-Ld×Id (Equation 7) The d-axis current value Id in the above (Equation 7) may be the d-axis current detection value Idc or the d-axis current command value Id*. Next, using the correspondence relationship between the magnet magnetic flux Φd0 and the magnet temperature, the magnet temperature estimated value is calculated from the magnet magnetic flux estimated value Φd0_est.

[0067] One example of a method for calculating a magnet temperature estimate from the magnet flux estimate value Φd0_est is to measure the magnet temperature in advance or calculate and store it by electromagnetic field simulation, and create a magnet temperature table using the magnet flux Φd0 as a parameter to use during estimation. Alternatively, the relationship between the magnet temperature and the magnet flux Φd0 may be expressed as a formula and used during estimation. In this embodiment, any calculation method is acceptable as long as the magnet temperature estimate can be calculated from the magnet flux estimate value Φd0_est.

[0068] As another method for calculating the magnet temperature estimated value in the magnet flux calculation unit 43a, the magnet temperature estimated value can be calculated directly from the d-axis magnetic flux estimated value Φd_est without calculating the magnet flux estimated value Φd0_est.

[0069] The d-axis magnetic flux Φd changes depending on the magnet temperature, the d-axis current value Id, and the q-axis current value Iq. Therefore, the magnet temperature estimation value is calculated using the correspondence relationship between the d-axis magnetic flux estimation value Φd_est, the d-axis current value Id, the q-axis current value Iq, and the magnet temperature.

[0070] As an example of a method for calculating the magnet temperature estimate from the d-axis magnetic flux estimate value Φd_est, the d-axis current value Id, and the q-axis current value Iq, there is a configuration in which the magnet temperatures for the d-axis magnetic flux Φd, the d-axis current value Id, and the q-axis current value Iq are measured in advance or calculated and stored by electromagnetic field simulation, and a magnet temperature table is created using the magnet magnetic flux Φd0, the d-axis current value Id, and the q-axis current value Iq as parameters, and used during estimation. In this embodiment, the calculation method is not important as long as the magnet temperature estimate can be calculated from the d-axis magnetic flux estimate value Φd_est.

[0071] (Configuration and Operation of Transient Component Calculation Unit) In order to estimate the magnet temperature with high accuracy in a transient state where the d-axis current or q-axis current is fluctuating, it is necessary to accurately calculate the q-axis magnetic flux differential value dΦq / dt, which is the transient component in the above (Equation 6). The q-axis magnetic flux differential value dΦq / dt can be expressed by the following (Equation 8) based on the above (Equation 5).

[0072] dΦq / dt = Lq × dIq / dt (Equation 8) The q-axis magnetic flux derivative dΦq / dt can be calculated by multiplying the derivative of the q-axis current value by the q-axis inductance. One possible method for calculating the derivative of the q-axis current value is to use only the q-axis current detection value and use the difference method, in which the difference between the past sampled value and the current value is divided by the time difference between the sampled values. With the difference method, because division is done by the time difference between the sampled values, which is a very small value, the impact of noise contained in the q-axis current detection value is significant. As a result, an error occurs in the q-axis magnetic flux derivative dΦq / dt, degrading the estimation accuracy of the transient state.

[0073] Therefore, in this embodiment, the difference ΔIq between the q-axis current command value Iq* and the q-axis current detection value Iqc is input to a transient component calculation section 42a, which calculates the q-axis magnetic flux differential value dΦq / dt.

[0074] FIG. 10 is a diagram showing an example of the configuration of the transient component calculation unit 42a.

[0075] 10, the transient component calculation unit 42a utilizes the fact that the q-axis current command value Iq* and the q-axis current detection value Iqc follow each other with a delay of the response time constant Tc. If the reciprocal of the response time constant Tc is set to the response angular frequency ωc and the response delay is approximated as a first-order lag, the transfer function of the q-axis current command value Iq* and the q-axis current detection value Iqc can be expressed by the following (Equation 9) using the Laplace operator s.

[0076] Iqc=1 / {1+s×(1 / ωc)}×Iq* (Equation 9) Furthermore, the following (Equation 10) can be obtained from the above (Equation 9).

[0077] s × Iqc = ωc × (Iq* - Iqc) ... (Equation 10) Since s × Iqc on the left side of the above equation (10) represents the derivative of the q-axis current detection value Iqc, the derivative value dIqc / dt of the q-axis current detection value Iqc can be calculated by multiplying the difference ΔIq between the q-axis current command value Iq* and the q-axis current detection value Iqc by the response angular frequency ωc.

[0078] The transient component calculation unit 42a calculates the q-axis magnetic flux differential value dΦq / dt by the following (Equation 11).

[0079] dΦq / dt=Lq×ωc×ΔIq (Equation 11) The q-axis magnetic flux differential value dΦq / dt can be calculated without performing an operation such as dividing by the time difference of the sampling values, which is a small value that causes the differential method to be significantly affected by noise.

[0080] (Effects of the Present Embodiment) The effects of the present embodiment configured as above will be described.

[0081] 11 to 13 are diagrams showing example waveforms of estimated magnet temperature values ​​when the handling of transient components is changed. Figures 11 to 13 show example waveforms of estimated magnet temperature values ​​when the torque is changed in steps from 0% to 100% under the condition that the magnet temperature is 100°C.

[0082] 11 shows an example of the waveform of the magnet temperature estimated value when a configuration that ignores transient components is adopted as a comparative example to the present invention. A configuration that ignores transient components corresponds, for example, to a case where the magnet temperature estimator 40a is operated with the q-axis magnetic flux derivative set to zero. In this case, a large estimated temperature error occurs when the torque changes from 0% to 100%.

[0083] 12 shows, as another comparative example to the present invention, an example waveform of the magnet temperature estimated value when the q-axis magnetic flux derivative, which is a transient component, is calculated from the q-axis current detection value based on the difference method. In this case, the estimated temperature error can be reduced when the torque changes from 0% to 100%, but the q-axis magnetic flux derivative is significantly affected by noise in the q-axis current detection value, resulting in a large error in the estimated temperature.

[0084] An example waveform of the magnet temperature estimation value in this embodiment is shown in Figure 13. As shown in Figure 13, in this embodiment, the q-axis magnetic flux derivative can be calculated with little effect from noise, so the estimated temperature error at the timing when the torque changes from 0% to 100% is small, and the estimated temperature error caused by noise can also be suppressed.

[0085] Second Embodiment A second embodiment of the present invention will be described with reference to FIGS.

[0086] This embodiment shows a case where a different configuration is used for calculating the transient component in the magnet temperature estimator 40 in the first embodiment. Note that in this embodiment, only the differences from the first embodiment will be described, and the same components as in the first embodiment will be given the same reference numerals and descriptions will be omitted as appropriate. (Magnet temperature estimator 40b) Figure 14 is a functional block diagram showing the processing contents of the magnet temperature estimator according to this embodiment.

[0087] As shown in FIG. 14, the transient component calculation unit 42b of the magnet temperature estimation unit 40b in this embodiment calculates the q-axis magnetic flux differential value using as input the difference ΔΦq between the q-axis magnetic flux command value Φq* and the q-axis magnetic flux detection value Φqc.

[0088] The q-axis magnetic flux command value calculation unit 44b calculates a q-axis magnetic flux command value Φq* from the d-axis current command value Id* and the q-axis current command value Iq*.

[0089] The q-axis magnetic flux detection value calculation unit 45b calculates a q-axis magnetic flux detection value Φqc from the d-axis current detection value Idc and the q-axis current detection value Iqc.

[0090] The q-axis magnetic flux command value calculation unit 44b and the q-axis magnetic flux detection value calculation unit 45b may be configured, for example, to calculate and store the q-axis magnetic flux Φq relative to the d-axis current value Id and the q-axis current value Iq in advance by measurement or electromagnetic field simulation, and create a table of the q-axis magnetic flux Φq using the d-axis current value Id and the q-axis current value Iq as parameters, and use this table during estimation. Alternatively, for example, the relationship between the q-axis magnetic flux Φq relative to the d-axis current value Id and the q-axis current value Iq may be expressed as a mathematical formula, and used during estimation. Note that in this embodiment, any calculation method is acceptable as long as the q-axis magnetic flux Φq can be calculated from the d-axis current value Id and the q-axis current value Iq.

[0091] (Transient Component Calculation Unit 42b) FIG. 15 is a functional block diagram showing the processing contents of the transient component calculation unit.

[0092] 15, the transient component calculation unit 42b calculates the q-axis magnetic flux differential value by utilizing the fact that the q-axis magnetic flux command value Φq* and the q-axis magnetic flux detection value Φqc follow each other with a delay of the response time constant Tc. If the reciprocal of the response time constant Tc is set to the response angular frequency ωc and the response delay is approximated as a first-order delay, the transfer function of the q-axis magnetic flux command value Φq* and the q-axis magnetic flux detection value Φqc can be expressed by the following (Equation 12) using the Laplace operator s.

[0093] Φqc=1 / {1+s×(1 / ωc)}×Φq* (Equation 12) Furthermore, from the above (Equation 12), the following (Equation 13) is obtained.

[0094] s×Φqc=ωc×(Φq*-Φqc) (Equation 13) Since s×Φqc on the left side of the above (Equation 13) represents the derivative of the q-axis magnetic flux detection value Φqc, the derivative value dΦqc / dt of the q-axis magnetic flux detection value Φqc can be calculated by multiplying the difference ΔΦq between the q-axis magnetic flux command value Φq* and the q-axis magnetic flux detection value Φqc by the response angular frequency ωc. The transient component calculation unit 42b calculates the q-axis magnetic flux derivative value dΦq / dt using the following (Equation 14).

[0095] dΦq / dt=ωc×ΔΦq (Equation 14) The other configurations are the same as those in the first embodiment.

[0096] (Effects of this embodiment) In this embodiment configured as above, it is possible to suppress the influence of noise, similarly to the first embodiment.

[0097] Furthermore, in this embodiment, the calculation load is small and the q-axis magnetic flux differential value dΦq / dt can be calculated. As in the first embodiment, the influence of noise can be suppressed.

[0098] This embodiment is particularly suitable when the voltage command value calculation unit 12 has the configuration based on the magnetic flux command value and the measured value shown in Fig. 7. The q-axis magnetic flux command value calculation unit 44b and the q-axis magnetic flux detected value calculation unit 45b in Fig. 14 can be omitted, and instead the q-axis magnetic flux command value Φq* and the q-axis magnetic flux detected value Φqc can be obtained from the q-axis magnetic flux command value calculation unit 81 and the q-axis magnetic flux detected value calculation unit 83 in Fig. 7. In other words, the additional configuration and calculation load are small, and it is possible to calculate the transient component.

[0099] Third Embodiment A third embodiment of the present invention will be described with reference to FIGS.

[0100] This embodiment shows a case where a different configuration is used for calculating the transient component in the magnet temperature estimation unit 40 in the first embodiment. Note that in this embodiment, only the differences from the first embodiment will be described, and the same members as in the first embodiment will be given the same reference numerals and descriptions thereof will be omitted as appropriate.

[0101] FIG. 16 is a diagram showing an example of the characteristics of the q-axis magnetic flux Φq and the q-axis inductance Lq with respect to the q-axis current value Iq.

[0102] In order to make motors more compact, the magnetic flux density is increasing. This trend is particularly noticeable in electric vehicles such as electric cars. In motors with high magnetic flux density, the influence of magnetic saturation on the magnetic flux and inductance characteristics relative to the current becomes more pronounced.

[0103] As shown in Figure 16, as the q-axis current value Iq increases from zero, the q-axis magnetic flux Φq initially increases linearly, but gradually the effects of magnetic saturation appear, the increase saturates, and the characteristic becomes nonlinear. Here, as shown in Figure 16, the slope of the q-axis magnetic flux Φq with respect to the q-axis current value Iq is the q-axis inductance Lq. Therefore, as the q-axis current value Iq increases from zero, the q-axis inductance Lq remains approximately constant or gradually decreases, and its value rapidly decreases above the current value at which the effects of magnetic saturation become significant.

[0104] In motors that are significantly affected by magnetic saturation, if the magnetic saturation characteristics of the q-axis magnetic flux Φq and the magnetic characteristics of the q-axis inductance described in Figure 16 are not taken into consideration when calculating the q-axis magnetic flux differential value dΦq / dt, which is the transient component, an error will occur in the calculated value of the transient component, and the accuracy of estimating the magnet temperature will deteriorate.

[0105] (Magnet Temperature Estimation Unit 40c) FIG. 17 is a functional block diagram showing the processing details of the magnet temperature estimation unit according to this embodiment.

[0106] In FIG. 17, in a magnet temperature estimation unit 40c according to this embodiment, a transient component calculation unit 42c receives the q-axis current command value Iq*, the d-axis current detection value Idc, and the q-axis current detection value Iqc as inputs, and calculates the q-axis magnetic flux differential value taking into account the influence of magnetic saturation.

[0107] (Transient Component Calculation Unit 42c) FIG. 18 is a functional block diagram showing the processing contents of the transient component calculation unit 42c.

[0108] As shown in FIG. 18 , the transient component calculation unit 42c, like the first embodiment, multiplies the difference ΔIq between the q-axis current command value Iq* and the d-axis current detection value Idc by the response angular frequency ωc to calculate the q-axis current differential value dIq / dt. The transient component calculation unit 42c also includes a q-axis inductance calculation unit 421c that receives the d-axis current detection value Idc and the q-axis current detection value Iqc as inputs, and calculates the q-axis inductance Lq taking magnetic saturation characteristics into account. The q-axis magnetic flux differential value dΦq / dt is calculated from the product of the q-axis inductance Lq and the q-axis current differential value dIq / dt. This allows accurate calculation of the transient component, even in motors that are significantly affected by magnetic saturation, thereby improving the accuracy of magnet temperature estimation during transients.

[0109] In one example of the configuration of the q-axis inductance calculation unit 421c, the q-axis inductance Lq for the d-axis current value Id and the q-axis current value Iq is measured or calculated and stored in advance by electromagnetic field simulation, and a table of the q-axis inductance Lq using the d-axis current value Id and the q-axis current value Iq as parameters is created and used during estimation. Alternatively, the relationship between the q-axis inductance Lq for the d-axis current value Id and the q-axis current value Iq may be expressed as a formula and used during estimation.

[0110] (Another Configuration Example of the Transient Component Calculation Unit: Transient Component Calculation Unit 42d) It is also possible to consider other configurations for the transient component calculation unit.

[0111] FIG. 19 is a diagram illustrating another example of the configuration of the transient component calculation unit.

[0112] In FIG. 19, a transient component calculation unit 42d calculates a q-axis current differential value dIq / dt from the q-axis current detection value Iqc using a differentiator, which is different from the transient component calculation unit 42c shown in FIG.

[0113] (Another Configuration Example of the Magnet Temperature Estimation Unit: Magnet Temperature Estimation Unit 40e) The magnet temperature estimation unit may also have another configuration.

[0114] FIG. 20 is a diagram showing another example of the configuration of the magnet temperature estimation unit.

[0115] In FIG. 20, the magnet temperature estimation unit 40e is equipped with a q-axis magnetic flux detection value calculation unit 45e that receives the d-axis current detection value Idc and the q-axis current detection value Iqc and outputs the q-axis magnetic flux detection value Φqc, and the transient component calculation unit 42e receives the q-axis magnetic flux detection value Φqc and calculates the q-axis magnetic flux differential value dΦq / dt, which is different from the magnet temperature estimation unit 40c shown in FIG.

[0116] (Transient Component Calculation Unit 42e) FIG. 21 is a functional block diagram showing the processing contents of the transient component calculation unit 42e.

[0117] As shown in Fig. 21, a transient component calculation unit 42e calculates a q-axis magnetic flux differential value dΦq / dt from the q-axis magnetic flux detection value Φqc using a differentiator. A q-axis magnetic flux detection value calculation unit 45e calculates the q-axis magnetic flux Φq in consideration of the magnetic saturation characteristics illustrated in Fig. 16.

[0118] In one example of the configuration of the q-axis magnetic flux detection value calculation unit 45e, the q-axis magnetic flux Φq relative to the d-axis current value Id and the q-axis current value Iq is measured or calculated and stored in advance by electromagnetic field simulation, and a table of the q-axis magnetic flux Φq using the d-axis current value Id and the q-axis current value Iq as parameters is created and used during estimation. Alternatively, the relationship between the q-axis magnetic flux Φq relative to the d-axis current value Id and the q-axis current value Iq may be expressed as a formula and used during estimation.

[0119] When the voltage command value calculation unit 12 has a configuration based on the magnetic flux command value and the measured value shown in FIG. 7, the q-axis magnetic flux detection value calculation unit 45e in FIG. 20 can be omitted, and instead the q-axis magnetic flux detection value Φqc can be obtained from the q-axis magnetic flux detection value calculation unit 83 in FIG. 7, thereby realizing the same function.

[0120] The other configurations are the same as those of the first embodiment.

[0121] (Effects of the Present Embodiment) The effects of the present embodiment configured as above will be described.

[0122] Figures 22 to 24 show example waveforms of estimated magnet temperature values ​​when the handling of transient components is changed. Figures 22 to 24 show example waveforms of estimated magnet temperature values ​​when the torque is changed in steps from 0% to 50%, and from 50% to 100%, at a magnet temperature of 100°C for a motor that is heavily affected by magnetic saturation.

[0123] 22 shows an example waveform of the magnet temperature estimated value when a configuration that ignores transient components is adopted as a comparative example to the present invention. A configuration that ignores transient components corresponds to a case where the magnet temperature estimator 40c operates with the q-axis magnetic flux derivative set to zero. In this case, large estimated temperature errors occur when the torque changes from 0% to 50% and when it changes from 50% to 100%.

[0124] Figure 23 shows an example waveform of the magnet temperature estimated value when a configuration that does not consider the effects of magnetic saturation is adopted as another comparative example of the present invention. A configuration that does not consider the effects of magnetic saturation corresponds to a case where the q-axis inductance Lq is constant in the magnet temperature estimator 40c. In this case, the estimated temperature error can be reduced when the torque changes from 0% to 50%, but a large estimated temperature error occurs when the torque changes from 50% to 100%. This is because, under conditions where the torque increases and the q-axis current also increases, the q-axis inductance decreases due to the effects of magnetic saturation. Even in such cases, if the transient component is calculated with the q-axis inductance Lq constant, the calculated value of the transient component will be excessive, resulting in an estimated temperature error.

[0125] Fig. 24 shows an example waveform of the magnet temperature estimation value in this embodiment. In this embodiment, the q-axis inductance Lq or the q-axis magnetic flux detection value Φqc is calculated taking into account the influence of magnetic saturation in accordance with the d-axis current value and q-axis current value, which change with torque, so as shown in Fig. 13, the estimated temperature error is small both at the timing when the torque changes from 0% to 50% and at the timing when the torque changes from 50% to 100%.

[0126] <Notes> The present invention is not limited to the above-described examples, and various modifications are possible without departing from the spirit of the present invention. For example, the above-described embodiments have been described in detail to clearly explain the present invention, and the present invention is not necessarily limited to those having all of the described configurations.

[0127] Furthermore, the above-described configurations, functions, processing units, processing procedures, etc. may be partly or entirely realized in hardware, for example, by designing them as integrated circuits, etc. Furthermore, the above-described configurations, functions, etc. may be realized in software, by a processor interpreting and executing a program that realizes each function.

[0128] Furthermore, the signal lines and information lines shown are those considered necessary for explanation, and do not necessarily represent all control lines and information lines. Furthermore, the above-mentioned configurations, functions, processing units, processing procedures, etc. do not necessarily need to be located in the same physical location; for example, some may be located on a network or in the cloud via communication, etc.

[0129] Also, while the motor has been described as having an inner rotor structure with the rotor placed inside the stator, it can of course also have an outer rotor structure. It also does not matter whether it is a radial gap motor or an axial gap motor. In addition to PMSM, a synchronous motor or SRM with a magnet can also be used.

[0130] 10...control unit, 11...current command value calculation unit, 12, 12a, 12b...voltage command value calculation unit, 13, 13a...coordinate conversion unit, 14...PWM signal generation unit, 15...angular velocity calculation unit, 20...electric motor (PM motor), 21...stator, 22...rotor, 25...winding (coil), 25...shunt resistor, 26...stator core (stator iron core), 27...permanent magnet, 28...winding, 28...fixed Child magnetic pole, 30... inverter, 32, 32a, 32b, 32c, 32d, 32e, 32f... switching elements, 34a, 34b, 34c, 34d, 34e, 34f... drive signals, 40, 40a, 40b, 40c, 40e... magnet temperature estimation unit, 41a... d-axis magnetic flux estimation unit, 42a, 42b, 42c, 42d, 42e... transient component calculation unit, 43a... magnet magnetic flux calculation unit, 4 4b...q-axis magnetic flux command value calculation unit, 45b, 45e...q-axis magnetic flux detection value calculation unit, 50...current detection unit, 60...angle sensor, 70...voltage sensor, 80...d-axis magnetic flux command value calculation unit, 81...q-axis magnetic flux command value calculation unit, 82...d-axis magnetic flux detection value calculation unit, 83...q-axis magnetic flux detection value calculation unit, 84...voltage command value calculation unit, 90...electrical angle, 90d...adder, 91f...subtractor, 92g, 92h, 9 2i, 92j, 92k... multipliers, 98a, 98b... filter circuits (low-pass filters), 100... motor control device, 114a... d-axis current controller, 114b... q-axis current controller, 120... DC voltage source, 123... gate driver circuit, 131... inverter module, 135... shunt resistor, 421c... q-axis inductance calculation unit, PMSM... permanent magnet synchronous motor

Claims

1. A motor control device that drives a motor using an inverter, comprising: a transient component calculation unit that calculates a q-axis magnetic flux derivative based on the difference between a q-axis magnetic flux command value and a q-axis magnetic flux detection value, or the difference between a q-axis current command value and a q-axis current detection value; a d-axis magnetic flux estimation unit that calculates a d-axis magnetic flux estimate based on the q-axis voltage value, the q-axis current, the electrical angular velocity, and the q-axis magnetic flux derivative; and a magnet temperature calculation unit that calculates a magnet temperature estimate based on the d-axis magnetic flux estimate.

2. A motor control device according to claim 1, characterized in that the transient component calculation unit calculates the q-axis magnetic flux differential value based on a value obtained by multiplying the difference between the q-axis magnetic flux command value and the q-axis magnetic flux detection value by the inverse of the response time constant of the q-axis magnetic flux detection value to the q-axis magnetic flux command value, or a value obtained by multiplying the difference between the q-axis current command value and the q-axis current detection value by the inverse of the response time constant of the q-axis current detection value to the q-axis current command value.

3. A motor control device as claimed in claim 1 or 2, comprising: a q-axis magnetic flux command value calculation unit which calculates the q-axis magnetic flux command value based on a d-axis current command value and the q-axis current command value; and a q-axis magnetic flux detection value calculation unit which calculates the q-axis magnetic flux detection value based on the d-axis current detection value and the q-axis current detection value.

4. A motor control device that drives a motor using an inverter, comprising: a transient component calculation unit that calculates a q-axis magnetic flux differential value taking into account magnetic saturation of the motor based on a d-axis current detection value, a q-axis current detection value, a q-axis current command value, or a q-axis magnetic flux detection value; a d-axis magnetic flux estimation unit that calculates a d-axis magnetic flux estimate value based on the q-axis voltage value, the q-axis current, the electrical angular velocity, and the q-axis magnetic flux differential value; and a magnet temperature calculation unit that calculates a magnet temperature estimate value based on the d-axis magnetic flux estimate value.

5. A motor control device as claimed in claim 4, wherein the transient component calculation unit comprises: a calculation means for calculating a q-axis current differential value from the difference between the q-axis current command value and the q-axis current detection value, or a q-axis current differentiator for differentiating the q-axis current detection value to calculate the q-axis current differential value; and a q-axis inductance calculation unit for calculating a q-axis inductance based on the d-axis current detection value and the q-axis current detection value, wherein the motor control device calculates the q-axis magnetic flux differential value by the product of the q-axis current differential value and the q-axis inductance, or calculates the q-axis magnetic flux differential value by a q-axis magnetic flux differentiator that differentiates the q-axis magnetic flux detection value.

6. A motor control device according to claim 4 or 5, further comprising a q-axis magnetic flux detection value calculation unit that calculates the q-axis magnetic flux detection value based on the d-axis current detection value and the q-axis current detection value.

7. A control method for a motor driven by an inverter, comprising: a transient component calculation unit that calculates a q-axis magnetic flux differential value based on the difference between a q-axis magnetic flux command value and a q-axis magnetic flux detection value, or the difference between a q-axis current command value and a q-axis current detection value; a step of calculating a d-axis magnetic flux estimate value based on the q-axis voltage value, the q-axis current, the electrical angular velocity, and the q-axis magnetic flux differential value; and a step of calculating a magnet temperature estimate value based on the d-axis magnetic flux estimate value.

8. A control method for a motor driven by an inverter, comprising the steps of: calculating a q-axis magnetic flux differential value taking into account magnetic saturation of the motor based on a d-axis current detection value, a q-axis current detection value, a q-axis current command value, or a q-axis magnetic flux detection value; calculating a d-axis magnetic flux estimate value based on the q-axis voltage value, the q-axis current, the electrical angular velocity, and the q-axis magnetic flux differential value; and calculating a magnet temperature estimate value based on the d-axis magnetic flux estimate value.

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