Switched inductor, resonance circuit, and RF power supply

Abstract

A switched inductor according to the present invention comprises a transformer and a switch unit. The switch unit is provided with: a semiconductor switch unit that switches the inductance of an inductor of a primary-side coil by means of connection and disconnection between both ends of a secondary-side coil of the transformer; a switch control unit that controls an opening / closing operation of a switching element of the semiconductor switch unit; and a bias unit that applies a bias voltage to the switching element of the semiconductor switch unit. The output capacity of the switching element is reduced by the application of the bias voltage, thereby increasing the self-resonant frequency of the switched inductor, improving the frequency characteristics of the inductor, and improving the action of the inductor in a high-frequency region.

Description

Switched inductor, resonant circuit, and RF power supply

[0001] The present invention relates to a switched inductor, a resonant circuit using a switched inductor, and an RF device.

[0002] Variable inductors that make inductance variable are used as circuit elements that constitute circuits such as resonant circuits, matching circuits, and resonant inverters, and are also used in RF power supplies that include these circuits.

[0003] A known variable inductor with variable inductance is a switched inductor, which varies the inductance between the ends of a coil by turning a switching element on and off. A known switched inductor has a configuration in which the inductance of the primary coil of a transformer is varied by opening and closing a loop conductor formed by a loop-shaped conductor including the secondary coil of the transformer with a switch.

[0004] The magnetic flux generated in the primary coil of the transformer causes an induced current to flow in the loop conductor of the closed loop formed when the switch is on. This induced current reduces the inductance between the ends of the primary coil of the transformer compared to when the switching element is off. (Patent Document 1) A switched inductor prevents overvoltage and overcurrent generated in a resonant circuit, a matching device, a load, etc. by gradually changing the inductive component through switch control.

[0005] JP 2012-60157 A

[0006] In high-power, high-frequency environments, such as those with 300 W or more and 1 MHz or more, a factor limiting the operation of a switched inductor is the output capacitance Coss of the switching element of the switched inductor. In a switched inductor, parasitic capacitances such as the output capacitance Coss lower the self-resonant frequency of the inductor, limiting operation in the high-frequency range. The self-resonant frequency (SRF) of an inductor is the maximum frequency at which the inductor's reactance first changes from positive inductance to negative capacitive as the frequency increases. A large output capacitance Coss of the transistor constituting the switching element lowers the self-resonant frequency, degrading the inductor's characteristics and limiting high-frequency operation in resonant inverters and RF power supplies. Therefore, conventional switched inductors and resonant inverters and RF power supplies equipped with switched inductors have the problem that the high-frequency characteristics of the switching element are limited by the output capacitance Coss of the transistor constituting the switching element.

[0007] The present invention aims to solve the above-mentioned conventional problems, reduce the output capacitance Coss of a switching element provided in a switched inductor, increase the self-resonant frequency of the inductor, improve the inductor characteristics, and enable high-frequency operation in a switched inductor, a resonant inverter, and an RF power supply.

[0008] The present invention relates to a switched inductor, a resonant inverter including the switched inductor as a component, and an RF power supply including the resonant inverter as a component. By including the switched inductor of the present invention, the resonant inverter and RF power supply can reduce the output capacitance Coss of the switching element of the switched inductor, increase the self-resonant frequency, improve the inductor characteristics, and enable high-frequency operation.

[0009] (Switched Inductor) The switched inductor of the present invention includes a transformer and a switch unit. Each end of the primary coil of the transformer constitutes each end of the switched inductor, and serves as an end of an inductor that constitutes a resonant circuit or the like. Each end of the secondary coil of the transformer is connected to each end of the switch unit.

[0010] The switch unit includes: (a) a semiconductor switch unit that disconnects both ends of the secondary coil of the transformer and switches the inductance of the inductor of the primary coil via the magnetic coupling of the transformer; (b) a switch control unit that controls the opening and closing operation of the switching element of the semiconductor switch unit; and (c) a bias unit that applies a bias voltage to the switching element of the semiconductor switch unit.

[0011] The switching element of the semiconductor switch unit is controlled by a switch control unit. When the switching element is in the off state, the inductor of the secondary coil and the switching element are in an open circuit, so the inductance L of the switched inductor becomes the inductance L1 of the primary coil of the transformer.

[0012] On the other hand, when the switching element is in the on state, the inductor of the primary coil and the switching element form a closed circuit, so the primary coil and secondary coil are magnetically coupled. When expressed as a coupling coefficient κ between the windings of the transformer due to magnetic coupling, the inductance L of the switched inductor is L1·(1−κ ２ ) As a result, the inductance L of the switched inductor is switched.

[0013] The output capacitance of a switching element is dependent on the voltage between the current input terminal and the current output terminal. Based on this output capacitance characteristic of the switching element, the present invention reduces the output capacitance of the switching element by applying a bias voltage to the switching element. By reducing the output capacitance of the switching element, the self-resonant frequency of the switched inductor increases, improving the frequency characteristics of its function as an inductor, and allowing it to be used as an inductor suitable for high-frequency ranges.

[0014] The bias section is composed of a series circuit of a constant voltage source and a resistor. The constant voltage source applies a bias voltage to the switching element via the resistor when the switching element is switched to the on state. The bias voltage is applied by connecting the positive voltage terminal of the constant voltage source to the input terminal of the switching element, and when the switching element is switched to the on state, a current flows from the current input terminal to the current output terminal of the switching element. In a configuration using a FET as the switching element, the bias voltage is applied to the drain terminal of the FET, and a current flows from the drain terminal to the source terminal.

[0015] The resistance of the bias section is set to be higher than the equivalent parasitic resistance of the transformer. Because the resistance of the bias section is significantly higher than the equivalent parasitic resistance of the transformer, the DC potential of the bias section acts as a bias potential for the voltage of the high-frequency signal of the transformer, and the potential of each current input terminal in both directions of the semiconductor switching section becomes substantially the same potential as or a potential close to the polarization voltage of the constant voltage source of the bias section.

[0016] The actual value of each parasitic capacitance of the series-connected switching elements of the semiconductor switching unit is closely related to the circuit operation. This is because, for example, the resistance of a 100 pF capacitor at 13.56 MHz is only about 117.4 Ω, and the contribution of parasitic capacitance is greater than the contribution of resistance. Here, the parasitic capacitance value of a 100 pF capacitor is a standard value for switching elements with high-end currents.

[0017] The bias section provides the following three significant effects: (a) The first effect is that it reduces the parasitic capacitance of the switching element. The reduction in parasitic capacitance is due to the effect of the bias voltage on the capacitance characteristics of the switching element, and the bias section allows the capacitors of the parasitic capacitance of the switching element to be considered as connected in series, thereby halving the capacitance.

[0018] Effect of bias voltage: Switching elements exhibit capacitance characteristics in which the output capacitance Coss decreases as the drain-source voltage Vds (V) increases. The output capacitance Coss of a transistor is a major component of the parasitic capacitance of the switching element to which it is connected in series, so applying a polarized voltage to the switching element to increase the drain-source voltage Vds (V) significantly reduces the output capacitance Coss.

[0019] Capacitance halving effect: The parasitic capacitance of the switching elements of the bidirectional semiconductor switching unit can be considered equivalent to a series connection of capacitors with parasitic capacitance when viewed from the output side of the transformer, since a high-resistance bias unit is connected between the transformer and the semiconductor switching unit. The bias unit reduces the effective parasitic capacitance of the switching elements of the semiconductor switching unit. For example, when the capacitances of two capacitors are equal, the capacitance is half that of each capacitor.

[0020] (b) As a second effect, the peak value of each switching element of the bidirectional semiconductor switch unit is approximately equal to the polarization voltage of the constant voltage source of the bias unit plus a voltage obtained by dividing the peak value of the high-frequency component at the output end of the transformer in half.

[0021] This is because the parasitic capacitance of each switching element of the bidirectional semiconductor switch section is divided into capacitive components, and because high-frequency components are differentially supplied to each switching element of the bidirectional semiconductor switch section with opposite polarities.

[0022] (c) Furthermore, as a third effect, waveform distortion occurring in each switching element is reduced, and unwanted harmonic components are suppressed.

[0023] The voltage and current of each switching element of the bidirectional semiconductor switch section suffer from significant waveform distortion due to the strong nonlinearity of each parasitic capacitance of each switching element, which causes unwanted harmonic components.

[0024] The bias section reduces the substantial parasitic capacitance of the switching elements due to the bias potential shown in the first effect, thereby reducing waveform distortion occurring in each switching element and suppressing unnecessary harmonic components.

[0025] The switched inductor has a number of configurations depending on the connection topology of the switching element or the constant voltage source.

[0026] First Configuration: The first configuration of the switched inductor comprises a semiconductor switch section made up of two switching elements, and a bias section having one series circuit of a constant voltage source and a resistor.

[0027] The semiconductor switch unit forms a bidirectional switch using a series connection circuit of two switching elements. The two switching elements are connected so that the current flows in opposite directions, and the connection point between the two switching elements is connected to a predetermined potential such as ground.

[0028] The bidirectional switch forms a current path in the secondary coil for current flowing in both directions through the primary coil of the transformer, forming inductance in both directions. The bias unit has a series circuit consisting of a constant voltage source and a resistor, and applies a bias voltage to the current input terminal of one of the two switching elements, and applies the bias voltage to the current input terminal of the other switching element via the secondary coil.

[0029] Second Configuration: The second configuration of the switched inductor comprises a semiconductor switch section made up of two switching elements, and a bias section having two series circuits each consisting of a constant voltage source and a resistor.

[0030] The semiconductor switch unit is similar to the first configuration, and a bidirectional switch is formed by a series connection circuit of two switching elements. The two switching elements are connected so that the current conducts in the opposite directions, and the connection point of the two switching elements is connected to a predetermined potential such as ground.

[0031] The bidirectional switch forms a current path in the secondary coil for current flowing in both directions through the primary coil of the transformer, thereby forming inductance in both directions.

[0032] The bias unit includes two series circuits each consisting of a constant voltage source and a resistor. The low-voltage sides of the two constant voltage sources are connected to the current output terminals of the two switching elements and to a predetermined potential including the ground potential, and the high-voltage sides of the two constant voltage sources are connected via the resistors to the current input terminals of the two switching elements and to the respective ends on the transformer side.

[0033] (Resonant Inverter) The resonant inverter of the present invention includes the switched inductor of the present invention as an inductor element, and changes the resonant frequency by switching the inductance of the switched inductor. Furthermore, the resonant inverter of the present invention applies a bias voltage to the switching element to reduce the output capacitance Coss of the switching element included in the switched inductor, thereby increasing the self-resonant frequency and improving the inductor characteristics, thereby enabling high-frequency operation.

[0034] The resonant inverter of the present invention comprises, as components, a square wave generating section and a resonant circuit to which a square wave signal from the square wave generating section is input.

[0035] The resonant circuit is a series LC resonant circuit or a parallel LC resonant circuit configured by the switched inductor and a capacitor of the present invention. The inductance of the resonant circuit is switched by switching the inductance of the switched inductor.

[0036] The resonant inverter of the present invention further comprises a Q-factor limiting circuit (QFL) as a component. The Q-factor (Quality Factor) is a parameter that indicates the quality of a resonant circuit and is expressed as the reciprocal of the loss factor of the ratio of the inductance according to frequency to the resistance component of the resonant circuit (R / 2πfL), i.e., Q = (1 / R) · √(L / C) / R = 2πfL / R. The higher the Q-factor, the lower the loss, resulting in suitable high-frequency characteristics for high-frequency inductance. On the other hand, a high Q-factor narrows the bandwidth, making the output of the resonant circuit unstable.

[0037] The resonant inverter of the present invention is provided with a Q-factor limiting circuit (QFL) to stabilize the output of the resonant circuit. Furthermore, the QFL of the resonant inverter of the present invention is configured to circulate the output energy of the Q-factor limiting circuit to the supply DC power supply when the voltage component increases excessively, thereby improving energy efficiency.

[0038] (RF Power Supply) The RF power supply of the present invention comprises a DC supply power supply, the resonant inverter of the present invention, and a control unit. The square wave generating unit of the resonant inverter generates a square wave from the DC voltage of the DC supply power supply by opening and closing a switching element.

[0039] The control unit has a first response control mode for controlling the power supply voltage of the DC power supply, and a second response control mode for controlling the resonant inverter.

[0040] In the first response control mode, the DC power supply is controlled based on the square wave voltage of the square wave generating unit. In the first response control mode, the DC power supply is controlled to stabilize the output power, the output voltage, or the output current.

[0041] The second response control mode controls the switching frequency and / or dead time for driving the switching element of the switched inductor based on any one or any combination of the current of the inductive element of the resonant circuit, the voltage of the capacitive element of the resonant circuit, and the time change (dV / dT) of the square wave voltage of the square wave generating unit. The second response control mode stabilizes the output power, output voltage, and output current by controlling the state of the switched inductor.

[0042] The response time τ2 to a change in current in the inductive element of the resonant circuit in the second response control mode is faster than the response time τ1 to a change in square wave voltage of the square wave generating unit in the first response control mode. The RF power supply of the present invention suppresses overvoltages and overcurrents generated in the resonant circuit and / or the load by using the first response control mode and the second response control mode.

[0043] The RF power supply of the present invention includes a zero voltage switching circuit (ZVS) consisting of a series circuit of an inductor and a capacitor between a square wave generating unit and a resonant circuit. The zero voltage switching circuit (ZVS) uses the operating voltage of the LC series circuit to discharge the charging voltage due to the output capacitance of the switching element constituting the square wave generating unit to zero volts, thereby realizing zero volt switching and reducing switching loss of the switching element.

[0044] As described above, the switched inductor, resonant inverter, and RF power supply of the present invention can reduce the output capacitance Coss of the switching element provided in the switched inductor, increase the self-resonant frequency, improve the characteristics as an inductor, and enable high frequency operation in the resonant inverter and RF power supply.

[0045] FIG. 1 is a diagram for explaining a schematic configuration of a switched inductor according to the present invention. FIG. 2 is a diagram for explaining a first configuration example of a switched inductor according to the present invention. FIG. 3 is a diagram for explaining a simulation of a bias voltage Vbias relative to a voltage Vds applied to a switching element. FIG. 4 is a diagram for explaining an operation example of the first configuration example of a switched inductor according to the present invention when the switching elements Q1 and Q2 are off. FIG. 5 is a diagram for explaining an operation example of the first configuration example of a switched inductor according to the present invention when the switching elements Q1 and Q2 are on. FIG. 6 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 7 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 8 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 9 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 10 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 11 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 12 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 13 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 14 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 15 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 16 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 17 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 18 is a diagram for explaining a second configuration example of a switched inductor according to the present invention. FIG. 19 is a diagram for explaining a second configuration example of a switched FIG. 1 is a diagram for explaining a third configuration example of a switched inductor of the present invention. FIG. 2 is a diagram for explaining an example where the midpoint of the semiconductor switch unit is at a predetermined potential. FIG. 3 is a diagram for explaining an example where the midpoint of the semiconductor switch unit is at a predetermined potential. FIG. 4 is a diagram for explaining an example where the midpoint of the semiconductor switch unit is at a predetermined potential. FIG. 5 is a diagram for explaining an example where the midpoint of the semiconductor switch unit is at a predetermined potential. FIG. 6 is a diagram for explaining a schematic configuration of a resonant inverter of the present invention. FIG. 7 is a diagram for explaining an example configuration of a resonant circuit of the present invention. FIG. 8 is a diagram for explaining an example configuration of a resonant circuit of the present invention. FIG. 9 is a diagram for explaining an example configuration of a resonant circuit formed by a series connection of an inductor L and a switched inductor SI. FIG. 10 is a diagram for explaining an example configuration of a resonant circuit formed by a series connection of an inductor L and a switched inductor SI. FIG. 11 is a diagram for explaining an example configuration of a resonant circuit formed by a series connection of an inductor L and a switched inductor SI.FIG. 1 is a diagram illustrating an example configuration of a resonant circuit formed by a series connection of an inductor L and a switched inductor SI. FIG. 2 is a diagram illustrating a first example configuration of an RF power supply of the present invention. FIG. 3 is a diagram illustrating a second example configuration of an RF power supply of the present invention. FIG. 4 is a diagram illustrating a fourth example configuration of an RF power supply of the present invention. FIG. 5 is a diagram illustrating a fifth example configuration of an RF power supply of the present invention.

[0046] (1) Switched Inductor (1-1) General Configuration of Switched Inductor The general configuration of the switched inductor of the present invention will be described with reference to Fig. 1. The switched inductor 1 of the present invention is composed of a transformer 2 and a switch section 3.

[0047] The transformer 2 has a primary coil 2-1 and a secondary coil 2-2, and each end of the primary coil 2-1 is connected to each end SI-1, SI-2 of the switched inductor 1, and each end of the secondary coil 2-2 is connected to each end 3-1, 3-2 of the switch section 3.

[0048] The switch unit 3 includes a semiconductor switch unit 3a, a switch control unit 3b, and a bias unit 3c. The semiconductor switch unit 3a connects and disconnects both ends of the secondary coil 2-2 of the transformer 2, and this connection and disconnection switches the inductance of the switched inductor 1 stepwise.

[0049] When both ends of the secondary coil 2-2 of the transformer 2 are open, the secondary coil 2-2 and the semiconductor switch unit 3a are in an open circuit, so the inductance L of the switched inductor is the inductance L1 of the primary coil 2-1 itself.

[0050] On the other hand, when both ends of the secondary coil 2-2 of the transformer 2 are closed, a current path is formed between the secondary coil 2-2 and the semiconductor switch unit 3a. With this current path formed, the primary coil 2-1 of the transformer 2 is magnetically coupled to the secondary coil 2-2. The inductance of the primary coil 2-1 changes due to the magnetic coupling with the secondary coil 2-2. If the coupling coefficient between the windings of the transformer is represented by κ, the inductance of the primary coil 2-1 changes due to the magnetic coupling with the secondary coil 2-2, as follows: L1·(1−κ ２The coupling coefficient κ is, for example, a value between 0.7 and 1.0, but is not limited to this value.

[0051] Since the inductance L of the switched inductor is the inductance of the primary coil 2-1, the inductance value changes between L1 and L1·(1−κ ２ ) and

[0052] The switch control unit 3b controls the opening and closing operations of the switching elements included in the semiconductor switch unit 3a. The switch control unit 3b controls the opening and closing operations of the switching elements that make up the semiconductor switch unit 3a based on ON / OFF signals input from outside. The bias unit 3c applies a bias voltage to the switching elements of the semiconductor switch unit 3a. The bias unit 3c is composed of a series circuit of a constant voltage source and a resistor.

[0053] The end of the series circuit on the constant voltage source side is connected to ground or a predetermined potential, while the end of the series circuit on the resistor side is connected to the connection point between the secondary coil 2-2 of the transformer 2 and the current input terminal of the switching element. The constant voltage source applies a bias voltage to the switching element via the resistor. The application of the bias voltage to the switching element reduces the output capacitance Coss of the switching element and increases the self-resonant frequency of the switched inductor.

[0054] FIG. 2 shows the capacitance characteristics of a switching element, showing the output capacitance Coss versus the drain-source voltage Vds (V) when the switching element is an FET. The horizontal axis represents the drain-source voltage Vds (V), and the vertical axis represents the capacitance of the output capacitance. The capacitance characteristics in FIG. 2 show that the output capacitance Coss decreases as the drain-source voltage Vds (V) increases, and the output capacitance Coss can be reduced by applying a bias voltage to the switching element to increase the drain-source voltage Vds (V). The bias voltage is, for example, a voltage exceeding 10% of the maximum drain-source voltage, preferably a voltage exceeding 50%.

[0055] 2, the capacitance is approximately 500 pF when no bias voltage is applied, and approximately 60 pF when a bias voltage is applied and the drain-source voltage Vds (V) is 400 V. In this example, the output capacitance Coss when a bias voltage is applied is reduced to approximately 1 / 8 of that when no bias voltage is applied.

[0056] By reducing the output capacitance of the switching element, the self-resonant frequency (SRF) of the switched inductor increases, the frequency range on the high frequency side in which it acts as an inductor is widened, and it can act as an inductor in the high frequency range.

[0057] (1-2) Configuration Examples of Switched Inductors (A) First Configuration Example: A first configuration example of a switched inductor will be described with reference to Figures 3 to 6. Figure 3 shows the first configuration example of a switched inductor.

[0058] The switched inductor 1A includes a transformer 2 and a switch unit 3A. The transformer 2 includes a primary coil 2-1 and a secondary coil 2-2, with the ends of the primary coil 2-1 connected to ends SI-1 and SI-2 of the switched inductor 1A and the ends of the secondary coil 2-2 connected to ends 3-1 and 3-2 of the switch unit 3A.

[0059] The transformer 2 has an inductor L1 in the primary coil 2-1 and an inductor L2 in the secondary coil 2-2, as well as parasitic capacitance and internal resistance. The parasitic capacitances include a capacitance C1 connected in parallel to the primary coil 2-1, a capacitance C2 connected in parallel to the secondary coil 2-2, and parasitic capacitors C3 and C4 between the primary coil 2-1 and the secondary coil 2-2. The internal resistances include a resistance R1 between the end SI-1 and one end of the primary coil 2-1, and a resistance R2 between the end 3-1 of the switch unit 3A and one end of the secondary coil 2-2.

[0060] The switch section 3A includes a semiconductor switch section 3Aa, a switch control section 3Ab, and a bias section 3Ac.

[0061] The semiconductor switch section 3Aa is composed of a series circuit of switching elements Q1 and Q2, with the current input terminal PA of switching element Q1 connected to the bias section 3Ac and terminal 3-1, and the current input terminal PB of switching element Q2 connected to terminal 3-2. The current output terminal of switching element Q1 and the current output terminal of switching element Q2 are connected at a connection point PC, which forms the midpoint of the series circuit of the switching elements and is also connected to ground or a predetermined potential. Figure 3 shows an example in which the connection point PC is grounded.

[0062] When switching element Q1 is in the on state, current flows from the current input terminal PA of switching element Q1 to the current output terminal, and is grounded via connection point PC, forming a current path. When switching element Q2 is in the on state, current flows from the current input terminal PB of switching element Q2 to connection point PC, and is grounded via connection point PC, forming a current path. In a configuration using FETs as switching elements, a bias voltage is applied to the drain terminal of the FET, and current flows from the drain terminal to the source terminal. Because the current directions of switching element Q1 and switching element Q2 are opposite, switching element Q1 and switching element Q2 form a bidirectional switch.

[0063] The switch control unit 3Ab includes a control circuit XQ1 that controls the opening and closing operation of the switching element Q1 and a control circuit XQ2 that controls the opening and closing operation of the switching element Q2. When the switching elements Q1 and Q2 are FETs, the control circuits XQ1 and XQ2 input gate signals to the gates of the switching elements Q1 and Q2. A common control signal is input to the control circuits XQ1 and XQ2, and the switching elements Q1 and Q2 simultaneously perform on / off operations.

[0064] The bias unit 3Ac is composed of a series circuit of a constant voltage source V1DC and a resistor R3. The positive voltage terminal of the constant voltage source V1DC is connected to a current input terminal PA on the input terminal side of the switching element Q1 via the resistor R3, and the constant voltage source V1DC applies a bias voltage V1 to the switching element Q1 via the resistor R3. Meanwhile, the constant voltage source V1DC also applies a bias voltage V1 to the current input terminal PB of the switching element Q2 via the resistor R3 and the secondary coil 2-2. As a result, the bias voltage V1 is applied to both the current input terminal PA of the switching element Q1 and the current input terminal PB of the switching element Q2.

[0065] The resistor R3 of the bias unit 3Ac has a higher resistance than the equivalent parasitic resistance of the transformer 2. By making the resistor R3 of the bias unit 3Ac significantly higher than the equivalent parasitic resistance of the transformer 2, the voltage V1 of the constant voltage source V1DC of the bias unit 3Ac acts at the current input terminal PA as a DC bias potential for the AC voltage of the high frequency signal of the transformer 2. The potentials of the current input terminals PA and PB in both directions of the semiconductor switch unit 3Aa become substantially the same potential as or close to the polarization voltage of the constant voltage source V1DC of the bias unit 3Ac.

[0066] As shown in the capacitance characteristics of Fig. 2, the bias unit 3Ac has the effect of reducing the parasitic capacitance of the switching elements Q1 and Q2 of the bidirectional semiconductor switch unit 3Aa by increasing the drain-source voltage Vds (V) of the switching elements Q1 and Q2 with the bias voltage. Note that the series connection of the parasitic capacitance Coss of the switching elements Q1 and Q2 acts as output capacitance.

[0067] Furthermore, because the high-resistance bias unit 3Ac is connected between the transformer 2 and the semiconductor switch unit 3Aa, the parasitic capacitances C5 and C6 of the switching elements Q1 and Q2 can be considered equivalent to series-connected capacitors having parasitic capacitance when viewed from the output side of the transformer 2. As a result, the bias unit 3Ac reduces the effective parasitic capacitance of the switching elements Q1 and Q2 of the semiconductor switch unit 3Aa. For example, when the two parasitic capacitances C5 and C6 are equal, the capacitance becomes half of each of the parasitic capacitances C5 and C6.

[0068] Therefore, the bias unit 3Ac reduces the output capacitance Coss of the switching elements Q1 and Q2 by applying a bias voltage to the switching elements Q1 and Q2 and connecting the capacitors in series, thereby increasing the self-resonant frequency of the switched inductor 1A.

[0069] A bias voltage V1 from a constant voltage source V1DC is applied to the current input terminal PA of the switching element Q1, along with the voltage VL of the secondary coil 2-2. Here, VL is the voltage generated across both ends of the secondary coil 2-2, V1 is the bias voltage of the constant voltage source V1DC, R1 and R2 are the DC resistances of the coil of the transformer 2, and since their resistance values ​​are in mΩ, their effect on bias operation can be ignored. Furthermore, R3 is the resistance of the bias unit 3Ac. Similar to the current input terminal PA, the bias voltage V1 is also applied to the current input terminal PB of the switching element Q2 through the secondary coil 2-2.

[0070] The actual value of the parasitic capacitance of each of the series-connected switching elements Q1 and Q2 of the semiconductor switch unit 3Aa is closely related to circuit operation. This is because, for example, the resistance of a 100 pF capacitor at 13.56 MHz is only approximately 117.4 Ω, and the contribution of parasitic capacitance is greater than the contribution of resistance. Here, the parasitic capacitance value of a 100 pF capacitor is a standard value for switching elements with high-end currents.

[0071] The effect of the bias voltage Vbias on the drain-source voltage Vds of the switching element is shown using the simulation in Figure 4. Figure 4 shows an example where the bias voltage Vbias is 400 V. In Figure 4, the horizontal axis represents the output voltage Vrf of the high-frequency component generated in the transformer, from zero voltage V0 to Vrf / 2, which is half the peak voltage, and the vertical axis represents the drain-source voltage Vds of the switching element. The solid line in Figure 4 represents the drain-source voltage Vds (= Vrf) when the bias voltage Vbias is not applied, and the dashed line represents the drain-source voltage Vds (= Vbias + Vrf / 2) obtained by simulation when the bias voltage Vbias is applied.

[0072] When the bias voltage Vbias is slightly larger than Vrf / 2, the peak voltage of the drain-source voltage Vds of a single transistor is approximately expressed as (Vbias + Vrf / 2) based on the relationship between the bias voltage Vbias and Vrf / 2. When the bias voltage Vbias=0, the peak voltage of the drain-source voltage Vds is equal to Vrf.

[0073] This is because the resistance of the constant voltage source of the bias section is high and is sufficiently larger than the impedance of the parasitic capacitance of each switching element, and further because the parasitic capacitance of each switching element of the bidirectional semiconductor switch section is capacitively divided and high-frequency components are differentially supplied to each switching element of the bidirectional semiconductor switch section with opposite polarities.

[0074] In the simulation of the drain-source voltage Vds when the bias voltage Vbias is 0 V in FIG. 4, one of the transistors in the bidirectional semiconductor switch unit operates in a diode mode, where it operates as a diode.

[0075] The voltage Vds applied to the current input terminal of the switching element corresponds to the voltage Vq1 in FIG. 3, the bias voltage Vbias corresponds to the bias voltage V1 in FIG. 3, and the output voltage Vrf corresponds to the voltage VL across the secondary coil 2-2 in FIG.

[0076] By setting the bias voltage V1 of the constant voltage source V1DC higher than the voltage (VL / 2) obtained by dividing the voltage VL of the secondary coil 2-2 in half, and by setting the resistance value of the resistor R3 to a resistance value sufficiently larger than the equivalent parasitic resistance of the transformer, the voltage Vq1 applied to the switching element Q1 is determined by the voltage (VL / 2) obtained by dividing the voltage VL of the secondary coil 2-2 in half and the bias voltage V1. The same applies to the current input terminal PB of the switching element Q2.

[0077] This indicates that when the transformer side voltage is in the low voltage range, the bias voltage Vbias is effectively applied to the switching element, and application of an excessive voltage to the switching element is avoided.

[0078] Furthermore, the bias section has the effect of reducing waveform distortion occurring in each switching element and suppressing unnecessary harmonic components. The voltage and current of each switching element in the bidirectional semiconductor switch section suffer from significant waveform distortion due to the strong nonlinearity of the parasitic capacitance of each switching element, which causes unnecessary harmonic components. The bias section reduces the effective parasitic capacitance of the switching elements due to the bias potential, thereby reducing waveform distortion occurring in each switching element and suppressing unnecessary harmonic components.

[0079] 5A and 5B show examples of operation when the switching elements Q1 and Q2 are turned off and on, respectively. Fig. 5A shows an example of operation when both the switching elements Q1 and Q2 of the semiconductor switch unit 3Aa are in the off state.

[0080] When both switching elements Q1 and Q2 are in the off state, the terminals 3-1 and 3-2 of the switch unit 3A are in an open state, and an open circuit is formed between both ends of the secondary coil 2-2. As a result, the inductance L of the switched inductor 1A is the inductance of the inductor L1 of the primary coil 2-1.

[0081] 5B and 5C show an example of operation when both the switching elements Q1 and Q2 of the semiconductor switch section 3Aa are in the ON state.

[0082] 5B shows a case where the voltage on the terminal 3-1 side of the switch unit 3A is higher than that on the terminal 3-2 side of the switch unit 3A. When the switching elements Q1 and Q2 are switched from the OFF state to the ON state, the bias voltage V1 of the constant voltage source V1DC is applied by the bias unit 3Ac to the current input terminal PA of the switching element Q1, and the bias voltage V1 of the constant voltage source V1DC of the bias unit 3Ac is applied to the current input terminal PB of the switching element Q2 via the secondary coil 2-2. As a result, the output capacitance Coss of the switching element Q1 is reduced compared to when no bias voltage is applied, and the self-resonant frequency (SRF) of the switched inductor is increased.

[0083] In the conductive switching elements Q1 and Q2, a current flows in both the drain-to-source direction and the source-to-drain direction. When the voltage at the end 3-1 of the switch unit 3A is higher than that at the end 3-2 of the switch unit 3A, a current path is formed for the current IA to flow from the end 3-1 through the conductive switching elements Q1 and Q2 to the end 3-2, and a current path is formed between both ends of the secondary coil 2-2.

[0084] When a current path is formed between both ends of the secondary coil 2-2, mutual coupling occurs between the primary coil 2-1 and the secondary coil 2-2, and the inductance of the primary coil 2-1 becomes L1·(1−κ ２ ) The inductance L of the switched inductor 1A is expressed by the inductance of the primary coil, so the inductance of the switched inductor 1A is L1·(1−κ ２ )

[0085] Operation Example of FIG. 5C: FIG. 5C shows an operation example when the voltage on the end 3-2 side of the switch section 3A is higher than that on the end 3-1 side of the switch section 3A, and the voltage direction is opposite to that of the operation example of FIG. 5B.

[0086] When the voltage on the end 3-2 side of the switch section 3A is higher than that on the end 3-1 side of the switch section 3A, a current path is formed for current IB to flow from the end 3-2 through the conductive switching element Q2 and switching element Q1 to the end 3-1.

[0087] As in the operation example of FIG. 5B, when a current path is formed between both ends of the secondary coil 2-2, mutual coupling occurs between the primary coil 2-1 and the secondary coil 2-2, and the inductance of the primary coil 2-1 becomes L1·(1−κ ２ ) The inductance of the switched inductor 1A is expressed by the inductance of the primary coil, so the inductance of the switched inductor 1A is L1·(1−κ ２ )

[0088] Fig. 6 shows the voltages and currents in the operation example of Fig. 5. Q1 and Q2 indicate the on / off states of the switching elements Q1 and Q2, VA indicates the potential at the current input terminal PA, VB indicates the potential at the current input terminal PB, and IA and IB indicate the currents at the current input terminals PA and PB.

[0089] VA and VB are in opposite phase to each other, and when the switching elements Q1 and Q2 are in the off state, the bias voltage Vbias increases, and when they are in the on state, voltages Va and Vb are generated due to the on resistance of the switching elements. Also, when the switching elements Q1 and Q2 are in the off state, IA and IB are currents due to the capacitance of the output capacitance Coss of the switching elements, and are 90° ahead in phase with the voltages VA and VB.

[0090] When the switching elements Q1 and Q2 are in the on state, the currents IA and IB flowing in opposite directions alternately according to the current direction of the primary coil 2-1, and mutual coupling occurs between the primary coil 2-1 and the secondary coil 2-2. As a result, the inductance of the primary coil 2-1 and, in turn, the inductance of the switched inductor 1A becomes L1·(1−κ ２ ) will be displayed.

[0091] (B) Second Configuration Example: A second configuration example of the switched inductor will be described with reference to Figures 7 to 9. Figures 7A and 7B show the second configuration example of the switched inductor.

[0092] The switched inductor 1B includes a transformer 2 and a switch unit 3B. The transformer 2 is the same as in the first example configuration, and therefore a description thereof will be omitted here.

[0093] The switch unit 3B includes a semiconductor switch unit 3Ba, a switch control unit 3Bb, and a bias unit 3Bc. The semiconductor switch unit 3Ba is configured as a series circuit of a switching element Q1 and a switching element Q2, with the current input terminal PA of the switching element Q1 connected to the bias unit 3Bc and terminal 3-1, and the current input terminal PB of the switching element Q2 connected to terminal 3-2. The current output terminal of the switching element Q1 and the current output terminal of the switching element Q2 are connected at a connection point PC, which forms the midpoint of the series circuit of the switching elements and is also connected to ground or a predetermined potential. Figures 7A and 7B show an example in which the connection point PC is grounded.

[0094] In an FET in the on state, current flows in both the drain-to-source direction and the source-to-drain direction. When switching element Q1 and switching element Q2 are in the on state, if end 3-1 has a higher voltage than end 3-2, current flows from end 3-1 through switching element Q1 to node PC, and then flows through switching element Q2 to end 3-2, forming a current path from end 3-1 to end 3-2. On the other hand, if end 3-2 has a higher voltage than end 3-1, current flows from end 3-2 through switching element Q2 to node PC, and then flows through switching element Q1 to end 3-1, forming a current path from end 3-2 to end 3-1.

[0095] In a configuration using FETs as switching elements, switching element Q1 and switching element Q2 form a bidirectional switch that forms a current path with a reversed current direction even when the voltage direction between terminal 3-1 and terminal 3-2 is reversed.

[0096] The switch control unit 3Bb includes a control circuit XQ1 that controls the opening and closing operation of the switching element Q1 and a control circuit XQ2 that controls the opening and closing operation of the switching element Q2. When the switching elements Q1 and Q2 are FETs, the control circuits XQ1 and XQ2 input gate signals to the gates of the switching elements Q1 and Q2. Because a common control signal is input to the control circuits XQ1 and XQ2, the switching elements Q1 and Q2 simultaneously perform on / off operations.

[0097] The bias unit 3Bc is composed of two series circuits: a series circuit of a constant voltage source V1DC and a resistor R3, and a series circuit of a constant voltage source V2DC and a resistor R4. The positive voltage terminal of the constant voltage source V1DC is connected to the current input terminal PA on the input terminal side of the switching element Q1 via the resistor R3, and the constant voltage source V1DC applies a bias voltage to the switching element Q1 via the resistor R3. The application of the bias voltage to the switching element Q1 reduces the output capacitance Coss of the switching element Q1 and increases the self-resonant frequency of the switched inductor 1A.

[0098] On the other hand, the positive voltage terminal of the constant voltage source V2DC is connected to the current input terminal PB on the input terminal side of the switching element Q2 via a resistor R4, and the constant voltage source V2DC applies a bias voltage to the switching element Q2 via the resistor R4. The application of the bias voltage to the switching element Q2 reduces the output capacitance Coss of the switching element Q2 and increases the self-resonant frequency of the switched inductor 1A.

[0099] While the bias unit 3Ac in the first configuration example is configured only with a series circuit of a constant voltage source V1DC and a resistor R3, the bias unit 3Bc in the second configuration example shown in Fig. 7A may be configured to include two series circuits each consisting of a constant voltage source V2DC and a resistor R4. The bias voltage can be applied to the switching element Q2 by a single series circuit of a constant voltage source V1DC and a resistor R3 as in the first configuration example, or by two series circuits, one consisting of a constant voltage source V1DC and a resistor R3 and the other consisting of a constant voltage source V2DC and a resistor R4 as in the second configuration example.

[0100] Furthermore, in the second configuration example, the constant voltage source V1DC and the constant voltage source V2DC can be configured as a single constant voltage source V1DC, as shown in Fig. 7B. In the configuration example shown in Fig. 7B, the constant voltage source V1DC is connected between the connection point of the switching element Q1 and the switching element Q2 and the connection point of the resistor R3 and the resistor R4.

[0101] In addition, in order to prevent a DC short circuit between the end 3-1 and the end 3-2 via the coil L2, a capacitor C is connected between the end 3-1 of the bias section 3Bc and the resistor R3, and between the end 3-2 of the bias section 3Bc and the resistor R4.

[0102] By setting the bias voltage V1 of the constant voltage source V1DC to a voltage that exceeds the voltage (VL / 2) obtained by dividing the voltage VL across the secondary coil 2-2 in half, and by setting the resistance values ​​of the resistors R3 and R4 to a resistance value that is sufficiently larger than the equivalent parasitic resistance of the transformer, the voltage Vq1 applied to the switching element Q1 is determined by the voltage (VL / 2) obtained by dividing the voltage VL across the secondary coil 2-2 in half and the bias voltage V1.

[0103] This indicates that when the transformer side voltage is in the low voltage range, the bias voltage Vbias is effectively applied to the switching element, and application of an excessive voltage to the switching element is avoided.

[0104] 8A, 8B, and 8C show examples of operation when the switching elements Q1 and Q2 are turned off and on. Fig. 8A shows an example of operation when both the switching elements Q1 and Q2 included in the semiconductor switch unit 3Ba are in the off state.

[0105] When both switching elements Q1 and Q2 are in the off state, the terminals 3-1 and 3-2 of the switch unit 3B are in an open state, and an open circuit is formed between both ends of the secondary coil 2-2. As a result, the inductance of the switched inductor 1A is the inductance of the inductor L1 of the primary coil 2-1.

[0106] 8B and 8C show an example of operation when both the switching elements Q1 and Q2 of the semiconductor switch section 3Ba are in the ON state.

[0107] 8B shows a case where the voltage on the end 3-1 side of the switch unit 3B is higher than that on the end 3-2 side of the switch unit 3B. When the switching elements Q1 and Q2 are switched from the OFF state to the ON state, the bias voltage V1 of the constant voltage source V1DC is applied by the bias unit 3Bc to the current input terminal PA of the switching element Q1, and the bias voltage V2 of the constant voltage source V2DC is applied by the bias unit 3Bc to the current input terminal PB of the switching element Q2. As a result, the output capacitance Coss of the switching elements Q1 and Q2 is reduced compared to when no bias voltage is applied, and the self-resonant frequency (SRF) of the switched inductor is increased.

[0108] When the voltage at end 3-1 of switch section 3B is higher than that at end 3-2 of switch section 3B, current IA flows from end 3-1 through conductive switching element Q1, connection point PC, and conductive switching element Q2 to end 3-2, forming a current path between both ends of secondary coil 2-2.

[0109] When a current path is formed between both ends of the secondary coil 2-2, mutual coupling occurs between the primary coil 2-1 and the secondary coil 2-2, and the inductance of the primary coil 2-1 becomes L1·(1−κ ２ ) The inductance of the switched inductor 1A is expressed by the inductance of the primary coil, so the inductance of the switched inductor 1A is L1·(1−κ ２ )

[0110] Operation example of FIG. 8C: FIG. 8C shows an operation example when the voltage on the end 3-2 side of the switch section 3A is higher than that on the end 3-1 side of the switch section 3A, and the voltage direction and current direction are opposite to those in the operation example of FIG. 8B.

[0111] When the voltage at end 3-2 is higher than that at end 3-1, current IB flows from end 3-2 through the conductive switching element Q2, connection point PC, and the conductive switching element Q1 toward end 3-1, forming a current path between both ends of the secondary coil 2-2.

[0112] As in the example of operation shown in FIG. 8B, when a current path is formed between both ends of the secondary coil 2-2, mutual coupling occurs between the primary coil 2-1 and the secondary coil 2-2, and the inductance of the primary coil 2-1 becomes L1·(1−κ ２ ) The inductance of the switched inductor 1A is expressed by the inductance of the primary coil, so the inductance of the switched inductor 1A is L1·(1−κ ２ )

[0113] Fig. 9 shows the voltages and currents in the operation example of Fig. 7. Q1 and Q2 indicate the on / off states of the switching elements Q1 and Q2, VA indicates the potential at the current input terminal PA, VB indicates the potential at the current input terminal PB, and IA and IB indicate the respective currents.

[0114] 9 are similar to those shown in Fig. 6, with VA and VB being in opposite phase to each other, and when the switching elements Q1 and Q2 are in the off state, the bias voltage Vbias increases, and when they are in the on state, voltages Va and Vb are generated due to the on-resistance of the switching elements. Also, when the switching elements Q1 and Q2 are in the off state, IA and IB are currents due to the capacitance of the output capacitance Coss of the switching elements, and are 90° ahead in phase with voltages VA and VB.

[0115] When the switching elements Q1 and Q2 are in the on state, the currents IA and IB flow in opposite directions alternately, causing mutual coupling between the primary coil 2-1 and the secondary coil 2-2, and the inductance of the primary coil 2-1, and therefore the inductance of the switched inductor 1A, becomes L1·(1−κ ２ ) will be displayed.

[0116] (D) Third Configuration Example: A third configuration example of the switched inductor will be described with reference to Fig. 10. A switched inductor 1C of the third configuration example has a configuration in which protective resistors R5 and R6 are connected in front of the semiconductor switch unit in the first configuration example.

[0117] The switched inductor 1C is similar to the switched inductor 1A of the first configuration example except for the protective resistors R5 and R6, so only the protective resistors R5 and R6 will be described here, and a description of the other configurations will be omitted.

[0118] A protective resistor R5 is connected between the end of resistor R3 opposite to constant voltage source V1DC and current input terminal PA, and a protective resistor R6 is connected between terminal 3-2 and current input terminal PB of switching element Q2. Protective resistors R5 and R6 protect switching elements Q1 and Q2 from surge pulse voltages generated in the secondary coil of transformer 2.

[0119] (E) Midpoint potential between switching elements Q1 and Q2 In the first to third configuration examples of the switched inductor, the midpoint potential of the connection point between switching element Q1 and switching element Q2 can be set to zero potential due to grounding or a predetermined potential. Since zero potential due to grounding is shown in each of the first to third configuration examples, an example of a predetermined potential will be described in Figure 11.

[0120] 11A and 11B show examples in which the midpoint potential of the connection point PC between the switching elements Q1 and Q2 is a predetermined potential Vc in the first and third configuration examples of the switched inductor.

[0121] 11A shows the current flow when the current input terminal PA of switching element Q1 is at a high voltage. When terminal 3-1 of switch unit 3 is at voltage VL relative to terminal 3-2, current IA flows from terminal 3-1 through switching elements Q1 and Q2 in a conducting state to terminal 3-2.

[0122] 11B shows the current flow when the current input terminal PB of switching element Q2 is at a high voltage. When terminal 3-2 of switch unit 3 is at voltage VL relative to terminal 3-1, current IB flows from terminal 3-2 through switching elements Q2 and Q1 in a conducting state to terminal 3-1.

[0123] In this example, in the switching elements Q1 and Q2 that constitute the bidirectional switch, the potential of the secondary coil 2-2 is floating, so current flows regardless of the magnitude of the predetermined potential Vc. Also, since the conduction condition of the switching elements is that the gate-source voltage Vgs of the switching elements Q1 and Q2 is greater than the threshold voltage Vth, when the gate-source voltage Vgs is the gate voltage Vg plus the predetermined potential Vc of the connection point PC (Vgs + Vc), the predetermined potential Vc is required to be a potential that satisfies (Vc > Vth - Vg).

[0124] 11C and 11D show an example in which the midpoint potential of the connection point PC between the switching elements Q1 and Q2 is a predetermined potential Vc in the second configuration example of the switched inductor.

[0125] 11C shows the current flow when the current input terminal PA of switching element Q1 is at a high voltage. When terminal 3-1 of switch unit 3 is at voltage VL relative to terminal 3-2, current IA flows from terminal 3-1 through switching elements Q1 and Q2 in a conductive state to terminal 3-2.

[0126] 11D shows the current flow when the current input terminal PB of switching element Q2 is at a high voltage. When the terminal 3-2 of switch unit 3 is at voltage VL relative to the terminal 3-1, current IB flows from terminal 3-2 to terminal 3-1 through the conducting switching elements Q2 and Q1.

[0127] In this example, in the switching elements Q1 and Q2 that constitute the bidirectional switch, the potential of the secondary coil 2-2 is floating, so that current flows regardless of the magnitude of the predetermined potential Vc. The conduction conditions of the switching elements require that the predetermined potential Vc be a potential that satisfies (Vc>Vth-Vg).

[0128] (2) Resonant Inverter (2-1) Schematic Configuration of Resonant Inverter The schematic configuration of the resonant inverter of the present invention will be described with reference to Fig. 12. The resonant inverter 10 of the present invention comprises a square wave generating unit 11 that generates a square wave from the DC voltage of a DC power supply 20, and a resonant circuit 12 that generates an AC signal from the square wave input from the square wave generating unit 11. The resonant circuit 12 comprises a switched inductor 1 of the present invention.

[0129] The resonant circuit 12 is composed of the switched inductor 1 of the present invention and a capacitor, and by switching the inductance of the switched inductor 1, the inductance of the resonant circuit 12 is switched in stages to change the resonant frequency.

[0130] Furthermore, the switched inductor 1 of the resonant inverter 10 of the present invention applies a bias voltage to the switching element, thereby reducing the output capacitance Coss of the switching element and increasing the self-resonant frequency, thereby improving the characteristics as an inductor and enabling high-frequency operation.

[0131] The resonant inverter 10 of the present invention further includes a Q-factor limiting circuit (QFL) 13. The Q-factor (Quality Factor) is a parameter that indicates the quality of an inductor, and is expressed as the reciprocal of the loss factor (Q = 2πfL / R) of the ratio of the inductance according to frequency to the inductor's resistance component (R / 2πfL). The higher the Q-factor, the less loss there is, resulting in high frequency characteristics suitable for high-frequency inductance. However, a high Q-factor narrows the bandwidth, making the output of the resonant circuit unstable.

[0132] The resonant inverter 10 of the present invention stabilizes the output of the resonant circuit 12 by providing a Q-factor limiting circuit (QFL) 13 in the resonant circuit 12. Furthermore, energy efficiency can be improved by circulating power from the resonant circuit 12 to the DC power supply.

[0133] The resonant inverter 10 of the present invention includes a control unit 30 that controls the square wave generating unit 11 , the resonant circuit 12 , and the switched inductor 1 .

[0134] The resonant inverter 10 of the present invention, together with the DC power supply 20 and the control unit 30, constitutes an RF power supply 100, which outputs a high-frequency signal generated in the resonant circuit 12. The high-frequency signal output from the RF power supply 100 is supplied to a load 300 via an output circuit 200. The output circuit 200 includes a directional coupler 201 that separates forward power directed toward the load 300 from reflected power reflected back from the load 300, and a matching device 202 that matches impedances between the RF power supply 100 and the load 300.

[0135] (2-2) Configuration Examples of Resonant Circuits Configuration examples of the resonant circuit of the present invention will be described with reference to Figures 13 and 14. The resonant circuit of the present invention can be configured by applying it to a series LC resonant circuit or a parallel LC resonant circuit.

[0136] 13A shows an example of application to a series LC resonant circuit. A resonant circuit 12s using a series LC resonant circuit is configured by connecting a switched inductor SI in series to an inductor L in a series circuit of an inductor L and a capacitor C.

[0137] 13B shows an example in which the present invention is applied to a parallel LC resonant circuit. A resonant circuit 12p using a parallel LC resonant circuit is configured by connecting a switched inductor SI in series with an inductor L in a parallel circuit of the inductor L and a capacitor C connected in parallel with a load. The switched inductor SI may be configured by connecting one switched inductor SI1 in series, or by connecting a parallel circuit of a switched inductor SI1 and a switched inductor SI2 in series.

[0138] FIG. 14 shows an example of the configuration of a resonant circuit made up of a series connection of an inductor L and a switched inductor SI.

[0139] 14A and 14B show examples of a configuration in which a switched inductor SI1 is connected between an inductor L10 and a capacitor C10 in a series LC resonant circuit of the inductor L10 and the capacitor C10. A resonant circuit 12A shown in Fig. 14A is an example of a configuration in which a parallel circuit of a capacitor C11 and a resistor R1 is connected in series to the capacitor C10, and a resonant circuit 12B shown in Fig. 14B is an example of a configuration in which a parallel circuit made of a capacitor C11, an inductor L11, and a resistor R1 is connected in series to the capacitor C10.

[0140] 14C is a parallel LC resonant circuit of an inductor L10 and a capacitor C10, and shows a configuration example in which a switched inductor SI1 is connected between the inductor L10 and the capacitor C10. In this circuit example, a parallel circuit of a capacitor C11 and a resistor R1 is connected to the connection point between the capacitor C10 and the switched inductor SI1.

[0141] A resonant circuit 12D shown in FIG. 14D has a configuration in which a parallel circuit of a capacitor C11, a resistor R1, and an inductor is connected in series to a series circuit of an inductor L10 and a capacitor C10, and the inductor of the parallel circuit is formed by a series circuit of a switched inductor SI1 and an inductor L11.

[0142] (3) RF Power Supply (3-1) Schematic Configuration of RF Power Supply The schematic configuration of the RF power supply 100 of the present invention is composed of a resonant inverter 10, a DC power supply 20, and a control unit 30 as shown in FIG. 12, and outputs a high-frequency signal obtained by resonating the DC voltage of the DC power supply 20 with the resonant circuit of the resonant inverter 10.

[0143] (3-2) Configuration Examples of RF Power Supply First Configuration Example: A first configuration example of the RF power supply will be described with reference to Fig. 15. An RF power supply 100A of the first configuration example includes a resonant inverter 10A, a DC power supply 20, and a control unit 30.

[0144] The resonant inverter 10A includes a square wave generating unit 11, a resonant circuit 12A, and a zero voltage switching (ZVS) circuit 14. The square wave generating unit 11 is configured as a half-bridge circuit of a switching element S1 and a switching element S2. One end of the switching element S1 is connected to a DC power supply 20, and one end of the switching element S2 is connected to a base, forming a closed circuit with a load 300. The midpoint of the half-bridge circuit, which is the connection point between the switching elements S1 and S2, is connected to one end of the resonant circuit 12A. The square wave generating unit 11 converts the DC voltage VCC of the DC power supply 20 into a square wave and supplies the square wave to the resonant circuit 12A. Note that C104 and C105 are parasitic capacitances of the switching elements S1 and S2.

[0145] The opening and closing operations of the switching elements S1 and S2 are controlled by gate signals GC1 and GC2 from the control unit 30. The switching elements S1 and S2 are alternately turned on and off by the gate signals GC1 and GC2, and convert the DC voltage VCC into a rectangular wave.

[0146] The resonant circuit 12A is composed of an LC series circuit of an inductor L3, a switched inductor SI1, and a capacitor C101, and converts the square wave input from the square wave generating unit 11 into a high-frequency signal through a resonant operation, and outputs the signal from the midpoint of the series circuit of the series-connected output capacitors C102A and C102B to the load 300 via the capacitor C107. Note that the output capacitor C102 is C102 = C102A + C102B. Here, an example of an inductive load including a resistance component and an inductance component is shown as the load 300.

[0147] The switched inductor SI1 is composed of an inductor L1 that forms a primary coil of a transformer, an inductor L2 that forms a secondary coil, and a switch SW1. The switch SW1 is composed of the switch unit 3 of the present invention.

[0148] The zero voltage switching circuit (ZVS) 14 is composed of an LC series circuit of an inductor L4 and a capacitor C103, and of the two ends of the series circuit, one end of the inductor L4 is connected to the midpoint of the half bridge circuit of the square wave generating unit 11, and one end of the capacitor C103 is connected to the base. The zero voltage switching circuit (ZVS) 14 discharges the charged voltage due to the parasitic capacitance of the switches SW1, SW1 that make up the square wave generating unit 11 to zero volts using the operating voltage of the LC series circuit, thereby achieving zero volt switching and reducing switching loss in the switching elements.

[0149] The control unit 30 measures the voltage and current of the resonant inverter 10A and controls the resonant inverter 10A via the response circuit 40. The control unit 30 has a first response control mode for controlling the DC voltage VCC of the DC power supply 20, which is the supply DC power source, and a second response control mode for controlling the resonant inverter 10A.

[0150] In the first response control mode, the DC power supply 20 is controlled based on the rectangular wave voltage of the rectangular wave generating unit 11. In the first response control mode, the DC power supply 20 is controlled to stabilize the output power, output voltage, or output current.

[0151] The second response control mode controls the switching frequency and / or dead time for driving the switching element of the switched inductor SI1 based on any one or any combination of the current Irt1 flowing through the inductive element of the resonant circuit 12A, the voltage Vrt1 of the capacitive element of the resonant circuit 12A, and the time change (dV / dT) of the rectangular wave voltage of the rectangular wave generating unit 11, thereby controlling the state of the switched inductor SI1 and stabilizing the output power, output voltage, and output current of the RF power supply 100A.

[0152] The response time τ1 of the square wave generating unit to a change in square wave voltage in the first response control mode is faster than the response time τ2 of the inductive element of the resonant circuit to a change in current in the second response control mode. The RF power supply of the present invention suppresses overvoltages and overcurrents generated in the resonant circuit and / or the load by using the first response control mode and the second response control mode.

[0153] First Response Control Mode: The response circuit 40 includes a voltage monitoring circuit 41monV and a slow response circuit 41con as circuits that perform the first response control mode.

[0154] The voltage monitoring circuit 41monV detects the output voltage of the DC power supply 20. The control unit 30 generates a control signal for controlling the DC voltage VCC of the DC power supply 20 based on the voltage of the DC power supply 20 monitored by the voltage monitoring circuit 41monV. The slow response circuit 41con controls the DC power supply 20 based on the control signal generated by the control unit 30.

[0155] Second response control mode: The response circuit 40 is equipped with the following monitoring circuits for performing the second response control mode: a current monitoring circuit 42monC that monitors the inductive element of the resonant circuit 12A, a voltage monitoring circuit 42monV that monitors the voltage Vrt1 of the capacitive element of the resonant circuit 12A, a voltage change monitoring circuit 42mondV that monitors the time change (dV / dT) of the rectangular wave voltage of the rectangular wave generating unit 11, and a high-speed response circuit 42con.

[0156] The control unit 30 generates a control signal to control the switched inductor SI1 based on the current Irt1 of the resonant circuit 12A monitored by the current monitoring circuit 42monC, the voltage Vrt1 of the resonant circuit 12A monitored by the voltage monitoring circuit 42monV, and the time change (dV / dT) of the rectangular wave voltage of the rectangular wave generating unit 11 monitored by the voltage change monitoring circuit 42mondV.

[0157] The high-speed response circuit 42con controls the switched inductor SI1 based on a control signal generated by the control unit 30. The control of the switched inductor SI1 involves adjusting the impedance by switching the inductor by switching the switch SW1, as well as controlling the switching frequency and / or dead time that drive the switching element of the switched inductor SI1. The second response control mode stabilizes the output power, output voltage, and output current by controlling the operating state of the switched inductor.

[0158] Second Configuration Example: A second configuration example of the RF power supply will be described with reference to Fig. 16. The RF power supply 100B of the second configuration example includes a resonant inverter 10B, a DC power supply 20, and a control unit 30. The resonant circuit 12B of the resonant inverter 10B includes a switched inductor SI1. The configurations of the control unit 30 and the response circuit 40 are similar to those of the RF power supply 100B of the first configuration example, and therefore will not be described here.

[0159] The RF power supply 100B includes the same components as the RF power supply 100A, and further includes a Q-factor limiting circuit (QFL) 13 in a resonant circuit 12B.

[0160] The Q-factor limiting circuit (QFL) 13 is connected between the base and the midpoint of the series circuit of output capacitors C102A and C102B connected to the output side of the resonant circuit 12B, via the capacitor C106. The Q-factor limiting circuit (QFL) 13 is connected in parallel with the output capacitor C102B, which is a capacitive reactance element on the output side of the resonant circuit 12B. The Q-factor limiting circuit (QFL) 13 is composed of a transformer TQFL1, a rectifier circuit PD1, and an inductor L7. The primary coil L5 of the transformer TQFL1 is connected between the capacitor C106 and the base, and the secondary coil L6 is connected to the input side of the rectifier circuit PD1. One output end of the rectifier circuit PD1 is connected to the DC power supply 20 via the inductor L7, and the other end is connected to the base.

[0161] The primary coil L5 and capacitor C106 form a series resonant circuit, and a Q factor limiting circuit (QFL) 13 circulates the voltage induced in the primary coil L5 of the transformer TQFL1 to the DC power supply 20, which is the supply DC power source. The Q factor limiting circuit (QFL) 13 reduces the Q value of the resonant circuit 12B and widens the half-value width, thereby suppressing excessive oscillation of the resonant circuit 12B, and improves energy efficiency by circulating power from the resonant circuit to the DC power supply 20, which is the supply DC power source.

[0162] Third Configuration Example: A third configuration example of an RF power supply will be described with reference to Fig. 17. An RF power supply 100C of the third configuration example has a configuration substantially similar to that of the second configuration example, and includes a resonant inverter 10C, a DC power supply 20, and a control unit 30, and a resonant circuit 12C of the resonant inverter 10C includes a switched inductor SI1. The configurations of the control unit 30 and the response circuit 40 are similar to those of the RF power supplies 100B and 100C of the first and second configuration examples, and descriptions of these parts will be omitted.

[0163] In the first and second configuration examples, the voltage monitoring circuit 42monV monitors the voltage Vrt1 of the capacitive element of the resonant circuit 12A, and the high-speed response circuit 42con uses this voltage Vrt1 as one of the control signals for the switched inductor SI1.

[0164] In the third configuration example, the current Iqfl1 flowing through the rectifier circuit PD1 of the Q-factor limiting circuit (QFL) 13 is monitored by a current monitoring circuit 42monC1 instead of the voltage Vrt1 of the resonant circuit 12A.

[0165] The current value of the current Iqfl1 flowing through the rectifier circuit PD1 of the Q factor limiting circuit (QFL) 13 is a value corresponding to the voltage generated in the primary coil L5 of the transformer TQFL1 of the Q factor limiting circuit (QFL) 13, and the voltage generated in the primary coil L5 is a value corresponding to the output voltage of the RF power supply 100C. Therefore, the output voltage of the RF power supply 100C can be monitored based on the current value of the current Iqfl1 instead of the voltage Vrt1 of the resonant circuit 12A.

[0166] Fourth Configuration Example: A fourth configuration example of an RF power supply will be described with reference to Fig. 18. An RF power supply 100D of the fourth configuration example includes two switched inductors SI1 and SI2 in which a resonant circuit 12D of a resonant inverter 10D is connected in series, and high-speed response circuits 42con1 and 42con2 that control the switched inductors SI1 and SI2. The other configurations are the same as those of the second configuration example.

[0167] The high-speed response circuits 42con1 and 42con2 control the switched inductors SI1 and SI2, respectively, and control the switching of the inductors, the switching cycle of the switching elements, and the dead time.

[0168] Fifth Configuration Example: A fifth configuration example of the RF power supply will be described with reference to Fig. 19. An RF power supply 100E of the fifth configuration example includes two switched inductors SI1 and SI2 connected in parallel to a resonant circuit 12E of a resonant inverter 10E, and high-speed response circuits 42con1 and 42con2 that control the switched inductors SI1 and SI2. The other configurations are the same as those of the second configuration example.

[0169] The high-speed response circuits 42con1 and 42con2 control the switched inductors SI1 and SI2, respectively, and control the switching of the inductors, the switching cycle of the switching elements, and the dead time.

[0170] The switched inductor of the present invention can be applied to a resonant circuit with a variable inductor, and the resonant inverter of the present invention can be applied to an inverter circuit with a variable inductor. The RF power supply of the present invention can be used in semiconductor manufacturing equipment, flat panel display manufacturing equipment such as liquid crystal panels and organic EL panels, solar panel manufacturing equipment, CO ２ The present invention can be applied to a DC power supply device used as a high frequency power supply that outputs high frequency waves in high frequency power supplies used in industrial applications such as laser processing machines.

[0171] 1, 1A, 1B, 1C Switched inductor 2 Transformer 2-1 Primary coil 2-2 Secondary coil 3, 3A, 3B, 3C Switch section 3-1, 3-2 End section 3Aa, 3Ba, 3Ca, 3a Semiconductor switch section 3Ab, 3Bb, 3Cb, 3b Switch control section 3Ac, 3Bc, 3Cc, 3c Bias section L4 Inductor 10, 10A, 10B, 10C, 10D, 10E Resonant inverter 11 Square wave generating section 12, 12A, 12B, 12C, 12D, 12E, 12p, 12s Resonant circuit 20 DC power supply 30 Control section 40 Response circuit 41con Slow response circuit 41monV Voltage monitoring circuit 42con, 42con1, 42con2 High-speed response circuit 42monC Current monitoring circuit 42monC1 Current monitoring circuit 42monV Voltage monitoring circuit 42mondV Voltage change monitoring circuit 100 RF power supply 100A, 100B, 100C, 100D, 100E RF power supply 200 Output circuit 202 Matching box 300 Load C, C1, C3, C4, C5, C6, C10, C11 Capacitors C101, C102, C103, C104, C105, C106, C107 Capacitors C102A, C102B Output capacitor Coss Output capacitance D1, D2 Diodes GC1, GC2 Gate signals IA, IB, Iqfl1, Irt1 Current L, L1, L2, L3, L7, L10, L11 Inductor L5 Primary coil L6 Secondary coil PA Current input terminal PA, PB Current input terminal PC Connection point PD1 Rectifier circuit Q, Q1, Q2, Q1a, Q1b Switching elements R1, R2, R3, R4 Resistors R5, R6 Protection resistors S1, S2 Switching elements SI, SI1, SI2 Switched inductors SI-1, SI-2 Ends SW1, SW2 Switches TQFL1 Transformers V1, V2 Bias voltages V1DC, V2DC Constant voltage sources XQ1, XQ2 Control circuit κ Coupling coefficient τ1 Response time τ2 Response time

Claims

1. A switched inductor comprising a transformer and a switch unit, wherein each end of the primary coil of the transformer is each end of the switched inductor, and each end of the secondary coil of the transformer is connected to each end of the switch unit, and the switch unit comprises: a semiconductor switch unit that disconnects both ends of the secondary coil of the transformer and switches the inductance of the inductor of the primary coil via the magnetic coupling of the transformer; a switch control unit that controls the opening and closing operation of a switching element of the semiconductor switch unit; and a bias unit that applies a bias voltage to the switching element of the semiconductor switch unit, and the bias unit is composed of a series circuit of a constant voltage source and a resistor, and applies a bias voltage to the switching element via the resistor, thereby reducing the output capacitance of the switching element and increasing the self-resonant frequency of the switched inductor.

2. The switched inductor according to claim 1, wherein the semiconductor switch section is a bidirectional switch formed by a series-connected circuit of two switching elements, and the bias section has one of the series circuits and applies a bias voltage to the current input terminal of one of the two switching elements.

3. The switched inductor according to claim 1, wherein the semiconductor switch section is a bidirectional switch formed by a series connection circuit of two switching elements, the bias section comprises a series circuit in which two constant voltage sources are connected with opposite polarity, the low voltage sides of the two constant voltage sources are connected to the current output terminals of the two switching elements and to a predetermined potential including ground potential, and the high voltage sides of the two constant voltage sources are connected via respective resistors to each current input terminal of the two switching elements and each end on the transformer side of the switch section.

4. The switched inductor according to claim 1, wherein the semiconductor switch section is a bidirectional switch formed by a series-connected circuit of two switching elements, the series circuit of the bias section comprises a constant voltage source connected between the connection point of the two switching elements and the connection point of two resistors, the low-voltage side of the constant voltage source is connected to the current output terminals of the two switching elements and to a predetermined potential including ground potential, and the high-voltage side of the constant voltage source is connected via each resistor to each current input terminal of the two switching elements and each end of the switch section on the transformer side.

5. A resonant inverter comprising a square wave generating unit and a resonant circuit to which a square wave signal from the square wave generating unit is input, wherein the resonant circuit is a series LC resonant circuit or a parallel LC resonant circuit configured with a switched inductor and a capacitor as set forth in any one of claims 1 to 4, and wherein the inductance of the switched inductor is switched to switch the inductance of the resonant circuit.

6. The resonant inverter according to claim 5, further comprising a Q-factor limiting circuit (QFL) connected in parallel with a capacitive reactance element at the output end of the resonant circuit, the Q-factor limiting circuit (QFL) circulating power from the resonant circuit to a supply DC power source to limit the Q-factor of the resonant circuit.

7. An RF power supply for outputting high frequency waves, comprising a supply DC power supply, a resonant inverter as defined in claim 5 or 6, and a control unit, wherein the square wave generating unit of the resonant inverter generates a square wave from the DC voltage of the supply DC power supply by opening and closing a switching element, and the control unit has a first response control mode for controlling the supply DC power supply and a second response control mode for controlling the resonant inverter.

8. The RF power supply according to claim 7, wherein the first response control mode controls the DC power supply based on the square wave voltage of the square wave generating section, and the second response control mode controls the switching frequency for driving the switching element of the switched inductor and / or the dead time based on any one or any combination of the current of the inductive element of the resonant circuit, the voltage of the capacitive element of the resonant circuit, and the time change (dV / dT) of the square wave voltage of the square wave generating section.

9. An RF power supply according to claim 7, further comprising a zero voltage switching circuit consisting of a series circuit of an inductor and a capacitor between said square wave generating section and said resonant circuit, and switching of said square wave generating section is performed at zero voltage.

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