Motor control device, motor control system, and motor control method

Abstract

In the present invention, a capacitor current detector detects the capacitor current of an inverter. A zero-phase voltage superimposing unit superimposes a zero-phase voltage on a voltage command value for controlling the phase current of a motor to calculate a new voltage command value. A phase current detector detects first phase current which is the phase current of the motor. A phase current calculator calculates second phase current, which is the phase current of the motor, from the capacitor current and the voltage command value obtained by superimposing the zero-phase voltage. A phase current replacement unit replaces the first phase current of at least one phase with the second phase current. A switching signal generator generates a switching signal for controlling a switching element from the voltage command value obtained by superimposing the zero-phase voltage.

Description

Motor control device, motor control system, and motor control method

[0001] The present disclosure relates to a motor control device, a motor control system, and a motor control method.

[0002] Conventionally, techniques for correcting offset errors in current detection values ​​have been disclosed to improve the current detection accuracy of motor control devices. For example, Patent Document 1 discloses an electric motor control device that corrects the detection value of a current sensor that detects a phase current of a target phase so that the terminal voltage of a smoothing capacitor matches a desired voltage value. Based on three phase voltage command signals generated from the detection values ​​of the current sensors, the target phase is identified as the phase of a second phase voltage command signal that has the largest difference from a first phase voltage command signal whose signal level is neither maximum nor minimum.

[0003] JP 2015-56919 A

[0004] In the motor control device described in Patent Document 1, a correction value calculated so that the inter-terminal voltage matches the desired voltage value is added to correct the detection value of the current sensor. Therefore, it is relatively easy to remove steady-state offset errors contained in the detection value. However, the period of error components contained in high-frequency components due to switching is shorter than the calculation period of the correction value. Because it is difficult to remove error components contained in the current value using the method described in Patent Document 1, current detection errors may increase.

[0005] An object of the present disclosure is to provide a motor control device, a motor control system, and a motor control method that solve the above-mentioned problems.

[0006] A first aspect is a motor control device comprising: a capacitor current detector that detects a capacitor current of an inverter; a zero-sequence voltage superimposer that calculates a new voltage command value by superimposing a zero-sequence voltage on a voltage command value for controlling a phase current of a motor; a phase current detector that detects a first phase current that is a phase current of the motor; a phase current calculator that calculates a second phase current that is a phase current of the motor from the capacitor current and the voltage command value on which the zero-sequence voltage is superimposed; a phase current replacer that replaces the first phase current of at least one phase with the second phase current; and a switching signal generator that generates a switching signal for controlling a switching element from the voltage command value on which the zero-sequence voltage is superimposed.

[0007] A second aspect is a motor control method in a motor control device, which detects a capacitor current of an inverter, calculates a new voltage command value by superimposing a zero-phase sequence voltage on a voltage command value for controlling a phase current of the motor, detects a first phase current which is the phase current of the motor, calculates a second phase current which is the phase current of the motor from the capacitor current and the voltage command value on which the zero-phase sequence voltage is superimposed, replaces the first phase current of at least one phase with the second phase current, and generates a switching signal for controlling a switching element from the voltage command value on which the zero-phase sequence voltage is superimposed.

[0008] According to the present disclosure, it is possible to suppress detection errors of the motor current.

[0009] FIG. 1 is a schematic block diagram showing an example configuration of a motor control system according to a first embodiment of the present disclosure. FIG. 2 is an explanatory diagram for describing an example of a procedure for generating a switching signal according to the first embodiment of the present disclosure. FIG. 3 is a diagram showing a first example of a voltage command value after superimposition of a zero-phase-sequence voltage. FIG. 4 is a diagram showing a second example of a voltage command value after superimposition of a zero-phase-sequence voltage. FIG. 5 is a diagram showing a first setting example of a switching signal based on a corrected voltage command value. FIG. 6 is a diagram showing a second setting example of a switching signal based on a corrected voltage command value. FIG. 7 is a diagram illustrating the relationship between a voltage vector and a capacitor current. FIG. 8 is a schematic block diagram showing an example configuration of a motor control system according to a second embodiment of the present disclosure. FIG. 9 is an explanatory diagram for describing a procedure for setting a detection timing adjustment value according to the second embodiment of the present disclosure. FIG. 10 is an explanatory diagram for describing the detection timing of a capacitor current in a calculation period according to a third embodiment of the present disclosure.

[0010] Hereinafter, embodiments of the present disclosure will be described with reference to the drawings. Elements that are common to or correspond to each other in the various drawings are denoted by the same reference numerals, and the description thereof will be incorporated unless otherwise specified. <First Embodiment> First, a first embodiment of the present disclosure will be described. FIG. 1 is a schematic block diagram showing an example configuration of a motor control system S1 according to this embodiment. The motor control system S1 includes a motor 1, a rotor position detector 2, a DC power supply 3, an inverter 4, and a controller 6.

[0011] Motor 1 consumes power from an AC current supplied from inverter 4 to rotate its rotor. Motor 1 is a three-phase AC motor. Motor 1 is a rotating machine equipped with windings and a rotor. Motor 1 is sometimes referred to as a rotating machine or an electric motor. An AC magnetic field is generated in the windings by electromagnetic induction. The rotor rotates due to the generated AC magnetic field. The rotation of motor 1 can be controlled on two rotating axes. Motor 1 may be, for example, a permanent magnet synchronous rotating machine, a wound-field synchronous rotating machine, an induction rotating machine, a synchronous reluctance motor, or the like. However, the following description will mainly focus on the case where motor 1 is a permanent magnet synchronous rotating machine.

[0012] The rotor position detector 2 detects the position of the rotor of the motor 1 as the rotor position θ. The rotor position detector 2 may include any of a resolver, an encoder, and an MR (Magneto-Resistance) sensor. The rotor position detector 2 notifies the controller 6 of the detected rotor position θ. Note that the rotor position detector 2 is not essential as long as the controller 6 can obtain the rotor position θ. The rotor position θ does not have to be an actually measured value, but may be an estimated value estimated from other information. The rotor position θ may also be an estimated value estimated by other equipment. For example, the rotor position θ is estimated based on position information of a sign attached to a known position of the motor rotor. The rotor position θ may be estimated by the measurement module using position information of the sign detected by a sensor that optically detects the sign.

[0013] The DC power supply 3 supplies the DC bus voltage V to the inverter 4 and the controller 6. dc The negative terminal of the DC power supply 3 is electrically connected to the potential reference point. The DC power supply 3 includes a device capable of supplying a DC voltage. The DC power supply 3 may include, for example, a battery, a DC-DC converter, a diode rectifier, a PWM rectifier, or the like. In the following description, the DC bus voltage V dc is sometimes called the "power supply voltage."

[0014] The inverter 4 is connected to the DC bus voltage V supplied from the DC power supply 3. dc and generates an AC current based on a switching signal input from the controller 6. The inverter 4 supplies the generated AC current to the motor 1. An example of the configuration of the inverter 4 will be described later.

[0015] The controller 6 receives a torque command T ref is input, and the voltage V Ru , V Rv , V Rw is input to the controller 6, and the rotor position θ is input from the rotor position detector 2. The controller 6 receives the torque command T ref , voltage across both ends V Ru , V Rv , V Rwand the switching signal G based on the rotor position θ up , G vp , G wp , G un , G vn , G wn The controller 6 generates the generated switching signal G up , G vp , G wp , G un , G vn , G wn is output to the inverter 4.

[0016] Next, a configuration example of the inverter 4 will be described. The inverter 4 has three upper arm switching elements S up , S vp , S wp and three lower arm switching elements S un , S vn , S wn and smoothing capacitor C inv and four current detection resistor elements R c , R u , R v , R w The current detection resistor element R c , R u , R v , R w is also called a shunt resistor. The indexes u, v, and w indicate the phases of the AC current generated in the motor 1. In this application, the upper arm switching element S up , S vp , S wp Some or all of the lower arm switching elements S un , S vn , S wn Some or all of this may be referred to as the "lower arm."

[0017] Smoothing capacitor C inv and the current detection resistor R c The upper arm switching element S up , lower arm switching element S un and current detection resistor element R uThe upper arm switching element S vp , lower arm switching element S vn and current detection resistor element R v and the upper arm switching element S wp , lower arm switching element S wn and current detection resistor element R w The pairs are connected in parallel with each other. inv and upper arm switching element S up , S vp , S wp A power supply voltage is applied to one end of each of the resistor elements R from a DC power supply 3. c , R u , R v , R w The other ends of the respective terminals are connected to a potential reference point.

[0018] Smoothing capacitor C inv The other end of the resistor R c The upper arm switching element S up , S vp , S wp The other end of each of the lower arm switching elements S un , S vn , S wn The lower arm switching element S un , S vn , S wn The other end of each resistor is connected to the current detection resistor R u , R v , R w is connected to.

[0019] Upper arm switching element S up , S vp , S wp and the lower arm switching element S un , S vn , S wn The base end of the controller 6 receives a switching signal G up , G vp , G wp , G un , G vn, G wn are input respectively. Switching signal G up , G vp , G wp , G un , G vn , G wn indicate an ON command or an OFF command, respectively. An ON command is a command to turn on, i.e., to close the switching element. An OFF command is a command to cut off, i.e., to open the switching element. An ON command and an OFF command are transmitted by potentials corresponding to values ​​1 and 0, respectively.

[0020] Upper arm switching element S up , S vp , S wp and the lower arm switching element S un , S vn , S wn are the switching signals G up , G vp , G wp , G un , G vn , G wn When the upper arm switching element for a certain phase closes both ends in response to an ON command and the corresponding lower arm switching element opens both ends in response to an OFF command, the power supplied from the DC power supply 3 along with the power supply voltage for that phase is output to the motor 1. When the upper arm switching element for a certain phase opens both ends in response to an OFF command and the lower arm switching element closes both ends in response to an ON command, the supply of power to the motor 1 for that phase is stopped and the potential at the other end of the upper arm switching element approaches the potential of the potential reference point. Therefore, the switching signal G up , G vp , G wp , G un , G vn , G wn The AC current I with periodically varying voltage for each phase is calculated based on u , I v , I w is supplied to the motor 1.

[0021] Current detection resistor element R u , Rv , R w At each end, a current i flows through the part. u , i v , i w The voltage V across the Ru , V Rv , V Rw Therefore, the AC current I u , I v , I w The voltage V Ru , V Rv , V Rw is applied to the motor 1. Current detection resistor R u , R v , R w If the resistance values ​​of each are known, the voltage across both ends V Ru , V Rv , V Rw are the currents i u , i v , i w or current I u , I v , I w These values ​​are proportional to the u, v, and w phases of the rotating machine, respectively, and can be regarded as detected values ​​of the u, v, and w phases of the rotating machine current. In this application, the current related to a phase on the three-phase coordinate system is sometimes called a "phase current," and the voltage related to that phase is sometimes called a "phase voltage." In addition, the current detection resistor element R c The voltage across the Rc The voltage across both ends V Rc is the current detection resistor element R c The current i c is proportional to the current i c is the smoothing capacitor C inv The capacitor current I c The current detection resistor element R c If the resistance value of is known, the voltage across V c is the capacitor current I c Since the amount is proportional to the capacitor current I c can be regarded as the detected value.

[0022] Next, a configuration example of the controller 6 will be described. The controller 6 illustrated in FIG. 1 includes a discrete time calculator and outputs a switching signal Gup , G vp , G wp , G un , G vn , G wn The discrete-time calculator is configured as a PWM controller that generates a PWM (Pulse Width Modulation) signal as a PWM signal. A PWM signal is a binary signal whose value changes periodically. The discrete-time calculator may be any device that has hardware capable of performing arithmetic processing, such as a microcomputer (microcomputer chip) or a DSP (Digital Signal Processor), and can provide the functions of a computer system. The discrete-time calculator reads a control program stored in advance in a non-transitory storage medium and executes arithmetic processing instructed by one or more commands written in the read control program, thereby realizing the functions of the following units. Note that the controller 6 may also be equipped with dedicated hardware to realize its functions.

[0023] The controller 6 is configured to include a speed calculator 11, a first current detector 12, a second current detector 13, a current substituter 14, a coordinate converter 15, a current command calculator 16, a voltage command calculator 17, a coordinate converter 18, a zero-phase voltage superimposer 19, a PWM signal generator 20, and a phase current calculator 23.

[0024] The speed calculator 11 time-differentiates the rotor position θ notified from the rotor position detector 2 to calculate the rotational angular velocity ω of the motor 1. The speed calculator 11 notifies the calculated rotational angular velocity ω to the current command calculator 16. The first current detector 12 calculates the voltage V Ru , V Rv , V Rw The rotating machine current I of each phase is calculated as the first phase current using us , I vs , I ws , i.e., phase current I us , I vs , I ws The first current detector 12 detects the detected phase current I us , I vs , I ws to the current substituter 14. The second current detector 13 detects the voltage V cUsing the capacitor current I c The second current detector 13 detects the detected capacitor current I c is output to the current substituter 14.

[0025] The current substituter 14 receives the phase current I as the first phase current from the first current detector 12. us , I vs , I ws and the phase current I is output as the second phase current from the phase current calculator 23. uc , I vc , I wc The current substituter 14 substitutes the first phase current with the second phase current for some or all of the three phases. When the first phase current is to be replaced for some of the phases, the current substituter 14 substitutes the voltage command value V u , V v , V w The current substituter 14 may determine the phase to be replaced with the second phase current as the substitution target phase based on, for example, the voltage command value V u , V v , V w The current substituter 14 selects the second phase current for the phase determined as the phase to be substituted from the input phase currents, and selects the first phase current for the other phases. The current substituter 14 selects the rotating machine current I u0 , I v0 , I w0 is output to the coordinate converter 15.

[0026] The coordinate converter 15 converts the rotating machine current I on the three-phase coordinate system input from the current converter 14. u0 , I v0 , I w0 is converted into coordinates based on the rotor position θ input from the rotor position detector 2, and the rotating machine current I d , I qThe indexes d and q respectively indicate the d-axis (direct axis) and q-axis (quadrature axis) of the motor 1. In other words, the two rotation axes refer to the d-axis and q-axis. The d-axis indicates the direction of the field flux of the motor 1. The q-axis indicates the direction of the rotor magnetic flux of the motor 1. The coordinate converter 15 converts the converted rotating machine current I d , I q is output to the voltage command calculator 17.

[0027] The current command calculator 16 receives a torque command T ref , the power supply voltage V supplied from the DC power supply 3 dc and the rotational angular velocity ω input from the velocity calculator 11, and a current command value I on the two rotation axes, which is a target value of the current to be supplied to the motor 1, is calculated using a known calculation method. dref , I qref The calculation method may be, for example, any of MTPA (Maximum Torque Per Ampere) control, MTPV (Maximum Torque Per Flux) control, and flux-weakening control. These calculation methods may be used depending on the range of the input value. The current command calculator 16 calculates the calculated current command value I dref , I qref is output to the voltage command calculator 17.

[0028] The voltage command calculator 17 calculates the current command value I dref , I qref , the rotating machine current I input from the coordinate converter 15 d , I q , DC bus voltage V supplied from DC power supply 3 dc and the rotational angular velocity ω input from the velocity calculator 11, the voltage command value V on the two rotational axes is calculated. d , V q The voltage command calculator 17 calculates the calculated voltage command value V d , V q is output to the coordinate converter 18.

[0029] The coordinate converter 18 converts the voltage command value V d , V qis converted into a coordinate system based on the rotor position θ input from the rotor position detector 2, and the voltage command value V u0 , V v0 , V w0 The coordinate converter 18 calculates the calculated voltage command value V u0 , V v0 , V w0 is output to the zero-phase voltage superimposer 19.

[0030] The zero-phase voltage superimposer 19 superimposes the voltage command value V u0 , V v0 , V w0 The new voltage command value V is obtained by superimposing the zero-phase voltage that is equal among the three phases on the u , V v , V w For example, the zero-phase voltage superimposer 19 calculates the voltage command value V u , V v , V w The zero-phase voltage is determined so that the maximum value of is equal to the power supply voltage. The power supply voltage is the DC bus voltage V supplied from the DC power supply 3. dc The zero-phase voltage superimposer 19 calculates the voltage command value V u , V v , V w to the current substituter 14, the PWM signal generator 20, and the phase current calculator 23. The zero-phase voltage superimposer 19 outputs the voltage command value V u , V v , V w The zero-phase sequence voltage may be determined so that the minimum value of is equal to the reference potential (i.e., 0 V). An example of the voltage command value after the zero-phase sequence voltage is superimposed will be described later.

[0031] The PWM signal generator 20 receives the voltage command value V u , V v , V w Based on the switching signal G up , G vp , G wp , G un , G vn , G wn The PWM signal generator 20 generates a PWM signal as the generated switching signal G up , Gvp , G wp , G un , G vn , G wn is output to the inverter 4. An example of the procedure for generating the switching signal will be described later.

[0032] The phase current calculator 23 calculates the capacitor current I c and the voltage command value V input from the zero-phase voltage superimposer 19 u , V v , V w Based on the second phase current I uc , I vc , I wc The phase current calculator 23 calculates the calculated phase current I uc , I vc , I wc is output to the current substituter 14. An example of calculating the phase current will be described later.

[0033] Next, an example of a procedure for generating a switching signal according to this embodiment will be described. Fig. 2 is an explanatory diagram for explaining an example of a procedure for generating a switching signal according to this embodiment. The PWM signal generator 20 has a carrier wave generator that generates a carrier triangular wave as a carrier wave. The waveform of the carrier wave has a predetermined period T c In the present application, this period T c is sometimes called a "carrier period." The carrier period may also be called a carrier period. Each carrier period T c The waveform in represents a time series of signal values ​​at each point in time within a period.

[0034] In the example of FIG. 2, the signal value of the carrier wave C reaches a minimum value of zero at time T1 and a maximum value of the power supply voltage V dc Time T1 corresponds to the start time of the carrier wave period. Time T2 corresponds to the time half a period after the start time (sometimes referred to as the "intermediate time" in this application). The signal value of the carrier wave C increases linearly from time T1 to time T2 for each carrier wave period, and decreases linearly from time T2 to time T1 of the next carrier wave period. The PWM signal generator 20 generates a voltage command value V u , Vv , V w and switching signal G based on carrier wave C up , G vp , G wp , G un , G vn , G wn Here, the PWM signal generator 20 generates a voltage command value V u , V v , V w and the signal value of the carrier wave C to determine which is larger. The PWM signal generator 20 uses the voltage command value V obtained for each phase u , V v , V w and the signal value of carrier wave C, and the values ​​mutually inverted between the upper and lower arms are set as the value of the switching signal corresponding to that pair of phase and arm.

[0035] Figure 2 shows the carrier wave C and the switching signal G up , G vp , G wp , G un , G vn , G wn are shown in that order from top to bottom. In each column, the vertical axis indicates the signal value, and the horizontal axis indicates the time. u , V v , V w is superimposed on the carrier wave C. The voltage command value V for the u phase u is greater than the signal value of the carrier wave C, the PWM signal generator 20 generates the upper arm switching signal G up and the value of the lower arm switching signal G un The values ​​of the voltage command value V are set to 1 (ON command) and 0 (OFF command), respectively. u is smaller than the signal value of the carrier wave C, the PWM signal generator 20 generates the upper arm switching signal G up and the value of the lower arm switching signal G un The values ​​of are set to 0 (OFF command) and 1 (ON command), respectively.

[0036] Voltage command value V for v-phase vis greater than the signal value of the carrier wave C, the PWM signal generator 20 generates the switching signal G vp , G vn The values ​​of the voltage command value V are set to 1 (ON command) and 0 (OFF command), respectively. v is smaller than the signal value of the carrier wave C, the PWM signal generator 20 generates the switching signal G vp , G vn The values ​​of V and V are set to 0 (OFF command) and 1 (ON command), respectively. w is greater than the signal value of the carrier wave C, the PWM signal generator 20 generates the switching signal G wp , G wn The values ​​of the voltage command value V are set to 1 (ON command) and 0 (OFF command), respectively. w is smaller than the signal value of the carrier wave C, the PWM signal generator 20 generates the switching signal G wp , G wn The values ​​of are set to 0 (OFF command) and 1 (ON command), respectively.

[0037] The PWM signal generator 20 may set the first and second short-circuit prevention periods to be periods within a predetermined range from the first switching time point at which the signal value of the carrier wave C for each phase switches from a state smaller than the voltage command value of that phase to a state larger than it, and from the second switching time point at which the signal value of the carrier wave C for each phase switches from a state larger than the voltage command value of that phase to a state smaller than it. The lengths of the first and second short-circuit prevention periods are each set to the carrier wave period T c The period may be sufficiently shorter than half of the first short circuit prevention period. The PWM signal generator 20 sets the values ​​of the upper and lower arm switching signals for each phase to 0 for the first short circuit prevention period and the second short circuit prevention period. This prevents the upper and lower arm switching elements of at least one phase in the inverter 4 from being short-circuited simultaneously across the two ends. This prevents abnormal current from passing through the current detection resistor element for that phase.

[0038] The switching signal generated by the above procedure is generated based on the voltage command value for each phase with a carrier period T cThe voltage corresponding to the value that varies with each switching element is also un , S vn , S wn The switching signal G un , G vn , G wn In this case, a pattern may occur in which the value of all three phases is 1. For example, this pattern occurs in period D in FIG. 2. The inverter 4 controls the output of current to the motor 1 in accordance with the value of the switching signal for each phase. The current output to the motor 1 has a carrier wave period T c It contains harmonic components with frequencies that are integer multiples of the carrier period T c In some cases, these harmonic components may cause unpleasant noise from the motor 1.

[0039] Therefore, the carrier period T c The carrier frequency, which is the reciprocal of c It is sufficient that the carrier frequency is set to a value outside the audible band. More preferably, the carrier frequency should be outside the audible band. In general, the upper and lower limits of the frequency band in which humans can easily hear sounds are lower than the upper limit (e.g., 20 kHz) and higher than the lower limit (e.g., 20 Hz) of the audible band. For example, when the motor 1 is used for steering assistance (i.e., power steering) of a vehicle, the carrier wave period T c The carrier period T should be set to 60 μs or less. c The switching frequency, which is the reciprocal of this, is 16.6 kHz or higher. Sounds with frequency components or harmonic components at this frequency are difficult or impossible for humans to hear. This reduces or eliminates the discomfort caused by abnormal noise.

[0040] Next, an example of a voltage command value after superimposing a zero-phase-sequence voltage will be described. FIG. 3 is a diagram showing a first example of a voltage command value after superimposing a zero-phase-sequence voltage. In FIG. 3, the vertical axis indicates the electrical angle (unit: deg (degrees)), and the vertical axis indicates the duty ratio (unit: %). The duty ratio is calculated as follows: c2, this period corresponds to the period during which the voltage command value exceeds the signal value of the carrier wave. Therefore, the larger the duty ratio, the larger the voltage command value.

[0041] In the example of Figure 3, the voltage command value V u , V v , V w It is assumed that the zero-phase voltage is determined so that the maximum value of is equal to the power supply voltage. u , d v , d w The maximum value of the duty ratio d is 100%. u , d v , d w The range of electrical angle where the duty ratio d of each phase is 100% is 120 degrees. u , d v , d w The minimum value of is greater than 0%.

[0042] 4 is a diagram showing a second example of a voltage command value after superimposing a zero-phase sequence voltage. In the example of FIG. 4, the voltage command value V u , V v , V w It is assumed that the zero-phase voltage is determined so that the minimum value of is equal to the reference potential. u , d v , d w The minimum value of the duty ratio d is 0%. u , d v , d w The range of electrical angle where the duty ratio d of each phase is 0% is 120 degrees. u , d v , d w The maximum value of is less than 100%.

[0043] Next, an example of setting a switching signal based on the corrected voltage command value will be described. FIG. 5 is a diagram showing a first example of setting a switching signal based on the corrected voltage command value. In the example of FIG. 5, similarly to the example of FIG. 3, the voltage command value V u , V v , Vw It is assumed that the maximum value of is equal to the power supply voltage. In this case, the voltage command value V u , V v , V w The upper arm switching signal G for the u phase that gives the maximum value of up The value of is always 1, and the lower arm switching signal G un The value of the lower arm switching signal G un , G vn , G wn At time T2, the values ​​of the lower arm switching signal G for the u phase are all 0. un The value of is 0, and the lower arm switching signal G vn , G wn The value of each is 1.

[0044] When the modulation factor is higher than a predetermined reference value of the modulation factor, the zero-phase voltage superimposer 19 superimposes the voltage command value V u , V v , V w The voltage command value V u , V v , V w The modulation factor corresponds to the ratio of the voltage command value to the maximum value of the carrier wave. In other words, the modulation factor is an index showing the utilization rate of the power supply voltage. The voltage command value V u , V v , V w With the exception of the phase in which the maximum value of is equal to the power supply voltage, the timing at which the value of the switching signal changes is separated from time T2, the timing at which the value of the carrier wave changes from increasing to decreasing. By separating these two timings, the concentration of switching noise caused by switching is alleviated. In turn, the reduction in noise components suppresses the deterioration of current detection accuracy. This switching noise is known as ringing, and mainly contains high-frequency components with frequencies higher than the carrier wave period.

[0045] Next, an example of setting a switching signal based on the corrected voltage command value will be described. FIG. 6 is a diagram showing a second example of setting a switching signal based on the corrected voltage command value. In the example of FIG. 6, the voltage command value V u , V v , V w It is assumed that the minimum value of is equal to the reference potential. In this case, the voltage command value V u , V v , V w The upper arm switching signal G for the w phase that gives the minimum value of wp The value of is always 0, and the lower arm switching signal G wn The value of is always 1. At time T1, the upper arm switching signal G for the w phase wn The value of is 0, and the upper arm switching signal G for the u and v phases vn , G wn At time T2, the value of the lower arm switching signal G un , G vn , G wn The value of each is 1.

[0046] Next, the relationship between the voltage vector and the capacitor current for each switching state will be explained. The switching state corresponds to a combination of the on and off states for each phase. In other words, the switching state is indicated by the value of the switching signal for the upper arm for each phase. The voltage vector is a vector whose elements are the voltages applied to the motor 1 from the inverter 4 for each phase. The voltage for each phase is either Hi or Lo. Hi corresponds to the power supply voltage. Lo corresponds to the reference potential. Therefore, there are eight types of voltage vectors. The eight types of voltage vectors are distinguished by notating them as voltage vector V0, etc. The capacitor current flows through the smoothing capacitor C inv is the current flowing through

[0047] FIG. 7 is a diagram illustrating the relationship between voltage vectors and capacitor currents. FIG. 7 shows the pairs of phase voltages and capacitor currents for each voltage vector. Of the eight voltage vectors, V0 and V7 are zero voltage vectors, and V1 to V6 are non-zero voltage vectors. A zero voltage vector is a vector that indicates a phase voltage that occurs when the switching state of all three phases is either the on state or the off state. A non-zero voltage vector is a voltage vector other than the zero voltage vector. In the example of FIG. 7, for a zero voltage vector, the capacitor current I c is the DC current I dc DC current I dc corresponds to the current supplied from the DC power source 3. For non-zero voltage vectors, the capacitor current I c is the DC current I dc When the voltage of one of the three phases is Hi, the capacitor current I c is the DC current I dc When the voltage of one of the three phases is Lo, the capacitor current I c is the DC current I dc and the phase current of that phase. Therefore, the phase current calculator 23 calculates the capacitor current I c The capacitor current I when a zero voltage vector occurs from c The phase current of that one phase can be obtained by dividing

[0048] Next, an example of calculating the phase current will be described. In the example of FIG. 5, at time T1, the voltages of all phases are Hi, so the voltage vector is V7. Therefore, the capacitor current I c (T1) is the DC current I dc At time T2, the u-phase voltage is Hi, and the v-phase voltage and w-phase voltage are both Lo. Therefore, the voltage vector is V1. The capacitor current I at time T2 c (T2) is I dc -I u Therefore, the phase current calculator 23 calculates the capacitor current I c (T2) to capacitor current Ic The difference value I obtained by subtracting (T1) u The phase current I uc The phase current calculator 23 can also calculate the phase current for phases other than the u-phase by performing the same procedure.

[0049] As described above, the current substituter 14 converts the voltage command value V u , V v , V w The phase that gives the maximum value of the voltage command value V is determined as the phase to be replaced from the first phase current to the second phase current. u , V v , V w Among them, the voltage command value V u The current substituter 14 determines the u-phase rotating machine current I us The rotating machine current I input from the phase current calculator 23 uc However, the current substituter 14 replaces the v-phase rotating machine current I vs and w-phase rotating machine current I ws In this case, the rotating machine current I of each phase output from the current replacer 14 to the coordinate converter 15 is maintained without being replaced. u0 , I v0 , I w0 I uc , I vs , I ws This becomes:

[0050] In a configuration in which a current detection resistor element is provided between the lower arm switching element and the negative pole of the DC power supply 3, such as the inverter 4 illustrated in FIG. 1, it is assumed that the zero-phase sequence voltage superimposer 19 superimposes a zero-phase sequence voltage so that the maximum value of the phase voltage command value becomes equal to the power supply voltage, thereby correcting the phase voltage command value. In this case, there may be a phase in which the current detection resistor element cannot detect the current. For example, in the u-phase illustrated in FIG. 5, the upper arm switching element S up are closed (ON state), and the lower arm switching element S unAt this time, the current detection resistor element R u Since no current supplied from the DC power supply 3 flows through the u-phase, the current related to the u-phase cannot be detected. Therefore, for a phase in which the current cannot be detected by the current detection resistive element, the phase current calculator 23 can calculate the phase current of that phase from the capacitor current using the above-mentioned method.

[0051] As described above, in this embodiment, the current detection resistor element R c is the capacitor current I c The phase current calculator 23 calculates the detected capacitor current I c The current I flows through motor 1 as the second phase current from uc , I vc , I wc The current substituter 14 calculates the current I detected from the motor 1 as the first phase current. us , I vs , I ws The current calculated for at least some of the phases of the smoothing capacitor C inv The position of the switching element S un , S vn , S wn The current detection resistor element R un , R vn , R wn than the switching element S un , S vn , S wn The switching timing is controlled based on the second phase current estimated from the capacitor current, instead of the first phase current, thereby reducing the influence of high-frequency noise caused by switching.

[0052] That is, a motor control device (e.g., controller 6) according to the present disclosure includes a capacitor current detector (e.g., second current detector 13) that detects the capacitor current of inverter 4, a zero-phase-sequence voltage superimposer 19 that superimposes a zero-phase voltage on a voltage command value for controlling the phase current of motor 1, a phase current detector (e.g., first current detector 12) that detects a first phase current, which is the phase current of the motor, a phase current calculator 23 that calculates a second phase current, which is the phase current of motor 1, from the capacitor current and the voltage command value, a phase current substituter (e.g., current substituter 14) that substitutes at least one first phase current with the second phase current, and a switching signal generator (e.g., PWM signal generator 20) that generates a switching signal for controlling switching elements from the voltage command value. Because the capacitor is located farther from the switching elements than the current detection resistor elements used to detect the first phase current, the influence of switching noise is mitigated. Therefore, by replacing the first phase current with the second phase current calculated from the capacitor current, switching noise, which is primarily comprised of high-frequency components, is reduced. Therefore, current detection errors are suppressed.

[0053] The phase current substituter may select the phase with the largest voltage command value as the phase to substitute the first phase current. In some cases, the inverter 4 may not be able to detect the first phase current of the phase with the largest voltage command value. By substituting the first phase current with the second phase current, it is possible to more reliably detect the currents of all three phases.

[0054] The zero-sequence voltage superimposer 19 may superimpose the zero-sequence voltage so that the maximum voltage command value for each phase is equal to the power supply voltage. At each of two detection times within the carrier wave period, the voltage vector supplied from the inverter 4 to the motor 1 is a zero voltage vector and a non-zero voltage vector. Therefore, the second phase current can be reliably calculated from the difference between the capacitor currents detected at each detection time.

[0055] Second Embodiment Next, a second embodiment of the present disclosure will be described. The following description will mainly focus on differences from the first embodiment. The description of the first embodiment will be used to refer to components common to the first embodiment. FIG. 8 is a schematic block diagram showing an example configuration of a motor control system S1 according to this embodiment. The controller 6 of the motor control system S1 according to this embodiment further includes a settling determiner 21 and a detection timing adjuster 22.

[0056] The detection timing adjuster 22 detects the voltage command value V u , V v , V w Based on this, the detection timing adjustment value t s1 , t s2 The detection timing adjuster 22 calculates the calculated detection timing adjustment value t s1 , t s2 to the phase current calculator 23. As will be described later, the phase current calculator 23 outputs the detection timing adjustment value t s1 , t s2 , the capacitor current I c This is used to identify the detection timing.

[0057] The detection timing adjuster 22 adjusts a period shorter than the period from the start time of the carrier wave period to the first switching timing by a detection timing adjustment value t s1 The first switching timing is when the value of the carrier wave C is equal to the voltage command value V u , V v , V w In the example of FIG. 9, the time when the value of the carrier wave C exceeds the voltage command value V w The time when this voltage command value V w is the voltage command value V u , V v , V w More specifically, as exemplified by equation (1), the detection timing adjuster 22 determines the period from the start time of the carrier wave period to the time preceding the first switching timing by a predetermined margin time α1 as the detection timing adjustment value ts1 This period corresponds to the period during which the inverter 4 outputs a zero vector to the motor 1.

[0058]

[0059] In formula (1), V min is the three-phase voltage command value V u , V v , V w The minimum value of min(V u , V v , V w ) indicates the power supply voltage V dc corresponds to the maximum value of the carrier wave. Therefore, the first term on the right side of equation (1) is the carrier wave period T c The margin time α1 indicates the length of the period from the start time to the first switching timing. A positive real number smaller than the period is set as the margin time α1.

[0060] The detection timing adjuster 22 adjusts a period shorter than the period from the midpoint of the carrier wave period to the first switching timing by a detection timing adjustment value t s2 The first switching timing is when the value of the carrier wave C is equal to the voltage command value V u , V v , V w In the example of FIG. 9, the maximum value of the carrier wave C corresponds to the earliest time when the voltage command value V u Then, the value of the carrier wave C is set equal to the voltage command value V v The time when the voltage command value V falls below this corresponds to the first switching timing. v is the voltage command value V u , V v , V w More specifically, as shown in equation (2), the detection timing adjuster 22 determines the period from the middle of the carrier wave period to the time preceding the first switching timing by a predetermined margin time α2 as the detection timing adjustment value t s2 This period corresponds to the period during which the inverter 4 outputs a non-zero vector to the motor 1.

[0061] ​

[0062] In equation (2), V mid is the three-phase voltage command value V u , V v , V w Median of (V u , V v , V w ) indicates the power supply voltage V dc corresponds to the maximum value of the carrier wave and is also equal to the maximum value of the voltage command. Therefore, the difference between the first and second terms on the right side of equation (2) is the carrier wave period T c The margin time α2 is set to a positive real number smaller than the period.

[0063] As shown in the bottom row of FIG. 9, the capacitor current I c Therefore, the capacitor current I c The convergence time t of the ringing included in stl The detection timing adjustment value t may be measured in advance. s1 , t s2 are convergence times t stl The margin times α1 and α2 are set in advance so that the above holds. As a result, the second current detector 13 detects the capacitor current I c can be detected.

[0064] The settling determiner 21 determines the voltage command value V u , V v , V w Based on this, the capacitor current I detected by the second current detector 13 at times T1 and T2 is c The time T1 is determined by the detection timing adjustment value t s1 Time T2 corresponds to the time after the detection timing adjustment value t s2 The settling decision unit 21 detects the capacitor current I at time T1. c Judgment result for f​s1 and time T ２ The capacitor current I detected at c Judgment result for f s2 The phase current calculator 23 is notified of the determination result f s1 , f s2 are expressed as a value indicating that the temperature has settled (for example, 1) or a value indicating that the temperature has not settled (for example, 0).

[0065] The settling determiner 21 calculates the period T from the last switching timing before time T1 to time T1. minV is the detection timing adjustment value t s1 The capacitor current I detected at time T1 is c During this period, the inverter 4 outputs a zero voltage vector to the motor 1. In the example of FIG. 9, the last switching timing before time T1 is when the value of the carrier wave C is equal to the voltage command value V w This corresponds to the time when the voltage command value V w is the three-phase voltage command value V u , V v , V w The settling decision unit 21 calculates the time period T minV The first term of equation (3) corresponds to the period from the time when the value of carrier wave C falls below the minimum value of the voltage command value in the carrier wave cycle immediately before the carrier wave cycle of interest to be processed, to the time when the value of carrier wave C next exceeds the minimum value of the voltage command value.

[0066]

[0067] The settling determiner 21 calculates the period T from the last switching timing before time T2 to time T2. midV is the detection timing adjustment value t s2 The capacitor current I detected at time T2 is c In the example of FIG. 9, the last switching timing before time T2 is the time when the value of the carrier wave C is equal to or greater than the voltage command value V v This corresponds to the time when the voltage command value V​v is the three-phase voltage command value V u , V v , V w The settling decision unit 21 calculates the time period T midV The difference between the first and second terms in equation (4) corresponds to the period from when the value of carrier wave C exceeds the median value of the voltage command value to when the value next falls below the minimum value of the voltage command value. During this period, a non-zero voltage vector is output from inverter 4 to motor 1.

[0068]

[0069] The phase current calculator 23 calculates the capacitor current I input from the second current detector 13 for each carrier wave period. c , the voltage command value V input from the zero-phase voltage superimposer 19 u , V v , V w , the detection timing adjustment value t input from the detection timing adjuster 22 s1 , t s2 , and the settling determination result f input from the settling determiner 21 s1 , f s2 Based on the phase current I uc , I vc , I wc The phase current calculator 23 calculates the capacitor current I c As the first detection timing, the detection timing adjustment value t s1 The phase current calculator 23 calculates the capacitor current I c As the second detection timing, the detection timing adjustment value t s2 Time T delayed by a period equivalent to ２ The following is established.

[0070] The phase current calculator 23 calculates the settling determination result f s1 indicates that the capacitor current I c Detects the capacitor current I c (T1) The newly detected capacitor current I c ​The phase current calculator 23 updates the settling determination result f s1 indicates that the capacitor current I c (T1) is not detected, and the already acquired capacitor current I c (T1).

[0071] The phase current calculator 23 calculates the settling determination result f s2 indicates that the capacitor current I c (T2) is detected and the already acquired capacitor current I c (T2) The newly detected capacitor current I c The phase current calculator 23 updates the settling determination result f s1 indicates that the capacitor current I c (T2) is not detected, and the already acquired capacitor current I c Then, the phase current calculator 23 stores the latest capacitor current I c (T1), I c Based on (T2), the phase current can be calculated using a method similar to that of the first embodiment.

[0072] As described above, the detection timing adjuster 22 according to this embodiment adjusts the capacitor current I c The first detection timing is adjusted to a time period from the start time of each carrier wave period to the first switching. The detection timing adjuster 22 adjusts the second detection timing to a time period from the intermediate time to the first switching. The settling determiner 21 adjusts the period T minV is the adjustment value t of the first detection timing s1 The capacitor current I detected at the first detection timing is determined based on whether it is greater than c The settling determiner 21 determines whether the period T (T1) from the last switching to the second detection timing has settled. midV is the second detection timing adjustment value t s2 The capacitor current I detected at the second detection timing is determined based on whether it is greater than c (T ２The phase current calculator 23 calculates the phase current using the capacitor current that has been determined to have settled.

[0073] This configuration allows the capacitor current I to be detected when the ringing with high frequency components has subsided. c The phase current is calculated using the c The capacitor current I is determined to have settled without using c The phase current is calculated using the above formula. Irregular switching based on the voltage command value stabilized based on the calculated phase current is suppressed. Ringing due to irregular switching is suppressed, and current detection errors are further suppressed.

[0074] That is, the motor control device (e.g., controller 6) according to this embodiment further includes a detection timing adjuster 22 that adjusts the detection timing of the capacitor current, and a settling determiner 21 that determines whether the capacitor current has settled. The detection timing adjuster 22 adjusts the detection timing based on a voltage command value. With this configuration, the influence of ringing contained in the capacitor current is suppressed, thereby reducing current detection errors. Furthermore, calculation of the second phase current when the influence of ringing remains is avoided, thereby reducing current detection errors due to ringing.

[0075] Furthermore, the detection timing adjuster 22 may adjust the detection timing of the capacitor current to the timing when the capacitor current is settling. With this configuration, the capacitor current is detected at the timing when the capacitor current is settling, thereby suppressing current detection errors due to ringing.

[0076] Alternatively, the settling determiner 21 may determine the output period of a zero voltage vector or a non-zero voltage vector based on the voltage command value, and determine that the capacitor current has settled if the output period exceeds a predetermined value. With this configuration, the settling of the capacitor current is determined based on the output period of the voltage vector. Therefore, whether the capacitor current has settled can be reliably determined through simple processing.

[0077] Third Embodiment Next, a third embodiment of the present disclosure will be described. The following description will mainly focus on differences from the second embodiment. The description of the second embodiment will be used for configurations common to the second embodiment. The following description is based on the premise that the motor control system S1 according to this embodiment has a configuration similar to that of the motor control system S1 shown in FIG. 8 .

[0078] The controller 6 according to this embodiment repeats its operation every calculation period. The calculation period is equal to the carrier wave period T c In contrast, in the first and second embodiments, the carrier wave period T c In the following description, of the two carrier wave periods included in one calculation period, the preceding one may be referred to as the first period, and the succeeding one may be referred to as the second period.

[0079] The PWM signal generator 20 generates voltage vectors indicating the same set of phase voltages in the first and second periods. That is, the PWM signal generator 20 outputs a PWM signal indicating the same switching state pattern twice for each calculation period to the inverter 4. The first current detector 12 detects the first phase current in the first period and does not detect the first phase current in the second period. In the example of FIG. 10 , the first current detector 12 detects the first phase current at the start time T 11 However, this does not apply when the voltage command value is corrected so that the maximum value is equal to the power supply voltage, as illustrated in Figures 3, 5, and 10. In this case, the first current detector 12 detects the first phase current at the intermediate time T 12 The first phase current is detected at

[0080] The second current detector 13 detects the capacitor current in the second period and does not detect the capacitor current in the first period. In the example of FIG. 10 , the second current detector 13 detects the capacitor current at time T 21 , T 12 The capacitor current is detected at each time T 21 is the detection timing adjustment value t calculated by the detection timing adjuster 22 from the start time of the second period. s1 This corresponds to the time delayed by the time T22 is the detection timing adjustment value t from the midpoint of the second period. s2 This corresponds to a time delayed by the time equivalent to

[0081] Other components of the controller 6, such as the speed calculator 11, the current substituter 14, the coordinate converter 15, the current command calculator 16, the voltage command calculator 17, the coordinate converter 18, the zero-phase voltage superimposer 19, the settling determiner 21, the detection timing adjuster 22, and the phase current calculator 23, perform the above operations for each calculation period.

[0082] Therefore, in this embodiment, the detection time of the capacitor current is variable within the second period, while the detection time of the phase current is set to a fixed time within the first period. By distributing the detection times of both in different periods, the influence of a detection phase difference is suppressed. Note that, although the above description exemplifies a case in which the first current detector 12 detects the first phase current in the first period and the second current detector 13 detects the capacitor current in the second period, this is not limiting. The first current detector 12 may detect the first phase current in the second period and the second current detector 13 may detect the capacitor current in the first period.

[0083] That is, in the motor control device (e.g., controller 6) according to this embodiment, a switching signal generator (e.g., PWM signal generator 20) generates a switching signal for each carrier wave period, and for each calculation period having two switching periods, a phase current detector (e.g., first current detector 12) detects the first phase current in one of the two switching periods, and a capacitor current detector (e.g., second current detector 13) detects the capacitor current in the other switching period. With this configuration, the detection timing of the first phase current is kept constant, and even if the detection timing of the capacitor current is variable, the influence of detection phase deviation is suppressed. Current detection errors are suppressed, and detection accuracy is improved.

[0084] Although the embodiments of the present disclosure have been described above, the present disclosure is not limited to the above embodiments. The drawings used in the above description are merely examples and are not limiting. The above embodiments may be freely modified without departing from the spirit of the present disclosure. For example, the first to third embodiments, or any two of them, may be combined or used separately as long as no contradiction occurs. Furthermore, some of the configurations may be omitted, converted or modified into other configurations, or separate configurations may be added.

[0085] In the above description, as illustrated in Fig. 3, the zero-phase-sequence voltage superimposer 19 mainly superimposes the zero-phase-sequence voltage so that the maximum value of the voltage command value is equal to the power supply voltage, thereby correcting the voltage command value of each phase. However, this is not limiting. As illustrated in Fig. 4, the zero-phase-sequence voltage superimposer 19 may also correct the voltage command value of each phase so that the minimum value of the voltage command value is equal to the reference potential. In this case, the current substituter 14 superimposes the voltage command value V u , V v , V w may be determined as the phase in which the first phase current is replaced with the second phase current. Furthermore, when the modulation factor is lower than a predetermined reference value of the modulation factor, the zero-phase sequence voltage superimposer 19 may correct the voltage command value of each phase so that the minimum value of the voltage command value becomes equal to the reference potential.

[0086] In calculating the phase current in a certain calculation period, the phase current calculator 23 calculates the capacitor current I c (T1), I c Instead of (T2), the capacitor current I detected in the calculation cycle immediately before or in a calculation cycle more recent than that c (T1), I c (T2) may be used. This allows the phase current to be calculated without significantly reducing the calculation accuracy even when the carrier wave period is not sufficiently small compared to the period of the phase current of the motor 1. In the calculation process in a certain calculation period, the current substituter 14, the settling determiner 21, the detection timing adjuster 22, and the phase current calculator 23 calculate the voltage command value V u , V v, V w Instead, the voltage command value V input in the immediately preceding calculation cycle or in a more recent calculation cycle is used. u , V v , V w may be used. As a result, even if the carrier wave period is not sufficiently small compared to the period of the phase current of the motor 1, the calculation processing of each section can be performed without a significant decrease in calculation accuracy. Note that the case where the carrier wave period is not sufficiently small compared to the period of the phase current of the motor 1 is, for example, when the period of the phase current of the motor 1 is 3 to 30 times, typically about 10 times, the carrier wave period.

[0087] 1 and 8 illustrate an example in which the controller 6 is one component in the motor control system S1, but this is not limiting. The controller 6 may be configured as a single motor control device, or may be configured integrally with another component, for example, the inverter 4.

[0088] In the above description, the motor 1 is used for assisting steering of a vehicle, but this is not a limitation. This embodiment can be applied to any device and for any purpose as long as it has a configuration including a motor and an inverter.

[0089] According to the steering control device, motor control device, motor control system, and motor control method disclosed herein, it is possible to suppress detection errors in the motor current.

[0090] S1...motor control system, 1...motor, 2...rotor position detector, 3...DC power supply, 4...inverter, 6...controller, 11...speed calculator, 12...first current detector, 13...second current detector, 14...current substituter, 15...coordinate converter, 16...current command calculator, 17...voltage command calculator, 18...coordinate converter, 19...zero-phase voltage superimposer, 20...PWM signal generator, 21...settling determiner, 22...detection timing adjuster, 23...phase current calculator

Claims

1. A motor control device comprising: a capacitor current detector that detects a capacitor current of an inverter; a zero-phase sequence voltage superimposer that calculates a new voltage command value by superimposing a zero-phase sequence voltage on a voltage command value for controlling a phase current of a motor; a phase current detector that detects a first phase current that is a phase current of the motor; a phase current calculator that calculates a second phase current that is a phase current of the motor from the capacitor current and the voltage command value on which the zero-phase sequence voltage is superimposed; a phase current replacer that replaces the first phase current of at least one phase with the second phase current; and a switching signal generator that generates a switching signal for controlling a switching element from the voltage command value on which the zero-phase sequence voltage is superimposed.

2. The motor control device according to claim 1, wherein the phase current substituter selects the phase for which the voltage command value is maximum as the phase for substituting the first phase current.

3. The motor control device according to claim 1, wherein the zero-phase sequence voltage superimposer superimposes the zero-phase sequence voltage so that the maximum value of the voltage command value for each phase becomes equal to the power supply voltage.

4. The motor control device according to claim 1, further comprising: a detection timing adjuster that adjusts the detection timing of the capacitor current; and a settling determiner that determines whether the capacitor current has settled, wherein the detection timing adjuster adjusts the detection timing based on the voltage command value.

5. The motor control device according to claim 4, wherein the detection timing adjuster adjusts the detection timing of the capacitor current to a timing when the capacitor current is settled.

6. The motor control device according to claim 4, characterized in that the settling determiner determines an output period of a zero voltage vector or a non-zero voltage vector based on the voltage command value, and determines that the capacitor current has settled if the output period exceeds a predetermined value.

7. The motor control device according to claim 4, wherein the phase current calculator calculates the second phase current when it is determined that the capacitor current has settled, and does not calculate the second phase current when it is determined that the capacitor current has not settled.

8. The motor control device according to claim 1, characterized in that the switching signal generator generates the switching signal for each carrier wave period, and for each calculation period having two switching periods, in one of the two switching periods, the phase current detector detects the first phase current, and in the other switching period, the capacitor current detector detects the capacitor current.

9. A motor control system comprising: the motor control device according to claim 1; and the motor.

10. A motor control method in a motor control device, comprising: detecting a capacitor current of an inverter; calculating a new voltage command value by superimposing a zero-phase sequence voltage on a voltage command value for controlling a phase current of the motor; detecting a first phase current which is the phase current of the motor; calculating a second phase current which is the phase current of the motor from the capacitor current and the voltage command value on which the zero-phase sequence voltage is superimposed; replacing the first phase current of at least one phase with the second phase current; and generating a switching signal for controlling a switching element from the voltage command value on which the zero-phase sequence voltage is superimposed.

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