High-frequency module and communication device equipped with same

Abstract

A high-frequency module (10) comprises: a terminal (T1) that receives a transmission signal; a terminal (T2) that is configured to be connectable to an antenna (ANT); a terminal (T3) that outputs a reception signal from the antenna; a transmission circuit (105); a reception circuit (106); and a hybrid coupler (130). The transmission circuit includes amplification circuits (140A, 140B). The reception circuit includes a reception line (155) for transmitting the signal received by the antenna to the terminal (T3). The hybrid coupler includes annularly connected lines (131-134). A connection node (N1) between the line (131) and the line (132) is connected to an output end of the amplification circuit (140A). A connection node (N2) between the line (131) and the line (133) is connected to an output end of the amplification circuit (140B). A connection node (N3) between the line (132) and the line (134) is connected to the terminal (T2). A connection node (N4) between the line (133) and the line (134) is connected to the reception line. A signal having a phase that is advanced by 90 degrees compared with that of a signal supplied to the connection node (N1) is supplied to the connection node (N2).

Description

High frequency module and communication device equipped with same

[0001] The present disclosure relates to a high-frequency module and a communication device equipped with the same, and more particularly to an isolation technique between a transmitting circuit and a receiving circuit in a communication circuit that uses a common radiating element for transmission and reception.

[0002] U.S. Patent No. 10,715,204 (Patent Document 1) discloses an electronic device that uses a common radiating element to transmit and receive radio frequency (RF) radio waves. The electronic device in Patent Document 1 is provided with a power amplifier for transmission and a low-noise amplifier for reception for a shared antenna, and is configured to switch between the transmission circuit and the reception circuit using a switch.

[0003] U.S. Pat. No. 1,071,5204

[0004] In the electronic device disclosed in Patent Document 1, a magnetic coupling transformer is configured to switch between transmission and reception coupling using a switch made of semiconductor. In such a configuration, even when the switch is in a non-conducting state, a portion of the high-frequency signal passes through the switch due to parasitic capacitance inevitably generated in the switch, which can reduce the isolation characteristics between the transmission circuit and the reception circuit. When the isolation characteristics between the transmission circuit and the reception circuit are reduced, transmission loss increases and can become a source of noise.

[0005] The present disclosure has been made to solve such problems, and its purpose is to improve the isolation characteristics between a transmitting circuit and a receiving circuit in a high-frequency module that transmits and receives using a common antenna.

[0006] A high-frequency module according to the present disclosure transmits and receives high-frequency signals via an antenna. The high-frequency module includes a first terminal for receiving a transmission signal, a second terminal connectable to the antenna, a third terminal for outputting a high-frequency signal received from the antenna, a transmission circuit, a reception circuit, and a hybrid coupler. The transmission circuit includes a first amplifier circuit and a second amplifier circuit for amplifying a signal received at the first terminal. The reception circuit includes a reception line for transmitting a signal received by the antenna to the third terminal. The hybrid coupler is connected to the transmission circuit, the reception circuit, and the second terminal. The hybrid coupler includes a first line, a second line, a third line, and a fourth line connected in a ring shape. A first connection node between the first line and the second line is connected to an output terminal of the first amplifier circuit. A second connection node between the first line and the third line is connected to an output terminal of the second amplifier circuit. A third connection node between the second line and the fourth line is connected to the second terminal. A fourth connection node between the third line and the fourth line is connected to the receiving line.

[0007] In the high-frequency module according to the present disclosure, by appropriately setting the phase of the output signal from each amplifier circuit and / or the operating state of each amplifier circuit during transmission and reception, it is possible to switch between transmission and reception without providing a switch for switching between transmission and reception. Therefore, in a high-frequency module that transmits and receives using a common antenna, it is possible to improve the isolation characteristics between the transmission circuit and the reception circuit.

[0008] 8 is a schematic configuration diagram of a communication device to which a high-frequency module according to a first embodiment is applied. FIG. 1 is a diagram for explaining the configuration of the power amplifier circuit in FIG. 1 during transmission. FIG. 2 is a diagram for explaining the configuration of the power amplifier circuit in FIG. 1 during reception. FIG. 3 is a diagram for explaining impedance in a first operating state. FIG. 4 is a diagram for explaining impedance in a second operating state. FIG. 5 is a diagram for explaining impedance in a third operating state. FIG. 6 is a diagram for explaining the relationship between output power and efficiency of the power amplifier circuit during transmission. FIG. 7 is a diagram for explaining the configuration of a power amplifier circuit in a high-frequency module according to a second embodiment and impedance in a first operating state. FIG. 8 is a diagram for explaining impedance in a second operating state of the power amplifier circuit of FIG. 8. FIG. 9 is a diagram for explaining impedance in a third operating state of the power amplifier circuit of FIG. 8.

[0009] Hereinafter, embodiments of the present disclosure will be described in detail with reference to the drawings. In the drawings, the same or corresponding parts are designated by the same reference numerals, and description thereof will not be repeated.

[0010] 1 is a schematic diagram of a communication device 1 to which a high-frequency module 10 according to an embodiment is applied. The communication device 1 is, for example, a mobile terminal such as a mobile phone, a smartphone, or a tablet, or a personal computer with a communication function.

[0011] 1 , a communication device 1 includes an antenna ANT, a high-frequency module 10, a baseband integrated circuit (BBIC) 20 constituting a baseband signal processing circuit, a radio frequency integrated circuit (RFIC) 30, and a power supply circuit 40. The high-frequency module 10 includes terminals T1 to T5, a bias control circuit 50, and a power amplifier circuit 100.

[0012] In general, the communication device 1 upconverts an intermediate frequency (IF) transmission signal TX transmitted from the BBIC 20 to a high frequency (radio frequency: RF) signal in the RFIC 30, amplifies the high frequency signal in the power amplifier circuit 100, and emits it from the antenna ANT. The communication device 1 also amplifies with low noise the high frequency signal received by the antenna ANT in the power amplifier circuit 100, downconverts the high frequency signal to an intermediate frequency (IF) signal in the RFIC 30, and transmits the intermediate frequency signal to the BBIC 20 as a received signal RX.

[0013] The RFIC 30 is an example of a signal processing circuit that processes a high-frequency signal. The RFIC 30 up-converts the intermediate frequency signal transmitted from the BBIC 20 into a high-frequency signal and outputs the generated high-frequency signal to the high-frequency module 10 via the terminal T1. The RFIC 30 also receives the high-frequency signal received by the antenna ANT from the high-frequency module 10 via the terminal T3.

[0014] The bias control circuit 50 receives a control signal CON from the RFIC 30 via a terminal T4. The bias control circuit 50 generates a bias signal BS based on the control signal CON and outputs it to the power amplifier circuit 100. The bias signal BS is a signal for controlling the magnitude and supply timing of the bias current of the amplifier included in the power amplifier circuit 100.

[0015] The power amplifier circuit 100 amplifies an input signal Tin received from the RFIC 30 via a terminal T1 to generate an output signal Tout, which is output to the antenna ANT. The power amplifier circuit 100 also amplifies a high-frequency signal Rin received by the antenna ANT to generate an output signal Rout, which is output to the RFIC 30 via a terminal T3.

[0016] The antenna ANT is, for example, a flat patch antenna. A terminal T2 of the high-frequency module 10 is connected to a feed point SP1 of the antenna ANT via a power supply line. The antenna ANT emits an output signal Tout, which is a high-frequency signal output from the high-frequency module 10, as a radio wave. The antenna ANT also receives radio waves from space, converts them into an electrical signal, and outputs the electrical signal to the RFIC 30.

[0017] The power supply circuit 40 is an example of a so-called digital tracker, and can supply power supply voltages Vcc of a plurality of different voltage levels to the power amplifier circuit 100. The power supply circuit 40 includes a multilevel power converter (MPC) 410, a power supply selection circuit 420, and a digital envelope tracker (digital ET) 430.

[0018] The MPC 410 includes a plurality of DC / DC converters, which are not shown in Fig. 1. The MPC 410 converts the battery voltage VB supplied from an external battery into a plurality of different voltage levels and supplies the voltages to the power supply selection circuit 420.

[0019] The digital ET 430 receives the I and Q waveform signals of the transmit signal from the BBIC 20 and tracks the envelope of the transmit signal in the digital ET mode. The digital ET 430 generates a selection signal SEL corresponding to the voltage level of the envelope of the transmit signal and outputs it to the power supply selection circuit 420.

[0020] The power supply selection circuit 420 selects a voltage corresponding to the selection signal SEL from the plurality of voltage levels supplied from the MPC 410, and supplies the selected voltage to the power amplifier circuit 100 via the terminal T5 as the power supply voltage Vcc.

[0021] (Detailed Configuration of Power Amplifier Circuit) Next, the detailed configuration of the power amplifier circuit 100 in the high-frequency module 10 will be described with reference to Fig. 2 and Fig. 3. Fig. 2 shows the configuration of the power amplifier circuit 100 during transmission, and Fig. 3 shows the configuration of the power amplifier circuit 100 during reception.

[0022] The power amplifier circuit 100 includes a transmission circuit 105, a reception circuit 106, and a hybrid coupler 130. The transmission circuit 105 includes a phase shift circuit 110 and amplifier circuits 140A and 140B. The reception circuit 106 includes a low noise amplifier (LNA) 150 and a reception line 155.

[0023] The phase-shift circuit 110 includes a branching circuit 115 and hybrid couplers 120A and 120B. The amplifier circuit 140A includes peak amplifiers 141A and 142A and a phase-shift line 135A. The amplifier circuit 140B includes a carrier amplifier 141B, a peak amplifier 142B and a phase-shift line 135B. The hybrid coupler 130 includes lines 131 to 134 connected in a ring shape.

[0024] The branch circuit 115 branches the input signal Tin received at the terminal T1 from the RFIC 30 into two paths with a phase difference, and outputs the branched signals to the hybrid couplers 120A and 120B, respectively. A signal whose phase is advanced by 90° relative to the signal supplied to the hybrid coupler 120A is supplied to the hybrid coupler 120B.

[0025] Each of the hybrid couplers 120A and 120B includes two input terminals and two output terminals. A signal from the branch circuit 115 is input to one input terminal of the hybrid coupler 120A. The other input terminal of the hybrid coupler 120A is connected to the ground potential GND. The two output terminals of the hybrid coupler 120A are output to the peak amplifiers 141A and 142A, respectively. A signal whose phase is advanced by 90° relative to the signal supplied to the peak amplifier 142A is supplied to the peak amplifier 141A.

[0026] A signal from the branch circuit 115 is input to one input terminal of the hybrid coupler 120B. The other input terminal of the hybrid coupler 120B is connected to the ground potential GND. The two output terminals of the hybrid coupler 120B are output to a carrier amplifier 141B and a peak amplifier 142B, respectively. A signal whose phase is advanced by 90° relative to the signal supplied to the peak amplifier 142B is supplied to the carrier amplifier 141B.

[0027] As described above, hybrid coupler 120B is supplied with a signal whose phase is 90° ahead of the signal supplied to hybrid coupler 120A, and as a result, peak amplifier 142B is supplied with a signal whose phase is 90° ahead of the signal supplied to peak amplifier 142A, and carrier amplifier 141B is supplied with a signal whose phase is 180° ahead of the signal supplied to peak amplifier 142A.

[0028] In the amplifier circuit 140A, one end of the phase-shift line 135A is connected to the output terminal of the peak amplifier 141A, and the other end is connected to the output terminal of the peak amplifier 142A. A connection node N5A between the output terminal of the peak amplifier 142A and the phase-shift line 135A is connected to a connection node N1 between the line 131 and the line 132 in the hybrid coupler 130. The amplifier circuit 140A also functions as a Doherty amplifier, with the peak amplifier 141A as a carrier amplifier and the peak amplifier 142A as a peak amplifier.

[0029] In the amplifier circuit 140B, one end of a phase-shift line 135B is connected to the output terminal of a carrier amplifier 141B, and the other end is connected to the output terminal of a peak amplifier 142B. A connection node N5B between the output terminal of the peak amplifier 142B and the phase-shift line 135B is connected to a connection node N2 between the line 131 and the line 133 in the hybrid coupler 130. The amplifier circuit 140B is a Doherty amplifier.

[0030] A connection node N3 between the line 132 and the line 134 of the hybrid coupler 130 is connected to the antenna ANT via a terminal T2. One end of a receiving line 155 of the receiving circuit 106 is connected to a connection node N4 between the line 133 and the line 134 of the hybrid coupler 130.

[0031] The other end of the reception line 155 is connected to the LNA 150. The reception line 155 transmits a signal received by the antenna ANT to the LNA 150. The LNA 150 amplifies the signal transmitted through the reception line 155 with low noise and outputs the amplified signal as an output signal Rout to the RFIC 30 via the terminal T3.

[0032] Each of the lines 131 to 134 of the hybrid coupler 130 and the phase-shift lines 135A and 135B of the amplifier circuit has a line length of approximately 1 / 4 wavelength of the center frequency of the high-frequency signal to be transmitted from the transmission circuit 105. Therefore, when the high-frequency signal passes through each of these lines, the phase of the high-frequency signal is delayed by approximately 90°.

[0033] The load impedance of the antenna ANT is Z ０ / 2, the impedance of the lines 131 and 134 of the hybrid coupler 130 is Z ０ and the impedance of the lines 132 and 133 is set to Z ０ The impedance of the phase shift lines 135A and 135B is set to 2Z / √2. ０ is set to

[0034] In the amplifier circuit 140A, when both the peak amplifiers 141A and 142A are driven, the signal amplified by the peak amplifier 141A is 90° ahead in phase with the signal amplified by the peak amplifier 142A. Therefore, the signal amplified by the peak amplifier 141A passes through the phase shift line 135A and is combined at the connection node N5A in phase with the signal amplified by the peak amplifier 142A.

[0035] Similarly, in amplifier circuit 140B, when both carrier amplifier 141B and peak amplifier 142B are driven, the signal amplified by carrier amplifier 141B is 90° ahead in phase with the signal amplified by peak amplifier 142B. Therefore, the signal amplified by carrier amplifier 141B passes through phase shift line 135B and is combined at connection node N5B in phase with the signal amplified by peak amplifier 142B. If the phase of the signal output from amplifier circuit 140A is 0°, the phase of the signal output from amplifier circuit 140B will be 90°.

[0036] The signal supplied from amplifier circuit 140A to connection node N1 passes through line 132 and therefore has a phase of −90° at connection node N3. The signal supplied from amplifier circuit 140B to connection node N2 is delayed in phase by 180° as it passes through lines 133 and 134, and therefore has a phase of −90° at connection node N3. Therefore, at connection node N3, the signal supplied from amplifier circuit 140A and the signal supplied from amplifier circuit 140B are combined in phase.

[0037] At connection node N4, the signal supplied from amplifier circuit 140A to connection node N1 has a phase of -180° after passing through lines 132 and 134. On the other hand, the signal supplied from amplifier circuit 140B to connection node N2 has a phase of 0° after passing through line 133. That is, at connection node N4, the signal supplied from amplifier circuit 140A and the signal supplied from amplifier circuit 140B are combined in opposite phases. As a result, the amplitudes of the two signals cancel each other out at connection node N4, resulting in a no-signal state for input signal Tin.

[0038] In this way, during transmission, the input signal Tin input from the RFIC 30 to the power amplifier circuit 100 is transmitted to the antenna ANT via the transmission circuit 105 and the hybrid coupler 130 (arrow AR1), but is not transmitted to the reception circuit 106. Therefore, isolation between the transmission circuit 105 and the reception circuit 106 is achieved.

[0039] 3, the amplifiers of the amplifier circuits 140A and 140B are deactivated by the bias control circuit 50. As a result, no signals are supplied from the amplifier circuits 140A and 140B to the hybrid coupler 130, and therefore the input signal Tin is not transmitted to the antenna ANT.

[0040] At this time, connection nodes N5A and N5B, which are the output terminals of amplifier circuits 140A and 140B, are shorted. Therefore, the impedance when looking at line 132 from connection node N3 and the impedance when looking at line 133 from connection node N4 are both open. As a result, the high-frequency signal received by antenna ANT is transmitted only to reception line 155 and LNA 150 of receiver circuit 106 via line 134 of hybrid coupler 130, as indicated by arrow AR2, and is not transmitted to transmitter circuit 105. Therefore, isolation between transmitter circuit 105 and receiver circuit 106 is achieved.

[0041] In recent years, development of communications in the so-called sub-terahertz frequency band exceeding 100 GHz has been progressing for the purpose of large-capacity, high-speed communications. However, signals in such frequency bands tend to have greater loss in the signal path than signals in millimeter waves and lower frequency bands. By adopting the configuration of power amplifier circuit 100 according to the first embodiment in a communication device using such a sub-terahertz frequency band, it is possible to achieve isolation between the transmitter circuit and the receiver circuit without using an active element such as a switch, even when a common antenna is used for transmission and reception. Therefore, it is possible to ensure isolation between the transmitter circuit and the receiver circuit while reducing transmission loss.

[0042] (Increasing the Back-Off Amount) Signals in higher frequency bands than conventionally used frequency bands, such as the sub-terahertz frequency band described above, tend to be more susceptible to attenuation than signals in lower frequency bands. Therefore, in a power amplifier circuit, it is necessary to make the power supply voltage follow changes in an input signal with a wide modulation bandwidth. To meet this requirement, a digital ET mode is generally used, in which the power supply voltage is set to multiple discrete, different voltage levels within one frame.

[0043] A typical amplifier has a tendency for its efficiency to decrease as the input signal decreases. Therefore, if the power level of the input signal decreases in digital ET mode, the amplifier's efficiency may decrease. On the other hand, an amplifier also has a tendency for its efficiency to increase as the load impedance increases. A Doherty amplifier is known as a configuration that takes advantage of this characteristic to increase amplification efficiency.

[0044] A Doherty amplifier generally has a configuration in which a carrier amplifier and a peaking amplifier are connected in parallel between an input terminal and an output terminal, and a phase-shift line functioning as an impedance inverter is arranged between the carrier amplifier and a combining node. When the output power is low, only the carrier amplifier operates, and when the output power exceeds a predetermined value, both the carrier amplifier and the peaking amplifier operate. When the peaking amplifier is deactivated, the load impedance of the carrier amplifier increases. Therefore, by switching between activation and deactivation of the peaking amplifier in response to changes in the power level of the input signal in digital ET mode, the Doherty amplifier can achieve a back-off of approximately 6 dB.

[0045] However, the back-off amount required for the power amplifier circuit may vary depending on, for example, the modulation scheme, the set voltage of the digital ET, and / or the target bandwidth, and therefore, depending on the specifications required for the device to which the power amplifier circuit is applied, a back-off amount greater than 6 dB may be required.

[0046] In the power amplifier circuit 100 according to the first embodiment, each of the amplifier circuits 140A and 140B has a Doherty amplifier configuration. Therefore, by setting the peak amplifier 142A in the amplifier circuit 140A and the peak amplifier 142B in the amplifier circuit 140B to a non-driven state, a back-off amount of 6 dB can be achieved.

[0047] When the peak amplifiers 142A and 142B are in a non-driven state, a Doherty amplifier is further configured by the carrier amplifier 141B of the amplifier circuit 140B, the peak amplifier 141A of the amplifier circuit 140A, and the line 131 of the hybrid coupler 130. Therefore, by putting the peak amplifier 141A of the amplifier circuit 140A in a non-driven state, a back-off amount of 15 dB can be achieved.

[0048] In the power amplifier circuit 100 of the first embodiment, each of the two amplifier circuits 140A, 140B is configured with a Doherty amplifier, and another Doherty amplifier is configured by the amplifier circuits 140A, 140B and the hybrid coupler 130. By configuring the Doherty amplifier in two stages in this way, a back-off amount greater than 6 dB can be achieved.

[0049] (Operation of the Transmitter Circuit) The operation of the transmitter circuit 105 during transmission will be described in detail below with reference to Figures 4 to 7. Figures 4 to 6 are diagrams for explaining the impedance within the circuit in different operating states of the transmitter circuit 105. Note that the phase shift circuit 110 in the transmitter circuit 105 is omitted from Figures 4 to 6.

[0050] The operating state of the transmitter circuit 105 changes depending on the power supply voltage Vcc. In the example of the first embodiment, the power supply voltage Vcc is switched between three stages, V1, V2, and V3 (V3<V2<V1), by the digital ET. When Vcc=V1, it is a first operating state, when Vcc=V2, it is a second operating state, and when Vcc=V3, it is a third operating state.

[0051] The first operating state (FIG. 4) corresponds to a state where the power level of the input signal Tin is greater than the first power value. In the first operating state, a large amount of power is required, so all of the peak amplifiers 141A, 142A, and 142B are driven, and amplification is performed using four amplifiers, including the carrier amplifier 141B.

[0052] The second operating state (FIG. 5) corresponds to a state in which the power level of the input signal Tin is a second power value that is smaller than the first power value. In the second operating state, the peak amplifiers 142A and 142B are deactivated, and the peak amplifier 141A is activated. That is, amplification is performed by the peak amplifier 141A of the amplifier circuit 140A and the carrier amplifier 141B of the amplifier circuit 140B.

[0053] The third operating state (FIG. 6) corresponds to a state where the power level of the input signal Tin is a third power value that is even lower than the second power value. In the third operating state, the peak amplifiers 141A, 142A, and 142B are in a non-driven state, and amplification is performed only by the carrier amplifier 141B.

[0054] 4 is a diagram for explaining the impedance in the power amplifier circuit 100 in the first operating state. As explained in FIG. 2, when the load impedance of the antenna ANT is Z ０ / 2, the impedance of the lines 131 and 134 of the hybrid coupler 130 is Z ０ and the impedance of the lines 132 and 133 is set to Z ０ The impedance of the phase shift lines 135A and 135B is set to 2Z / √2. ０ is set to

[0055] Since the signal path branches in two directions at the connection node N3, the load impedance seen from the branch path to the line 132 and the load impedance seen from the branch path to the line 134 are both Z ０ This becomes:

[0056] The impedance of the line 132 is Z ０ / √2, the impedance when looking at the line 132 from the connection node N1 is Z ０ Since the signal path branches in two directions at the connection node N1, the load impedance seen from the branch path to the amplifier circuit 140A and the load impedance seen from the branch path to the line 131 are both Z ０ This becomes:

[0057] In the amplifier circuit 140A, the signal path branches into two directions at the connection node N5A, and the impedance of the phase shift line 135A is 2Z. ０ Therefore, the impedance when looking at the load side from the output terminals of the peak amplifiers 141A and 142A is 2Z ０ This becomes:

[0058] The impedance of the line 134 of the hybrid coupler 130 is Z ０ Therefore, the impedance when looking at the line 134 from the connection node N4 is also Z ０ Here, as described above, since no signal is transmitted from the connection node N4 to the receiving line 155 and the state is essentially open, the load impedance when the connection node N4 is viewed from the branch path to the line 133 is Z ０ It will remain as it is.

[0059] As a result, similar to the path on the amplifier circuit 140A side, the impedance when looking at the load side from the output terminals of the carrier amplifier 141B and the peak amplifier 142B is 2Z. ０ This becomes:

[0060] 5 is a diagram illustrating the impedance within the power amplifier circuit 100 in the second operating state. In the second operating state, the peak amplifier 142A in the amplifier circuit 140A and the peak amplifier 142B in the amplifier circuit 140B are in a non-driven state. That is, the impedance when the peak amplifier 142A is viewed from the connection node N5A and the impedance when the peak amplifier 142B is viewed from the connection node N5B are in an open state.

[0061] As a result, the impedance seen from the branch path to the phase shift line 135A toward the load side and the impedance seen from the branch path to the phase shift line 135B toward the load side are both Z ０ Then, the impedance of the phase shift lines 135A and 135B is 2Z ０ Therefore, the impedance when looking at the load side from the output terminals of the peak amplifier 141A and the carrier amplifier 141B is 4Z ０ This becomes:

[0062] In this way, in the second operating state, the number of amplifiers is halved compared to the first operating state, and the load impedance of the operating amplifiers is doubled, resulting in a back-off amount of 6 dB.

[0063] 6 is a diagram illustrating the impedance within the power amplifier circuit 100 in the third operating state. In the third operating state, in addition to the second operating state, the peak amplifier 141A of the amplifier circuit 140A is deactivated. This causes the impedance of the peak amplifier 141A seen from the phase-shift line 135A to be open, resulting in the impedance of the connection node N1 being shorted. Therefore, the impedance of the line 131 seen from the connection node N2 and the impedance of the line 132 seen from the connection node N3 are both open.

[0064] Then, the impedance of the branch path from the connection node N3 to the line 134 is Z ０ / 2, so the impedance at the connection node N4 is 2Z ０ The impedance at the connection nodes N2 and N5B is Z ０ The impedance of the phase shift line 135B is 2Z / 4. ０ Therefore, the impedance when looking at the load side from the output terminal of the carrier amplifier 141B is 16Z. ０ That is, in the third operating state, the impedance of the carrier amplifier 141B is four times as large as that in the second operating state.

[0065] In this way, in the third operating state, the number of amplifiers is halved compared to the second operating state, and the load impedance of the operating amplifiers is four times greater, so an additional 9 dB of back-off can be obtained, resulting in a total back-off of 15 dB compared to the first operating state.

[0066] FIG. 7 is a diagram for explaining the relationship between the output power and efficiency during transmission of the power amplifier circuit 100. In FIG.

[0067] In FIG. 7, the horizontal axis represents the power level of the output signal Tout output from the terminal T2 to the antenna ANT, and the vertical axis represents the efficiency of the power amplifier circuit 100.

[0068] 7, the dashed line LN2 is a graph showing the results when the peak amplifiers 141A, 142A, and 142B and the carrier amplifier 141B are driven in a first operating state in which the four class AB amplifiers corresponding to these amplifiers are driven in average power tracking (APT) mode. The APT mode is a mode in which the power supply voltage level is set based on the average output power over a predetermined period of time.

[0069] The dashed-dotted line LN3 is a graph showing a case where switching is performed between the first operating state and a second operating state in which the peak amplifier 141A and the carrier amplifier 141B are driven and the peak amplifiers 142A and 142B are deactivated. The dashed-two-dotted line LN4 is a graph showing a case where switching is performed between the first operating state and a third operating state in which only the carrier amplifier 141B is driven and the peak amplifiers 141A, 142A, and 142B are deactivated. The solid line LN1 is a graph showing a case where switching is performed between the first operating state, the second operating state, and the third operating state according to the power level of the input signal Tin as in the first embodiment.

[0070] In the third operating state, the region until the amplifier reaches saturation is designated as RG3, in the second operating state, the region until the amplifier reaches saturation is designated as RG2, and in the first operating state, the region until the amplifier reaches saturation is designated as RG1. The power supply voltage Vcc in the first operating state is set to V1, the power supply voltage Vcc in the second operating state is set to V2, and the power supply voltage Vcc in the third operating state is set to V3 (V1>V2>V3).

[0071] In region RG1, the parallel operation of the two amplifiers included in each of the two Doherty amplifiers of amplifier circuits 140A and 140B improves efficiency compared to the case of a class AB amplifier (dashed line LN2).

[0072] In the region RG2 where the power level of the input signal Tin drops from the first power value to the second power value, as described in FIG. 5, the peak amplifier 142A in the amplifier circuit 140A and the peak amplifier 142B in the amplifier circuit 140B are stopped, and therefore the impedance of the peak amplifier 142A and the carrier amplifier 141B in the driven state increases, and the efficiency of each amplifier increases.

[0073] In the region RG3 where the power level of the input signal Tin drops to the third power value, the peak amplifier 141A of the amplifier circuit 140A is stopped, and the impedance of the carrier amplifier 141B in the driven state further increases, thereby improving the efficiency of the carrier amplifier 141B.

[0074] Therefore, in the power amplifier circuit 100 of the first embodiment, a back-off amount of 15 dB can be achieved by stopping some of the amplifiers as the power level of the input signal Tin decreases.

[0075] As described above, by adopting a configuration in which two amplifier circuits are connected to the two input terminals of a hybrid coupler and the two output terminals of the hybrid coupler are connected to an antenna and a receiving circuit, respectively, it is possible to achieve isolation between the transmitting circuit and the receiving circuit while preventing loss reduction due to switches.

[0076] Furthermore, by configuring each of the two amplifier circuits of the transmission circuit as a Doherty amplifier and stopping some of the amplifiers in the amplifier circuits in response to a decrease in the power level of the input signal, a back-off amount greater than 6 dB can be achieved.

[0077] The "amplifier circuit 140A," the "amplifier circuit 140B," and the "LNA 150" in the first embodiment correspond to the "first amplifier circuit," the "second amplifier circuit," and the "third amplifier circuit," respectively, in the present disclosure. The "terminals T1 to T3" in the first embodiment correspond to the "first terminal," the "second terminal," and the "third terminal," respectively, in the present disclosure. The "lines 131 to 134" in the first embodiment correspond to the "first line" to the "fourth line," respectively, in the present disclosure. The "connection node N1" to the "connection node N4" in the first embodiment correspond to the "first connection node" to the "fourth connection node," respectively, in the present disclosure. The "peak amplifier 141A," the "peak amplifier 142A," the "carrier amplifier 141B," and the "peak amplifier 142B" in the first embodiment correspond to the "first amplifier" to the "fourth amplifier," respectively, in the present disclosure. The "phase shift line 135A" and the "phase shift line 135B" in the first embodiment correspond to the "first phase shift line" and the "second phase shift line" in the present disclosure, respectively.

[0078] [Embodiment 2] In the first embodiment, a configuration was described in which two amplifier circuits of a transmission circuit are directly connected to the input terminals of a hybrid coupler. In the second embodiment, a configuration will be described in which two amplifier circuits of a transmission circuit are connected to a hybrid coupler via a transformer. The first embodiment corresponds to a configuration in which signals from the two amplifier circuits are combined using currents. On the other hand, the configuration using a transformer in the second embodiment corresponds to a configuration in which two signals are combined using voltages.

[0079] 8 to 10 are diagrams illustrating the configuration of the power amplifier circuit 100A in the high-frequency module according to embodiment 2 and the impedance in each operating state. Figures 8 to 10 correspond to Figures 4 to 6 in embodiment 1. Note that the phase shift circuit 110 is also omitted in Figures 8 to 10.

[0080] 8, a power amplifier circuit 100A is configured such that amplifier circuits 145A and 145B and phase shift lines 136A and 136B are provided instead of amplifier circuits 140A and 140B in power amplifier circuit 100 of the first embodiment.

[0081] The amplifier circuit 145A includes peak amplifiers 141A and 142A, a phase shift line 137A, and a transformer TR1. The amplifier circuit 145B includes a carrier amplifier 141B, a peak amplifier 142B, a phase shift line 137B, and a transformer TR2.

[0082] Each of the transformers TR1 and TR2 is an isolated transformer in which the winding ratio between the primary winding and the secondary winding is set to 1:m. One end of the primary winding of the transformer TR1 is connected to the output terminal of the peak amplifier 141A. The other end of the primary winding of the transformer TR1 is connected to the output terminal of the peak amplifier 142A via a phase shift line 137A. One end of the secondary winding of the transformer TR1 is connected to a connection node N1 in the hybrid coupler 130 via a phase shift line 136A. The other end of the secondary winding of the transformer TR1 is connected to the ground potential GND.

[0083] One end of the primary winding of the transformer TR2 is connected to the output terminal of the carrier amplifier 141B. The other end of the primary winding of the transformer TR2 is connected to the output terminal of the peak amplifier 142B via a phase shift line 137B. One end of the secondary winding of the transformer TR2 is connected to the connection node N2 of the hybrid coupler 130 via a phase shift line 136B. The other end of the secondary winding of the transformer TR2 is connected to the ground potential GND.

[0084] Each of the phase shift lines 136A, 136B, 137A, and 137B has a line length of approximately 1 / 4 wavelength of the center frequency of the high frequency signal to be transmitted. ０ / 2, both are Z ０ is set to

[0085] In the first operating state (FIG. 8), the amplifiers in the amplifier circuits 145A and 145B are in a driving state. In this state, the load impedance of the branch line from the connection node N1 to the phase shift line 136A and the load impedance of the branch line from the connection node N2 to the phase shift line 136B are both Z ０ Therefore, the impedance when looking at the load side from the secondary windings of the transformers TR1 and TR2 is also Z０ Since the winding ratio of the transformers TR1 and TR2 is 1:m, the impedance when looking at the load side from the output terminal of each amplifier in the amplifier circuits 145A and 145B is Z ０ / 2m ２ This becomes:

[0086] In the second operating state (FIG. 9), the peak amplifier 142A of the amplifier circuit 145A and the peak amplifier 142B of the amplifier circuit 145B are in a non-driven state. As a result, the impedance of the output terminals of the peak amplifiers 142A and 142B is in an open state, and therefore the connection node between the phase shift line 137A and the primary winding of the transformer TR1 and the connection node between the phase shift line 137B and the primary winding of the transformer TR2 are in a short state. Therefore, the impedance when looking at the load side from the output terminal of the peak amplifier 141A in the amplifier circuit 145A and the output terminal of the carrier amplifier 141B in the amplifier circuit 145B is Z ０ / m ２ This becomes:

[0087] Therefore, in the second operating state, the number of amplifiers is halved compared to the first operating state, and the load impedance of the operating amplifiers is doubled, so that a back-off of 6 dB can be achieved.

[0088] In the third operating state (FIG. 10), the peak amplifier 141A of the amplifier circuit 145A is further deactivated from the second operating state. That is, the entire amplifier circuit 145A is deactivated, and the impedance at the output terminal of the amplifier circuit 145A is open. As a result, the impedance at the connection node N1 is shorted, and the impedance when looking at the line 131 from the connection node N2 and the impedance when looking at the line 132 from the connection node N3 are both open.

[0089] In this state, as in the first embodiment, the load impedance at the connection node N2 is Z ０ / 4, the load impedance at the secondary winding of the transformer TR2 is 4Z / 4 due to the phase shift line 136B. ０As a result, the impedance when looking at the load side from the output terminal of the carrier amplifier 141B is 4Z. ０ / m ２ This is four times as much as in the second operating state.

[0090] Therefore, in the third operating state, the number of amplifiers is halved compared to the second operating state, and the load impedance of the operating amplifiers is four times greater, so an additional 9 dB back-off can be achieved, resulting in a total back-off of 15 dB.

[0091] As described above, even in a configuration in which signals are combined using voltages using a transformer as in Embodiment 2, by connecting two amplifier circuits to two input terminals of a hybrid coupler and connecting the two output terminals of the hybrid coupler to an antenna and a receiver circuit, respectively, it is possible to achieve isolation between the transmitter circuit and the receiver circuit while preventing loss reduction due to switches. Furthermore, by configuring each amplifier circuit with a Doherty amplifier and further configuring a Doherty amplifier with two amplifier circuits and a hybrid coupler, it is possible to achieve a back-off amount greater than 6 dB.

[0092] The "phase shift line 136A," "phase shift line 136B," "phase shift line 137A," and "phase shift line 137B" in the second embodiment correspond to the "third phase shift line" to the "sixth phase shift line" in the present disclosure, respectively. The "transformer TR1" and the "transformer TR2" in the second embodiment correspond to the "first transformer" and the "second transformer" in the present disclosure, respectively.

[0093] The embodiments disclosed herein should be considered to be illustrative in all respects and not restrictive. The scope of the present invention is defined by the claims, not by the description of the above embodiments, and is intended to include all modifications within the meaning and scope of the claims.

[0094] 1 Communication device, 10 High frequency module, 20 BBIC, 30 RFIC, 40 Power supply circuit, 50 Bias control circuit, 100, 100A Power amplifier circuit, 105 Transmitting circuit, 106 Receiving circuit, 110 Phase shift circuit, 115 Branch circuit, 120A, 120B, 130 Hybrid coupler, 131 to 134 Lines, 135A to 137A, 135B to 137B Phase shift lines, 140A, 140B, 145A, 145B Amplifying circuit, 141A, 142A, 142B Peak amplifier, 141B Carrier amplifier, 150 LNA, 155 Receiving line, 410 MPC, 420 Power supply selection circuit, 430 Digital ET, ANT Antenna, GND Ground potential, N1 to N4, N5A, N5B Connection node, RG1 to RG3 areas, SP1 power supply point, T1 to T5 terminals, TR1, TR2 transformers, VB battery voltage, Vcc power supply voltage.

Claims

1. A high-frequency module that transmits and receives high-frequency signals via an antenna, comprising: a first terminal that receives a transmission signal; a second terminal configured to be connectable to the antenna; a third terminal that outputs the high-frequency signal received from the antenna; a transmission circuit including a first amplifier circuit and a second amplifier circuit that amplify the signal received at the first terminal; a reception circuit including a reception line that transmits the signal received by the antenna to the third terminal; and a hybrid coupler connected to the transmission circuit, the reception circuit, and the second terminal, wherein the hybrid coupler includes a first line, a second line, a third line, and a fourth line connected in a ring shape, a first connection node between the first line and the second line being connected to an output terminal of the first amplifier circuit, a second connection node between the first line and the third line being connected to an output terminal of the second amplifier circuit, a third connection node between the second line and the fourth line being connected to the second terminal, and a fourth connection node between the third line and the fourth line being connected to the reception line.

2. The high frequency module according to claim 1, wherein, when transmitting a signal from the antenna, the transmitting circuit supplies to the second connection node a signal whose phase is 90° ahead of the signal supplied to the first connection node, and when receiving a signal from the antenna, the transmitting circuit is set to a non-driven state.

3. The high frequency module according to claim 1 or 2, wherein the receiving circuit further includes a third amplifier circuit that amplifies the signal from the receiving line and outputs the amplified signal to the third terminal.

4. The radio frequency module according to any one of claims 1 to 3, wherein the first amplifier circuit is a Doherty amplifier including a first amplifier, a second amplifier, and a first phase shift line connected between the output terminal of the first amplifier and the output terminal of the second amplifier; the second amplifier circuit is a Doherty amplifier including a third amplifier, a fourth amplifier, and a second phase shift line connected between the output terminal of the third amplifier and the output terminal of the fourth amplifier; the output terminal of the second amplifier is connected to the first connection node; and the output terminal of the fourth amplifier is connected to the second connection node.

5. The amplifier further comprises a third phase-shift line connected between the output terminal of the first amplifier circuit and the first connection node, and a fourth phase-shift line connected between the output terminal of the second amplifier circuit and the second connection node, wherein the first amplifier circuit comprises a first amplifier, a second amplifier, a fifth phase-shift line, and a first transformer, wherein the second amplifier circuit comprises a third amplifier, a fourth amplifier, a sixth phase-shift line, and a second transformer, wherein one end of a primary winding of the first transformer is connected to the output terminal of the first amplifier, and the other end is connected to the output terminal of the second amplifier via the fifth phase-shift line, wherein one end of a secondary winding of the first transformer is connected to the third phase-shift line, and the other end is connected to ground potential, wherein one end of a primary winding of the second transformer is connected to the output terminal of the third amplifier, and the other end is connected to the output terminal of the fourth amplifier via the sixth phase-shift line, 4. The high-frequency module according to claim 1, wherein one end of a secondary winding of the second transformer is connected to the fourth phase-shift line, and the other end is connected to the ground potential.

6. The high-frequency module according to claim 4 or claim 5, wherein the transmission circuit further includes a phase-shift circuit that applies a phase difference to the transmission signal received at the first terminal and supplies the signal to the first amplifier circuit and the second amplifier circuit, and wherein, when transmitting a signal from the antenna, the phase-shift circuit supplies to the first amplifier and the fourth amplifier a signal whose phase is 90° ahead of the signal supplied to the second amplifier, and supplies to the second amplifier a signal whose phase is 180° ahead of the signal supplied to the second amplifier.

7. The radio frequency module according to claim 6, wherein when the power level of the transmission signal is equal to or greater than a first threshold, the first amplifier, the second amplifier, the third amplifier, and the fourth amplifier are driven; when the power level of the transmission signal is less than the first threshold and equal to or greater than a second threshold, the first amplifier and the third amplifier are driven, while the second amplifier and the fourth amplifier are deactivated; and when the power level of the transmission signal is less than the second threshold, the third amplifier is driven, while the first amplifier, the second amplifier, and the fourth amplifier are deactivated.

8. A high-frequency module according to any one of claims 4 to 7, wherein when a signal is received from the antenna, the first amplifier to the fourth amplifier are all in a non-driven state.

9. A communication device comprising: a high-frequency module according to any one of claims 1 to 8; the antenna; and a signal processing circuit configured to process the transmission signal supplied to the high-frequency module and the reception signal received by the antenna.

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