Current source type LCCL double-frequency resonance system and parameter matching method thereof

Abstract

The invention discloses a current source type LCCL double-frequency resonance system and a parameter matching method thereof, which are suitable for synchronous double-frequency induction quenching of complex workpieces such as gears. The system is composed of two current type resonant inverters working at a low frequency and a high frequency respectively and a dual-frequency LCCL resonant cavity, and dual-frequency current superposition is realized through parallel output. The double-frequency LCCL resonant cavity has two parallel resonance points, a mode based on simplified topology is adopted to simplify parameter design, and meanwhile design errors are maintained at a low level. A power matching coefficient is introduced, the ratio of high-frequency output power to low-frequency output power is adjusted by adjusting the value of the power matching coefficient, and the problem of double-frequency power unbalance caused by frequency variation characteristics of load equivalent resistance and the like is solved. The superposed double-frequency current forms a low-frequency envelope embedded high-frequency oscillation composite waveform on the induction heating coil, cooperative control of tooth root deep quenching (low frequency) and tooth surface hard quenching (high frequency) is achieved, and the method has the advantages of being simple in parameter design, high in power decoupling performance, high in operation stability and the like.

Description

TECHNICAL FIELD

[0001] The application relates to the field of induction heating, and in particular to a current source type LCCL double-frequency resonant system and a parameter matching method thereof. BACKGROUND

[0002] Induction heating is a technology for realizing non-contact heating by exciting eddy current in conductive materials through an alternating magnetic field, and is widely applied to industrial fields such as metal heat treatment, welding and smelting. Compared with traditional heating methods, induction heating has the advantages of high efficiency, fast heating, energy saving and environmental protection, high automation and the like, and has important application value in surface quenching of parts such as gears and shafts. In the application of surface quenching of mechanical parts such as gears, sprockets and springs with complex surface geometries, in order to simultaneously consider the heating depth and hardening effect of different parts such as tooth tips and tooth roots, a single-frequency induction heating scheme has been difficult to meet the requirements. Research has found that low-frequency (Low Frequency, LF) current can effectively penetrate the tooth root area, helping to improve fatigue strength, while high-frequency (High Frequency, HF) current is suitable for rapid heating of the tooth surface to enhance its wear resistance. If only low-frequency heating is used, the whole tooth body may be overheated, affecting its toughness; if only high-frequency heating is used, the tooth root area is difficult to effectively heat. Therefore, the simultaneous action of low-frequency and high-frequency induction currents is an ideal solution for surface quenching of complex profile workpieces.

[0003] Most of the current mainstream double-frequency induction heating systems adopt a voltage source type inverter (Voltage-Source Inverter, VSI) structure, and realize the injection and control of low-frequency and high-frequency power through multi-resonant circuits, PWM modulation or multi-level technology. However, the structure of the voltage source type inverter has problems such as parasitic oscillation prone to occur in the system, complex control, and difficulty in constant output current. Especially in the aspect of multi-frequency simultaneous output control, the system faces challenges in high coupling degree, steady-state and dynamic control, etc. In comparison, the current source type induction heating system naturally has the advantages of suppressing short-circuit current and controllable output current waveform in theory. However, at present, the research on the current source type double-frequency induction heating topology structure is still relatively scarce, and most of the existing work focuses on the voltage source type structure, and the current source type double-frequency induction heating system lacks systematic research on structure design, current decoupling, power control, load resonant tuning and the like. SUMMARY

[0004] Based on the above background, the application develops a current source type double-frequency induction heating system which is compact in structure, frequency-independent controllable, flexible in power distribution and suitable for induction heat treatment of complex tooth-shaped workpieces, and has important significance for improving equipment performance and reducing operation and maintenance costs.

[0005] As an aspect of the present application, a current source type LCCL dual-frequency resonant system is provided, comprising a first current source type resonant inverter, a second current source type resonant inverter and a dual-frequency LCCL resonant cavity.

[0006] The first current source type resonant inverter is a low-frequency (LF) inverter operating at a low-frequency resonant frequency, and the second current source type resonant inverter is a high-frequency (HF) inverter operating at a high-frequency resonant frequency.

[0007] The dual-frequency LCCL resonant cavity has two parallel resonant frequency points and has a first input port and a second input port, the first current source type resonant inverter is connected to the first input port, and the second current source type resonant inverter is connected to the second input port.

[0008] Preferably, the first current source type resonant inverter and the second current source type resonant inverter each comprise a first IGBT tube, a second IGBT tube, a third IGBT tube, a fourth IGBT tube, and four diodes connected in series with the first IGBT tube, the second IGBT tube, the third IGBT tube and the fourth IGBT tube, respectively.

[0009] The first IGBT tube and the third IGBT tube constitute a first series branch, the second IGBT tube and the fourth IGBT tube constitute a second series branch, and the first series branch and the second series branch are connected in parallel.

[0010] Preferably, an input inductance Lm is connected in series at the input side of the first current source type resonant inverter and the second current source type resonant inverter.

[0011] Preferably, the dual-frequency LCCL resonant cavity comprises a first inductance L1, a second inductance L2, a first capacitance C1, a second capacitance C2 and an internal resistance R.

[0012] The first input port and the second input port share a first common terminal and a second common terminal, a first terminal of the first inductance L1 is connected to the first common terminal, and a second terminal of the first inductance L1 is connected to the second common terminal.

[0013] A first terminal of the first capacitance C1 is connected to the first common terminal, a second terminal of the first capacitance C1 is connected to a first terminal of the second capacitance C2, and a second terminal of the second capacitance C2 is connected to the second common terminal.

[0014] A first terminal of the second inductance L2 is connected to the second terminal of the first capacitance C1 and the first terminal of the second capacitance C2, a second terminal of the second inductance L2 is connected to a first terminal of the internal resistance R, and a second terminal of the internal resistance R is connected to the second common terminal.

[0015] Preferably, the two parallel resonance frequency points of the dual-frequency LCCL resonant cavity are ω1 and ω2, respectively.

[0016]

[0017] Preferably, the system has three working modes, namely a low-frequency working mode, a high-frequency working mode, and a dual-frequency superposition working mode.

[0018] Preferably, the dual-frequency superposition working mode is as follows:

[0019] The working frequency of the first current-mode resonant inverter corresponds to the low-frequency resonance frequency point of the dual-frequency LCCL resonant cavity, and the working period is T1 = 1 / f1. The working frequency of the second current-mode resonant inverter corresponds to the high-frequency resonance frequency point of the dual-frequency LCCL resonant cavity, and the working period is T2 = 1 / f2.

[0020] In each low-frequency resonance period T1, the controller switches the diagonal switch tubes of the first current-mode resonant inverter at a frequency of f1. In the first half of the resonance period, the first IGBT tube and the fourth IGBT tube are turned on, and the second IGBT tube and the third IGBT tube are turned off. In the second half of the resonance period, the second IGBT tube and the third IGBT tube are turned on, and the first IGBT tube and the fourth IGBT tube are turned off.

[0021] At the same time, the controller switches the diagonal switch tubes of the second current-mode resonant inverter at a frequency of f2, thereby embedding N high-frequency resonance periods T2 into a low-frequency resonance period T1, where f2 = Nf1.

[0022] As another aspect of the present application, a parameter matching method for a current source type LCCL dual-frequency resonant system is further provided, comprising the following steps:

[0023] S1: Set the target values of the two parallel resonance frequency points ω1 and ω2 of the dual-frequency LCCL resonant cavity, measure the inductance value L2 of the second inductance L2 of the dual-frequency LCCL resonant cavity, and determine the load equivalent resistance R ω1 and the load equivalent resistance R ω2 when the target value of the parallel resonance frequency point ω2 is taken.

[0024] S2: Define the ratio of the impedance of the second capacitor C2 to the equivalent impedance of the second inductance L2 and the internal resistance R as a, and define the ratio of the impedance of the first capacitor C1 to the equivalent impedance of the second capacitor C2, the second inductance L2, and the internal resistance R as b.

[0025] Let N = ω2 / ω1, the relationship between the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 can be determined by a, b, and N:

[0026]

[0027] S3: Topology simplification is performed on the dual-frequency LCCL resonant cavity to obtain simplified expressions ω1' and ω2' of the two parallel resonance frequency points ω1 and ω2 of the dual-frequency LCCL resonant cavity.

[0028] S4: A power matching coefficient β = L1 / L2 is defined, where L1 is the inductance value of the first inductor L1 of the dual-frequency LCCL resonant cavity, and L2 is the inductance value of the second inductor L2 of the dual-frequency LCCL resonant cavity. The power matching coefficient β is calculated based on the maximum output power P1 at low frequency and the maximum output power P2 at high frequency, and then the inductance value L1 of the first inductor L1 is obtained.

[0029] S5: Based on the relationship between the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 in the above step S2 and ω1' and ω2' in the above step S3, a relationship between a, b, N, L1, and L2 is obtained where ω2' = N*ω1';

[0030] S6: The capacitance value C1 of the first capacitor C1 is obtained based on L1, L2, and the simplified expression of ω1'.

[0031] S7: Based on the relationship between a, b, N, L1, and L2 in the above step S5, a set of values of a and b is determined, and the capacitance value C2 of the second capacitor C2 is obtained by substituting the capacitance value C1 calculated in the above step S6 into the relationship between the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 in the above step S2.

[0032] Preferably, the calculation method of a and b in the above step S2 is specifically:

[0033] Neglecting the resistance value of the internal resistance R, the expression of a and b is obtained

[0034] Preferably, the above step S3 is specifically:

[0035] In the low-frequency state, |a| >> 1 is taken, and the simplified expression of the parallel resonance frequency point ω1 is obtained

[0036] In the high-frequency state, |b| >> 1 is taken, and the simplified expression of the parallel resonance frequency point ω2 is obtained

[0037] Compared with the prior art, the present application has the following technical effects:

[0038] 1. The present application provides a current source type LCCL dual-frequency resonant system, which is composed of two current type resonant inverters and a dual-frequency LCCL resonant cavity, can output current including high-frequency component and low-frequency component on an induction heating coil, and realizes efficient induction heat treatment of complex tooth-shaped workpieces.

[0039] 2. The present application provides a parameter matching method for the current source type LCCL dual-frequency resonant system, which simplifies the high-frequency and low-frequency topologies of the LCCL dual-frequency resonant cavity by adding constraint conditions, and further simplifies the resonant frequency formula for actual engineering application, so that the design error is extremely low.

[0040] 3. The present application can adjust the high-low frequency output power ratio by adjusting the power matching coefficient value, and solve the problem of dual-frequency power imbalance caused by the equivalent resistance frequency variation characteristics of the load and other reasons.

[0041] 4. The LF and HF current type resonant inverters of the present application work at two parallel resonant frequency points of the dual-frequency LCCL resonant cavity, do not interfere with each other, realize frequency layer injection and energy separation control, and linearly superimpose low-frequency and high-frequency currents. BRIEF DESCRIPTION OF DRAWINGS

[0042] Figure 1 Two typical resonant inverter structures in the prior art;

[0043] Figure 2 The current source type LCCL dual-frequency resonant system architecture of the present application;

[0044] Figure 3 Two simplified topological structures of the dual-frequency LCCL resonant cavity of the present application;

[0045] Figure 4 The complete topological structure of the dual-frequency LCCL resonant cavity of the present application and the comparison diagram of the amplitude-frequency characteristics of the two simplified topological structures;

[0046] Figure 5 The superimposed current waveform diagram output by the present application in the dual-frequency superimposed working mode;

[0047] Figure 6 The current waveform of the induction heating coil of the present application and its FFT analysis. DETAILED DESCRIPTION

[0048] The technology described below can have various modifications and can be various embodiments, and is described in detail below with reference to specific embodiments in conjunction with the accompanying drawings. However, this does not mean that the technology described below is limited to specific embodiments. It should be understood that the present application includes all similar modifications, equivalents and alternatives without departing from the spirit and technical scope of the technology described below.

[0049] Figure 1 Two typical resonant inverter structures in the prior art are voltage source type series resonant inverters and current source type parallel resonant inverters according to different resonant load structures, where L is an inductive coil, C is a compensation capacitor, and R is an equivalent resistance of a workpiece to be heated. The voltage source type series resonant inverter is generally powered by a voltage source, and a large capacitor is connected in parallel at the input side. The inverter input is a constant voltage. As long as a certain regular switching voltage is provided for switches S1-S4, and they are alternately switched regularly, a square wave voltage can be output at the inverter output. The frequency depends on the switching frequency of the switching device, and the amplitude is related to the input voltage of the inverter. When working at the resonant point, the load impedance is minimum. The current source type parallel resonant inverter is generally powered by a current source, and a large inductor is connected in series at the input side of the inverter. The output of the power supply is a constant current. Unlike series resonance, parallel resonance amplifies the equivalent resistance. Simple LC parallel resonance makes the impedance maximum at the resonant point, which is Q 2 times the equivalent resistance of the load, where Q = ωL / R is the quality factor of the inductor L. The current type resonant inverter has lower requirements for the withstand voltage and current level of the power device, does not require short circuit protection, and has relatively simple load matching, which reduces the requirements for the power device and makes the circuit run more stably, and has high practical value.

[0050] Figure 2 The current source type LCCL dual-frequency resonant system architecture of the present application is composed of two current type resonant inverters and a dual-frequency LCCL resonant cavity. The dual-frequency LCCL resonant cavity contains two capacitors, two inductors and an internal resistance. L1 is defined as the resonant cavity power matching inductor, C1 and C2 are defined as the constraint capacitors for topology simplification, L2 is the inductance of the inductive heating coil, and R is the equivalent resistance of the coil.

[0051] The two current type resonant inverters each include four IGBT tubes and four fast recovery diodes connected in series with the four IGBT tubes, respectively. The two current type resonant inverters work in low frequency LF and high frequency HF modes, respectively, and the outputs of the two current type resonant inverters are simultaneously connected across the input side of the dual-frequency LCCL resonant cavity, which is equivalent to connecting the dual-frequency LCCL resonant cavity after parallel connection of the two current source type inverters. The dual-frequency LCCL resonant cavity has two parallel resonant frequency points, which are:

[0052]

[0053]

[0054] Figure 3 For the two simplified topologies of the dual-frequency LCCL resonant cavity, the expressions of the two parallel resonant frequency points ω1, ω2 of the above dual-frequency LCCL resonant cavity are complex, the parameter design is difficult, and it is difficult to be practically applied to engineering practice. Therefore, a and b are defined as simplified coefficients (a and b can be positive or negative) to represent the impedance size relationship between each branch inside the resonant cavity. Among them, in the L2-R branch, R is much smaller than L2 impedance and can be ignored.

[0055] Therefore, the impedance size of the C1 branch is The impedance size of the C2 branch is The impedance size of the L2-R branch is The impedance size of the C2 / / (L2-R) parallel branch is

[0056] When the system is at the low-frequency parallel resonant frequency point ω1, the topology is simplified, and a is defined to represent the multiple relationship of the impedance of the C2 branch and the L2-R branch, that is:

[0057]

[0058] When the system is at the high-frequency parallel resonant frequency point ω2, the topology is simplified, and b is defined to represent the multiple relationship of the impedance of the C1 branch and the C2 / / (L2-R) parallel branch, that is:

[0059]

[0060] For convenience, it is directly defined that That is, a and b can be positive or negative, and only represent the multiple relationship of the branch impedance.

[0061] Accordingly, the relationship of C1 and C2 can be obtained:

[0062]

[0063] Where N = ω2 / ω1, representing the multiple relationship between high frequency and low frequency, C1 and C2 are the capacitance values of the resonant capacitors C1 and C2, respectively.

[0064] For the case at low frequency, when |a| >> 1 is satisfied, the impedance on the C2 branch is much larger than the impedance on the (L2-R) branch, and the current flowing through the C2 branch is extremely small, so the C2 branch can be approximated as an open circuit, and the dual-frequency LCCL resonant cavity can be simplified as an LCL series-parallel resonant. At this time, the resonant cavity works at the low-frequency resonant frequency point, and its resonant frequency ω1 can be simplified as:

[0065]

[0066] The quality factor at low frequency is defined as β = L1 / L2, where ω1' is the resonant frequency of the simplified topology at low frequency, L1 is the matching inductance value, L2 is the inductance value of the induction heating coil, R ω1 is the equivalent resistance of the load at the resonant frequency ω1', and β is the power matching coefficient. The equivalent load impedance expression of the simplified topology at ω1' is:

[0067]

[0068] The output power at low frequency is thus where R1 = Q1 2 β 2 R ω1 , i is the imaginary unit, and I d is the magnitude of the square wave current output by the inverter.

[0069] For the case at high frequency, when |b| » 1 is satisfied, the impedance on C1 is much smaller than that on the (L2-R) / / C2 branch, and the voltage divided on C1 is extremely small, so the C1 branch can be approximated as a short circuit, and the dual-frequency LCCL resonant cavity can be simplified as an LC parallel resonant. At this time, the resonant cavity works at the high-frequency resonant frequency point, and its resonant frequency ω2 can be simplified as:

[0070]

[0071] The quality factor at high frequency is defined as β = L1 / L2, where ω2' is the resonant frequency of the simplified topology at high frequency, L1 is the matching inductance value, L2 is the inductance value of the induction heating coil, R ω2 is the equivalent resistance of the load at the resonant frequency ω2', and β is the power matching coefficient. The equivalent load impedance expression of the simplified topology at ω2 is:

[0072]

[0073] The output power at high frequency is thus where R2 = Q2 2 R ω2 , i is the imaginary unit, and I d is the magnitude of the square wave current output by the inverter.

[0074] On the one hand, the equivalent resistance R ω1 of the induction heating coil L2 when working at low frequency is ω2The difference is large, on the other hand, due to the above two kinds of simplified resonant cavity equivalent resistance R amplification factor is not consistent, can lead to the output power difference is large when working at high and low frequency, at this time can be through the power matching coefficient β on the high frequency output power and low frequency output power matching, namely adjusting The value of the high and low frequency full load output power ratio.

[0075] The resonant frequency formula of the two simplified topologies is: Where ω2'=N*ω1', C1, C2 relationship formula Can be obtained

[0076]

[0077] According to the above derivation, take a group of a, b value, and then you can easily find out all the parameters of the dual-frequency LCCL resonant cavity.

[0078] As Figure 4 The amplitude-frequency characteristic comparison chart of the complete topology structure and the two simplified topologies of the dual-frequency LCCL, the resonant frequency points and the characteristics of the parameter design by the simplified topology are compared with the complete topology, which verifies the rationality of the parameter design method based on the topology simplification.

[0079] Then a parameter design example is used to illustrate the parameter matching method of the application.

[0080] The default HF inverter and LF inverter output current size is consistent:

[0081] First, determine the target design frequency f1=8khz and f2=120khz, then N=15, the actual inductance value of the induction heating coil L2 is 12.20μH, the load equivalent resistance R ω1 At 8khz, R ω2 At 120khz, the equivalent resistance R

[0082] Further, the target values of the two parallel resonant frequency points ω1 and ω2 are 2pi*f1 and 2pi*f2 respectively.

[0083] Second, design the low frequency maximum output power P1 and the high frequency maximum output power P2 close in size, that is, R2 / R1≈1, R1=Q1 2 β 2 R ω1 And R2=Q2 2 R ω2 , calculation can get β=6.71, so L1=βL2=81.84μH;

[0084] Third, N, L1, L2 value is brought into Get We can choose a = 196 and b = -197;

[0085] Fourth step, substitute the L1 and L2 values ​​into... The result is C1 = 4.2087 μF;

[0086] Fifth step, substitute a = 196, b = -197 into the equation. We can obtain C1 = 25.391C2, and further derive C2 = 0.1658μF;

[0087] In summary, the parameters of the dual-frequency LCCL resonator are:

[0088] L1=81.84μH, L2=12.20μH, C1=4.2087μF, C2=0.1658μF.

[0089] Under the above parameters, the simplified topological resonant frequencies are 8000Hz and 120000Hz, respectively. The two parallel resonant frequencies of the complete dual-frequency LCCL topology are 7997.33Hz and 119996Hz. It can be seen that the difference between the two cases is extremely small, with errors both less than 0.035%. Observation Figure 4 The full topology and the simplified low-frequency topology were found to have almost identical amplitude-frequency characteristics at low frequencies. (See...) Figure 4 (a); The amplitude-frequency characteristics at high frequencies are almost identical to those of the simplified high-frequency topology, see Figure 4 (b). Therefore, the topology simplification method based on the present invention is effective for the parameter design of dual-frequency LCCL resonators.

[0090] Figure 5 , 6 All results are simulation results of the current-source type LCCL dual-frequency resonant system of this invention. Since it is difficult to reflect the relationship between the equivalent resistance and frequency in the simulation, R is taken as... ω1 =R ω2 =0.5Ω, and the other inductor and capacitor parameters are the same as the previous design parameters.

[0091] In the dual-frequency superposition working mode, the system uses two sets of current-source inverters connected to the input of the LCCL dual-frequency resonant cavity at the same time. The two sets of inverters operate at the two parallel resonant frequencies of the dual-frequency LCCL resonant cavity, respectively injecting low-frequency and high-frequency components into the resonant cavity, and finally forming a dual-frequency composite current on the load.

[0092] The specific operation mode is as follows: assuming that the two resonant frequencies of the dual-frequency LCCL resonant cavity are low frequency f1 and high frequency f2, and Nf1=f2; the working frequency of the LF inverter is f1, and the working frequency of the HF inverter is f2; in each complete low-frequency period T1=1 / f1, N high-frequency periods T2=1 / f2 are contained. The low-frequency part is powered by the LF inverter, and in each low-frequency resonant period T1=1 / f1, the LF inverter works in the traditional current source inverter mode, and the controller opens its diagonal switch tubes in turn at the frequency of f1. In the first half resonant period, the switch tube Sa1 and the switch tube Sa4 are turned on, the switch tube Sa2 and the switch tube Sa3 are turned off, the current is positive, and the resonant cavity is injected clockwise; in the second half resonant period, the switch tube Sa2 and the switch tube Sa3 are turned on, the switch tube Sa1 and the switch tube Sa4 are turned off, the current is reversed, and the process realizes a complete low-frequency resonant period T1=1 / f1, generating a low-frequency sinusoidal current component. The high-frequency part is powered by the HF inverter, which also works in the traditional current source inverter mode, that is, the action mode of the switch tubes Sb1-Sb4 is the same as that of the switch tubes Sa1-Sa4. With the frequency f2=Nf1, N high-frequency current periods are embedded into the low-frequency period, and the total current in the resonant cavity is the linear superposition of the low-frequency sinusoidal current and the high-frequency sinusoidal current.

[0093] As shown in Figure 5 , with the low-frequency resonant period T1 as the system period, in the dual-frequency superposition output, the superimposed current of the inverter output is a square wave current with 0, 2I d jumping in the first half resonant period with f2 as the period, and a square wave current with 0, -2I d jumping in the second half resonant period with f2 as the period. The total current in the resonant cavity is the linear superposition of the low-frequency current and the high-frequency current. In this operation mode, the LF and HF inverters work at different resonant frequencies of the resonant cavity and do not interfere with each other, realizing frequency layer injection and energy separation control.

[0094] Figure 6 (a) shows the current waveform of the induction heating coil L2, and the low-frequency current constitutes the envelope of the current waveform, while the high-frequency current oscillates rapidly inside the envelope. In this operation mode, the two inverters drive different frequency resonant points without interference, realizing frequency layer injection and energy separation control; Figure 6 (b) shows the FFT analysis of the superimposed current waveform on the induction heating coil L2, and it is found that there are mainly two frequency components, i.e. the low-frequency resonant frequency f1=8 kHz and the high-frequency resonant frequency f2=120 kHz designed in the above example.

[0095] Although the present application has been described in detail with particular references to illustrative embodiments thereof, it should be understood that modifications and changes can be made to such embodiments without departing from the spirit and scope of the application. The above description is intended to be illustrative and not restrictive. Other changes and modifications which are obvious to one having ordinary skill in the art are intended to be within the scope of the present application.

Claims

1. A current-source LCCL dual-frequency resonant system, comprising a first current-source resonant inverter, a second current-source resonant inverter, and a dual-frequency LCCL resonant cavity, characterized in that: the first current-source resonant inverter is a low-frequency (LF) inverter operating at a low-frequency resonant frequency, and the second current-source resonant inverter is a high-frequency (HF) inverter operating at a high-frequency resonant frequency; the dual-frequency LCCL resonant cavity has two parallel resonant frequency points and has a first input port and a second input port, the first current-source resonant inverter is connected to the first input port, and the second current-source resonant inverter is connected to the second input port.

2. The current-source LCCL dual-frequency resonant system of claim 1, wherein, The first current-source resonant inverter and the second current-source resonant inverter each include a first IGBT tube, a second IGBT tube, a third IGBT tube, a fourth IGBT tube, and four diodes connected in series with the first IGBT tube, the second IGBT tube, the third IGBT tube, and the fourth IGBT tube, respectively; the first IGBT tube and the third IGBT tube form a first series branch, and the second IGBT tube and the fourth IGBT tube form a second series branch, and the first series branch is connected in parallel with the second series branch.

3. The current-source LCCL dual-frequency resonant system of claim 2, wherein, An input inductor Lm is connected in series at the input side of the first current-source resonant inverter and the second current-source resonant inverter.

4. The current-source LCCL dual-frequency resonant system of claim 3, wherein, The dual-frequency LCCL resonant cavity includes a first inductor L1, a second inductor L2, a first capacitor C1, a second capacitor C2, and an internal resistor R; the first input port and the second input port share a first common terminal and a second common terminal, a first terminal of the first inductor L1 is connected to the first common terminal, and a second terminal of the first inductor L1 is connected to the second common terminal; a first terminal of the first capacitor C1 is connected to the first common terminal, a second terminal of the first capacitor C1 is connected to a first terminal of the second capacitor C2, and a second terminal of the second capacitor C2 is connected to the second common terminal; a first terminal of the second inductor L2 is connected to the second terminal of the first capacitor C1 and the first terminal of the second capacitor C2, a second terminal of the second inductor L2 is connected to a first terminal of the internal resistor R, and a second terminal of the internal resistor R is connected to the second common terminal.

5. The current-source LCCL dual-frequency resonant system of claim 4, wherein, The two parallel resonant frequency points of the dual-frequency LCCL resonant cavity are ω1 and ω2, respectively:

6. The current-source LCCL dual-frequency resonant system of claim 5, wherein, The system has three operating modes, namely a low-frequency operating mode, a high-frequency operating mode, and a dual-frequency superposition operating mode.

7. The current-source LCCL dual-frequency resonant system of claim 6, wherein, The dual-frequency superposition operating mode is specifically: the operating frequency of the first current-source resonant inverter corresponds to the low-frequency resonant frequency point of the dual-frequency LCCL resonant cavity, and the operating period is T1 = 1 / f1, and the operating frequency of the second current-source resonant inverter corresponds to the high-frequency resonant frequency point of the dual-frequency LCCL resonant cavity, and the operating period is T2 = 1 / f2; In each low-frequency resonance period T1, the controller switches the diagonal switch tubes of the first current-mode resonant inverter in turn at the frequency of f1, in the first half of the resonance period, the first IGBT tube and the fourth IGBT tube are turned on, and the second IGBT tube and the third IGBT tube are turned off, in the second half of the resonance period, the second IGBT tube and the third IGBT tube are turned on, and the first IGBT tube and the fourth IGBT tube are turned off; At the same time, the controller switches the diagonal switch tubes of the second current-mode resonant inverter in turn at the frequency of f2, thereby embedding N high-frequency resonance periods T2 into the low-frequency resonance period T1, where f2=Nf1.

8. A method for parameter matching of the current source type LCCL dual-frequency resonant system according to any one of claims 1-7, characterized in that, The method comprises the following steps: S1: set the target values of the two parallel resonance frequency points ω1, ω2 of the dual-frequency LCCL resonant cavity, measure the inductance value L2 of the second inductance L2 of the dual-frequency LCCL resonant cavity, and determine the load equivalent resistance R when the target value of the parallel resonance frequency point ω1 is taken ω1 and the load equivalent resistance R when the target value of the parallel resonance frequency point ω2 is taken ω2 ; S2: defining the ratio of the impedance of the second capacitor C2 to the equivalent impedance of the second inductor L2 and the internal resistance R as a, and defining the ratio of the impedance of the first capacitor C1 to the equivalent impedance of the second capacitor C2, the second inductor L2 and the internal resistance R as b; Let N=ω2 / ω1, the relationship between the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 can be determined by a, b, and N: S3: simplifying the topology of the dual-frequency LCCL resonant cavity to obtain the simplified expressions ω1' and ω2' of the two parallel resonance frequency points ω1 and ω2 of the dual-frequency LCCL resonant cavity; S4: defining the power matching coefficient β=L1 / L2, where L1 is the inductance value of the first inductor L1 of the dual-frequency LCCL resonant cavity, and L2 is the inductance value of the second inductor L2 of the dual-frequency LCCL resonant cavity; calculating the power matching coefficient β based on the low-frequency maximum output power P1 and the high-frequency maximum output power P2, and then obtaining the inductance value L1 of the first inductor L1; S5: Based on the relationship between the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 in the above step S2 and ω1' and ω2' in the above step S3, a relationship formula between a, b, N, L1, and L2 is obtained where ω2' = N * ω1'. S6: obtaining the capacitance value C1 of the first capacitor C1 based on L1, L2 and the simplified expression of ω1'; S7: determining a set of values of a and b based on the relationship between a, b, N, L1 and L2 in step S5, combining the capacitance value C1 calculated in step S6, and substituting the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 into the relationship between the capacitance value C1 of the first capacitor C1 and the capacitance value C2 of the second capacitor C2 in step S2 to obtain the capacitance value C2 of the second capacitor C2.

9. The parameter matching method according to claim 8, characterized in that, The calculation method of a and b in step S2 is as follows: Neglecting the resistance value of the internal resistance R, the expressions for a and b are obtained 10. The parameter matching method of claim 9, wherein, Step S3 is as follows: In the low frequency state, taking |a|»1, a simplified expression of the parallel resonance frequency point ω1 can be obtained In the high frequency state, taking |b|»1, a simplified expression of the parallel resonance frequency point ω2 can be obtained

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