Millimeter wave broadband tunable filter

Abstract

The millimeter wave broadband tunable filter comprises an orthogonal coupler I, two paths of core circuits, an orthogonal coupler II and an LO matching circuit, wherein each core circuit comprises a double-balanced mixer I, a baseband amplifier and a double-balanced mixer II; the quadrature coupler is based on a transformer structure, and adopts differential input and output to realize a phase quadrature distribution function, so that a single-phase 50% duty ratio clock signal is alternatively used, and the generation difficulty of a non-overlapping high-frequency local oscillator signal is reduced; the baseband amplifier adopts a two-stage structure, the first stage is a differential amplifier structure and provides high gain, the second stage adopts a source follower structure and ensures good frequency response (wide bandwidth), and the baseband amplifier adjusts the cut-off frequency of the amplifier by controlling the size of injection current Iss and realizes filtering bandwidth control. The gain of the baseband amplifier effectively compensates the insertion loss of the filter.

Description

TECHNICAL FIELD

[0001] The present application relates to the technical field of radio frequency integrated circuits, and in particular to a wideband tunable filter suitable for millimeter wave frequency bands. BACKGROUND

[0002] With the development of next-generation communication systems such as 5G / 6G, high-frequency operation of millimeter waves and large-scale MIMO technology have become key technologies to achieve high-speed and large-capacity concurrent communication. A direct digital radio frequency modulator (DDRM) generates a radio frequency signal through a digital circuit, breaking through the limitation of the sampling frequency (f s ) and having a maximum operating frequency of up to 40 GHz, so it is widely used in millimeter wave modulators, significantly reducing system costs. However, the application of DDRM in millimeter wave circuits introduces a large amount of spurious signals, which seriously affects the performance of subsequent circuits, and high-performance millimeter wave filters are urgently needed to suppress spurious signals and ensure stable system operation. Note that traditional microwave filters use microstrip lines or waveguide structures to achieve flexibility in filtering, and cannot adjust the bandwidth, which is not suitable for the current hardware reconfigurable requirements of 5G / 6G.

[0003] In recent years, active filters have been increasingly favored due to their reconfigurable flexibility and good compatibility with silicon technology. International literature

S. Hari, C. J. Ellington and B. A. Floyd, "A 6-31GHz Tunable Reflection-Mode N-Path Filter," 2021 IEEE Radio Frequency Integrated Circuits Symposium (RFIC), Atlanta, GA, USA, 2021, pp. 143-146

[0004] Based on the above-mentioned technical problems, this application discloses a millimeter-wave broadband tunable filter, including: an orthogonal coupler I, a core circuit containing two channels, each including a double-balanced mixer I, a baseband amplifier, and a double-balanced mixer II, an orthogonal coupler II, and an LO matching circuit.

[0005] The output of the quadrature coupler I is divided into two paths, which are respectively connected to the input of the double balanced mixer I in the two core circuits;

[0006] Inside each core circuit, the output of double-balanced mixer I is connected to the input of the baseband amplifier, and the output of the baseband amplifier is connected to the input of double-balanced mixer II; the output of double-balanced mixer II in both core circuits is connected to the input of quadrature coupler II.

[0007] The output of the LO matching circuit is connected to the local oscillator signal input of the double-balanced mixer I and the double-balanced mixer II in the two core circuits, respectively.

[0008] The orthogonal coupler I and orthogonal coupler II have the same structure, both including: capacitor C1, capacitor C2, capacitor C3, capacitor C4, transformer TR1, and transformer TR2;

[0009] The LO matching circuit includes: resistor R1, resistor R2, resistor R3, inductor L1, capacitor C5, and capacitor C6;

[0010] The double-balanced mixer I and the double-balanced mixer II have the same structure, both including: NMOS transistor M1, NMOS transistor M2, NMOS transistor M3, and NMOS transistor M4;

[0011] The baseband amplifier includes: NMOS transistor M5, NMOS transistor M6, PMOS transistor M7, PMOS transistor M8, PMOS transistor M9, and NMOS transistor M1. 10 NMOS transistor M 11 NMOS transistor M 12 NMOS transistor M 13 NMOS transistor M 14 NMOS transistor M 15 Resistors R4, R5, R6, R7, and error amplifier EA.

[0012] Preferably, the orthogonal coupler I adopts a transformer-based differential structure, with input terminals IN+ and IN-, and output terminals CPL+, CPL-, THRU+, and THRU-. CPL+ and CPL- are connected to one path, and THRU+ and THRU- are connected to two paths. Port IN+ is formed by connecting one end of capacitor C1 to port a of transformer TR1; port IN- is formed by connecting one end of capacitor C3 to port a' of transformer TR2; and port CPL+ is formed by connecting the other end of capacitor C1 to port b of transformer TR1. CPL- is formed by connecting the other end of capacitor C3 to port b' of transformer TR2. Port THRU+ is formed by connecting one end of capacitor C2 to port c of transformer TR1. Port THRU- is formed by connecting one end of capacitor C4 to port c' of transformer TR2. Port ISO+ is formed by connecting the other end of capacitor C2 to port d of transformer TR1. Port ISO- is formed by connecting the other end of capacitor C4 to port d' of transformer TR2. A 100-ohm resistor is connected between ISO+ and ISO- to form a differential isolation port.

[0013] Preferably, the input terminals of the LO matching circuit are LO+ and LO-, and the output terminals are S0 and S1. The local oscillator signal enters the LO matching circuit via LO+ and LO-, and is output from S0 and S1 and connected to the S0 and S1 ports of the double-balanced mixer. The LO+ port is formed by connecting one end of resistor R1, one end of inductor L1, and one end of capacitor C5. The LO- port is formed by connecting the other end of resistor R1, the other end of inductor L1, and one end of capacitor C6. The S0 port is formed by connecting one end of resistor R2 and the other end of capacitor C5. The S1 port is formed by connecting one end of resistor R3 and the other end of capacitor C6. The other end of resistor R2 and the other end of resistor R3 are connected to form the bias voltage input port V. LO .

[0014] Preferably, the input terminals of the dual-balanced mixer I are differential input ports 1 and 2, wherein port 1 is connected to the CPL+ port of the quadrature coupler I, and port 2 is connected to the CPL- port. The output terminals of the dual-balanced mixer I are differential output ports 3 and 4. Port 1 is formed by connecting the drains of NMOS transistors M1 and M2. Port 2 is formed by connecting the drains of NMOS transistors M3 and M4. The gates of NMOS transistors M1 and M4 are connected to the local oscillator input port S0. The gates of NMOS transistors M2 and M3 are connected to the local oscillator input port S1. Port 3 is formed by connecting the sources of NMOS transistors M1 and M3. Port 4 is formed by connecting the sources of NMOS transistors M2 and M4.

[0015] Preferably, the baseband amplifier adopts a two-stage structure. The first stage is a classic differential amplifier structure, and the second stage is a source follower structure. The cutoff frequency of the amplifier is adjusted by controlling the magnitude of the injected current Iss, thereby achieving filter bandwidth control. In addition, a common-mode feedback loop and a negative feedback loop are introduced to ensure a stable frequency response of the amplifier. The amplifier input terminals are IN+ and IN-, where the IN+ port is connected to port 3 of a dual-balanced mixer I, and the IN- port is connected to port 4. The output terminals are OUT+ and OUT-. The IN+ port is formed by connecting the gate of NMOS transistor M9 and one end of resistor R4, and the IN- port is formed by connecting NMOS transistor M9 and R4. 10 The gate of the NMOS transistor is connected to one end of resistor R5, and the other end of resistor R4 is connected to the other end of resistor R5 to form port Vbias. The source of the NMOS transistor M9 and the NMOS transistor M... 10 The source and NMOS transistor M 12 The drain of NMOS transistor M9 and the gate of NMOS transistor M5 are connected to the drain of PMOS transistor M7. 10 The drain and NMOS transistor M 13 The gate of the NMOS transistor is connected to the drain of the PMOS transistor M8. 12 The gate and NMOS transistor M 11 The gate of the NMOS transistor M 11 The drain connection forms port Iss, and the NMOS transistor M 12 The source and NMOS transistor M 11 The source of NMOS transistor M5 is grounded. The source of NMOS transistor M5 and the drain of NMOS transistor M6 are connected to one end of resistor R7 to form the output port OUT+. The source of NMOS transistor M6 is grounded. The drain of NMOS transistor M5, the source of PMOS transistor M7, the source of PMOS transistor M8, and the source of NMOS transistor M7 are grounded. 13 The drain connection forms the port VDD. The gates of PMOS transistor M7 and PMOS transistor M8 are connected to the output terminal out of the error amplifier. The NMOS transistor M... 13 The source and NMOS transistor M 15 The drain of the NMOS transistor M is connected to one end of resistor R6 to form the output port OUT-. 15 The gate and NMOS transistor M 14 The gate and drain of the NMOS transistor M6 are connected to the gate of the NMOS transistor M6 to form port Ief. 14 The source and NMOS transistor M 15The source of the resistor is grounded, and the other end of the resistor R6 and the other end of the resistor R7 are connected as the in+ port of the error amplifier. The in- port of the error amplifier is connected to VDD / 2.

[0016] Preferably, the input terminals of the dual-balanced mixer II are differential input ports 3 and 4, wherein port 3 is connected to the OUT+ port of the baseband amplifier and port 4 is connected to the OUT- port, and the output terminals of the dual-balanced mixer II are differential output ports 1 and 2.

[0017] Preferably, the input terminals of the quadrature coupler II are IN+, IN-, CPL+, and CPL-, wherein the IN+ port is connected to port 1 of one double-balanced mixer II, the IN- port is connected to port 2, the CPL+ port is connected to port 1 of two double-balanced mixer II, and the CPL- port is connected to port 2. The output terminals of the quadrature coupler II are differential output ports THRU+ and THRU-, wherein the THRU+ port serves as the output port OUT+ of the filter, and the THRU- port serves as the output port OUT- of the filter.

[0018] Compared with the prior art, the technical solution of this application has the following technical effects:

[0019] This invention replaces the phase-shifting design of the traditional local oscillator path with the phase-shifting and phase-conversion function of the radio frequency signal path through the transformer differential structure of the quadrature coupler. It can be directly driven by a single-phase 50% duty cycle clock signal without the need for complex non-overlapping high-frequency local oscillator signals, which greatly reduces the difficulty of generating clock signals in the millimeter-wave band. At the same time, by taking advantage of the compatibility of continuous wave clock sources, it significantly improves the maximum operating frequency of the filter, breaks through the frequency limit of traditional active filters, and better adapts to the high-frequency operation requirements of 5G / 6G millimeter-wave communication.

[0020] The baseband amplifier of this invention adopts a two-stage structure design. By controlling the magnitude of the injected current Iss, the cutoff frequency is adjusted, thereby flexibly controlling the filtering bandwidth. Compared with the traditional filter scheme that relies on a fixed capacitor to determine the bandwidth, this design can achieve dynamic bandwidth adjustment without changing the hardware, perfectly meeting the current communication system's requirement for hardware reconfigurability. At the same time, the differential amplifier in the two-stage structure provides high gain, and the source follower ensures wide bandwidth response, ensuring the stability of frequency response during bandwidth adjustment.

[0021] The baseband amplifier of this invention effectively compensates for the insertion loss of the filter by introducing gain, which solves the problem of excessive insertion loss caused by the lack of gain structure in traditional active filters. By biasing the transistor in the subthreshold operating region, the gain is independent of the injected current, and the gain is kept stable during bandwidth adjustment. This ensures that the energy loss of the signal is minimized during the filtering process, thereby improving the signal transmission efficiency and overall performance of the millimeter-wave communication system.

[0022] The orthogonal coupler of this invention employs a transformer-capacitor coupling network, resulting in lower insertion loss on silicon substrates compared to traditional microstrip line structures. Furthermore, it achieves excellent input-output matching through a port reflection signal cancellation mechanism. The parameter configuration of the LO matching circuit ensures maximum local oscillator signal transmission and avoids reflection interference. In addition, the baseband amplifier incorporates common-mode feedback and negative feedback loops to further ensure frequency response stability. The overall structure exhibits excellent compatibility with silicon process technology, facilitating integration into millimeter-wave communication equipment.

[0023] The above description is only an overview of the technical solution of this application. In order to better understand the technical means of this application and implement it in accordance with the contents of the specification, and to make the above and other objects, features and advantages of this application more obvious and understandable, the preferred embodiments of this application are described in detail below with reference to the accompanying drawings.

[0024] The above and other objects, advantages and features of this application will become more apparent to those skilled in the art from the following detailed description of specific embodiments in conjunction with the accompanying drawings. Attached Figure Description

[0025] To more clearly illustrate the technical solutions in the embodiments of this application or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort. In all drawings, similar elements or parts are generally identified by similar reference numerals. In the drawings, the elements or parts are not necessarily drawn to scale.

[0026] Based on the description of the figures and their corresponding technical content in the document, the titles of the figures are as follows:

[0027] Figure 1 This is a simplified structural schematic diagram of a millimeter-wave broadband tunable filter according to the present invention;

[0028] Figure 2 This is a detailed structural schematic diagram of a millimeter-wave broadband tunable filter according to the present invention;

[0029] Figure 3 This is a schematic diagram of an orthogonal coupler circuit for a millimeter-wave broadband tunable filter according to the present invention.

[0030] Figure 4 This is a schematic diagram of the LO matching circuit of a millimeter-wave broadband tunable filter according to the present invention.

[0031] Figure 5This is a circuit diagram of a double-balanced mixer for a millimeter-wave broadband tunable filter according to the present invention.

[0032] Figure 6 This is a circuit diagram of a baseband amplifier for a millimeter-wave broadband tunable filter according to the present invention.

[0033] Figure 7 The gain S of a millimeter-wave broadband tunable filter according to the present invention 21 Resulting image;

[0034] Figure 8 The circuit input reflection coefficient S of a millimeter-wave broadband tunable filter according to the present invention 11 Resulting image;

[0035] Figure 9 The present invention relates to a millimeter-wave broadband tunable filter with gain S. 21 Relationship between bandwidth and baseband injection current. Detailed Implementation

[0036] To make the objectives, technical solutions, and advantages of the embodiments of this application clearer, the technical solutions of the embodiments of this application will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of this application, not all embodiments. In the following description, specific details such as specific configurations and components are provided merely to help fully understand the embodiments of this application. Therefore, those skilled in the art should understand that various changes and modifications can be made to the embodiments described herein without departing from the scope and spirit of this application. In addition, for clarity and brevity, descriptions of known functions and structures are omitted in the embodiments.

[0037] It should be understood that the phrase "an embodiment" or "this embodiment" throughout the specification means that a specific feature, structure, or characteristic related to the embodiment is included in at least one embodiment of this application. Therefore, "an embodiment" or "this embodiment" appearing throughout the specification does not necessarily refer to the same embodiment. Furthermore, these specific features, structures, or characteristics can be combined in any suitable manner in one or more embodiments.

[0038] Furthermore, reference numerals and / or letters may be repeated in different examples within this application. Such repetition is for the purpose of simplification and clarity and does not in itself indicate a relationship between the various embodiments and / or settings discussed.

[0039] In this article, the term "and / or" is merely a description of the relationship between related objects, indicating that three relationships can exist. For example, A and / or B can mean: A exists alone, B exists alone, and A and B exist simultaneously. The term " / and" in this article describes another type of relationship between related objects, indicating that two relationships can exist. For example, A / and B can mean: A exists alone, and A and B exist alone. In addition, the character " / " in this article generally indicates that the related objects before and after it are in an "or" relationship.

[0040] In this article, the term "at least one" is merely a description of the relationship between related objects, indicating that there can be three relationships. For example, "at least one of A and B" can mean: A exists alone, A and B exist simultaneously, or B exists alone.

[0041] It should also be noted that, in this document, relational terms such as "first" and "second" are used only to distinguish one entity or operation from another, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Furthermore, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion.

[0042] Example 1

[0043] This embodiment mainly describes a millimeter-wave broadband tunable filter, such as... Figures 1-2 As shown, it includes: quadrature coupler I, two core circuits each containing a double-balanced mixer I, a baseband amplifier, and a double-balanced mixer II, quadrature coupler II, and LO matching circuit;

[0044] The output of the quadrature coupler I is divided into two paths, which are respectively connected to the input of the double balanced mixer I in the two core circuits;

[0045] Inside each core circuit, the output of double-balanced mixer I is connected to the input of the baseband amplifier, and the output of the baseband amplifier is connected to the input of double-balanced mixer II; the output of double-balanced mixer II in both core circuits is connected to the input of quadrature coupler II.

[0046] The output of the LO matching circuit is connected to the local oscillator signal input of the double-balanced mixer I and the double-balanced mixer II in the two core circuits, respectively.

[0047] like Figure 3 As shown, the orthogonal coupler I and orthogonal coupler II have the same structure, both including: capacitor C1, capacitor C2, capacitor C3, capacitor C4, transformer TR1, and transformer TR2;

[0048] like Figure 4As shown, the LO matching circuit includes: resistor R1, resistor R2, resistor R3, inductor L1, capacitor C5, and capacitor C6.

[0049] like Figure 5 As shown, the double-balanced mixer I and the double-balanced mixer II have the same structure, both including: NMOS transistor M1, NMOS transistor M2, NMOS transistor M3, and NMOS transistor M4;

[0050] like Figure 6 As shown, the baseband amplifier includes: NMOS transistor M5, NMOS transistor M6, PMOS transistor M7, PMOS transistor M8, PMOS transistor M9, and NMOS transistor M1. 10 NMOS transistor M 11 NMOS transistor M 12 NMOS transistor M 13 NMOS transistor M 14 NMOS transistor M 15 Resistors R4, R5, R6, R7, and error amplifier EA.

[0051] Furthermore, the orthogonal coupler I adopts a transformer-based differential structure, with input terminals IN+ and IN-, and output terminals CPL+, CPL-, THRU+, and THRU-. CPL+ and CPL- are connected to one path, and THRU+ and THRU- are connected to two paths. Port IN+ is formed by connecting one end of capacitor C1 to port a of transformer TR1; port IN- is formed by connecting one end of capacitor C3 to port a' of transformer TR2; and port CPL+ is formed by connecting the other end of capacitor C1 to port b of transformer TR1. CPL- is formed by connecting the other end of capacitor C3 to port b' of transformer TR2. Port THRU+ is formed by connecting one end of capacitor C2 to port c of transformer TR1. Port THRU- is formed by connecting one end of capacitor C4 to port c' of transformer TR2. Port ISO+ is formed by connecting the other end of capacitor C2 to port d of transformer TR1. Port ISO- is formed by connecting the other end of capacitor C4 to port d' of transformer TR2. A 100-ohm resistor is connected between ISO+ and ISO- to form a differential isolation port.

[0052] Furthermore, the input terminals of the LO matching circuit are LO+ and LO-, and the output terminals are S0 and S1. The local oscillator signal enters the LO matching circuit via LO+ and LO-, and is output from S0 and S1 and connected to the S0 and S1 ports of the double-balanced mixer. The LO+ port is formed by connecting one end of resistor R1, one end of inductor L1, and one end of capacitor C5. The LO- port is formed by connecting the other end of resistor R1, the other end of inductor L1, and one end of capacitor C6. The S0 port is formed by connecting one end of resistor R2 and the other end of capacitor C5. The S1 port is formed by connecting one end of resistor R3 and the other end of capacitor C6. The other end of resistor R2 and the other end of resistor R3 are connected to form the bias voltage input port V. LO .

[0053] Furthermore, the input terminals of the dual-balanced mixer I are differential input ports 1 and 2, wherein port 1 is connected to the CPL+ port of the quadrature coupler I, and port 2 is connected to the CPL- port. The output terminals of the dual-balanced mixer I are differential output ports 3 and 4. Port 1 is formed by connecting the drains of NMOS transistor M1 and NMOS transistor M2. Port 2 is formed by connecting the drains of NMOS transistor M3 and NMOS transistor M4. The gates of NMOS transistor M1 and NMOS transistor M4 are connected to the local oscillator input port S0. The gates of NMOS transistor M2 and NMOS transistor M3 are connected to the local oscillator input port S1. Port 3 is formed by connecting the sources of NMOS transistor M1 and NMOS transistor M3. Port 4 is formed by connecting the sources of NMOS transistor M2 and NMOS transistor M4.

[0054] Furthermore, the baseband amplifier adopts a two-stage structure. The first stage is a classic differential amplifier structure, and the second stage is a source follower structure. The cutoff frequency of the amplifier is adjusted by controlling the magnitude of the injected current Iss, thereby achieving filter bandwidth control. In addition, a common-mode feedback loop and a negative feedback loop are introduced to ensure a stable frequency response of the amplifier. The amplifier input terminals are IN+ and IN-, where the IN+ port is connected to port 3 of a dual-balanced mixer I, and the IN- port is connected to port 4. The output terminals are OUT+ and OUT-. The IN+ port is formed by connecting the gate of NMOS transistor M9 and one end of resistor R4, and the IN- port is formed by connecting NMOS transistor M9 and R4. 10 The gate of the NMOS transistor is connected to one end of resistor R5, and the other end of resistor R4 is connected to the other end of resistor R5 to form port Vbias. The source of the NMOS transistor M9 and the NMOS transistor M... 10 The source and NMOS transistor M 12 The drain of NMOS transistor M9 and the gate of NMOS transistor M5 are connected to the drain of PMOS transistor M7.10 The drain and NMOS transistor M 13 The gate of the NMOS transistor is connected to the drain of the PMOS transistor M8. 12 The gate and NMOS transistor M 11 The gate of the NMOS transistor M 11 The drain connection forms port Iss, and the NMOS transistor M 12 The source and NMOS transistor M 11 The source of NMOS transistor M5 is grounded. The source of NMOS transistor M5 and the drain of NMOS transistor M6 are connected to one end of resistor R7 to form the output port OUT+. The source of NMOS transistor M6 is grounded. The drain of NMOS transistor M5, the source of PMOS transistor M7, the source of PMOS transistor M8, and the source of NMOS transistor M7 are grounded. 13 The drain connection forms the port VDD. The gates of PMOS transistor M7 and PMOS transistor M8 are connected to the output terminal out of the error amplifier. The NMOS transistor M... 13 The source and NMOS transistor M 15 The drain of the NMOS transistor M is connected to one end of resistor R6 to form the output port OUT-. 15 The gate and NMOS transistor M 14 The gate and drain of the NMOS transistor M6 are connected to the gate of the NMOS transistor M6 to form port Ief. 14 The source and NMOS transistor M 15 The source of the resistor is grounded, and the other end of the resistor R6 and the other end of the resistor R7 are connected as the in+ port of the error amplifier. The in- port of the error amplifier is connected to VDD / 2.

[0055] Furthermore, the input terminals of the dual-balanced mixer II are differential input ports 3 and 4, wherein port 3 is connected to the OUT+ port of the baseband amplifier and port 4 is connected to the OUT- port. The output terminals of the dual-balanced mixer II are differential output ports 1 and 2.

[0056] Furthermore, the input terminals of the quadrature coupler II are IN+, IN-, CPL+, and CPL-, wherein the IN+ port is connected to port 1 of one double-balanced mixer II, the IN- port is connected to port 2, the CPL+ port is connected to port 1 of two double-balanced mixer II, and the CPL- port is connected to port 2. The output terminals of the quadrature coupler II are differential output ports THRU+ and THRU-, wherein the THRU+ port serves as the output port OUT+ of the filter, and the THRU- port serves as the output port OUT- of the filter.

[0057] This embodiment details the use of orthogonal couplers to achieve phase shift conversion of two-phase to four-phase RF signals, thereby avoiding the local oscillator phase shift characteristics of traditional solutions. This allows the use of a conventional local oscillator clock signal with a 50% duty cycle, reduces the difficulty of generating high-frequency non-overlapping clock local oscillator circuits, significantly improves the center operating frequency of the filter, and breaks through the operating frequency limitations of traditional active filters. By using a baseband amplifier with an adjustable cutoff frequency to replace the fixed parallel capacitor in the traditional core circuit, flexible control of the millimeter-wave band filter bandwidth is achieved. Furthermore, the introduction of the gain of the baseband amplifier also reduces the overall insertion loss of the active filter.

[0058] Based on Embodiment 1, this embodiment describes in detail the specific implementation process and technical effects of a millimeter-wave broadband tunable filter, specifically as follows:

[0059] The quadrature coupler of this application is based on a transformer structure and uses differential input and output to realize the function of a four-phase distributor. By transferring the I / Q generation function of the local oscillator (LO) path to the radio frequency (RF) signal path, a single-phase 50% duty cycle clock signal can be used instead (instead of a four-phase 25% duty cycle clock signal), simplifying the generation of non-overlapping high-frequency signals. The 50% duty cycle clock signal allows the use of continuous wave (CW) as a clock source, making it easier to generate high-frequency clock signals, thereby increasing the maximum operating frequency of the filter.

[0060] The baseband amplifier in this application adopts a two-stage structure. The first stage is a differential amplifier structure, providing high gain, and the second stage adopts a source follower structure, ensuring good frequency response (wide bandwidth). The cutoff frequency f of the baseband amplifier is... c The formula is: Where C0 is the total parasitic capacitance of the transistor at the output node of the first-stage differential amplifier, and I... ss The tail current source control current, λ is the channel length modulation coefficient of the transistor; by controlling the injected current I... ss The cutoff frequency of the amplifier can be adjusted by the magnitude of the current, thereby achieving flexible control of the filter bandwidth. On the other hand, it is desirable for the gain to remain stable and unaffected by bandwidth changes. This requires that the gain expression be independent of the bias current, which can be achieved by biasing the transistor in the subthreshold operating region. Specifically, the transistor current I in the subthreshold region... D The formula is: V T Where is the threshold voltage, n is the subthreshold inversion coefficient, and I0 is the reference current related to transistor process and device size. Figure 6 I D Replace with I ss The formula for finding the transderivative is: g m To achieve the transconductance of the transistor, g was realized. m ∝Iss The linear relationship; furthermore, the amplifier differential pair output node impedance r0 is: Where λ n λ is the channel length modulation factor of the NMOS transistor. p Let be the channel length modulation coefficient, and the overall voltage gain be: Where Av represents the voltage gain of the baseband amplifier, λ np To comprehensively consider the equivalent parameters of the channel length modulation coefficient of NMOS and PMOS transistors, the gain and Ig were achieved. ss The independent relationship; note that the baseband amplifier in the weak inversion region can provide a certain gain to compensate for the insertion loss of the microwave active filter, which shows an advantage in gain compared with the traditional structure.

[0061] like Figure 3 The orthogonal coupler in the design is implemented using a transformer-capacitor coupling network. This structure exhibits lower insertion loss characteristics on silicon substrates compared to traditional microstrip line structures. Its design is based on the following formula: Where Z0 is the characteristic impedance, ω0 is the operating angular frequency, and under good operating conditions, the power at the thru and CPL ports must be equal, and the phase difference must satisfy 90°. Simulations show that when the coupling coefficient k = 0.55, the thru and CPL ports of the transformer capacitor-coupled network can obtain relatively flat IQ amplitude and phase responses. At this time, the corresponding L... ac，bd 434pH, C 1，2 20fF. Note that the THRU and CPL ports of the quadrature coupling network are connected to the same two-layer active circuit. Although the overall input port impedance of the mixer and baseband amplifier is high, failing to meet the matching condition, on the quadrature coupler side, the reflected signals from the THRU and CPL ports satisfy the cancellation condition of being 180° out of phase and having the same amplitude. This results in a zero reflection coefficient at the IN port, achieving a good input matching level. Similarly, a good port matching level can be achieved at the output port. Additionally, as... Figure 4 As shown, the parameters of the local oscillator port matching circuit of the LO matching circuit are configured as follows: R2-3: 8kΩ, C5-6: 0.5pF; R1: 160Ω, L1: 310pH, in order to ensure maximum transmission of the local oscillator signal and avoid reflection.

[0062] This embodiment is designed and implemented using a 40nm SOI CMOS process. At a 1.2V supply voltage, the power consumption is approximately 13mW, and the applied local oscillator AC signal is 10dBm. The design dimensions of the core transistor are shown in [reference needed]. Figure 5 and Figure 6 Mark; and Figure 6 The operational amplifier in the circuit uses common-mode feedback to ensure that the output level is stable at 0.6V, and the input level is provided by Vbias: 0.6V; the local oscillator bias level is V...LO The specific simulation results obtained are as follows: 0.85V. Figure 7 The S values ​​of the circuit at local oscillator input frequencies of 35~65GHz are given. 21 The parameter simulation results show that the bandwidth BW is set to 600MHz, and the control current I is... SS =100uA, simulation results show that at the center frequency S 21 The parameter is approximately 0dB; it should be emphasized that the upper limit of the center operating frequency of this circuit is limited by the operating frequency of the transformer in the quadrature coupler, such as... Figure 8 As shown, the S values ​​of the circuit at local oscillator input frequencies of 35~65GHz are given. 11 The parameter simulation results show that S... 11 All values ​​are less than -10dB, indicating good matching; simulations show that the circuit noise figure is approximately 13dB across the entire frequency band. Figure 9 The bandwidth and control current I of the filter at a 50GHz local oscillator input are given. SS Simulation results of the variation relationship show that the bandwidth tuning range of the filter is 0.1-0.9 GHz, corresponding to I SS 10-200uA. And note S 21 amplitude and injection current I SS The change is irrelevant.

[0063] Table 1 provides a comparison of the technical specifications of this technology and existing technologies, specifically:

[0064] Table 1 Performance Comparison

[0065]

[0066] The existing technology is described in [S. Hari, CJ Ellington and BAFloyd, "A 6-31GHz Tunable Reflection-Mode N-Path Filter," 2021 IEEE Radio Frequency Integrated Circuits Symposium (RFIC), Atlanta, GA, USA, 2021, pp. 143-146]. Therefore, this technology has significant advantages in center frequency, bandwidth tuning range, insertion loss, and power consumption. Overall, by utilizing the phase-shifting function of the RF link and using a conventional local oscillator signal with a 50% duty cycle to drive the N-path filter, this invention provides a filter circuit that can operate in the 35-65GHz frequency band. It features wideband tunability, high center frequency, low insertion loss, and low power consumption, and can be widely used in millimeter-wave communication equipment.

[0067] This embodiment details how the filter transfers the I / Q generation function of the local oscillator path to the radio frequency path through an orthogonal coupler, which can use a single-phase 50% duty cycle clock signal to simplify high-frequency signal generation and improve the operating frequency; the two-stage baseband amplifier has an adjustable cutoff frequency, which enables flexible bandwidth control and stable gain compensation for insertion loss; the overall structure is well-matched, with low insertion loss and low power consumption, making it suitable for 5G / 6G millimeter-wave communication requirements.

[0068] The above are merely preferred embodiments of the present invention and are not intended to limit the scope of protection of the present invention. For those skilled in the art, the present invention can have various modifications and variations. Any changes, modifications, substitutions, integrations, and parameter changes made to these embodiments within the spirit and principles of the present invention, without departing from the principles and spirit of the present invention, through conventional substitutions or to achieve the same function, fall within the scope of protection of the present invention.

Claims

1. A millimeter-wave broadband tunable filter, characterized in that, include: The core circuits include quadrature coupler I, two channels each containing double-balanced mixer I, baseband amplifier, double-balanced mixer II, quadrature coupler II, and LO matching circuit; The output of the quadrature coupler I is divided into two paths, which are respectively connected to the input of the double balanced mixer I in the two core circuits; Inside each core circuit, the output of double-balanced mixer I is connected to the input of the baseband amplifier, and the output of the baseband amplifier is connected to the input of double-balanced mixer II. The outputs of the dual-balanced mixer II in both core circuits are connected to the input of the quadrature coupler II. The output of the LO matching circuit is connected to the local oscillator signal input of the double-balanced mixer I and the double-balanced mixer II in the two core circuits, respectively. The orthogonal coupler I and orthogonal coupler II have the same structure. Includes: capacitors C1, C2, C3, and C4; transformer TR1; and transformer TR2. The LO matching circuit includes: resistor R1, resistor R2, resistor R3, inductor L1, capacitor C5, and capacitor C6; The double-balanced mixer I and the double-balanced mixer II have the same structure, both including: NMOS transistor M1, NMOS transistor M2, NMOS transistor M3, and NMOS transistor M4; The baseband amplifier includes: NMOS transistor M5, NMOS transistor M6, PMOS transistor M7, PMOS transistor M8, PMOS transistor M9, and NMOS transistor M1. 10 NMOS transistor M 11 NMOS transistor M 12 NMOS transistor M 13 NMOS transistor M 14 NMOS transistor M 15 Resistors R4, R5, R6, R7, and error amplifier EA.

2. The millimeter-wave broadband tunable filter according to claim 1, characterized in that, The orthogonal coupler I adopts a transformer-based differential structure. Its input terminals are IN+ and IN-, and its output terminals are CPL+, CPL-, THRU+, and THRU-. CPL+ and CPL- are connected to one path, and THRU+ and THRU- are connected to two paths. Port IN+ is formed by connecting one end of capacitor C1 to port a of transformer TR1. Port IN- is formed by connecting one end of capacitor C3 to port a' of transformer TR2. Port CPL+ is formed by connecting the other end of capacitor C1 to port b of transformer TR1. Port CP... L- is formed by connecting the other end of capacitor C3 to port b' of transformer TR2. Port THRU+ is formed by connecting one end of capacitor C2 to port c of transformer TR1. Port THRU- is formed by connecting one end of capacitor C4 to port c' of transformer TR2. Port ISO+ is formed by connecting the other end of capacitor C2 to port d of transformer TR1. Port ISO- is formed by connecting the other end of capacitor C4 to port d' of transformer TR2. A 100-ohm resistor is connected between ISO+ and ISO- to form a differential isolation port.

3. The millimeter-wave broadband tunable filter according to claim 1, characterized in that, The input terminals of the LO matching circuit are LO+ and LO-, and the output terminals are S0 and S1. The local oscillator signal enters the LO matching circuit through LO+ and LO-, and is output from S0 and S1 and connected to the S0 and S1 ports of the double-balanced mixer. The LO+ port is formed by connecting one end of resistor R1, one end of inductor L1, and one end of capacitor C5. The LO- port is formed by connecting the other end of resistor R1, the other end of inductor L1, and one end of capacitor C6. The S0 port is formed by connecting one end of resistor R2 and the other end of capacitor C5. The S1 port is formed by connecting one end of resistor R3 and the other end of capacitor C6. The other ends of resistor R2 and resistor R3 are connected to form the bias voltage input port V. LO .

4. The millimeter-wave broadband tunable filter according to claim 1, characterized in that, The input terminals of the dual-balanced mixer I are differential input ports 1 and 2, where port 1 is connected to the CPL+ port of quadrature coupler I, and port 2 is connected to the CPL- port. The output terminals of the dual-balanced mixer I are differential output ports 3 and 4. Port 1 is formed by connecting the drains of NMOS transistors M1 and M2. Port 2 is formed by connecting the drains of NMOS transistors M3 and M4. The gates of NMOS transistors M1 and M4 are connected to the local oscillator input port S0. The gates of NMOS transistors M2 and M3 are connected to the local oscillator input port S1. Port 3 is formed by connecting the sources of NMOS transistors M1 and M3. Port 4 is formed by connecting the sources of NMOS transistors M2 and M4.

5. A millimeter-wave broadband tunable filter according to claim 1, characterized in that, The baseband amplifier employs a two-stage structure. The first stage is a classic differential amplifier, and the second stage uses a source follower structure. The cutoff frequency of the amplifier is adjusted by controlling the magnitude of the injected current Iss, thereby achieving filter bandwidth control. Furthermore, common-mode feedback loops and negative feedback loops are introduced to ensure a stable frequency response. The amplifier's input terminals are IN+ and IN-. The IN+ port is connected to port 3 of a dual-balanced mixer I, and the IN- port is connected to port 4. The output terminals are OUT+ and OUT-. Port IN+ is formed by connecting the gate of NMOS transistor M9 and one end of resistor R4, and port IN- is formed by connecting NMOS transistor M... 10 The gate of the NMOS transistor M9 and one end of the resistor R5 are connected together to form a port Vbias. The other end of the resistor R4 is connected to the other end of the resistor R5. The source of the NMOS transistor M9 and the source of the NMOS transistor M... 10 The source and NMOS transistor M 12 The drain of NMOS transistor M9 and the gate of NMOS transistor M5 are connected to the drain of PMOS transistor M7. 10 The drain and NMOS transistor M 13 The gate of the NMOS transistor is connected to the drain of the PMOS transistor M8. 12 The gate and NMOS transistor M 11 The gate of the NMOS transistor M 11 The drain connection forms port Iss, and the NMOS transistor M 12 The source and NMOS transistor M 11 The source of NMOS transistor M5 is grounded. The source of NMOS transistor M5 and the drain of NMOS transistor M6 are connected to one end of resistor R7 to form the output port OUT+. The source of NMOS transistor M6 is grounded. The drain of NMOS transistor M5, the source of PMOS transistor M7, the source of PMOS transistor M8, and the source of NMOS transistor M7 are grounded. 13 The drain connection forms the port VDD. The gates of PMOS transistor M7 and PMOS transistor M8 are connected to the output terminal out of the error amplifier. The NMOS transistor M... 13 The source and NMOS transistor M 15 The drain of the NMOS transistor M is connected to one end of resistor R6 to form the output port OUT-. 15 The gate and NMOS transistor M 14 The gate and drain of the NMOS transistor M6 are connected to the gate of the NMOS transistor M6 to form port Ief. 14 The source and NMOS transistor M 15 The source of the resistor is grounded, and the other end of the resistor R6 and the other end of the resistor R7 are connected as the in+ port of the error amplifier. The in- port of the error amplifier is connected to VDD / 2.

6. A millimeter-wave broadband tunable filter according to claim 1, characterized in that, The input terminals of the dual-balanced mixer II are differential input ports 3 and 4, where port 3 is connected to the OUT+ port of the baseband amplifier and port 4 is connected to the OUT- port. The output terminals of the dual-balanced mixer II are differential output ports 1 and 2.

7. A millimeter-wave broadband tunable filter according to claim 1, characterized in that, The input terminals of the quadrature coupler II are IN+, IN-, CPL+, and CPL-. The IN+ port is connected to port 1 of one double-balanced mixer II, the IN- port is connected to port 2, the CPL+ port is connected to port 1 of two double-balanced mixer II, and the CPL- port is connected to port 2. The output terminals of the quadrature coupler II are differential output ports THRU+ and THRU-. The THRU+ port serves as the output port OUT+ of the filter, and the THRU- port serves as the output port OUT- of the filter.

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