An ultrafast commutated square wave ac welding power source

Abstract

This invention provides an ultra-fast commutation square-wave AC welding power supply, comprising: a rectifier bridge circuit for rectifying AC to DC; a filter and voltage regulator circuit for eliminating high-order ripple and obtaining smooth DC; a full-bridge inverter circuit for converting DC to high-frequency AC; a high-frequency transformer for isolation and voltage reduction; a secondary full-bridge rectifier circuit for converting high-frequency AC to DC; a coupling inductor for coupling; a half-bridge inverter circuit for converting DC to AC square waves; and a constant voltage clamping circuit; the load is connected between the midpoint of the half-bridge inverter circuit and the center tap of the high-frequency transformer. This power supply enables faster commutation of square-wave AC current and energy recovery during the commutation process, thereby improving the stability of the AC welding process and the conversion efficiency of the square-wave AC welding power supply.

Description

Technical Field

[0001] This invention relates to the field of welding power source technology, and more specifically, to a square wave AC welding power source with ultra-fast commutation. Background Technology

[0002] AC arc welding power supplies are commonly used in tungsten inert gas (TIG) welding and plasma arc welding of magnesium, aluminum, and their alloys because they remove oxides from the workpiece during the current output period when the electrode is positive (cathode cleaning). In AC TIG welding, the arc current crosses zero twice during each cycle, which can lead to arc interruption. To ensure continuous arc combustion, the power supply must provide a sufficiently high re-ignition voltage or a sufficiently high current commutation rate.

[0003] Regarding the former, Chinese invention patent "A Digital Variable Polarity Gas Shielded Welding Power Supply for Thin Plates" (Publication No.: CN114226918A) proposes an auxiliary arc stabilization circuit connected in parallel to the output terminal. This circuit provides a stable and adjustable high-voltage pulse during welding current commutation, ensuring reliable reignition of the welding arc. Regarding the latter, Chinese invention patent "AC Commutation Arc-Maintaining Circuit and AC Welding Power Supply" (Publication No.: CN114070117A) proposes an AC commutation arc-maintaining circuit connected in parallel to the main circuit. During current commutation, the energy stored in the capacitor is instantaneously superimposed onto the pulse current in the main circuit through a current-limiting resistor or inductor, achieving a sufficiently high current conversion rate. However, these methods all increase system complexity and can induce strong electromagnetic interference (EMI).

[0004] Traditional square-wave AC welding power supply topologies mainly consist of a high-frequency (20kHz) full-bridge inverter and a low-frequency (from DC to several hundred Hz) half-bridge inverter. The high-frequency full-bridge inverter is used to regulate the current amplitude (here, it can be assumed that the arc welding power supply is current-driven), and the low-frequency half-bridge inverter is used to switch the polarity of the output current. However, traditional AC arc welding power supply topologies struggle to achieve an ideal output current waveform because the arc welding power supply load is inductive, and the output current commutation can only occur at the arc voltage. V arc The process proceeds slowly in an exponential manner.

[0005] Existing research on AC arc welding power supplies has not considered the significant impact of the output cable inductance (in μH) on the polarity switching rate of the output current. Therefore, a new square-wave AC welding power supply needs to be designed, replacing the filter inductor in the traditional topology with a coupling inductor. The new topology was theoretically analyzed, and a novel constant-voltage clamping circuit was proposed to increase the voltage between the two electrodes when the current crosses zero and to improve the commutation rate of the output current. Summary of the Invention

[0006] To overcome the shortcomings and deficiencies of the existing technology, the purpose of this invention is to provide an ultra-fast commutation square wave AC welding power supply; this power supply can achieve faster commutation of square wave AC current and energy recovery during the commutation process, thereby improving the stability of the AC welding process and the conversion efficiency of the square wave AC welding power supply.

[0007] To achieve the above objectives, the present invention provides a square-wave AC welding power supply with ultra-fast commutation, comprising: A rectifier bridge circuit used to rectify alternating current into direct current; A filter and voltage regulator circuit used to eliminate high-order ripple and obtain smooth DC current; A full-bridge inverter circuit used to convert direct current into high-frequency alternating current; High-frequency transformers used for isolation and voltage reduction; Secondary full-bridge rectifier circuit used to convert high-frequency alternating current into direct current; Coupling inductors used for coupling; A half-bridge inverter circuit used to convert direct current into an alternating square wave; And constant voltage clamping circuit; The load is connected between the midpoint of the half-bridge inverter circuit and the center tap of the high-frequency transformer.

[0008] Preferably, the full-bridge inverter circuit is composed of switching transistors Q1 to Q4; wherein, switching transistors Q1 and Q3 are the upper and lower transistors of one inverter half-bridge; and switching transistors Q2 and Q4 are the upper and lower transistors of another inverter half-bridge. The secondary full-bridge rectifier circuit is composed of rectifier diodes D5 to D8; wherein, rectifier diodes D5 and D7 are the upper and lower transistors of one rectifier half-bridge; rectifier diodes D6 and D8 are the upper and lower transistors of the other rectifier half-bridge. The coupled inductor includes winding L1 and winding L2; The half-bridge inverter circuit consists of switching transistors Q5 and Q6; wherein, switching transistor Q5 is the upper transistor of the half-bridge inverter circuit, and switching transistor Q6 is the lower transistor of the half-bridge inverter circuit. The constant voltage clamping circuit includes clamping diodes D9 to D12, clamping capacitor C7, and a step-up / step-down module connected in parallel across the clamping capacitor C. The secondary winding of the coupling inductor L1 is connected to the collector of the switching transistor Q5 and the anode of the clamping diode D9; the secondary winding of the coupling inductor L2 is connected to the emitter of the switching transistor Q6 and the cathode of the clamping diode D12; the emitter of the switching transistor Q5 and the collector of the switching transistor Q6 are connected, and the connection point is connected to the cathode of the clamping diode 10 and the anode of the clamping diode D11; the transformer center tap, the connection point of the emitter of the switching transistor Q5 and the collector of the switching transistor Q6 are respectively connected to the two ends of the load through the output cable; The cathodes of clamping diode D9 and clamping diode D11 are connected to the upper end of clamping capacitor C7; the anodes of clamping diode D10 and clamping diode D12 are all grounded.

[0009] Preferably, the switching transistors Q1 to Q4 of the full-bridge inverter circuit operate using a phase-shift controlled switching sequence.

[0010] Preferably, the process of the welding power source's output current changing from positive to negative is performed sequentially by executing the following six switching modes: In switching mode 1, switch Q5 is turned on and switch Q6 is turned off; the secondary side of the high-frequency transformer forms a circuit through rectifier diode D5, coupled inductor winding L1, switch Q5, output cable and load; the load operates in positive polarity. In switching mode two, switch Q5 is off and switch Q6 is on; the secondary side of the high-frequency transformer is input to the coupling inductor winding L1 through rectifier diode D5 to generate current. i L1 Current i L1 A portion of the circuit is formed via clamping diode D9, clamping capacitor C7, clamping diode D10, output cable, and load; current i L1 Another portion passes through the coupling inductor winding L2, causing the current in the coupling inductor winding L2 to... i L2 The load operating mode is established starting with clamping diode D9, clamping capacitor C7, and clamping diode D12; the load operating mode switches from positive to 0A. In switching mode 3, switch Q5 is turned off, and switch Q6 remains on. The current in the coupling inductor winding L2 is split into two branches. One branch passes through the load, the output cable, and switch Q6, while the other branch passes through the coupling inductor winding L1, clamping diode D9, clamping capacitor C7, and clamping diode D12. The current then returns to the coupling inductor winding L2. The load operating mode switches from 0A to negative polarity. Switching mode four: Switch Q6 remains on, and switch Q5 remains off; high-frequency transformer secondary rectified voltage V S After rectification by rectifier diodes D6 and D8, a closed circuit is formed through coupling inductor winding L2, switching transistor Q6, output cable, and load; the load operates in negative polarity. Switching mode 5: Switch Q6 is on, switch Q5 is off; current in coupling inductor winding L2... i L2A portion of the current forms a loop through the load, output cable, clamping diode D11, clamping capacitor C7, and clamping diode D12. The other portion is coupled to the coupling inductor winding L1 through the mutual inductance of the coupling inductor, causing the current in the coupling inductor winding L1 to... i L1 The load operating mode begins to be established via the coupling inductor winding L1, clamping diode D9, clamping capacitor C7, clamping diode D12, and coupling inductor winding L2; the load operating mode switches from negative polarity to 0A. In switching mode six, switch Q5 is turned on and Q6 is turned off; the current in the coupling inductor winding L1 is divided into two branches. One branch passes through the coupling inductor winding L1, switch Q5, output cable and load, while the other branch passes through the coupling inductor winding L1, switch Q5, clamping diode D11, clamping capacitor C7, clamping diode D12, coupling inductor winding L2, and then returns to the coupling inductor winding L1; the load operating mode switches from 0A to positive polarity.

[0011] Preferably, the buck-boost module includes switching transistors Q7 to Q10 and a filter inductor Lc; The drain of switching transistor Q7 is connected to the upper end of clamping capacitor C7; the source of switching transistor Q8 and the lower end of switching transistor Q10 are connected to the source clamping capacitor C7; the source of switching transistor Q7 and the drain of switching transistor Q8 are connected, and the connection point is connected to the drain of switching transistor Q9 and the drain of switching transistor Q10 respectively through filter inductor Lc; the source of switching transistor Q9 is connected to the filter and voltage regulator circuit.

[0012] Preferably, two sets of complementary drive signals are used to drive the switching transistors Q7 to Q10; one set of drive signals drives the switching transistors Q7 and Q10 to conduct with the same duty cycle, and the other set of drive signals drives the switching transistors Q8 and Q9 to conduct with the same duty cycle. When the voltage VC1 of clamping capacitor C7 is detected to be greater than the set value, the duty cycle of switching transistors Q7 and Q10 is increased; when the voltage VC1 of clamping capacitor C7 is detected to be less than the set value, the duty cycle of switching transistors Q7 and Q10 is decreased.

[0013] Preferably, the switching transistors Q1 to Q4 of the full-bridge inverter circuit are each equipped with parallel body diodes and are each connected in parallel with a capacitor.

[0014] Compared with the prior art, the present invention has the following advantages and beneficial effects: 1. The ultra-fast commutation AC welding power supply topology proposed in this invention can realize faster commutation of square wave AC current and energy recovery during the commutation process, thereby improving the stability of AC welding process and the conversion efficiency of square wave AC welding power supply. 2. The present invention introduces a coupling inductor and a low-frequency half-bridge inverter on the secondary side. The coupling inductor provides energy through the magnetic field during commutation, which makes the current commutation faster and is conducive to the re-ignition and continuous combustion of the arc, thereby improving the stability of the AC welding process. The constant voltage clamping circuit generates a stable high voltage, which helps to ensure that the current can be quickly commutated when crossing zero, and improves the dynamic response of the welding machine. 3. The energy released by the coupling inductor in this invention is absorbed and recovered by the constant voltage clamping circuit, reducing energy loss during commutation and improving the conversion efficiency of the welding power supply. The constant voltage clamping circuit provides high voltage clamping while limiting voltage spikes in the switching transistors of the half-bridge inverter, reducing the risk of overvoltage and improving system reliability and service life. Attached Figure Description

[0015] Figure 1 This is a topology diagram of an ultra-fast commutation square wave AC welding power supply; Figure 2 This is the timing diagram of the full-bridge inverter circuit for an ultra-fast commutation square wave AC welding power supply. Figure 3 This is a working mode analysis diagram of an ultra-fast commutating square wave AC welding power supply with positive polarity output; Figure 4 This is a working mode analysis diagram of an ultra-fast commutation square wave AC welding power supply switching from positive polarity to 0A. Figure 5 This is a working mode analysis diagram of an ultra-fast commutation square wave AC welding power supply switching from 0A to negative polarity. Figure 6 This is a working mode analysis diagram of an ultra-fast commutating square wave AC welding power supply with negative polarity output; Figure 7 This is a working mode analysis diagram of an ultra-fast commutation square wave AC welding power supply switching from negative polarity to 0A. Figure 8 This is a working mode analysis diagram of an ultra-fast commutation square wave AC welding power supply switching from 0A to positive polarity. Detailed Implementation

[0016] The present invention will now be described in further detail with reference to the accompanying drawings and specific embodiments.

[0017] Example

[0018] This embodiment describes an ultra-fast commutation square wave AC welding power supply, which includes a rectifier bridge circuit, a filter and voltage regulator circuit, a full-bridge inverter circuit, a high-frequency transformer, a secondary full-bridge rectifier circuit, a coupling inductor, a half-bridge inverter circuit, and a constant voltage clamping circuit.

[0019] The rectifier bridge circuit rectifies the AC power input from the mains into DC power with higher ripple; the filter and voltage regulator circuit eliminates high-order ripple, resulting in smooth DC power; the full-bridge inverter circuit converts the DC power into high-frequency AC power; after being isolated and stepped down by the high-frequency transformer, the high-frequency AC power outputs a square-wave AC current capable of ultra-fast commutation under the combined action of the secondary full-bridge rectifier circuit, coupling inductor, half-bridge inverter circuit, and constant voltage clamping circuit. The load is connected between the midpoint of the half-bridge inverter circuit and the center tap of the high-frequency transformer.

[0020] like Figure 1 As shown, the full-bridge inverter circuit consists of switching transistors Q1 to Q4. Switches Q1 and Q3 form the upper and lower transistors of one inverter half-bridge; switches Q2 and Q4 form the upper and lower transistors of the other inverter half-bridge. Each of the switching transistors Q1 to Q4 in the full-bridge inverter circuit has a parallel diode and a capacitor connected in parallel. The connection point of switching transistors Q1 and Q3 is connected to the primary input terminal of the high-frequency transformer through a DC blocking capacitor Cb, and the connection point of switching transistors Q2 and Q4 is connected to the primary input terminal of the high-frequency transformer.

[0021] The secondary full-bridge rectifier circuit consists of rectifier diodes D5 to D8; rectifier diodes D5 and D7 form the upper and lower transistors of one rectifier half-bridge; rectifier diodes D6 and D8 form the upper and lower transistors of the other rectifier half-bridge. The coupling inductors include windings L1 and L2. The half-bridge inverter circuit consists of switching transistors Q5 and Q6; Q5 is the upper transistor of the half-bridge inverter circuit, and Q6 is the lower transistor. The constant voltage clamping circuit includes clamping diodes D9 to D12, clamping capacitor C7, and a step-up / step-down converter connected in parallel across clamping capacitor C.

[0022] The secondary winding of coupling inductor L1 is connected to the collector of switching transistor Q5 and the anode of clamping diode D9; the secondary winding of coupling inductor L2 is connected to the emitter of switching transistor Q6 and the cathode of clamping diode D12; the emitter and collector of switching transistor Q5 are connected, and the connection point is connected to the cathode of clamping diode 10 and the anode of clamping diode D11; the transformer center tap, the connection point of the emitter of switching transistor Q5 and the collector of switching transistor Q6 are respectively connected to the two ends of the load through the output cable (equivalent resistance is Ro, equivalent inductance is Lo); the cathodes of clamping diode D9 and clamping diode D11 are connected to the upper end of clamping capacitor C7; the anodes of clamping diode D10 and clamping diode D12 and the lower end of clamping capacitor C7 are all grounded.

[0023] The step-up / step-down module includes switching transistors Q7 to Q10 and a filter inductor Lc; the drain of switching transistor Q7 is connected to the upper end of clamping capacitor C7; the source of switching transistor Q8 and switching transistor Q10 are connected to the lower end of source clamping capacitor C7; the source of switching transistor Q7 and the drain of switching transistor Q8 are connected, and the connection point is connected to the drain of switching transistor Q9 and the drain of switching transistor Q10 respectively through the filter inductor Lc; the source of switching transistor Q9 is connected to the filter and voltage regulator circuit.

[0024] Common control strategies for full-bridge inverter circuits include finite bipolar and phase-shift control strategies. This embodiment analyzes the operation of the full-bridge inverter circuit with phase-shift control for the switching timing of transistors Q1 to Q4.

[0025] like Figure 2 As shown, Switching Phase 1 (t0~t1): During t0~t1, both switching transistors Q1 and Q4 are turned on. At this time, the grid voltage, after rectification and filtering, provides energy to the subsequent circuits. Ignore the on-state voltage drop of the switching transistors. The voltage across the high-frequency transformer is equal to the bus voltage, with positive at the top and negative at the bottom. The primary current Ip rises at a certain slope through switching transistor Q1, DC blocking capacitor Cb, the high-frequency transformer, and switching transistor Q4. The secondary current Is = Ip / n flows through rectifier diode D5 and coupling inductor winding L1 to provide energy to the load.

[0026] Phase Two (t1~t2): At time t1, switch Q1 is turned off. During this time, capacitors C3 and C5, connected in parallel with switches Q1 and Q3, provide energy to the load; C3 is charged and C5 is discharged. During this period, the leakage inductance of the high-frequency transformer (the resonant inductance is the leakage inductance Lk of the high-frequency transformer) and the coupling inductance of the secondary winding are in series, and the coupling inductance is very large. Therefore, the primary current can be considered approximately constant, and similarly, the secondary current is also approximately constant. Due to the presence of capacitors C3 and C5 in parallel with the switches, switch Q1 is turned off with zero voltage. The flow directions of the primary and secondary currents are the same as in the previous period.

[0027] Phase 3 (t2~t3): At time t2, the voltage across capacitor C5 drops to zero, and the parallel diode D3 of switch Q3 naturally conducts and freewheels. After this, switch Q3 is turned on with zero voltage. At this time, the primary current Ip flows through switch Q3, parallel diode D3, DC blocking capacitor Cb, high-frequency transformer, and switch Q4, freewheeling and slowly decreasing. The voltage across the high-frequency transformer is opposite to the current flowing through it, short-circuiting the high-frequency transformer and ceasing energy transfer. The current flowing through rectifier diode D5 slowly decreases in the original direction, while rectifier diode D6 conducts and freewheels.

[0028] Phase 4 (t3~t4): At time t3, switch Q4 is turned off. Due to the presence of capacitors C4 and C6 connected in parallel with switches Q2 and Q4, switch Q4 is turned off at zero voltage. At this time, the primary current freewheeling circuit shifts from switch Q4 to capacitors C4 and C6 connected in parallel with switches Q2 and Q4. C4 is discharged and C6 is charged. Since the high-frequency transformer is short-circuited, only the leakage inductance of the high-frequency transformer participates in the charging and discharging process of capacitors C4 and C6, so the primary current drops significantly. At this time, rectifier diodes D5 and D6 conduct simultaneously. The high-frequency transformer is short-circuited and no energy is transferred. The current in the coupling inductor winding L1 freewheels through rectifier diodes D5 and D6.

[0029] Phase 5 (t4~t5): At time t4, the voltage across capacitor C4 drops to zero, and the parallel diode D2 of switch Q2 naturally conducts and freewheels. At this time, switch Q2 is turned on, and Q2 is turned on with zero voltage. Although switch Q2 is turned on at this time, no current flows through switch Q2. Ip still flows through switch Q2 and parallel diode D2, and decreases linearly under the influence of the bus voltage. When the primary current drops to zero, current ip begins to flow through switch Q2 and rises in reverse. The current of secondary rectifier diode D5 decreases linearly, and the current of D6 increases linearly. In this phase, since secondary rectifier diodes D5 and D6 are turned on simultaneously, no energy transfer still occurs.

[0030] The above describes the work situation for half a cycle; the work situation for the other half of the cycle is reversed and symmetrical.

[0031] The working principle of the half-bridge inverter circuit is introduced below.

[0032] The instantaneous commutation of the square wave current can be considered as charging the equivalent inductance of the output cable, and its rate of rise is determined by the magnitude of the voltage u applied across it. That is:

[0033] Where L is the equivalent inductance of the output cable; The current rise rate.

[0034] Assuming the voltage across clamping capacitor C7 remains approximately constant at VC1, and the coupling inductors L1 and L2 are equal and strongly coupled, ignoring the on-state voltage drop, the process of the output current commutating from positive to negative can be divided into the following switching modes (independent of the switching stages described above): Switching Mode 1 (T0~T1): During times T0~T1, such as Figure 3As shown, switch Q5 is on and switch Q6 is off; clamping diodes D9 to D12 are all reverse-biased and cut off; the secondary side of the high-frequency transformer forms a circuit through rectifier diode D5, coupling inductor winding L1, switch Q5, output cable, and load; the load operates in positive polarity; coupling inductor winding L1 is used as a general filter inductor, and the current in coupling inductor winding L1 is... i L1 ≈ Output current i o It fluctuates around the set value with relatively small ripple; the current in the coupled inductor winding L2 is approximately zero.

[0035] Switching mode two (T1~T2): At time T1, as follows Figure 4 As shown, switch Q5 is off and switch Q6 is on. Since the current in the coupled inductor winding L1 cannot change abruptly, the original forward current needs to find a new freewheeling path. At this time, clamping diodes D9, D10, and D12 are forward-biased and D11 is reverse-biased. The secondary side of the high-frequency transformer is input to the coupled inductor winding L1 through rectifier diode D5 to generate current. i L1 Current i L1 A portion of the circuit is formed via clamping diode D9, clamping capacitor C7, clamping diode D10, output cable, and load; current i L1 Another portion passes through the coupling inductor winding L2, causing the current in the coupling inductor winding L2 to... i L2 The clamping process begins with clamping diode D9, clamping capacitor C7, and clamping diode D12; the load operating mode switches from positive to 0A. Since the voltage across clamping capacitor C7 is approximately a constant clamping voltage... V C1 The output terminal is equivalent to having a constant reverse high voltage applied. V AB Output current i o exist V AB Under the influence of the coupling inductor, the voltage decreases rapidly with an approximately linear slope. The energy stored in the coupling inductor winding L1 and the equivalent inductance Lo of the output cable is partially transferred to the coupling inductor winding L2 and partially transferred to the clamping capacitor C7, causing the voltage of the clamping capacitor C7 to increase slightly.

[0036] Switching mode 3 (T2~T3): At time T2, such as Figure 5As shown, switch Q5 is off, while switch Q6 remains on; clamping diodes D9 and D12 are forward-biased, while D10 and D11 are reverse-biased; the current in the coupling inductor winding L2 is split into two branches, one of which passes through the load, output cable, and switch Q6, and the other passes through the coupling inductor winding L1, clamping diode D9, clamping capacitor C7, and clamping diode D12; then it returns to the coupling inductor winding L2; the load operating mode switches from 0A to negative polarity. Under constant clamping voltage... V C1 Under this action, the output current increases rapidly from zero to the negative current value with an approximately linear slope. i 1. The electric arc maintains a high voltage excitation throughout the zero crossing process, which is conducive to the re-ignition and stable combustion of the arc.

[0037] Switching mode four (T3~T4): After time T3, such as Figure 6 As shown, the output current has been established as a stable negative current, with switch Q6 remaining on and switch Q5 remaining off; clamping diodes D9 to D12 are all reverse-biased and out of the main current loop; the secondary rectified voltage V of the high-frequency transformer... S After rectification by rectifier diodes D6 and D8, a closed loop is formed through the coupling inductor winding L2, the switching transistor Q6, the output cable, and the load. The load operates in negative polarity. The coupling inductor winding L2 is used as a filter inductor, and the output current... i o The voltage changes slowly near the set negative value. After a slight change in the previous switching mode, the voltage of clamping capacitor C7 is restored and maintained at the preset clamping voltage through energy exchange between the switching transistors Q7-Q10 of the subsequent buck-boost module, the filter inductor Lc, and the bus capacitor. V C1 This prepares for the next current commutation from negative to positive.

[0038] Switching mode five (T4~T5): such as Figure 7 As shown, switch Q6 is on and switch Q5 is off. Since the current in L2 cannot change abruptly, the original reverse current needs to find a new freewheeling path. At this time, clamping diodes D9, D11, and D12 are forward-biased and D10 is reverse-biased. The current in the coupling inductor winding L2... i L2 A portion of the current forms a loop through the load, output cable, clamping diode D11, clamping capacitor C7, and clamping diode D12. The other portion is coupled to the coupling inductor winding L1 through the mutual inductance of the coupling inductor, causing the current in the coupling inductor winding L1 to... i L1The coupling inductor winding L1, clamping diode D9, clamping capacitor C7, clamping diode D12, and coupling inductor winding L2 begin to establish; the load operating mode switches from negative polarity to 0A; since the voltage across clamping capacitor C7 is approximately constant... V C1 The output terminal is equivalent to having a constant reverse high voltage applied. V AB Output current i o exist V AB Under the influence of the coupling inductor, the voltage decreases rapidly with an approximately linear slope. The energy stored in the coupling inductor winding L2 and the equivalent resistance Lo of the output cable is partially transferred to the coupling inductor winding L1 and partially transferred to the clamping capacitor C7, causing the voltage of the clamping capacitor C7 to increase slightly.

[0039] Switching Mode 6 (T5~T6): At time T5, such as Figure 8 As shown, the output current i o As the current decreases to zero, the current in the coupled inductor winding L2 continues to decay, while the current in the coupled inductor winding L1 continues to rise. During this stage, switch Q5 is turned on, Q6 is turned off, clamping diodes D11 and D12 are forward-biased, and D9 and D10 are reverse-biased. The current in the coupled inductor winding L1 splits into two branches. One branch passes through the coupled inductor winding L1, switch Q5, output cable, and load. The other branch passes through the coupled inductor winding L1, switch Q5, clamping diode D11, clamping capacitor C7, clamping diode D12, and coupled inductor winding L2, before returning to the coupled inductor winding L1. The load operating mode switches from 0A to positive polarity. Under the action of constant clamping voltage VC1, the output current rapidly increases from zero to the forward current value with an approximately linear slope. i 1. The arc maintains a high voltage excitation throughout the zero-crossing process, which is beneficial for the arc to reignite and burn stably. The circuit enters a new positive steady-state output stage at time T6, corresponding to the operating state of the aforementioned switching mode one.

[0040] The constant voltage clamping circuit, controlled independently of other power supply modules, primarily maintains the stability of the clamping capacitor voltage. Switches Q7 through Q10 are driven by two sets of complementary drive signals. One set of signals turns on Q7 and Q10 with the same duty cycle, while the other set turns on Q8 and Q9 with the same duty cycle. When the clamping capacitor C7 voltage VC1 is detected to be greater than a set value, the duty cycle of switches Q7 and Q10 is increased; when the clamping capacitor C7 voltage VC1 is detected to be less than a set value, the duty cycle of switches Q7 and Q10 is decreased.

[0041] The above embodiments are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. Any changes, modifications, substitutions, combinations, or simplifications made without departing from the spirit and principle of the present invention shall be considered equivalent substitutions and shall be included within the protection scope of the present invention.

Claims

1. A square wave AC welding power supply with ultra-fast commutation, characterized in that: include: A rectifier bridge circuit used to rectify alternating current into direct current; A filter and voltage regulator circuit used to eliminate high-order ripple and obtain smooth DC current; A full-bridge inverter circuit used to convert direct current into high-frequency alternating current; High-frequency transformers used for isolation and voltage reduction; Secondary full-bridge rectifier circuit used to convert high-frequency alternating current into direct current; Coupling inductors used for coupling; A half-bridge inverter circuit used to convert direct current into an alternating square wave; And constant voltage clamping circuit; The load is connected between the midpoint of the half-bridge inverter circuit and the center tap of the high-frequency transformer.

2. The ultra-fast commutation square wave AC welding power supply according to claim 1, characterized in that: The full-bridge inverter circuit consists of switching transistors Q1 to Q4; wherein, switching transistors Q1 and Q3 are the upper and lower transistors of one inverter half-bridge; and switching transistors Q2 and Q4 are the upper and lower transistors of another inverter half-bridge. The secondary full-bridge rectifier circuit is composed of rectifier diodes D5 to D8; wherein, rectifier diodes D5 and D7 are the upper and lower transistors of one rectifier half-bridge; rectifier diodes D6 and D8 are the upper and lower transistors of the other rectifier half-bridge. The coupled inductor includes winding L1 and winding L2; The half-bridge inverter circuit consists of switching transistors Q5 and Q6; wherein, switching transistor Q5 is the upper transistor of the half-bridge inverter circuit, and switching transistor Q6 is the lower transistor of the half-bridge inverter circuit. The constant voltage clamping circuit includes clamping diodes D9 to D12, clamping capacitor C7, and a step-up / step-down module connected in parallel across the clamping capacitor C. The secondary winding of the coupling inductor L1 is connected to the collector of the switching transistor Q5 and the anode of the clamping diode D9; the secondary winding of the coupling inductor L2 is connected to the emitter of the switching transistor Q6 and the cathode of the clamping diode D12; the emitter of the switching transistor Q5 and the collector of the switching transistor Q6 are connected, and the connection point is connected to the cathode of the clamping diode 10 and the anode of the clamping diode D11; the transformer center tap, the connection point of the emitter of the switching transistor Q5 and the collector of the switching transistor Q6 are respectively connected to the two ends of the load through the output cable; The cathodes of clamping diode D9 and clamping diode D11 are connected to the upper end of clamping capacitor C7; the anodes of clamping diode D10 and clamping diode D12 are all grounded.

3. The ultra-fast commutation square wave AC welding power supply according to claim 2, characterized in that: The switching transistors Q1 to Q4 of the full-bridge inverter circuit operate using a phase-shift controlled switching sequence.

4. The ultra-fast commutation square wave AC welding power supply according to claim 3, characterized in that: The process of the welding power supply's output current changing from positive to negative involves sequentially executing the following six switching modes: In switching mode 1, switch Q5 is turned on and switch Q6 is turned off; the secondary side of the high-frequency transformer forms a circuit through rectifier diode D5, coupled inductor winding L1, switch Q5, output cable and load; the load operates in positive polarity. In switching mode two, switch Q5 is off and switch Q6 is on; the secondary side of the high-frequency transformer is input to the coupling inductor winding L1 through rectifier diode D5 to generate current. i L1 Current i L1 A portion of the circuit is formed via clamping diode D9, clamping capacitor C7, clamping diode D10, output cable, and load; current i L1 Another portion passes through the coupling inductor winding L2, causing the current in the coupling inductor winding L2 to... i L2 The load operating mode is established starting with clamping diode D9, clamping capacitor C7, and clamping diode D12; the load operating mode switches from positive to 0A. In switching mode 3, switch Q5 is turned off, and switch Q6 remains on. The current in the coupling inductor winding L2 is split into two branches. One branch passes through the load, the output cable, and switch Q6, while the other branch passes through the coupling inductor winding L1, clamping diode D9, clamping capacitor C7, and clamping diode D12. The current then returns to the coupling inductor winding L2. The load operating mode switches from 0A to negative polarity. Switching mode four: Switch Q6 remains on, and switch Q5 remains off; high-frequency transformer secondary rectified voltage V S After rectification by rectifier diodes D6 and D8, a closed circuit is formed through coupling inductor winding L2, switching transistor Q6, output cable, and load; the load operates in negative polarity. Switching mode 5: Switch Q6 is on, switch Q5 is off; current in coupling inductor winding L2... i L2 A portion of the current forms a loop through the load, output cable, clamping diode D11, clamping capacitor C7, and clamping diode D12. The other portion is coupled to the coupling inductor winding L1 through the mutual inductance of the coupling inductor, causing the current in the coupling inductor winding L1 to... i L1 The load operating mode begins to be established via the coupling inductor winding L1, clamping diode D9, clamping capacitor C7, clamping diode D12, and coupling inductor winding L2; the load operating mode switches from negative polarity to 0A. In switching mode six, switch Q5 is turned on and Q6 is turned off; the current in the coupling inductor winding L1 is divided into two branches. One branch passes through the coupling inductor winding L1, switch Q5, output cable and load, while the other branch passes through the coupling inductor winding L1, switch Q5, clamping diode D11, clamping capacitor C7, clamping diode D12, coupling inductor winding L2, and then returns to the coupling inductor winding L1; the load operating mode switches from 0A to positive polarity.

5. The ultra-fast commutation square wave AC welding power supply according to claim 2, characterized in that: The buck-boost module includes switching transistors Q7 to Q10 and a filter inductor Lc; The drain of switching transistor Q7 is connected to the upper end of clamping capacitor C7; the source of switching transistor Q8 and the lower end of switching transistor Q10 are connected to the source clamping capacitor C7; the source of switching transistor Q7 and the drain of switching transistor Q8 are connected, and the connection point is connected to the drain of switching transistor Q9 and the drain of switching transistor Q10 respectively through filter inductor Lc; the source of switching transistor Q9 is connected to the filter and voltage regulator circuit.

6. The ultra-fast commutation square wave AC welding power supply according to claim 5, characterized in that: Two sets of complementary drive signals are used to drive the switching transistors Q7 to Q10; one set of drive signals drives the switching transistors Q7 and Q10 to conduct with the same duty cycle, and the other set of drive signals drives the switching transistors Q8 and Q9 to conduct with the same duty cycle. When the voltage VC1 of clamping capacitor C7 is detected to be greater than the set value, the duty cycle of switching transistors Q7 and Q10 is increased; when the voltage VC1 of clamping capacitor C7 is detected to be less than the set value, the duty cycle of switching transistors Q7 and Q10 is decreased.

7. The ultra-fast commutation square wave AC welding power supply according to claim 2, characterized in that: The switching transistors Q1 to Q4 of the full-bridge inverter circuit are each equipped with parallel body diodes and are each connected in parallel with a capacitor.

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