Spread spectrum signal acquisition method that eliminates the effects of high speed data symbols

Abstract

This invention discloses a method for acquiring spread spectrum signals to eliminate the influence of high-speed data symbols, comprising the following steps: When down-converting the input signal obtained from the ADC sampling receiver, a fixed frequency offset is artificially introduced; a local pseudo-code sequence is correlated with the down-converted signal; the correlated signal is low-pass filtered; the I-path and Q-path of the filtered signal are multiplied, and the result is input to the FFT calculation module; the highest energy spectral line is found in the spectrum output by the FFT module, and the Doppler frequency offset of the input signal is calculated if the energy of this spectral line exceeds a threshold, then pseudo-code locking and carrier locking are performed; it is determined whether pseudo-code locking and carrier locking are completed: if locked, the acquisition process ends; if not locked, the phase of the local pseudo-code sequence is adjusted and the process returns to the down-conversion step. This invention can eliminate the influence of high-speed data symbol flipping on spread spectrum signal acquisition, is applicable to different data rates, and significantly enhances the communication capability of spread spectrum communication systems.

Description

Technical Field

[0001] This invention relates to the field of spread spectrum communication, and more specifically to a method for acquiring spread spectrum signals that eliminates the influence of high-speed data symbols. Background Technology

[0002] Spread spectrum communication features anti-interference, concealment, and multiple access capabilities, and is widely used in military, inter-satellite, and space-to-ground communication systems. With the development of these systems, the amount of communication data has increased accordingly, placing increasingly higher demands on the data rate of spread spectrum communication. Taking space-to-ground spread spectrum telemetry and control communication systems as an example, early telemetry and control systems had relatively low data rates, typically between 100 bps and 4 kbps, while currently, the data rates of commonly used telemetry and control transponders on the market have generally reached 16 kbps.

[0003] Spread spectrum signal acquisition is the primary step in spread spectrum reception. Its main purpose is to rapidly achieve precise synchronization between the local pseudocode and the received pseudocode, and to lock the carrier Doppler frequency offset within a two-dimensional space of pseudocode phase and carrier Doppler frequency offset. Currently, the mainstream spread spectrum signal acquisition method first correlates the local pseudocode with the spread spectrum input signal converted to baseband (despreading the input signal can be achieved when the local pseudocode and the input pseudocode are perfectly aligned). Then, the correlated signal is sampled at a certain time interval (denoted as Ts), and an FFT operation is performed to find the highest energy spectral line in the signal spectrum. If the energy of the highest energy spectral line in the signal spectrum exceeds a threshold, it can be preliminarily determined that the local pseudocode and the received pseudocode have achieved coarse synchronization. Simultaneously, the frequency corresponding to the highest energy spectral line is the carrier Doppler frequency offset of the input signal. This result can be used to assist the carrier tracking loop in tracking and locking the carrier of the input signal. The signal sampled within the Ts time interval and input to the FFT operation module is crucial. If the data symbols flip within the Ts interval, the energy of the highest spectral line in the FFT input spectrum will decrease, thereby reducing the acquisition probability of the pseudocode and carrier Doppler. If the data rate is high enough that multiple flips occur in the Ts interval, it will cause the FFT input spectrum to be distorted, which will lead to a loss of correspondence between the highest energy spectral line and the carrier Doppler frequency offset, making the acquisition process difficult to complete.

[0004] Current spread spectrum signal acquisition methods are mostly based on the assumption that data symbols remain unchanged within the sampling interval. Therefore, these methods are mainly suitable for low communication rates. This is exemplified by the papers "Block Acquisition of Weak GPS Signals in a Software Receiver" (author: MLPsiaki) and "A Weak Signal Acquisition Method in a Direct-Sequence Spread Spectrum System Based on FFT" (author: Li Dafeng), both mentioning a data rate of 50 bps. Even with currently available telemetry and control transponders, with a pseudocode rate of 10.23 Mbps and a code sequence period of 1023 chips, the communication rate reaches its limit at 16 kbps. If the communication rate is further increased, for example to 32 kbps or 64 kbps, existing methods struggle to acquire spread spectrum signals. Summary of the Invention

[0005] This invention provides a spread spectrum signal acquisition method that eliminates the influence of high-speed data symbols, solving the technical problem mentioned above that traditional acquisition methods cannot support high-speed spread spectrum communication. Specifically, the technical solution is as follows: A method for acquiring spread spectrum signals that eliminates the influence of high-speed data symbols includes the following steps: When down-converting the receiver input signal obtained by the ADC sampling, a fixed frequency offset is artificially introduced; Correlate the local pseudocode sequence with the down-converted signal; Perform low-pass filtering on the correlated signal; The I-path and Q-path of the filtered signal are multiplied, and the result of the multiplication is input into the FFT operation module; Find the highest energy spectral line in the spectrum output by the FFT module: if the energy of the spectral line exceeds the threshold, calculate the Doppler frequency offset of the input signal, and then proceed to pseudocode locking and carrier locking; if the energy of the spectral line does not exceed the threshold, adjust the phase of the local pseudocode sequence and return to the downconversion step. Determine whether pseudocode locking and carrier locking are completed: if locked, the acquisition process ends; if not locked, adjust the phase of the local pseudocode sequence and return to the downconversion step.

[0006] Furthermore, the receiver's desired Doppler tracking range is -F. d ≤ f d ≤ +F d The artificially introduced fixed frequency offset is +F. d .

[0007] Furthermore, the expression for the input signal is: , in, The value is ±1, which represents the bipolar digital signal mapped to ±1 after the pseudocode and data are XORed. Indicates the frequency of the input signal; The expression for the local oscillator signal of the down-conversion is: , in F represents the frequency of the local oscillator signal. d This indicates the magnitude of a fixed frequency offset introduced artificially. The signal expression after downconversion is: , in, Let the input signal represent the true carrier Doppler. After low-pass filtering the correlated signal, then... , in, This is the signal frequency offset.

[0008] Furthermore, if the local pseudocode is... If the pseudocode phase is aligned or nearly aligned, then... After near-despreading, the spectral width of the correlated signal depends primarily on the data rate. If the local pseudocode and If the pseudocode phase is not aligned, the correlated signal will still be in a spread spectrum state, and its spectral width mainly depends on the pseudocode rate. The signal expression after correlation is: , in, This represents the local pseudocode signal represented by bipolar code.

[0009] Furthermore, the low-pass filtering of the correlated signal is used to limit the signal energy input to the FFT module when the local pseudocode sequence is not aligned with the pseudocode sequence in the input signal; The bandwidth of the low-pass filter used is 2 to 10 times the data rate.

[0010] Furthermore, the multiplication of the I and Q paths is used to eliminate modulated data symbols in the signal, resulting in a single-tone signal that is independent of data symbols and pseudo-code signals and only related to the carrier frequency offset. : , The frequency of the single-tone signal is equal to twice the carrier frequency offset of the down-converted signal.

[0011] Furthermore, if the frequency corresponding to the highest energy spectral line is found through FFT operation... If the spectral line energy exceeds the threshold, the carrier Doppler frequency offset can be calculated using the following formula: , Calculated It is used as an auxiliary carrier tracking loop to lock onto the carrier in the input signal.

[0012] Furthermore, the threshold is in a fixed proportion to the total energy of the FFT output spectrum for each time, and the proportion is set according to the requirements of the false alarm probability and the missed alarm probability.

[0013] Furthermore, the phase adjustment step of the local pseudocode sequence is one-quarter of a chip.

[0014] Furthermore, the pseudo-code locking and carrier locking employ a carrier tracking loop and a pseudo-code tracking loop. After waiting for a preset time, it is determined whether the carrier tracking loop and the pseudo-code tracking loop have successfully locked. If both are locked, the spread spectrum signal acquisition process ends successfully; otherwise, the local pseudo-code sequence phase is adjusted and the process returns to the down-conversion step.

[0015] The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols provided by this invention eliminates the impact of high-speed data symbol flipping on FFT spectrum analysis by multiplying the I and Q paths, solving the limitation of traditional methods that require data symbols to remain unchanged within the sampling interval, thus enabling spread spectrum communication systems to support rates of 32kbps, 64kbps and higher. It also solves the frequency sign ambiguity problem caused by the symmetry of the real signal spectrum by artificially introducing a fixed frequency offset; and effectively suppresses signal energy when pseudo-code is misaligned by using a low-pass filter, avoiding invalid lock-in judgments and significantly shortening the acquisition time. The overall scheme is simple to implement, possesses both high data rate adaptability and fast acquisition capability, and significantly enhances the communication capability of spread spectrum communication systems. Attached Figure Description

[0016] To more clearly illustrate the technical solutions in the embodiments of this application or the prior art, the drawings used in the description of the embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are only some embodiments of this application. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0017] Figure 1 This is a flowchart illustrating the implementation of the present invention; Figure 2 A block diagram of a traditional spread spectrum signal acquisition system; Figure 3 This is a block diagram of the spread spectrum signal acquisition system of the present invention; Figure 4 The input signal spectrum is shown in a specific embodiment of the present invention. Figure 5 This is the spectrum of the down-conversion signal in a specific embodiment of the present invention; Figure 6 This is the spectrum of the relevant signal when the local pseudocode is aligned with the input signal pseudocode in a specific embodiment of the present invention; Figure 7 This is the spectrum of the relevant signal when the local pseudocode and the input signal pseudocode are not aligned in a specific embodiment of the present invention; Figure 8 This is the spectrum of the related signal after passing through a low-pass filter when the local pseudocode is aligned with the input signal pseudocode in a specific embodiment of the present invention; Figure 9 This is the spectrum of the related signal after passing through a low-pass filter when the local pseudocode and the input signal pseudocode are not aligned in a specific embodiment of the present invention; Figure 10 This is the spectrum calculation result of the FFT output when the local pseudocode is aligned with the input signal pseudocode in a specific embodiment of the present invention; Figure 11 This is the spectrum calculation result of the FFT output when the local pseudocode and the input signal pseudocode are not aligned in a specific embodiment of the present invention; Figure 12 This is the ratio between the highest spectral energy of the FFT output spectrum and the total spectral energy under different pseudocode phase deviations without low-pass filtering in a specific embodiment of the present invention. Figure 13 This refers to the ratio between the highest spectral energy of the FFT output spectrum and the total spectral energy under different pseudocode phase deviations when low-pass filtering is used in a specific embodiment of the present invention. Detailed Implementation

[0018] The embodiments of this application are described in detail below. Examples of the embodiments are shown in the accompanying drawings, wherein the same or similar reference numerals denote the same or similar elements or elements having the same or similar functions throughout. The embodiments described below with reference to the accompanying drawings are exemplary and intended to explain this application, and should not be construed as limiting this application.

[0019] like Figure 1 The diagram illustrates a spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to this application, comprising the following steps: S1) When down-converting the receiver input signal obtained by the ADC sampling, a fixed frequency offset is artificially introduced so that the frequency of the down-converted signal is constrained to between 0 and 2Fd, where Fd is the upper limit of the Doppler tracking range expected by the receiver. S2) Correlate the local pseudocode sequence with the down-converted signal; S3) Perform low-pass filtering on the correlated signal; S4) Multiply the I-path and Q-path of the filtered signal and input the result of the multiplication into the FFT operation module; S5) Find the highest energy spectral line in the spectrum output by the FFT module: if the energy of the spectral line exceeds the threshold, calculate the Doppler frequency offset of the input signal, and then enter pseudocode locking and carrier locking; if the energy of the spectral line does not exceed the threshold, adjust the phase of the local pseudocode sequence and return to the downconversion step. S6) Determine whether pseudocode locking and carrier locking are completed: if locked, the acquisition process ends; if not locked, adjust the phase of the local pseudocode sequence and return to the downconversion step.

[0020] In step S1 above, the signal acquired by the ADC is input to the receiver. Then, during the down-conversion processing of the input signal within the receiver, a fixed frequency offset is actively introduced. The magnitude of this artificially introduced fixed frequency offset is related to the receiver's desired Doppler tracking range; that is, if the receiver's desired Doppler tracking range is -Fd ≤ fd ≤ +Fd, then the magnitude of the artificially introduced fixed frequency offset is +Fd. d This makes the original distribution in ±F d The true carrier Doppler frequency offset f within the range d Moved to 0-2F d The positive frequency range is defined, meaning the frequency of the down-converted signal is constrained to between 0 and 2Fd. This ensures that the effective spectral lines in the real signal spectrum after I-channel and Q-channel multiplication always lie in the positive frequency region, facilitating direct extraction of frequency information without needing to distinguish between positive and negative signs. This step solves the symmetry problem of the real signal spectrum generated after subsequent I-channel and Q-channel multiplication, making the frequency sign distinguishable and ensuring the correct calculation of Doppler frequency offset. Thus, the quantitative relationship between fixed frequency offset and Doppler tracking range is clarified, ensuring the accuracy and consistency of frequency shifting and simplifying the subsequent frequency offset calculation logic.

[0021] Specifically, the expression for the input signal in the above steps is as follows: , in, The value is ±1, representing a bipolar digital signal mapped to ±1 after the pseudocode and data are XORed. This indicates the frequency of the input signal.

[0022] The expression for the local oscillator signal of the down-conversion is: , in Fd represents the frequency of the local oscillator signal, and Fd represents the magnitude of the artificially introduced fixed frequency deviation.

[0023] The signal expression after downconversion is: , in, Representing the true carrier Doppler of the input signal, low-pass filtering is applied to the correlated signal, which can ignore the sum-frequency components. Therefore: , As can be seen from this formula, after a fixed frequency offset is artificially introduced into the downconversion, the signal frequency offset becomes... .

[0024] Step S2) above uses the local pseudo-code sequence to correlate with the down-converted signal to achieve pseudo-code despreading. The locally generated pseudo-code sequence is correlated with the down-converted signal. When the local pseudo-code is phase-aligned or nearly aligned with the pseudo-code in the input signal, the signal is despread, and the spectral width changes from the pseudo-code rate to the data rate level. When they are not aligned, the signal remains spread, and the spectral width still depends on the pseudo-code rate. In other words, if the local pseudo-code and the input signal are phase-aligned... If the pseudocode phase is aligned or nearly aligned, then... After near-despreading, the spectral width of the correlated signal depends primarily on the data rate; if the local pseudocode and If the pseudocode phase is not aligned, the correlated signal will still be in a spread spectrum state, and its spectral width will mainly depend on the pseudocode rate. This method clarifies the signal spectral differences between the aligned and unaligned pseudocode states, providing a basis for selecting the low-pass filter bandwidth and ensuring that the filter can effectively distinguish between the two states.

[0025] Specifically, the signal expression after correlation is as follows: , in, This represents the local pseudocode signal represented by bipolar code.

[0026] Step S3) above performs low-pass filtering on the correlated signal to achieve energy control. In other words, low-pass filtering of the correlated signal limits the signal energy input to the FFT module when the local pseudocode sequence is misaligned with the pseudocode sequence in the input signal. The bandwidth of the low-pass filter used is 2 to 10 times the data rate to filter the despread signal. If the correlated signal is close to... If the despread version is used, then the low-pass filter will hardly attenuate the input signal, and the spectrum of the output signal will be consistent with the spectrum of the input signal. That is, when the pseudocode is aligned, the signal energy passes through with almost no attenuation; if the correlated signal is close to... In a spread spectrum version, the portion of the signal outside the low-pass filter's passband will be significantly attenuated, resulting in an output signal with energy much lower than the input signal. In other words, when the pseudocode is misaligned, most of the spread spectrum signal's energy is filtered out. This mechanism effectively distinguishes between aligned and misaligned pseudocodes, avoiding the triggering of complex locking decisions when the pseudocode is misaligned, thus significantly improving acquisition efficiency. This means that regardless of whether the pseudocode is aligned or not, a single tone can be obtained. The result of performing an FFT operation on the signal is almost always a spectrum of a single-tone signal, in which the spectral energy of the single-tone signal is likely to exceed the threshold, thereby triggering the process in step six to determine whether carrier locking and pseudocode locking have been completed.

[0027] Determining carrier lock and pseudocode lock typically takes a considerable amount of time. If this process is performed after every FFT operation, the entire spread spectrum signal acquisition process will take a very long time, potentially tens of seconds. Therefore, introducing a low-pass filter in step three limits the signal energy supplied to the FFT module when the pseudocode is misaligned. This avoids triggering the carrier and pseudocode lock determination process in this situation, reducing the overall time required for spread spectrum signal acquisition.

[0028] Step S4) above multiplies the I-path and Q-path of the filtered signal and inputs the result to the FFT calculation module. This step directly multiplies the I-path (in-phase component) and Q-path (quadrature component) of the filtered complex signal, using trigonometric functions to eliminate the influence of the modulated data symbol d(t) and pseudocode p(t) in the signal, resulting in a single-tone signal that is independent of both the data symbol and the pseudocode, and only related to the carrier frequency offset. The frequency of this single-tone signal is twice the carrier frequency offset after down-conversion, making it suitable for any data rate, including high-speed scenarios such as 32kbps and 64kbps.

[0029] Specifically, the I-channel and Q-channel multiplication is used to eliminate modulated data symbols in the signal, resulting in a single-tone signal that is independent of data symbols and pseudo-code signals and depends only on the carrier frequency offset. : .

[0030] Additionally, it should be noted that the complex signal... The product of the I-path and Q-path is obtained It is a real signal. Existing spread spectrum signal acquisition techniques typically target complex signals. The Doppler frequency offset of the input signal is obtained by performing an FFT operation. Under the assumption that "the data symbols remain unchanged during the sampling time interval," the complex signal... In reality, it is also a single-tone complex signal. The spectrum of a single-tone complex signal always has only one spectral line, and the magnitude and sign of the frequency can be easily determined through the spectrum output by the FFT. However, the spectrum of a single-tone real signal always has two symmetrical spectral lines, so only the absolute value of the single-tone signal frequency can be obtained from the spectrum, but the sign of the frequency cannot be determined, and therefore it is impossible to distinguish the relative strength of the input signal carrier frequency compared to the local carrier frequency. In step one, this application artificially introduces a fixed frequency offset +F when down-converting the input signal. d , will be in ±F d between f d Transformed into 0 to 2F d between f d + F d Based on this premise, We only need to focus on the positive frequency portion of the spectrum, thus solving the above problem.

[0031] In step S5 above, the highest-energy spectral line is found in the spectrum output by the FFT module. If the energy of this spectral line exceeds a threshold, the Doppler frequency offset of the input signal is calculated, and then pseudocode locking and carrier locking are performed. If the energy of this spectral line does not exceed the threshold, the phase of the local pseudocode sequence is adjusted, and the process returns to the down-conversion step to re-execute the acquisition process. This step enables frequency offset detection and threshold determination. An FFT operation is performed on the real signal obtained by multiplying the I and Q channels, and the highest-energy spectral line is searched in the output spectrum. If the energy of this spectral line exceeds a preset threshold, the carrier Doppler frequency offset is calculated based on the spectral line frequency fm. The process then proceeds to the lock determination phase; if the threshold is not exceeded, the pseudocode is determined to be misaligned, and the process returns to readjust the pseudocode phase. Specifically: If the frequency corresponding to the highest energy spectral line is found through FFT calculation... If the spectral line energy exceeds the threshold, the carrier Doppler frequency offset can be calculated using the following formula: Calculated It is used as an auxiliary carrier tracking loop to lock onto the carrier in the input signal.

[0032] Furthermore, the threshold is a fixed proportion to the total energy of the FFT output spectrum for each iteration, and this proportion is set according to the requirements of the false alarm probability and the false alarm probability. In this step, the energy threshold compared to the highest energy spectral line is not a fixed value; it is set to a fixed proportion to the total energy of the FFT output spectrum for each iteration. If this proportion is too high, the energy threshold is too high, which will increase the false alarm probability of the spread spectrum signal. If the proportion is too low, the energy threshold is too low, which will increase the false alarm probability of the spread spectrum signal. Therefore, this proportion needs to be set according to specific circumstances.

[0033] In step S6 above, it is determined whether pseudocode locking and carrier locking are completed: if locked, the acquisition process ends; if not locked, the local pseudocode sequence phase is adjusted and the process returns to the downconversion step.

[0034] Specifically, the carrier tracking loop and pseudocode tracking loop are enabled. After a preset time, it is determined whether the carrier tracking loop and pseudocode tracking loop have successfully locked. If both are locked, the spread spectrum signal acquisition process ends successfully; otherwise, the local pseudocode sequence phase is adjusted, and the process returns to the downconversion operation in step S1) to re-execute the acquisition process until acquisition is completed or the maximum number of attempts is reached. This method improves the reliability of acquisition confirmation by replacing simple correlation peak determination with tracking loop lock determination. The closed-loop feedback mechanism ensures automatic searching even when the pseudocode phase is not aligned until acquisition is completed, realizing a fully automatic and highly reliable spread spectrum signal acquisition process, thereby completing the acquisition closed-loop control. Furthermore, the phase adjustment step of the local pseudocode sequence is one-quarter of a chip. This step size is chosen to maintain a balance between acquisition accuracy and acquisition speed: too large a step size may skip the correct phase, leading to missed acquisition; too small a step size will increase the search time. A one-quarter chip step size can control the number of searches within a reasonable range while ensuring the coarse synchronization accuracy of the pseudocode. By optimizing the pseudocode phase search step size, the search space size can be effectively controlled, the number of iterations required for acquisition can be reduced, and the acquisition efficiency can be improved while ensuring the pseudocode acquisition accuracy.

[0035] This application eliminates the impact of spreading pseudocode symbols and high-speed data symbol flipping on FFT spectrum analysis by multiplying the I and Q paths of the signal. This solves the limitation of traditional methods requiring data symbols to remain unchanged within the sampling interval, enabling spread spectrum communication systems to support rates of 32kbps, 64kbps, and higher. By artificially introducing a fixed frequency offset, it resolves the frequency sign ambiguity problem caused by the symmetry of the real signal spectrum. Furthermore, a low-pass filter effectively suppresses signal energy when the pseudocode is misaligned, avoiding invalid lock-on decisions and significantly shortening the acquisition time. The overall scheme is simple to implement, possesses both high data rate adaptability and fast acquisition capability, and significantly enhances the communication capabilities of spread spectrum communication systems.

[0036] In contrast. Figure 2 A block diagram of a traditional spread spectrum signal acquisition system is given. Figure 3 A block diagram of the spread spectrum signal acquisition system given in this invention is provided, and... Figure 3 The improvements made by this invention to the original spread spectrum signal acquisition system are marked with dashed boxes.

[0037] The following are specific embodiments of this solution. First, the settings of the relevant parameters in the embodiments are explained as follows: Local carrier frequency ; Input signal carrier Doppler ; Artificially introduced fixed frequency offset This means that in this example, the receiver's Doppler tracking range is ±50 kHz; Pseudocode rate ; Data rate ; The passband bandwidth of a low-pass filter ; Sampling rate of digital systems .

[0038] Figure 4 The spectrum of the input signal is shown. The specific operation steps are as follows: Step 1: Down-convert the input signal and introduce a fixed frequency offset. . Figure 5 The spectrum after downconversion and the introduction of a fixed frequency offset is given.

[0039] Step 2: Correlate the local pseudocode sequence with the down-converted signal. Figure 6 The spectrum of the correlated signal after aligning the local pseudocode with the input signal pseudocode is given. Figure 7 The spectrum of the correlated signal when the local pseudocode and the input signal pseudocode are aligned is given. It can be seen that when the local pseudocode and the input signal pseudocode are aligned, the correlated signal is equivalent to the data signal modulated at a frequency of... On the carrier wave, its spectrum is equivalent to the data signal spectrum shifted 52kHz to the right. The main lobe width of the spectrum (the frequency interval between the two zeros on either side of the main lobe) is equal to the data rate. It is twice that of the input signal, i.e., 128kHz. Even when the local pseudocode and the input signal pseudocode are not aligned, the correlated signal still closely resembles the spread spectrum, with a spectrum similar to that of the spread spectrum signal and a very large bandwidth.

[0040] Step 3: Perform low-pass filtering on the correlated signal. Figure 8 The spectrum of the correlated signal after passing through a low-pass filter is given when the local pseudocode is aligned with the input signal pseudocode. Figure 9 The spectrum of the correlated signal after passing through a low-pass filter is given when the local pseudocode and the input signal pseudocode are not aligned.

[0041] Step 4: Multiply the I-channel and Q-channel of the filtered signal, and then input the result into the FFT module to calculate the spectrum. Figure 10 The spectrum of the FFT output is given when the local pseudocode is aligned with the input signal pseudocode. Figure 11 The spectrum of the FFT output is given when the local pseudocode is not aligned with the input signal pseudocode. This is to illustrate the necessity of low-pass filtering in step three. Figure 12The ratio of the highest spectral energy to the total spectral energy of the FFT output spectrum under different pseudocode phase deviations without low-pass filtering is given. Figure 13 The ratio of the highest spectral line energy to the total spectral energy of the FFT output spectrum under different pseudocode phase deviations with low-pass filtering is given. (Comparison) Figure 12 and Figure 13 It can be observed that without a low-pass filter, the multiplication of the I and Q paths of the signal eliminates the polarity of the pseudocode and data in the signal. This results in the ratio of the highest spectral energy to the total spectral energy of the FFT output spectrum under different pseudocode phase delays being very close. This means that the FFT calculation stage loses its function of verifying whether the pseudocode has achieved coarse synchronization. If the above steps are performed every time the local pseudocode phase is moved, and then the carrier tracking loop and pseudocode tracking loop are locked, the entire spread spectrum signal acquisition process will take a very long time. However, after introducing a low-pass filter, the FFT calculation stage can not only calculate the Doppler frequency offset of the input signal, but also verify whether the pseudocode has achieved coarse synchronization. This is because only when the pseudocode achieves coarse synchronization can the energy of the highest spectral line in the FFT output spectrum exceed the threshold, allowing the process to proceed to step seven.

[0042] Step 5: Locate the highest-energy spectral line in the FFT module's output spectrum. If the energy of this spectral line exceeds a threshold, calculate the Doppler of the input signal and simultaneously enable the carrier tracking loop and pseudocode tracking loop; if the energy of this spectral line does not exceed the threshold, adjust the phase of the local pseudocode sequence and return to the down-conversion step to re-execute the acquisition process. Figure 10 As shown, within the positive frequency range, the frequency corresponding to the highest energy spectral line is... Based on this, the carrier frequency of the input signal can be calculated. There is an error between the actual carrier Doppler at 2 kHz, which is related to the frequency resolution of the FFT input spectrum. However, this error is much smaller than the acquisition and tracking bandwidth of the carrier tracking loop, so it will not adversely affect carrier tracking. Figure 3 As shown, This will be used to assist carrier tracking. Next, proceed to step six, waiting for the carrier tracking loop and pseudocode tracking loop to lock. If the highest energy spectral line in the FFT module output spectrum does not exceed the threshold, proceed to step one and re-execute the capture process.

[0043] Step Six: After calculating the input signal carrier Doppler, the carrier tracking loop and pseudo-code tracking loop begin operation. In this step, a period of time is waited before determining whether the carrier tracking loop and pseudo-code tracking loop have successfully locked. If locked, successful acquisition of the spread spectrum signal is achieved; otherwise, the local pseudo-code phase is adjusted, and then the process is repeated in Step One. The adjustment step for the local pseudo-code phase is one-quarter of a chip, or 0.25 chips.

[0044] The foregoing has shown and described the basic principles, main features, and advantages of the present invention. Those skilled in the art should understand that the above embodiments do not limit the present invention in any way, and all technical solutions obtained by equivalent substitution or equivalent transformation fall within the protection scope of the present invention.

Claims

1. A method for acquiring spread spectrum signals to eliminate the influence of high-speed data symbols, characterized in that, Includes the following steps: When down-converting the receiver input signal obtained by the ADC sampling, a fixed frequency offset is artificially introduced; Correlate the local pseudocode sequence with the down-converted signal; Perform low-pass filtering on the correlated signal; The I-path and Q-path of the filtered signal are multiplied, and the result of the multiplication is input into the FFT operation module; Find the highest energy spectral line in the spectrum output by the FFT module: if the energy of the spectral line exceeds the threshold, calculate the Doppler frequency offset of the input signal, and then proceed to pseudocode locking and carrier locking; if the energy of the spectral line does not exceed the threshold, adjust the phase of the local pseudocode sequence and return to the downconversion step. Determine whether pseudocode locking and carrier locking are completed: if locked, the acquisition process ends; if not locked, adjust the phase of the local pseudocode sequence and return to the downconversion step.

2. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 1, characterized in that, The receiver's desired Doppler tracking range is -F. d ≤ f d ≤ +F d The artificially introduced fixed frequency offset is +F. d .

3. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 2, characterized in that, The expression for the input signal is: ， in, The value is ±1, which represents the bipolar digital signal mapped to ±1 after the pseudocode and data are XORed. Indicates the frequency of the input signal; The expression for the local oscillator signal of the down-conversion is: ， in F represents the frequency of the local oscillator signal. d This indicates the magnitude of a fixed frequency offset introduced artificially. The signal expression after downconversion is: ， in, Let the input signal represent the true carrier Doppler. After low-pass filtering the correlated signal, then... ， in, This is the signal frequency offset.

4. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 3, characterized in that, If the local pseudocode and If the pseudocode phase is aligned or nearly aligned, then... After near-despreading, the spectral width of the correlated signal depends primarily on the data rate. If the local pseudocode and If the pseudocode phase is not aligned, the correlated signal will still be in a spread spectrum state, and its spectral width mainly depends on the pseudocode rate. The signal expression after correlation is: ， in, This represents the local pseudocode signal represented by bipolar code.

5. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 1, characterized in that, The low-pass filtering of the correlated signal is used to limit the signal energy input to the FFT module when the local pseudocode sequence is not aligned with the pseudocode sequence in the input signal. The bandwidth of the low-pass filter used is 2 to 10 times the data rate.

6. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 1, characterized in that, The multiplication of the I and Q paths is used to eliminate modulated data symbols in the signal, resulting in a single-tone signal that is independent of data symbols and pseudocode signals and is only related to the carrier frequency offset. : ， The frequency of the single-tone signal is equal to twice the carrier frequency offset of the down-converted signal.

7. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 6, characterized in that, If the frequency corresponding to the highest energy spectral line is found through FFT calculation... If the spectral line energy exceeds the threshold, the carrier Doppler frequency offset can be calculated using the following formula: , Calculated It is used as an auxiliary carrier tracking loop to lock onto the carrier in the input signal.

8. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 1, characterized in that, The threshold is a fixed proportion to the total energy of the output spectrum of each FFT, and the proportion is set according to the requirements of the false alarm probability and the missed alarm probability.

9. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 1, characterized in that, The phase adjustment step of the local pseudocode sequence is one-quarter of a chip.

10. The spread spectrum signal acquisition method for eliminating the influence of high-speed data symbols according to claim 1, characterized in that, The pseudocode locking and carrier locking employ a carrier tracking loop and a pseudocode tracking loop. After waiting for a preset time, it is determined whether the carrier tracking loop and the pseudocode tracking loop have been successfully locked. If both are locked, the spread spectrum signal acquisition process ends successfully; otherwise, the phase of the local pseudocode sequence is adjusted, and the process returns to the downconversion step.

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