A design method for a broadband power amplifier covering the VLF-VHF frequency band

Abstract

The present invention discloses a method for designing a broadband power amplifier covering the VLF-VHF frequency band, which mainly solves the problem of narrow frequency coverage of existing RF solid-state broadband power amplifier design methods. The present invention combines the traditional real frequency method with the transmission line transformer matching circuit design method, proposes and applies a transmission line circuit model that takes both low frequency and high frequency into consideration, making it possible to simulate and design a power amplifier covering the VLF-VHF frequency band. At the same time, impedance transformation and port processing are used in the design process to transform balanced circuits and unbalanced circuits, solving the problem that the traditional real frequency method only considers unbalanced circuits, so that this method can be used for the design of broadband power amplifiers for balanced and unbalanced circuits at the same time. Therefore, it is suitable for promotion and application.

Description

Technical Field

[0001] The invention relates to the technical field of amplifiers, and in particular to a design method for a broadband power amplifier covering a VLF-VHF frequency band. Background Art

[0002] RF power amplifiers provide RF energy for communication and test systems. Systems such as long-range early warning, underwater acoustic communication and interference, and RF testing require high-power signals with frequencies starting from a few kHz, ranging from tens of watts to several kilowatts. How to obtain high-power output covering a few kHz to hundreds of MHz has attracted attention in the industry, and broadband power amplifiers based on solid-state devices are an important member of this group. Therefore, there is a significant demand for broadband power amplifiers that can support a few kHz to hundreds of MHz at the same time. Conventional RF solid-state broadband power amplifiers can cover MHz to GHz and GHz to tens of GHz in a single segment, but it is difficult to go down to kHz; while audio solid-state amplifiers can cover tens of Hz to tens of kHz, and it is difficult to go up to MHz.

[0003] The great challenge of broadband power amplifier design is bandwidth expansion. Its essence is broadband impedance matching: matching the impedance of the power device with the system impedance within the entire working bandwidth. The system impedance is usually 50Ω, and the impedance of the power device is often a complex number R+jX, which is much smaller than the system impedance and changes with frequency. The larger the bandwidth to be matched, the worse the optimal performance that can be obtained within the band. The matching degree is described by the Bode-Fanno method as follows:

[0004]

[0005] Where Γ(ω) is the reflection coefficient of the matching network, ω is the angular frequency, and τ is the time constant of the load network. This shows that under the same load, the wider the bandwidth, the worse the reflection coefficient.

[0006] Around the challenge of broadband matching, theoretical methods such as numerical fitting method, prototype topology method, and real frequency method have been developed, among which the real frequency method has been widely used. It directly uses the measured power tube impedance or S parameters, quantifies the effect of impedance conversion through the transmission gain function TPG, obtains the matching network polynomial, and then performs microwave circuit synthesis to obtain the actual matching circuit. Since the real frequency method does not have any restrictions when calculating the polynomial expression of the matching circuit, it can ensure that the obtained circuit is globally optimal. However, the real frequency method is usually based on the electrical length and frequency to fit the target impedance, so when the frequency gradually decreases, the circuit will become too large due to the physical size and is no longer realistic. The design of low-frequency audio solid-state power amplifiers is often based on the transformer principle, but the parasitic parameters between the turns of the transformer will produce resonance. When the frequency increases to the MHz level, the transformer theory can no longer guide practice.

[0007] Therefore, there is an urgent need for a method to guide the design of broadband power amplifiers covering a frequency range from a few kHz to hundreds of MHz. Summary of the invention

[0008] The object of the present invention is to provide a method for a VLF-VHF broadband power amplifier, mainly to solve the problem of narrow frequency coverage range in existing radio frequency solid-state broadband power amplifier design methods.

[0009] To achieve the above purpose, the technical solution adopted by the present invention is as follows:

[0010] A design method for a broadband power amplifier covering a VLF-VHF frequency band, the amplifier comprising a power tube group, and an input matching circuit and an output matching circuit connected to both ends of the power tube group; the input matching circuit and the output matching circuit have the same circuit topology structure and the circuit forms of the two are arranged symmetrically about the power tube group; the input matching circuit is composed of a circuit based on a coaxial transmission line transformer and a circuit based on a capacitor-loaded microstrip line; the design method comprises the following steps:

[0011] (S1) obtaining the S parameters of the power tube;

[0012] (S2) determining the target impedance, gain, and power of the amplifier;

[0013] (S3) determining the coaxial transmission line transformer and its transformation ratio according to the actual bandwidth and impedance transformation span;

[0014] (S4) verifying the low-frequency and high-frequency performance of the coaxial transmission line transformer;

[0015] (S5) determining whether the low-frequency and high-frequency performances of the coaxial transmission line transformer meet the matching requirements of the amplifier, if not, returning to step (S3), if so, proceeding to the next step;

[0016] (S6) confirming the circuit impedance of the capacitively loaded microstrip line at the rear end of the coaxial transmission line transformer;

[0017] (S7) applying a real frequency method to obtain a polynomial expression for a matching network of a circuit based on a capacitor-loaded microstrip line;

[0018] (S8) taking into account the order and electrical length coefficient of the matching network, using a network synthesis method to obtain lumped parameter matching component values, and then applying Norton transformation to convert the lumped parameter components into distributed parameter components;

[0019] (S9) Inspecting the physical feasibility of the obtained lumped parameter and distributed parameter circuits, if it is feasible, proceeding to the next step, if it is not feasible, returning to step (S8);

[0020] (S10) The amplifier is brought into the power amplifier model for simulation verification to determine whether the power efficiency index meets the design requirements. If it does, the design is completed. If not, return to step (S8).

[0021] Further, in the step (S4), the low-frequency and high-frequency performance verification is performed to verify the equivalent impedance Z generated by the core characteristics of the coaxial transmission line transformer. sc , the calculation formula is:

[0022]

[0023] Where ω is the angular frequency, which is related to the operating frequency, L0 is the inductance of the coaxial line itself, μ is the real part of the equivalent complex permeability, and μ″ is the imaginary part of the equivalent complex permeability.

[0024] Furthermore, in the step (S6), the designed target impedance polynomial is:

[0025] F B (jω)=R B (ω)+jX B (ω); (1)

[0026] Where R B (ω) is composed of n line segments, representing the real part of impedance that varies with frequency, X B (ω) is the imaginary part of impedance.

[0027] Further, in step (S2), the target gain of the designed amplifier satisfies:

[0028]

[0029] From this we get R B (ω) expression, and then obtain X through Hilbert transform relationship B (ω), and then the matching network F of the target impedance is obtained by formula (1) B The complete expression of (ω).

[0030] Compared with the prior art, the present invention has the following beneficial effects:

[0031] The present invention combines the traditional real frequency method with the transmission line transformer matching circuit design method, proposes and applies a transmission line circuit model that takes both low and high frequencies into account, making it possible to simulate and design a power amplifier covering the VLF to VHF frequency band. At the same time, impedance transformation and port processing are used in the design process to transform balanced circuits and unbalanced circuits, solving the problem that the traditional real frequency method only considers unbalanced circuits, so that this method can be used for the design of broadband power amplifiers for balanced and unbalanced circuits at the same time. BRIEF DESCRIPTION OF THE DRAWINGS

[0032] Figure 1 Schematic diagram of the process designed for the present invention.

[0033] Figure 2 A schematic diagram of the structure of the amplifier designed for the present invention.

[0034] Figure 3 It is the equivalent circuit diagram of the coaxial transmission line in the design of the present invention.

[0035] Figure 4 It is a schematic diagram of converting a balanced circuit into an unbalanced circuit in the design of the present invention.

[0036] Figure 5 Schematic diagram of the simulation results of the amplifier designed for the present invention. DETAILED DESCRIPTION

[0037] The present invention is further described below in conjunction with the accompanying drawings and embodiments. The embodiments of the present invention include but are not limited to the following embodiments.

[0038] Example

[0039] The invention discloses a design method for a broadband power amplifier covering a VLF-VHF frequency band. The amplifier comprises a power tube group, and an input matching circuit and an output matching circuit connected to two ends of the power tube group; the circuit topology structures of the input matching circuit and the output matching circuit are the same and the circuit forms of the two are arranged symmetrically about the power tube group; the input matching circuit is composed of a circuit based on a coaxial transmission line transformer and a circuit based on a capacitor-loaded microstrip line.

[0040] The broadband power amplifier of the present invention has a bandwidth coverage range of kHz to 100 MHz. The left side is the input matching circuit, the middle is the power tube, and the right side is the output matching circuit. The input matching circuit and the output matching circuit are similar in form, but the input matching focuses on gain matching, while the output matching focuses on power and efficiency matching, with different emphases. Specifically, in the design, it is reflected in the different target impedances selected. And according to different matching bandwidths, there are different orders, Figure 1 The example is a fourth-order transformer. If the actual order is higher, there will be more components. The coaxial transmission line transformer needs to consider its high-frequency and low-frequency performance separately, and the microstrip line part should be designed using the real frequency method.

[0041] like Figure 1The specific process of the design method of the present invention is shown. First, the S parameters of the power tube are obtained by using the manufacturer's device data, simulation model, and Load-Pull test on the actual object. Determine the target impedance, gain, and power of the power amplifier. The output impedance of the power tube varies greatly throughout the frequency band. The gain and power values ​​required by the design need to be as high and flat as possible. Therefore, it is necessary to try and select multiple frequency points within the band to finally determine a suitable series of target impedances. According to the actual bandwidth and impedance transformation span, select a suitable transmission line transformation ratio. For example, generally speaking, 9:1 and 4:1 transmission line transformers are simple and commonly used transformers. The 9:1 transformer has a greater impedance transformation capability, while the 4:1 transformer has a wider transformation bandwidth.

[0042] Coaxial transmission line transformers face leakage current at low frequencies and parasitic capacitance, joint discontinuity and other effects at high frequencies. When the operating frequency is reduced to the point where the equivalent electrical length of the coaxial line is comparable to its physical length, the current in the coaxial line shell is directly shunted to the ground by the skin; as the frequency gradually increases, the impedance generated by the distributed capacitance between the coaxial line shell and the ground gradually decreases, shunting the signal of the outer conductor to the ground. Therefore, special attention should be paid to verifying the availability at high and low frequencies.

[0043] In actual design, the coaxial line model in the transmission line is as follows Figure 3 As shown in the figure, L p is the equivalent parallel inductance. C p is the equivalent parallel capacitance, Z sc It is the equivalent impedance generated by the core characteristics. This equivalent impedance needs to take into account the complex permeability, so its value will change with frequency, and can be calculated by the following formula:

[0044]

[0045] Where ω is the angular frequency, which is related to the operating frequency, L0 is the inductance of the coaxial line itself, μ is the real part of the equivalent complex permeability, and μ″ is the imaginary part of the equivalent complex permeability.

[0046] When applying this model to the design and simulation of transmission line transformers, special attention should be paid to the performance verification of high and low frequencies. And judge whether the performance of high and low frequencies meets the matching requirements. If it does, proceed to the next step; if not, return to adjust the transmission line circuit. The length, thickness, and winding method of the transmission line will affect its equivalent inductance and parasitic capacitance. Therefore, when the design does not meet the requirements, it can be adjusted and verified from the above multiple angles.

[0047] Then the rear-end impedance of the coaxial transmission line is calculated. The impedance matching circuit at the rear end of the transmission line is often a balanced circuit, which needs to be converted into an unbalanced circuit, such as Figure 4 The impedance transformation formula is:

[0048]

[0049] In the specific transformation, in order to facilitate design and calculation, the impedance and electrical length of the microstrip line are usually kept unchanged, and only the capacitance and target impedance value are converted.

[0050] The real frequency method is applied to obtain the polynomial expression of the matching network. Assuming that the target transmission power gain TPG is determined to be T0(ω), the design goal of the real frequency method is to make the gain T(ω) of the design network as close as possible to the target gain T0(ω). Generally, the matching network polynomial is obtained by least squares line segment approximation.

[0051] For the target impedance, the frequency-dependent polynomial is:

[0052] F L (jω)=R L (ω)+jX L (ω)

[0053] The matching network polynomial that needs to be designed can be written as follows:

[0054] F B (jω)=R B (ω)+jX B (ω)

[0055] Since the multinomial expression F of the matching network is now B (ω) has no definite form, which ensures that the optimized network is not restricted by the topological structure, so this matching network is globally optimal. B (ω) is composed of n line segments, representing the real part of impedance that varies with frequency, X B (ω) is the imaginary part of impedance. Take n points in the entire frequency band, corresponding to ω n , are the frequency cutoff points, and the real frequency method will perform calculations at these points.

[0056] Based on the above polynomial, the transmission power gain TPG of the circuit can be expressed as:

[0057]

[0058] The design goal is to make T(ω) as close to T0(ω) as possible. When the following equation is satisfied:

[0059]

[0060] It is considered that the design requirements are met and R B (ω) expression, and then obtain X through Hilbert transform relationship B (ω). So far we have obtained the matching network FB (ω) is a complete expression that is globally optimal.

[0061] Taking into account the order and electrical length coefficient of the matching network, the network synthesis method is used to obtain the lumped parameter matching component values, and then the Norton transform is applied to convert the lumped parameter components into distributed parameter components. In the design process, the larger the order of the selected matching network, the wider its bandwidth and the smaller the fluctuation within the frequency band, but the greater the complexity of the matching network, the greater the insertion loss caused by the matching components; the electrical length coefficient will affect the fluctuation amplitude and physical size of the frequency response, and unreasonable physical size will lead to difficulties in selecting boards, processing or installation. Usually, an initial value is determined first according to the actual design limitations, and then gradually adjusted after subsequent calculation verification. At the same time, if the final circuit is a balanced circuit, the synthesized unbalanced circuit must be converted into a balanced circuit.

[0062] Inspect the physical feasibility of the lumped parameter and distributed parameter circuits of the power tube obtained, mainly involving whether the electrical parameters and physical dimensions are too large or too small. If it can be achieved, proceed to the next step. If it cannot be achieved, the network matching needs to be re-performed.

[0063] Put the amplifier into the power amplifier model simulation verification. Add additional circuits such as feeding, biasing, and stability network to the final matching circuit, simulate the final complete circuit, and analyze it from multiple aspects such as output power, gain, in-band fluctuation, efficiency, and stability. Determine whether indicators such as power efficiency meet the requirements. If they meet, complete the design. If not, re-match the network.

[0064] like Figure 2 As shown, a VLF-VHF broadband power amplifier designed by the method, wherein the input matching circuit includes a coaxial transmission line transformer T1, an N-order microstrip line circuit based on capacitor loading connected to two ports on one side of the output end of the coaxial transmission line transformer T1, and coaxial transmission line transformers T2 and T3 connected to the output end of the microstrip line circuit; wherein N is an even number; an input port of the coaxial transmission line transformer T2 is connected to an input port of the coaxial transmission line transformer T3; the output ports of the coaxial transmission line transformers T2 and T3 are connected to a power tube group, and one input end of the coaxial transmission line transformer T1 is used as a radio frequency input, and the other input end is grounded.

[0065] The present embodiment is a 4th-order microstrip line circuit, which includes microstrip lines TL1 and TL2 connected to two ports of the output end of the coaxial transmission line transformer T1, a loading capacitor C1 connected between the other ends of the microstrip lines TL1 and TL2, a microstrip line TL3 having one end connected to the common end of the microstrip line TL1 and the loading capacitor C1, a microstrip line TL4 having one end connected to the common end of the microstrip line TL2 and the loading capacitor C1, and a loading capacitor C2 connected between the other ends of the microstrip lines TL3 and TL4; wherein an input port of the coaxial transmission line transformer T2 is connected to the common end of the microstrip line TL3 and the loading capacitor C2, and an input port of the coaxial transmission line transformer T3 is connected to the common end of the microstrip line TL4 and the loading capacitor C2.

[0066] In this embodiment, the power tube group is composed of two drain-connected and grounded power tubes M1 and M2; one output port of the coaxial transmission line transformers T2 and T3 is connected to the gate of the power tube M1, and the other output port of the coaxial transmission line transformers T2 and T3 is connected to the gate of the power tube M2; correspondingly, one output port of the two coaxial transmission line transformers corresponding to the coaxial transmission line transformers T2 and T3 of the output matching circuit is connected to the source of the power tube M1, and the other output port is connected to the source of the power tube M2. The gain simulation results of the 9k~400MHz power amplifier designed by this method are shown in the figure. Figure 5 In the range of 9kHz to 400MHz, the flatness is within 3dB, which meets the requirements of practical applications.

[0067] The above embodiment is only one of the preferred implementation modes of the present invention and should not be used to limit the protection scope of the present invention. Any changes or modifications that are made to the main design concept and spirit of the present invention and have no substantive significance, and the technical problems they solve are still consistent with the present invention, should be included in the protection scope of the present invention.

Claims

1. A method for designing a broadband power amplifier covering the VLF-VHF frequency band, wherein the amplifier comprises a power tube group, and an input matching circuit and an output matching circuit connected to both ends of the power tube group; the circuit topology structures of the input matching circuit and the output matching circuit are the same and the circuit forms of the two are arranged symmetrically about the power tube group; the input matching circuit is composed of a circuit based on a coaxial transmission line transformer and a circuit based on a capacitor-loaded microstrip line, characterized in that: The design method comprises the following steps: (S1) obtaining the S parameters of the power tube; (S2) Determine the target impedance, gain, and power of the amplifier; wherein the target gain of the designed amplifier satisfies: From this we get R B (ω) expression, and then obtain X through Hilbert transform relationship B (ω); (S3) determining the coaxial transmission line transformer and its transformation ratio according to the actual bandwidth and impedance transformation span; (S4) verifying the low-frequency and high-frequency performance of the coaxial transmission line transformer; wherein the low-frequency and high-frequency performance verify the equivalent impedance Z generated by the magnetic core characteristics of the coaxial transmission line transformer sc , the calculation formula is: Where ω is the angular frequency, which is related to the operating frequency, L0 is the inductance of the coaxial line itself, μ is the real part of the equivalent complex permeability, and μ′ is the imaginary part of the equivalent complex permeability. (S5) determining whether the low-frequency and high-frequency performances of the coaxial transmission line transformer meet the matching requirements of the amplifier, if not, returning to step (S3), if so, proceeding to the next step; (S6) confirming the circuit impedance of the capacitively loaded microstrip line at the rear end of the coaxial transmission line transformer; (S7) Apply the real frequency method to obtain the polynomial expression of the matching network of the circuit based on the capacitor loaded microstrip line: F B (jω)=R B (ω)+jX B (ω); (1) Where R B (ω) is composed of n line segments, representing the real part of impedance that varies with frequency, X B (ω) is the imaginary part of impedance; combined with formula (2), the matching network F of the target impedance is obtained B The complete expression of (ω); (S8) taking into account the order and electrical length coefficient of the matching network, using a network synthesis method to obtain lumped parameter matching component values, and then applying Norton transformation to convert the lumped parameter components into distributed parameter components; (S9) Inspecting the physical feasibility of the obtained lumped parameter and distributed parameter circuits, if it is feasible, proceeding to the next step, if it is not feasible, returning to step (S8); (S10) The amplifier is brought into the power amplifier model for simulation verification to determine whether the power efficiency index meets the design requirements. If it does, the design is completed. If not, return to step (S8).

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