Isolated DC / DC Converter with Wide Output Voltage Range and Its Control Method

Abstract

This case discloses an isolated multi-stage DC / DC resonant converter. Compared with the output voltage range and the device switching frequency range of the prior art, this case has a wide range of output voltages under the condition of a narrow device switching frequency range. The control circuit selects one of three modulation methods to operate the primary-side switching device of the resonant converter at any given time according to at least one of the output voltage, the output current, the input signal, and one or more external control signals. The three modulation methods and their selected device switching frequencies generate different voltage waveforms to the primary-side transformer coupled to the secondary-side transformer, thereby providing different output voltages.

Description

Technical Field

[0001] This case relates to a power converter, particularly an isolated DC / DC converter applicable to a wide output voltage range and its control method. More specifically, this case relates to controlling a resonant DC / DC converter using different modulation methods to adjust the output voltage within an extremely wide voltage range. Background Art

[0002] Many power conversion applications (such as battery charging for electric vehicles (EVs)) require adjusting the output voltage within a wide voltage range. For example, a typical EV battery charging circuit has two conversion stages, namely a front-end AC / DC converter and an isolated DC / DC converter directly connected to the battery. The front-end AC / DC converter can provide a fixed DC bus voltage or a variable DC bus voltage. The DC / DC converter needs to provide an adjustable voltage under various load current conditions and battery states and within a wide voltage range. For example, a typical battery for a conventional EV has a voltage range of 240 volts to 460 volts. However, the output battery voltage range for some high-end electric vehicles, multi-purpose electric vehicles, and electric buses or articulated vehicles is between 500 volts and 950 volts. Therefore, a DC / DC converter that can provide an adjustable output voltage with an extremely wide output voltage range is needed to meet the charging requirements under different battery voltage levels.

[0003] Due to the high efficiency of the LLC resonant converter topology and the simplified structure achieved through magnetic component integration, soft switching can be realized on both the primary side switch and the secondary side switch, and it is applicable to a wide voltage range. Therefore, the LLC resonant converter topology is widely used in isolated DC / DC converters.

[0004] Figure 1A Fig. is the circuit structure diagram of a traditional full-bridge LLC resonant converter under closed-loop voltage control. Figure 1B is Figure 1A The switching control signal of the resonant converter shown and the timing diagram of the primary side full-bridge output voltage V AB . By controlling the switching frequency of the primary side switch, the output voltage V o can be adjusted. When the LLC resonant converter operates at the resonant frequency (f r ), or when the DC voltage gain M is equal to the turns ratio N P / N S , the LLC resonant converter has the highest efficiency, where the resonant frequency (f r ) is determined by the resonant inductor L r and the resonant capacitor C r , and the DC voltage gain M is equal to V o / V in . When the switching frequency (fsw ) When it is greater than the resonant frequency f r , the DC voltage gain M decreases. Conversely, when the switching frequency f sw is less than the resonant frequency f r , the DC voltage gain M increases. However, the conversion efficiency will decrease as the switching frequency f sw deviates from the resonant frequency f r . To achieve a wide output voltage range, the LLC resonant converter operates in a very wide frequency range, so its conversion efficiency will inevitably be reduced. In addition, the maximum and minimum DC voltage gains of the LLC resonant converter are determined by circuit parameters (such as the ratio of the magnetizing inductance L- m to the resonant inductance L r ) and the load conditions. Therefore, the LLC resonant converter cannot achieve a very wide output voltage range under all load conditions.

[0005] For applications with a wide output voltage range, such as battery charging applications, the LLC circuit parameters must be carefully balanced between the conversion efficiency and the output voltage range. In the prior art, many techniques have been developed to achieve a wide output voltage range, such as:

[0006] (a) The literature titled "Design Methodology of LLC Resonant Converters for Electric Vehicle Battery Chargers", authored by J. Deng et al., published in the IEEE Transactions on Vehicular Technology, vol. 63, no. 4, pp. 1581 - 1592, with a publication date of May 2014;

[0007] (b) The literature titled "An LLC Resonant DC - DC Converter for Wide Output Voltage Range Battery Charging Applications", authored by F. Musavi et al., published in the IEEE Transactions on Power Electronics, vol. 28, no. 12, pp. 5437 - 5445, with a publication date of December 2013;

[0008] (c) The literature titled "A Design Procedure for Optimizing the LLC Resonant Converter as a Wide Output Range Voltage Source" by R. Beiranvand et al., published in the IEEE Transactions on Power Electronics, vol. 27, no. 8, pp. 3749 - 3763, on August 2012; and

[0009] (d) The literature titled "Optimal Design Methodology for LLC Resonant Converter in Battery Charging Applications Based on Time-Weighted Average Efficiency" by Z. Fang et al., published in the IEEE Transactions on Power Electronics, vol. 30, no. 10, pp. 5469 - 5483, on October 2015.

[0010] However, the technologies described in the above-mentioned literatures can only optimize the conversion circuit in traditional EV charging applications to achieve a limited output voltage range (e.g., between 200 volts and 500 volts).

[0011] To achieve a wider output voltage range with good conversion efficiency, the prior art has proposed modification methods for the circuit structure and control method of the traditional LLC resonant converter. Taking the literature titled "Multimode Optimization of the Phase-Shifted LLC Series Resonant Converter" by U. Mumtahina and P. J. Wolfs, published in the IEEE Transactions on Power Electronics, vol. 33, no. 12, pp. 10478 - 10489, in December 2018 as an example, it discloses an LLC resonant converter that combines traditional frequency control and phase-shift control methods to achieve a wider output voltage range. Figure 2A FIG. is the circuit structure diagram of the full-bridge LLC resonant converter under closed-loop voltage control in this literature, Figure 2B is Figure 2A the switching control signal of the resonant converter and the primary-side full-bridge output voltage VAB A timing diagram under phase shift control. This document describes that a phase shift can be provided between two pairs of gate signals of two primary side switch legs. By using phase shift control and switching frequency control simultaneously, the LLC resonant converter can operate at a lower switching frequency in step-down operation.

[0012] U.S. Patent US9,263,960B2, authored by M. Jovanovic and B. Irving, with the invention title "Power Converters for Wide Input or Output Voltage Range and Control Methods Thereof", was announced on February 16, 2016. This patent discloses a topology transformation control method for a full-bridge LLC circuit operating in a full-bridge or half-bridge topology. Figure 3A It is the circuit structure diagram of a full-bridge LLC resonant converter under topology transformation control in this invention. Figure 3B For Figure 3A The timing diagram of the switching control signal of the shown resonant converter in the topology transformation transition interval from full-bridge to half-bridge. In this patent, the circuit topology adapts to the control signal, and the control signal responds reversely to the input or output operating conditions.

[0013] U.S. Patent Application US2015 / 0229225A1, authored by Y. Jang and M. Jovanovic, with the invention title "Resonant Converter and Control Methods Thereof", and the application date is August 13, 2015. This patent discloses a control method for a series resonant converter (SRC), which combines a variable frequency control method and a delay time control method. Figure 4 It is the circuit structure diagram of the series resonant converter in this invention. In this invention, the variable frequency control method is applied to the primary side switch, and the delay time control method is applied to the secondary side switch. Accordingly, this patent can boost the output voltage, thereby achieving a wider output voltage range under the control of a narrower switching frequency range.

[0014] To enable a traditional LLC resonant converter to achieve a wider output voltage range, many variations of topology structures and control methods have also been proposed in the prior art. However, these variations all have their drawbacks, such as being difficult to implement specifically, an increase in the number of components, or unwanted dynamic changes. Additionally, these variations still cannot achieve a sufficiently wide output voltage range, such as an EV fast charger that requires an output voltage range of 200 volts to 1000 volts.

[0015] In the process of achieving a wide output voltage range, one of the difficulties is that the input voltage of the LLC resonant converter must be high enough to avoid generating an undesired high DC gain. Since each switching device on the primary side needs to be able to block the entire input voltage (such as the LLC resonant converter shown in Figure 1A ), high-voltage components need to be used to cope with the higher input voltage, which increases the cost.

[0016] To achieve a wider output voltage range, the input voltage of the LLC resonant converter must be high enough to avoid generating an undesired high DC gain. Therefore, a three-level topology has been disclosed. Each switching device in this three-level topology only blocks half of the input voltage, so it is more attractive than the Figure 1A full-bridge topology in. For example, the three-level topology disclosed in the literature titled "The three-level ZVS PWM converter - a new concept in high voltage DC-to-DC conversion", whose authors are J.R. Pinheiro and I. Barbi, was published in the Proceedings of the 1992 International Conference on Industrial Electronics, Control, Instrumentation, and Automation, San Diego, CA, USA, 1992, pp. 173 - 178 vol.1. The three-level topology of the LLC converter is disclosed in the literature titled "Three-level LLC series resonant DC / DC converter", whose authors are Y. Gu, et al., and was published in IEEE Transactions on Power Electronics, vol. 20, no. 4, pp. 781 - 789, in July 2005. The LLC converter in this literature can achieve zero-voltage switching (ZVS) of the switches without an additional auxiliary circuit.

[0017] The three-level serial half bridge (SHB) topology (also known as the stacked buck topology) has been disclosed in the document titled "DC-DC converter: four switches Vpk = Vin / 2, capacitive turn-off snubbing, ZV turn-on", authored by I. Barbi, et al., published in IEEE Transactions on Power Electronics, vol. 19, no. 4, pp. 918-927, in July 2004. In this document, two clamping diodes are removed from the traditional three-level topology in the previously mentioned document.

[0018] A two-level modulation method for half the input voltage of a DC / DC converter has been disclosed in the document titled "Asymmetrical Duty Cycle-Controlled LLC Resonant Converter With Equivalent Switching Frequency Doubler", authored by S. Zong, et al., published in IEEE Transactions on Power Electronics, vol. 31, no. 7, pp. 4963-4973, in July 2016. This document halves the switching frequency of the primary side switches to reduce drive losses.

[0019] For applications with a wide input voltage range, the two-stage modulation method for the full input voltage and the two-stage modulation method for the half input voltage can be applied to the SHB LLC converter in the following documents: (a) the document titled "Wide input voltage range compensation in DC / DC resonant architectures for on-board traction power supplies", authored by A. Coccia, et al., published in the 2007 European Conference on Power Electronics and Applications in 2007 AD; (b) the document titled "Variable Frequency Multiplier Technique for High-Efficiency Conversion Over a Wide Operating Range", authored by W. Inam, et al., published in IEEE Journal of Emerging and Selected Topics in Power Electronics, vol. 4, no. 2, pp. 335-343 in June 2016 AD; and (c) the document titled "Research on Dual-Operation Mode of 3-level LLC resonant converter", authored by A. Z. Li, et al., published in the 2015 9th International Conference on Power Electronics and ECCE Asia (ICPE-ECCE Asia) in 2015 AD.

[0020] To achieve a wider output voltage range, at the maximum operating frequency, the three-stage modulation method reduces the voltage gain. Figure 5A and Figure 5BThey are respectively the circuit structure diagram of the SHB LLC converter and the timing diagram of the gate signal of the SHB LLC converter, which shows that the three-level modulation method is also an effective method for regulating the output voltage. However, in the three-level modulation method, the current flowing through the primary-side power switch is not balanced, which may lead to thermal imbalance on the device, shorten the life of the device, and even cause device damage. In order to balance the current in the power switch, the literature titled "Periodically Swapping Modulation (PSM) Strategy for Three-Level (TL) DC / DC Converters With Balanced Switch Currents" discloses a periodically swapping modulation method, whose authors are D. Liu, et al., published in IEEE Transactions on Industrial Electronics, vol. 65, no. 1, pp. 412-423, and the publication date is January 2018 AD. Figure 6A and Figure 6B They are respectively the circuit structure diagram of the DC / DC converter in this literature and the timing diagram of its gate signal. However, in this literature, the gate signals of each half-bridge are not complementary, which will increase the complexity of the gate drive circuit. In addition, the body diode of the MOSFET will conduct current under the influence of the unoptimized gate signal, resulting in increased conduction loss and reduced system efficiency.

[0021] Therefore, how to develop a power converter that can improve the above-mentioned prior art is an urgent need at present. Summary of the Invention

[0022] The object of this case is to provide an isolated DC / DC converter applicable to a wide output voltage range and its control method to solve the problems encountered in the above-mentioned conventional technologies.

[0023] According to a preferred embodiment of the present case, a resonant DC / DC converter (such as a series half-bridge LLC resonant converter (SHB LLC resonant converter)) can receive an input signal from a fixed or variable voltage source. The resonant DC / DC converter includes: (a) a primary side circuit, which includes: (1) a first pair of series switch devices and a second pair of series switch devices connected in series, each switch device being controlled by a switching control signal, wherein the input signal is provided across the first pair of switch devices and the second pair of switch devices connected in series; (2) an LC resonant circuit electrically connected to a first electrical node of the first pair of switch devices and a second electrical node of the second pair of switch devices; and (3) an isolation transformer having a first winding and a second winding, wherein the first winding is connected to a third electrical node and a fourth electrical node of the LC resonant circuit; (b) a secondary side circuit connected in parallel to the second winding of the isolation transformer, wherein the secondary side circuit includes a filter capacitor that provides an output voltage or an output current to a load; and (c) a control circuit, wherein based on one or more of the output voltage, output current, input signal, and one or more external control signals, the control circuit selects one of a plurality of modulation methods at any time and provides a switching control signal to control the switch devices of the primary side circuit to operate in the selected modulation method.

[0024] Therefore, the high-efficiency method of the present case enables the SHB LLC resonant converter to have a wide range of adjustable output voltages under the condition of a narrow switching frequency range. The present case can use both modulation method control and device switching frequency control. During operation, the modulation method can be selected according to one or more control signals provided by a controller or an external command. The device switching frequency can be determined, for example, under the closed-loop control of the modulated output voltage. According to the selected modulation method and device switching frequency, the controller generates a control signal that drives the primary side switch devices of the SHB LLC resonant converter.

[0025] In another preferred embodiment of the present case, in order to obtain an adjustable voltage within a high output voltage range, the SHB LLC resonant converter operates in a symmetric modulation mode, and its device switching frequency is adjusted under closed-loop control. In order to obtain an adjustable voltage within a low output voltage range, the SHB LLC resonant converter operates in an asymmetric modulation mode and adopts device switching frequency control. In order to obtain a stable voltage within an extremely low output voltage range, under the three-level modulation mode of the embodiment of the present invention, the SHB LLC resonant converter operates at the maximum allowable switching frequency, and its duty cycle is adjusted under closed-loop control. Therefore, the SHB LLC resonant converter operates with higher efficiency within a relatively narrow device switching frequency range, and at the same time has an extremely wide output voltage range under different load conditions. In addition, the method of the present case has less limitation on specific circuit parameters (such as the ratio of L m and L r while achieving the required minimum and maximum DC voltage gains. The method of the present case is applicable to resonant converters with different secondary-side topologies and secondary-side control methods. The method of the present case can operate under conditions of fixed or variable input voltage.

[0026] In another preferred embodiment of the present case, the LLC SHB resonant converter transitions from a two-level asymmetric modulation mode to a three-level modulation mode and vice versa. When the ratio of the input voltage to the output voltage is within a lower range, that is, when the input voltage is low or when the output voltage is high, a three-level topology is used. Alternatively, when the input-output voltage ratio is within a higher range, that is, when the input voltage is high or the output voltage is low, the modulation mode is changed to a two-level asymmetric modulation mode. The transition between the above two modulation modes is achieved through trajectory control and / or pulse width modulation (PWM) control of four switching devices. Specifically, if trajectory control is used, when transitioning between the two-level asymmetric modulation mode and the three-level modulation mode, an additional voltage level of the resonant cavity voltage is inserted between the two modulation modes, and the duration of the additional voltage level is preset by the modulation transition controller. Specifically, if PWM control is used, when transitioning from the three-level modulation mode to the two-level asymmetric modulation mode, the duty cycle of every other pulse of one of the first pair of switching devices is reduced from 50% to 0%. The duty cycle of every other pulse of one of the second pair of switching devices is reduced from 50% to 0%.

[0027] The present invention can be better understood through the following detailed description in conjunction with the drawings. Description of the Drawings

[0028] Figure 1A FIG. is a circuit structure diagram of a conventional full-bridge LLC resonant converter under closed-loop voltage control, Figure 1B is Figure 1A the switching control signal and the primary-side full-bridge output voltage V- of the resonant converterAB Timing diagram of

[0029] Figure 2A is the circuit structure diagram of a full-bridge LLC resonant converter under closed-loop voltage control, Figure 2B is Figure 2A the switch control signal of the resonant converter of AB and the timing diagram of the primary-side full-bridge output voltage V under phase-shift control.

[0030] Figure 3A is the circuit structure diagram of a full-bridge LLC resonant converter under topological transformation control, Figure 3B is Figure 3A the timing diagram of the switch control signal of the resonant converter of

[0031] Figure 4 is the circuit structure diagram of a traditional full-bridge series resonant converter under frequency and delay time control.

[0032] Figure 5A is the circuit structure diagram of a traditional series half-bridge LLC resonant converter, Figure 5B is Figure 5A the switch control signals S1 to S4 of the resonant converter of AB and the timing diagram of the primary-side output voltage V.

[0033] Figure 6A is the circuit structure diagram of a traditional series half-bridge LLC resonant converter, Figure 6B is Figure 6A the switch control signals of the switching devices S1 to S4 of the resonant converter of AB and the timing diagram of the primary-side output voltage V under periodic switching three-level control.

[0034] Figure 7A is the circuit structure diagram of an exemplary series half-bridge SHB LLC resonant converter 700 with a full-wave secondary-side diode rectifier. Figure 7B is Figure 7A the switch control signals S1 to S4 of the resonant converter of AB and the timing diagram of the phase-leg output voltage V under symmetric modulation. Figure 7C is Figure 7A the switch control signals of the switching devices S1 to S4 of the resonant converter of AB and the timing diagram of the phase-leg output voltage V under asymmetric modulation.

[0035] Figure 8 is the circuit structure diagram of an SHB LLC resonant converter 800 according to a preferred embodiment of the present case. The SHB LLC resonant converter 800 includes a control circuit 801, and the control circuit 801 is based on the output voltage Vo (and an optional output current I o ) outputs switching control signals for switching devices S1 to S4.

[0036] Figure 9A is a timing diagram of the switching control signals of switching devices S1 to S4 and the phase leg output voltage V when the control method switches from the symmetric modulation mode to the asymmetric modulation mode. AB of. Figure 9B is a timing diagram of the switching control signals of switching devices S1 to S4 and the phase leg output voltage V when the control method switches from the asymmetric modulation mode to the symmetric modulation mode. AB of.

[0037] Figure 10A is a circuit structure diagram of the series half-bridge LLC resonant converter 1000. Figure 10B is for Figure 10A the switching control signals of the primary side switching devices S1 to S-4 and the primary side output voltage V of the resonant converter under the three-level modulation mode. AB of.

[0038] Figure 11 is for Figure 10A the SHB LLC resonant converter 1000 under the switching period T SW during the key waveform schematic diagram. Figure 11 shows the signal transitions at times t1, t2, t3, t4, t5, t6, t7, t8, t9, t10, t11, t12, t13, t14, t15, and t16.

[0039] Figures 12A to 12P is a circuit structure diagram showing the sixteen operating stages of the SHB LLC resonant converter 1000 under the three-level modulation mode of the embodiment of the present case, which are time periods (t0, t1), (t1, t2), (t2, t3), (t3, t4), (t4, t5), (t5, t6), (t6, t7), (t7, t8), (t8, t9), (t9, t10), (t10, t11), (t11, t12), (t12, t13), (t13, t14), (t14, t15), and (t15, t16).

[0040] Figure 13 is for the primary side switching devices S1 to S4 of the embodiment of the present case under the three-level modulation mode without considering any dead time, and the signal schematic diagrams of the currents I1 to I4 flowing through the primary side switching devices S1 to S4 during the switching period T SW during.

[0041] Figure 14Circuit structure diagram of the SHB LLC resonant converter 1400 according to another preferred embodiment of the present case, which includes a control circuit 1401. The control circuit 1401 outputs a switching control signal to the primary side switching devices S1 to S4 according to the output voltage V o (and optionally the output current I o ).

[0042] Figure 15A Shows the switching control signal of the primary side switching devices S1 to S4 and the phase leg output voltage V AB when the control method is switched from the symmetric modulation mode to the three-level modulation mode.

[0043] Figure 15B Shows the switching control signal of the primary side switching devices S1 to S4 and the phase leg output voltage V AB when the control method is switched from the three-level modulation mode to the symmetric modulation mode.

[0044] Figure 16A Circuit structure diagram of the SHB LLC resonant converter 1600 having a full-bridge synchronous rectifier on the secondary side. Figure 16B Circuit structure diagram of the SHB LLC resonant converter 1650 having a center-tapped transformer TR and synchronous rectifiers S5 and S6 on the center side.

[0045] Figure 17A Circuit structure diagram of the SHB LLC resonant converter 1700 having a variable DC input voltage source 1702. Figure 17B Circuit structure diagram of the SHB LLC resonant converter 1750 having DC input voltages V in1 and V in2 .

[0046] Figure 18A , Figure 18B , Figure 18C and Figure 18D respectively show the example circuit structure diagram of the series half-bridge LLC resonant converter 1800 having a modulation transition controller 1801 in another embodiment of the present case, the timing diagram of the switching devices S1 to S4 during the transition from the three-level modulation mode to the two-level modulation mode, the timing diagram of the switching devices S1 to S4 during the transition from the two-level modulation mode to the three-level modulation mode, and the trajectories of the converter when operating in the two-level modulation mode and the three-level modulation mode respectively before implementing the modulation mode transition.

[0047] Figure 19A , Figure 19B and Figure 19C The figures respectively show the timing diagrams of different embodiments of the switching devices S1 to S4 during the transition from the three-level modulation mode to the two-level modulation mode.

[0048] Figure 20A and Figure 20B and Figure 20C respectively show the timing diagrams of different embodiments of the switching devices S1 to S4 during the transition from the two-level modulation mode to the three-level modulation mode.

[0049] Figure 21A and Figure 21B and Figure 21C show the exemplary circuit structure diagram of the series half-bridge LLC resonant converter 2100 with the modulation mode transition controller 2101 in another embodiment of this case, the timing diagram of the switching devices S1 to S4 during the transition from the three-level modulation mode to the two-level modulation mode, and the timing diagram of the switching devices S1 to S4 during the transition from the two-level modulation mode to the three-level modulation mode.

[0050] Among them, the reference numerals are explained as follows:

[0051] V in 、V in1 、V in2 : Input voltage

[0052] C in 、C in1 、C in2 : Input capacitance

[0053] V o : Output voltage

[0054] I o 、i Lm 、i Lr 、I1, I2, I3, I4: Current

[0055] S1, S2, S3, S4, S5, S6, S7, S8: Switching devices

[0056] D1, D2, D3, D4: Diode rectifiers

[0057] C o : Output capacitance

[0058] N p : Primary side winding

[0059] N s : Secondary side winding

[0060] TR: Transformer

[0061] C r : Resonant capacitance

[0062] L r : Resonant inductance

[0063] L m: Excitation inductance

[0064] A: First phase-leg output terminal

[0065] B: Second phase-leg output terminal

[0066] V AB : Phase-leg output voltage

[0067] T SW : Switching period

[0068] f SW : Device switching frequency

[0069] 700, 800, 1000, 1400, 1600, 1650, 1700, 1750, 1800, 2100: Power converter

[0070] 801, 1401, 1601, 1651, 1701, 1751, 1801, 2101: Control circuit

[0071] 1702: Input voltage source

[0072] V ctrl : Signal

[0073] P, N, O: Input terminals

[0074] σ, θ: Time periods Detailed implementation manners

[0075] Some typical embodiments embodying the features and advantages of this case will be described in detail in the following description. It should be understood that this case can have various variations in different aspects, all of which do not depart from the scope of this case, and the descriptions and illustrations therein are essentially for illustrative purposes rather than for limiting this case.

[0076] To avoid a high cost due to a high input voltage, a multi-stage topology can be used to maintain the same device rated voltage for the primary-side bridge arm. Figure 7A Shows an SHB topology (also known as "stacked buck topology") including an LLC resonant converter 700, and the LLC resonant converter 700 has a series half-bridge on its primary side. As Figure 7A shown, the series half-bridge includes four serially connected switching devices S-1 to S4, and the switching devices S1 to S4 are configured to block half of the input voltage across the input terminals P and N. The input terminals P and N are provided at both ends of the serially connected input capacitors C in1 and C in2 and at both ends of the switching devices S1 and S4. The input capacitors C in1 and C in2The common terminal is simultaneously the common terminal between the switching devices S2 and S3. Thus, the input capacitor C in1 is connected in parallel with the switching devices S-1 and S2, and the input capacitor C in2 is connected in parallel with the switching devices S3 and S4. On the secondary side, a full-wave diode rectifier D1 to D4 is provided. The filter capacitor C o is provided on the secondary side and configured to provide an output voltage V o or an output current I o .

[0077] The first phase-leg output terminal A is located at the common terminal between the switching devices S1 and S2, and the second phase-leg output terminal B is located at the common terminal between the switching devices S3 and S4. A resonant circuit formed by an isolated transformer TR and a series-connected resonant inductor L r , a resonant capacitor C r and a transformer exciting inductor L m is connected between the first phase-leg output terminal A and the second phase-leg output terminal B. The primary side winding of the isolated transformer TR is connected in parallel with the exciting inductor L m . The secondary side winding of the isolated transformer TR can be a center-tapped winding with two rectifying elements or a single winding with a full-bridge rectifier. The switching devices S1 to S4 block the allocated input voltage in one direction, but each switching device can conduct current bidirectionally. The switching devices S1 to S4 can be, for example but not limited to, semiconductor switches such as MOSFETs, IGBTs, BJTs or other semiconductor switches. Alternatively, a synchronous rectifier can replace the diode rectifier circuit in the secondary side to reduce conduction losses.

[0078] Figure 7B and Figure 7C respectively show Figure 7A the "symmetric" and "asymmetric" modulation modes of the SHB LLC resonant converter 700. Each modulation mode can control the switching control signals to operate the switching devices S1 to S4. In any modulation mode, the switching control signals of the switching devices S1 and S2 are complementary, and the switching control signals of the switching devices S3 and S4 are also complementary. In fact, a small dead time can be inserted between each pair of complementary switching control signals to avoid phase-leg shoot-through. Figure 7B is the timing diagram of the switching control signals for the switching devices S1 to S4 and the phase-leg output voltage V AB under symmetric modulation. The switching control signals of the switching devices S1 and S4 have a duty cycle of 50% and are in-phase signals with each other. Similarly, in addition to the switching control signals of the switching devices S1 and S4 being complementary to the switching control signals of the switching devices S2 and S3, the switching control signals for operating the switching devices S2 and S3 also have a duty cycle of 50% and are in-phase signals with each other. Thus, the phase-leg output voltage V AB(i.e., the leg output voltage) switches between V and 0 at a duty cycle of 50% at the device switching frequency. in and 0.

[0079] Figure 7C are the switching control signals for the switching devices S1 to S4 and the phase leg output voltage V AB under asymmetric modulation. As Figure 7C shown, the switching control signals of the switching devices S1 and S4 are phase-shifted 180° from each other and have a duty cycle of 25%, while the switching control signals of the switching devices S2 and S3 are phase-shifted 180° from each other and have a duty cycle of 75%. Through asymmetric modulation, when the switching devices S1 and S3 are off, the voltage across the input capacitor C in1 is used as the phase leg output voltage V AB . When the switching devices S2 and S4 are off, the voltage across the input capacitor C in2 is used as the phase leg output voltage V AB . When the switching devices S2 and S3 are off, the phase leg output voltage V AB is zero volts. Therefore, the phase leg output voltage V AB switches between 0.5V SW and 0 volts at twice the device switching frequency f in at a duty cycle of 50%. In either modulation method, the device switching frequency f SW can be a control variable for achieving different control objectives.

[0080] Figure 7B and Figure 7C shown symmetric and asymmetric modulation methods generate different phase leg output voltages V r -C r -L m across the series-connected resonant circuit, where the phase leg output voltage V AB includes different DC and AC components. During the operation of the LLC resonant converter 700, the resonant capacitor C AB blocks the DC component, such that only the AC component appears across the primary side winding of the isolation transformer TR. When the device switching frequency f r equals the resonant frequency SW , the output voltage V equals the average value of the product of the rectified AC component of the phase leg output voltage V o and the transformer voltage gain. Since the transformer voltage gain turns ratio is AB , the output voltage V can be obtained from O , where N and N S and N P are the number of turns of the secondary side winding and the primary side winding, respectively. is the average value after rectification of the AC component of the phase leg output voltage V AB .

[0081] In the case of adopting the symmetric modulation method, the AC component of the phase leg output voltage V AB is a bipolar square wave with a duty cycle of 50%. This bipolar square wave has an amplitude of SW at the device switching frequency f . Therefore, the DC voltage gain is r at the resonant frequency f . And in the case of adopting the asymmetric modulation method, the AC component of the phase leg output voltage V AB is a bipolar square wave with a duty cycle of 50%. This bipolar square wave has an amplitude of SW at twice the device switching frequency f . Therefore, the DC voltage gain is r at the resonant frequency f . Therefore, the DC voltage gain of the symmetric modulation method at the resonant frequency f r is twice that of the asymmetric modulation method at the resonant frequency f r . And the voltage gain of the LLC resonant converter 700 can be adjusted by the switching frequency control.

[0082] As described above, when the LLC resonant converter 700 operates at an operating point equal to or close to the resonant frequency f r , the maximum efficiency can be achieved. And in order to obtain a wider output voltage range, the traditional control changes the device switching frequency f SW to adjust the DC voltage gain. However, the switching frequency control will cause the operating point to deviate from the maximum efficiency. In addition, for a very wide output voltage range, with given fixed circuit parameters, it may not be possible to achieve the required DC voltage gain even by using the above control method.

[0083] The inventors of this case realized that the SHB LLC resonant converter can combine different modulation methods to efficiently obtain different required DC voltage gains at or close to the resonant frequency (i.e., in a relatively narrow input switching frequency range) to provide a wider output voltage range. In addition, the control method of the embodiments of this case can achieve a wider output voltage range and circuit gain even when the circuit parameter values are fixed. Figure 8 FIG. is the circuit structure diagram of the SHB LLC resonant converter 800 of the preferred embodiment of this case. The SHB LLC resonant converter 800 includes a control circuit 801. The control circuit 801 generates a switching control signal for the switching devices S1 to S4 according to the output voltage V o (and optionally the output current I o ).

[0084] In the SHB LLC resonant converter 800, the control circuit 801 can use the output voltage V o as its primary control objective. The output current I o can also be used as a separate control objective or a feedback signal representative of the load condition. The reference value representative of each control objective can be generated internally by the control circuit 801 or from an external source. Based on the difference between the sensed voltage V o (or the sensed current I o ) and the corresponding reference value, the control circuit 801 switches between multiple modulation schemes to control the switching devices S1 to S4 of the primary side pins in the SHB LLC resonant converter 800. One or more control objectives can be used to determine other control parameter values, such as the device switching frequency f SW and phase shift in the selected modulation scheme. The switching control signals for operating the switching devices S1 to S4 on the primary side are generated based on the selected modulation scheme and other control parameter values.

[0085] In Figure 8 , the turns ratio of the isolation transformer TR (which is only a scaling factor in this case) is set to 1 to simplify the detailed description. As Figure 7B and Figure 7C shown, at a frequency equal to or close to the device switching frequency f SW , the symmetric modulation scheme provides an output voltage V o that is twice the output voltage V o provided by the asymmetric modulation scheme. Specifically, under symmetric modulation, the pin output voltage V r at the resonant frequency f AB is 0.5V in , so the output voltage V- o is also 0.5V in . In some embodiments, the resonant parameters of the isolation transformer TR are such that when the device switching frequency f sw varies from 0.5f r to 1.4fr, the output voltage V o varies from 0.3V in to 0.8V in . Similarly, under asymmetric modulation, the pin output voltage V AB at the resonant frequency fr is 0.25V in , so the output voltage V o is also 0.25V in . In the same embodiment, the resonant parameters of the isolation transformer TR are such that when the device switching frequency f SW varies from 0.25f r to 0.8fr When, the output voltage V o varies from 0.125V in to 0.4V in . It should be noted that the device switching frequency f SW is only half of the frequency of the phase-leg output voltage V AB . Accordingly, an embodiment of the present case provides a control method that combines selectable modulation methods and frequency control to achieve a predetermined output voltage range. (The output voltage adjustment ranges of different modulation methods may overlap each other). The selection of the modulation method in the embodiments disclosed in the present case can be based on control parameters such as, for example, voltage control targets, load conditions, and allowable operating frequency ranges. When one or more conditions change such that the preferred modulation method is different from the current modulation method, the control circuit 801 can switch from the current modulation method to the preferred modulation method.

[0086] In the above embodiment, the SHB LLC resonant converter 800 has an output voltage range V in from 0.125V in to 0.8V o , and the device switching frequency f SW is in the range of 0.25f r to 1.6f r . The ratio of the maximum DC gain to the minimum DC gain is 6.4, and the maximum device switching frequency f SW is only 1.6f r . For low output voltages, there is no need to significantly increase the device switching frequency f SW as in traditional frequency control to increase the loss of the switches. (In fact, when the double-frequency asymmetric modulation method is used for low output voltages, the device switching frequency f SW even decreases accordingly). In the method of the embodiments disclosed in the present case, especially under extremely low output voltages and / or extremely light load conditions, the maximum device switching frequency f SW is significantly reduced. By adjusting the circuit parameter values, a higher ratio of the maximum DC gain to the minimum DC gain can be obtained within the same narrow device switching frequency range, which cannot be achieved in a traditionally controlled SHB LLC resonant converter.

[0087] In some embodiments, Figure 9A is the timing diagram of the switching control signals of the switching devices S1 to S4 and the phase-leg output voltage V AB when the control method switches from the symmetric modulation method to the asymmetric modulation method.

[0088] Figure 9B is the timing diagram of the switching control signals of the switching devices S1 to S4 and the phase-leg output voltage V ABTiming diagram. Figure 9A Shows the transition from the symmetric modulation mode (interval 901) to the asymmetric modulation mode (interval 903) when the output voltage control target changes from a high voltage to a low voltage within the output voltage range. The transition interval 902 between the modulation modes can be implemented in different ways, such as frequency shift, phase shift, duty cycle shift, or any combination thereof. Similarly, Figure 9B Shows the transition from the asymmetric modulation mode (interval 904) to the symmetric modulation mode (interval 906) when the output voltage control target changes from a low voltage to a high voltage within the output voltage range. The transition interval 905 between the modulation modes can be implemented in the same way as in the Figure 9A transition but in the reverse order. For example, control parameters such as frequency shift, phase shift, duty cycle shift, etc., or any combination thereof. Of course, any other suitable transition control method can also be used.

[0089] Three-level modulation

[0090] According to another preferred embodiment of the present case, Figure 10B Shows the three-level modulation mode of the switching signals of the primary side switching devices S1, S2, S3, and S4 for operating the Figure 10A SHB LLC resonant converter 1000. Among them, the switching signals of the switching devices S1 and S2 are complementary, and the switching signals of the switching devices S3 and S4 are also complementary. In fact, a short dead time is inserted between each pair of complementary switching signals to avoid shoot-through. In this embodiment, based on the output voltage or output current, the control circuit generates the first, second, third, and fourth switching control signals S1 to S4 according to the first modulation mode. In the first modulation mode, (1) the first switching control signal S1 and the second switching control signal S2 are substantially complementary, and the third switching control signal S3 and the fourth switching control signal S4 are substantially complementary; (2) the first, second, third, and fourth switching control signals S1, S2, S3, and S4 have the same switching period T SW ; (3) in the switching period T SW each switching control signal has two rising edges and two falling edges, and the first rising edge of the first switching control signal S1 lags behind the first rising edge of the fourth switching control signal S4 by a first predetermined time, and the second rising edge of the first switching control signal S1 leads the second rising edge of the fourth switching control signal S4 by a second predetermined time; and (4) the control circuit can change the switching period, the first predetermined time, and the second predetermined time.

[0091] Figure 10B Is the signal timing diagram of the primary side switching devices S1 to S4 and the SHB phase leg output voltage V AB In the switching period T SWAmong them, the primary-side switching devices S1, S2, S3, and S4 are each turned on twice and turned off twice. Each time they are turned on, the primary-side switching device S1 or S4 has a fixed on-period, and the on-period lasts for 25% of the switching period T SW so that the cumulative on-period is a total of 50% of the switching period T SW . As Figure 10B shown, during the switching period T SW , the first rising edge of the primary-side switching device S1 lags behind the second rising edge of the primary-side switching device S4, and the second rising edge of the primary-side switch S1 leads the second rising edge of the primary-side switch S4. As Figure 10B shown, both the leading time and the lagging time are φ. In addition, the first rising edge of the primary-side switching device S1 leads the second rising edge of the primary-side switching device S4 by 1 / 2T SW . Operating the primary-side switching devices S1, S2, S3, and S4 in this way, the phase-leg output voltage V AB has voltages in three different stages. In this embodiment, (1) when neither the first switching device S1 nor the fourth switching device S4 is turned on, the voltages on the first and second electrical nodes become the first voltage level; (2) when either the first switching device S1 or the fourth switching device S4 is turned on, the voltage between the first and second electrical nodes reaches the second voltage level; and (3) when both the first switching device S1 and the fourth switching device S4 are turned on, the voltage on the first electrical node reaches the third voltage level. The first voltage level can be the ground voltage, the second voltage level is substantially half of the input signal, and the third voltage level is substantially the input signal. When the primary-side switching devices S1 and S4 are both in the on state, the phase-leg output voltage V AB is equal to V in . When the primary-side switching devices S -2 and S4 are both in the on state, or when the primary-side switching devices S1 and S3 are both in the on state, the phase-leg output voltage V AB is equal to 0.5V in . When the primary-side switching devices S2 and S3 are both in the on state, the phase-leg output voltage V AB is equal to zero volts.

[0092] Therefore, the waveform of the phase-leg output voltage V AB is periodic, and its period is half of the switching period T SW . Since the time period φ corresponds to the duration when the primary-side switching devices S2 and S4 or the primary-side switching devices S1 and S3 are both in the on state of the primary-side switching device, the time period φ determines the duration when the phase-leg output voltage V AB is equal to 1 / 2V in . The switching frequency f SW(Switching period T SW ) and time period φ are control variables used to set different control objectives.

[0093] Key waveforms and stage analysis

[0094] In this detailed description, to simplify the analysis of the operation, as Figure 10A shown, it is assumed that the ripple voltages of the input capacitor C in and the output capacitor C o can be ignored, and thus they can be represented by constant voltage sources V in and V o respectively. And the semiconductor circuit elements in this article are all assumed to have zero resistance (i.e., be short - circuited) when they are in the on - state. However, the output capacitance of the primary - side switching device cannot be ignored.

[0095] Figure 11 is the key waveform diagram of the SHB LLC resonant converter 1000 in FIG. 10. Figure 11 Shows the switching control voltages S1, S2, S3, and S4 (i.e., the positive voltages indicating that the primary - side switching devices S1 to S4 are in the on - state), the primary - side output voltage V AB , the resonant current i Lr of the resonant inductor Lr, the current i Lm in the magnetizing inductor Lm, and the voltages V S1 and V S2 across the primary - side switching devices S1 and S - 2. Figure 11 Shows the signal transitions at times t1, t2, t3, t4, t5, t6, t7, t8, t9, t10, t11, t12, t13, t14, t15, and t16.

[0096] Combined with Figure 11 , Figures 12A - 12P shows the sixteen operation stages of the SHB LLC resonant converter 1000 under the three - level modulation method of the embodiment of this case, which are time periods (t0, t1), (t1, t2), (t2, t3), (t3, t4), (t4, t5), (t5, t6), (t6, t7), (t7, t8), (t8, t9), (t9, t10), (t10, t11), (t11, t12), (t12, t13), (t13, t14), (t14, t15), and (t15, t16). In Figures 12A to 12P , the switching devices in the off - state are represented by dashed lines (e.g., Figure 12A the primary - side switching devices S2 and S3 in the off - state in S1 , C S2 , CS3 and C S4 is represented by

[0097] As Figure 12A shown, during the time period (t0, t1), the resonant current i Lr increases and flows into the resonant inductor Lr, transformer TR, exciting inductor Lm, resonant capacitor Cr, and primary side switching devices S1 and S4. The phase leg output voltage V AB is equal to the input voltage V in .

[0098] As Figure 12B shown, during the time period (t1, t2), after the primary side switching device S1 is turned off, the capacitor C S1 is charged to 1 / 2V in , and the capacitor C CS2 is completely discharged (i.e., becomes zero volts). The current i Lr starts to decrease. And due to the charging and discharging in the capacitors C S1 and C S2 , the phase leg output voltage V AB drops to 1 / 2V in .

[0099] As Figure 12C shown, during the time period (t2, t3), the primary side switching device S2 conducts at zero volts (i.e., conducts under ZVS conditions without generating switching losses). The phase leg output voltage V AB remains at 1 / 2V in , and the current i Lr continues to decrease.

[0100] As Figure 12D shown, during the time period (t3, t4), after the primary side switching device S4 is turned off, the capacitor C S4 is charged to 1 / 2V in , and the capacitor C S3 is completely discharged. Due to the charging and discharging in the capacitors C S4 and C S3 , the phase leg output voltage V AB becomes zero volts. The current i Lr continues to decrease.

[0101] As Figure 12E shown, during the time period (t4, t5), the primary side switching device S3 conducts under ZVS conditions. The phase leg output voltage V AB remains at zero volts, and the current i Lr continues to decrease.

[0102] As Figure 12FAs shown, during the time period (t5, t6), after the primary side switching device S3 is turned off, the capacitor C S3 is charged to 1 / 2V in , and the capacitor C S4 is completely discharged. Due to the charging and discharging of the capacitors C S3 and C S4 , the phase leg output voltage V AB rises to 1 / 2V in , and the current i Lr starts to increase.

[0103] As Figure 12G shown, during the time period (t6, t7), the primary side switching device S4 is turned on under ZVS conditions. The phase leg output voltage V AB remains at 1 / 2V in , and the current i Lr continues to increase.

[0104] As Figure 12H shown, during the time period (t7, t8), after the primary side switching device S2 is turned off, the capacitor C S2 is charged to 1 / 2V in , and the capacitor C S1 is completely discharged. Due to the charging and discharging of the capacitors C S2 and C S1 , the phase leg output voltage V AB becomes V in . The current i Lr continues to increase.

[0105] As Figure 12I shown, during the time period (t8, t9), the primary side switching device S1 conducts under ZVS conditions. The phase leg output voltage V AB remains at 1 / 2V in , and the current i Lr continues to increase.

[0106] As Figure 12J shown, during the time period (t9, t10), after the primary side switching device S4 is turned off, the capacitor C S4 is charged to 1 / 2V in , the capacitor C S3 is completely discharged. The current i Lr starts to decrease. Due to the charging and discharging of the capacitors C S4 and C S3 , the phase leg output voltage V AB drops to 1 / 2V in .

[0107] As Figure 12KAs shown, during the time period (t10, t11), the primary side switching device S3 conducts under ZVS conditions. The phase leg output voltage V AB remains at 1 / 2V in , and the current i Lr continues to decrease.

[0108] As Figure 12L shown, during the time period (t11, t12), after the primary side switching device S1 turns off, the capacitor C S1 charges to 1 / 2V in , and the capacitor C S2 fully discharges. Due to the charging and discharging in the capacitors C S1 and C S2 , the phase leg output voltage V AB becomes zero volts. The current i Lr continues to decrease.

[0109] As Figure 12M shown, during the time period (t12, t13), the primary side switching device S2 conducts under ZVS conditions. The phase leg output voltage V AB remains at zero volts, and the current i Lr continues to decrease.

[0110] As Figure 12N shown, during the time period (t13, t14), after the primary side switching device S2 turns off, the capacitor C S2 charges to 1 / 2V in , and the capacitor C S1 fully discharges. Due to the charging and discharging in the capacitors C S2 and C S1 , the phase leg output voltage V AB rises to 1 / 2V in , and the current i Lr starts to increase.

[0111] As Figure 12O shown, during the time period (t14, t15), the primary side switching device S1 conducts under ZVS conditions. The phase leg output voltage V AB remains at 1 / 2V in , and the current i Lr continues to increase.

[0112] As Figure 12P shown, during the time period (t15, t16), after the primary side switching device S3 turns off, the capacitor C S3 charges to 1 / 2V in , and the capacitor C S4 fully discharges. Due to the charging and discharging in the capacitors C S3 and C S4During charging and discharging, the phase-leg output voltage V AB becomes V in . The current i Lr continues to increase.

[0113] Since all primary-side switching devices are turned on under ZVS conditions, the three-level modulation method of the embodiments of the present case significantly reduces the total switching loss.

[0114] Advantages of the three-level modulation method of the embodiments of the present case:

[0115] A. Reducing the output voltage

[0116] In the embodiments of the present case, the three-level modulation method generates a controllable phase-leg output voltage V r -C r -L m across the series L AB , and the phase-leg output voltage V AB includes DC and AC components. In this embodiment, during the operation of the LLC resonant converter, the resonant capacitor C r blocks the DC component, so that only the AC component appears across the primary-side winding of the transformer TR. When the device switching frequency f SW is equal to the resonant frequency and the duration φ = 0, the output voltage V o is equal to the average value of the product of the rectified AC component of the phase-leg output voltage V AB and the voltage gain of the transformer TR. Since the voltage gain of the transformer TR, i.e., the turns ratio therefore the output voltage V o can be obtained from , where N S and N P are the number of turns of the secondary-side winding and the primary-side winding respectively, is the average value of the AC component of the phase-leg output voltage V AB after rectification. For the expected value of the time period φ, if the switching frequency is fixed, the average voltage of the rectified AC component decreases. Therefore, the output voltage V o also decreases. Therefore, the three-level modulation method of the embodiments of the present case can be used to adjust the output voltage V o without affecting the switching frequency f r .

[0117] B. Balancing the current stress in each primary-side switching device

[0118] The "on" resistance of the power device causes conduction loss. In Figure 10ASHB LLC resonant converter 1000, the primary - side switching devices S1 to S4 cause conduction losses when conducting. FIG. 13 shows a three - level modulation method in an embodiment of this case, and without considering the influence of dead - time, the control switch voltages S1 to S4 of the primary - side switching devices S1 to S4, and during the switching period T SW The signal schematic diagrams of the currents I1 to I4 flowing through the primary - side switching devices S1 to S4. The currents I1 to I4 have the same root - mean - square (RMS) value, thus balancing the thermal stress among the primary - side switching devices. Therefore, these devices have a longer service life, thereby increasing the reliability of the SHB LLC resonant converter 1000.

[0119] Voltage balance of DC bus capacitors

[0120] Ideally, in the three - level modulation method of the embodiment of this case, when the top capacitor and the bottom capacitor (for example, capacitor C - in1 and C in2 ) transfer the same power to the resonant circuit, the voltages of their DC bus capacitors will automatically balance without any control. However, in any actual SHB LLC resonant converter, parameter mismatches of the equivalent series resistance (ESR) and the capacitance of the DC bus capacitors, as well as timing mismatches of the switching control or gate signals, are inevitable. These mismatches will cause voltage imbalance of the DC bus capacitors.

[0121] The embodiment of this case provides a method that can balance the voltages of the DC bus capacitors and keep the SHB LLC resonant converter within a safe range. When the voltage of capacitor C in1 is much greater than the voltage of capacitor C in2 , the switching control signals of the primary - side switching devices S1 and S2 are both delayed by a short time, and the switching control signals of the primary - side switching devices S3 and S4 are also advanced by the same time. Similarly, when the voltage of capacitor C in2 is greater than the voltage of capacitor C in1 , the switching control signals of the primary - side switching devices S1 and S2 are both advanced by a short time, and the switching control signals of the primary - side switching devices S3 and S4 are also delayed by the same time. This short time preferably should not exceed a preset limit, such as but not limited to 5% of the switching period T SW to avoid any adverse effects on the normal operation of the SHB LLC resonant converter.

[0122] By combining pulse - frequency modulation (PFM) with the three - level modulation method, operation with a wide output voltage range is achieved.

[0123] As described above, when the SHB LLC resonant converter is at the resonant frequency f rat or very close to the resonant frequency f r When operating at the operating point, the maximum conversion efficiency can be achieved. And in order to obtain a wider output voltage range, traditional control changes the device switching frequency f SW to adjust the DC voltage gain. However, the switching frequency control will cause the operating point to deviate from the maximum efficiency. In addition, for a very wide output voltage range, due to fixed circuit parameters, even adjusting the switching frequency control cannot obtain the required DC voltage gain.

[0124] The inventors of the present case realized that the SHB LLC resonant converter can combine different modulation methods to efficiently obtain different required DC voltage gains at or near the resonant frequency (i.e., in a relatively narrow input switching frequency range) to provide a wider output voltage range. In addition, the three-level modulation method disclosed in the embodiments of the present case can achieve a wider output voltage range and circuit gain even when the circuit parameter values are fixed. Figure 14 FIG. 9 is a schematic diagram of an SHB LLC resonant converter 1400 according to a preferred embodiment of the present case. The SHB LLC resonant converter 1400 includes a control circuit 1401. The control circuit 1401 generates a switching control signal for switching devices S1 to S4 according to the output voltage V o (and optionally the output current I o ).

[0125] In the SHB LLC resonant converter 1400, the control circuit 1401 can use the output voltage V o as its main control target. The output current I o can also be used as a separate control target or a feedback signal representing the load condition. The reference value representing each control target can be generated internally by the control circuit 1401 or from an external source. According to the difference between the sensed voltage V o (or the sensed current I o ) and the corresponding reference value, the control circuit 1401 switches between multiple modulation methods to control the switching devices S1 to S4 of the primary side pins in the SHB LLC resonant converter 1400. One or more control targets can be used to determine other control parameter values, such as the device switching frequency f SW and phase shift in the selected modulation method. The switching control signal used to operate the switching devices S1 to S4 on the primary side is generated according to the selected modulation method and other control parameter values.

[0126] The traditional symmetric frequency modulation method changes the switching frequency f r near the resonant frequency f SWTo provide the desired output voltage gain. The three - level modulation method of the embodiments of the present case provides an additional output voltage gain at the maximum allowable switching frequency by controlling the dead - time of the primary - side switches. Therefore, the embodiments of the present case provide a control method that combines selectable modulation methods and frequency control to achieve a preset output voltage range. (The output voltage adjustment ranges of different modulation methods may overlap.) The selection of the modulation method in the embodiments of the present case can be based on control parameters such as voltage control targets, load conditions, and allowable operating frequency ranges, for example.

[0127] Modulation method transition

[0128] Figure 15A shows the control signals for the primary - side switching devices S1 to S4 and the phase - leg output voltage V when the SHB LLC resonant converter 1400 switches from the symmetric modulation method to the three - level modulation method. AB . In some embodiments, such as Figure 15B , Figure 15B shows the control signals for the primary - side switching devices S1 to S4 and the phase - leg output voltage V when the SHB LLC resonant converter 1400 switches from the three - level modulation method to the symmetric modulation method. AB . As Figure 15A shown, when the control target of the phase - leg output voltage V AB changes from a high voltage to a low voltage V O , the symmetric modulation method (interval 1501) switches to the three - level modulation method (interval 1502) and no transition interval is required. Similarly, as Figure 15B shown, when the control target of the phase - leg output voltage V AB changes from a low voltage to a high voltage V O , the three - level modulation method (interval 1503) switches to the symmetric modulation method (interval 1504) and no transition interval is required either. In some embodiments, the symmetric modulation method includes variable - frequency modulation. In other embodiments, the symmetric modulation method includes fixed - frequency modulation.

[0129] Topology extension

[0130] The embodiments of the present case are equally applicable to SHB LLC resonant converters with other secondary - side topologies to provide a wide output voltage range with the same narrow device switching frequency. According to another preferred embodiment of the present case, Figure 16A shows an SHB LLC resonant converter 1600 with a full - bridge synchronous rectifier on the secondary side. Figure 16B shows an SHB LLC resonant converter 1650 with a center - tapped transformer TR on the center side and synchronous rectifiers S5 and S6 on the secondary side. In the SHB LLC resonant converter 1600, the control circuit 1601 can be incorporated inFigure 10B the method discussed in, i.e., in the traditional secondary-side control method, signals are generated to operate the switching devices S1 to S4 on the primary side and the switching devices S5 to S8 on the secondary side. In the SHB LLC resonant converter 1650, the control circuit 1651 can combine the information about Figure 10B the method discussed in, i.e., in the traditional secondary-side control method, signals are generated to operate the switching devices S1 to S4 on the primary side and the switching devices S5 to S8 on the secondary side. The embodiments of this case are equally applicable to the switching devices S5 to S8 on the secondary side of the SHB LLC resonant converter 1650, or the switching devices S5 and S6 on the secondary side of the SHB LLC resonant converter 1650 are replaced by diodes.

[0131] The embodiments of this case are also applicable to the SHB LLC resonant converter with a variable DC input voltage or with two different input voltages, such as Figure 17A and Figure 17B shown, Figure 17A shows the SHB LLC resonant converter 1700 with a variable DC input voltage source 1702. Figure 17B shows the SHB LLC resonant converter 1750 with DC input voltages V in1 and V in2 respectively controlled according to an embodiment of the DC converter. An embodiment of this case is as Figure 17A shown, the variable DC input voltage source 1702 can be controlled by the signal V ctrl , and this signal can be generated internally or provided externally. Combining the control method of the signal V ctrl with any method of the present invention discussed above can further expand the output voltage range or further reduce the device switching frequency range of the SHB LLC resonant converter 1700.

[0132] In another embodiment of this case, by understanding that for the same transformer turns ratio Np / Ns and the same values of the resonant cavity elements L r , C r and L m, the DC voltage gain in the three - level modulation mode is usually higher than that in the two - level asymmetric half - input voltage modulation mode. Thereby, the efficiency of the SHB LLC resonant converter operating within a relatively wide input or output voltage range can be improved. Therefore, in some usage scenarios, in some embodiments, the three - level modulation mode can be more suitable for applications with higher DC voltage gain, while the two - level asymmetric half - input voltage modulation mode is more suitable for applications with lower DC voltage gain. Since the converter can operate in one modulation mode and transition to another modulation mode, optimized performance can be obtained through appropriate control in the SHB LLC resonant converter operating within a relatively wide input or output voltage range.

[0133] To illustrate in a relatively simple form, a controller can be utilized to achieve an interactive transition between the three - level modulation mode and the two - level asymmetric half - input voltage modulation mode. This is to briefly turn off the switching device to transition between the above - mentioned modulation modes. Since there is no power transfer between the input and output terminals during the transition, under the above - mentioned transition method, the output variable (usually the output voltage or output current) drops (i.e., during the conversion, the value of the output variable drops below a steady - state value). Under this transition method, to reduce the drop amplitude of the output variable and maintain the output variable within a specific range during the topology transition, energy storage can be increased in the output filter. On the other hand, a controller that can transition between the two modulation modes without turning off the switching device can also be adopted. Since the two modulation modes have completely different switching patterns, an instantaneous modulation mode transition will cause severe oscillations and voltage spikes in the system, thereby damaging the converter.

[0134] According to another embodiment, the present case provides a trajectory - control - based SHB LLC resonant converter, which can transition the modulation mode without turning off the switching device (i.e., during the transition, all switching devices maintain the modulation state), and will not cause oscillations during or after the transition, while keeping all system variables within the specified range without any additional components.

[0135] Figure 18A The SHB LLC resonant converter 1800 showing another embodiment of the present case can transition the modulation mode between the three - level modulation mode and the two - level asymmetric half - input voltage modulation mode, and can improve or even avoid the problems of circuit oscillations and parameter peaks in the system. Figure 18A The SHB LLC resonant converter 1800 operating in the three - level modulation mode or the two - level asymmetric half - input voltage modulation mode is shown. In Figure 18A As shown in the circuit diagram, the transition between the two modulation modes is achieved by using trajectory control. For example, in this embodiment, Figure 18BShows the switching signals of switching devices S1 and S4 during the transition from the three-level modulation mode to the two-level asymmetric half-input voltage modulation mode. During the operation of the three-level modulation mode, the switching devices S1 to S4 operate at a fixed switching frequency and a variable phase shift angle When the transition starts, the controller 1801 first generates the last 1 / 2V AB voltage level after the high voltage level of V in and sets it as the transition starting point. Then, the controller 1801 generates a specific gate signal to form a zero voltage level of V AB and its duration is θ. After the zero voltage level, the controller 1801 generates the two-level modulation mode gate signal, so the transition is completed. Figure 18C Shows the switching signals of switching devices S1 and S4 during the transition from the two-level asymmetric half-input voltage modulation mode to the three-level modulation mode. As Figure 18C shown, when the transition starts, the controller 1801 first generates the last 1 / 2V AB voltage level after the high voltage level of V in in the two-level modulation mode and sets it as the transition starting point. Then, the controller 1801 generates a specific gate signal to form a zero voltage level of V AB and its duration is σ. After the zero voltage level, the controller 1801 generates the three-level modulation mode gate signal, so the transition is completed.

[0136] The output voltage during the transition is determined by the resonator energy and is usually represented by the trajectories of the normalized resonant voltage V CRN and the normalized resonant current i LRN . Figure 18D Shows the trajectories in the three-level modulation mode and the trajectories in the two-level modulation mode. Among them, the dark trajectory represents the trajectory in the two-level modulation mode, and the light trajectory represents the trajectory in the three-level modulation mode. A part of the trajectory of the three-level modulation mode overlaps with a part of the trajectory of the two-level modulation mode, which means that the origin and radius of the above two parts are the same. In other words, the resonator energy stored in the resonant element is the same. Therefore, the transition period should start from the place where the trajectories start to overlap, so that the energy mismatch of the resonator before and after the transition is small, and thus optimized transition performance can be obtained.

[0137] As Figure 18B and Figure 18C shown, before the transition, the switching devices S2 and S4 are turned on to form the last 1 / 2V in voltage level of the three-level modulation mode. After the transition, the switching devices S2 and S3 are turned on to form the first zero voltage level of the two-level modulation mode. However, other methods that can generate the same V ABFIG. 19 shows another possible gate signal during the transition from the three-level modulation mode to the two-level modulation mode. FIG. 20 shows another possible gate signal during the transition from the two-level modulation mode to the three-level modulation mode.

[0138] According to another embodiment of the present invention, a method for providing a modulation mode transition without interrupting a switching device maintains an output variable within a specified range without increasing the storage capacity in an output filter.

[0139] Figure 21A , Figure 21B and Figure 21C A SHB LLC resonant converter according to another embodiment of the present invention is shown, which transitions between a three-level modulation mode and a two-level asymmetric half-input voltage modulation mode without turning off the switching device. Figure 21A A SHB LLC resonant converter 2100 is shown that can operate in a three-level modulation scheme or a two-level asymmetric half-input voltage modulation scheme. Figure 21A The transition between the two modulation modes in the circuit structure diagram is achieved by PWM signal control of four switch devices, and the PWM control signal is controlled by the controller 2101. For example, Figure 21B The switching signals of the switching devices S1 to S4 during the transition period from the three-level modulation mode to the two-level asymmetrical half input voltage modulation mode are shown. During the operation in the three-level modulation mode, the switching devices S1 and S4 operate at a variable switching frequency and a 50% duty cycle. During the transition period, the switching devices S2 and S3 continue to operate at a variable switching frequency and complement the switching devices S1 and S4, respectively, and the PWM conversion control monotonically reduces the duty cycle of one of every two pulses of the switching device S1 from 50% to 0%, and monotonically reduces the duty cycle of two pulses of every two pulses of the switching device S4 from 50% to 0%. In addition, the narrow pulse of the switching device S1 is phase-shifted by 180 degrees compared to the narrow pulse of the switching device S4, and the non-variable pulse of the switching device S1 is phase-shifted by 180 degrees compared to the non-variable pulse of the switching device S4. In this embodiment, since the duty cycle of the switching devices S1 and S4 becomes 25%, the LLC resonant converter continues to operate in the two-level asymmetrical half input voltage modulation mode during the transition period with variable frequency control. Figure 21C FIG. 2 shows the switching signals of the switching devices S1 to S4 during the transition from the two-level asymmetrical half-input voltage modulation mode to the three-level modulation mode. Figure 21C As shown, the modulation of the switch devices S1 and S4 transitions from the two-level asymmetric half-input voltage modulation mode to the three-level modulation mode, and the transition direction is opposite to the transition from the three-level modulation mode to the two-level asymmetric half-input voltage modulation mode. Figure 21CAmong them, the duty cycle of every other pulse of the switching devices S1 and S4 monotonically increases from 0% to 50%, and the equivalent duty cycle of S1 and S4 increases from 25% to 50%.

[0140] In some embodiments, according to at least one of the power supply voltage, the load voltage, and the load current, the controller selects to control the operation of the power converter in a three-level or two-level modulation mode. In some embodiments, the controller controls the operation of the switching device in at least one of a variable switching frequency, a constant-frequency pulse-width modulation mode, and a constant-frequency phase-shift control.

[0141] In some embodiments, during the modulation mode transition, the change in the duty cycle of the switching devices S1 and S4 depends on the time required for the average voltage of the resonant capacitor Cr to change between the three-level modulation mode and the two-level asymmetric half-input voltage modulation mode. Specifically, when transitioning from the three-level modulation mode to the two-level asymmetric half-input voltage modulation mode, the average voltage of the resonant capacitor changes from Vin / 2 to Vin / 4. Correspondingly, when transitioning from the two-level asymmetric half-input voltage modulation mode to the three-level modulation mode, the average voltage of the resonant capacitor changes from Vin / 4 to Vin / 2. If the topology transition time is too short or occurs suddenly (worst case), the voltage on the resonant capacitor Cr does not have enough time to fully charge to the appropriate value before the topology transition is completed. In the above cases, the instantaneous volt-second imbalance on the transformer may cause the transformer core to saturate or the output voltage to exceed the desired range.

[0142] As Figure 21B and Figure 21C shown, the duty cycle of the switching devices S1 and S4 can be reduced by simultaneously delaying the conduction edge and the leading turn-off edge. However, other PWM control methods can also achieve the above effects. For example, the duty cycle can be reduced only by delaying the conduction edge or the turn-off edge.

[0143] Generally speaking, the modulation mode transition is extremely important for input or output voltage changes or output current changes. Therefore, in order to achieve the transition control of the modulation mode, it is necessary to sense the input voltage, the output voltage, or the output current. The topology transition can be initiated by a system controller (e.g., a power management controller), or can be autonomously initiated by the power converter controller. The controller with modulation mode transition control can be implemented by an analog circuit and a digital circuit. However, because it is easier to implement digital adaptive control and complex timing requirements, in some embodiments, the digital method is a better choice.

[0144] The modulation mode transition of the SHB LLC resonant converter according to the embodiments of the present case can also be applied to other resonant converters, including conventional full-bridge LLC resonant converters. The modulation mode transition occurs between the full-bridge two-level modulation mode and the half-bridge two-level modulation mode. In addition, in the embodiments of the present case, the controller or similar or equivalent technical terms can be equivalent to or regarded as the control circuit. Correspondingly, the control circuit or similar or equivalent technical terms can also be equivalent to or regarded as the controller.

[0145] It should be noted that the above are only the preferred embodiments proposed for the purpose of illustrating the present case. The present case is not limited to the described embodiments. The scope of the present case is determined by the appended claims. And the present case can be variously modified by those skilled in the art, but all are not beyond the scope of protection of the appended claims.

Claims

1. A power converter that receives an input signal from a voltage source and provides an output voltage or an output current to at least one load, the power converter comprising: A primary side circuit, comprising: A first pair of switching devices and a second pair of switching devices connected in series, wherein each of the switching devices is controlled by a corresponding switching control signal to provide the input signal across the series-connected first pair of switching devices and the second pair of switching devices; An LC resonant circuit electrically connected between a first electrical node of the first pair of switching devices and a second electrical node of the second pair of switching devices; and An isolation transformer having a first winding and a second winding, wherein the first winding is connected between a third electrical node and a fourth electrical node of the LC resonant circuit; A secondary side circuit connected in parallel to the second winding of the isolation transformer, wherein the secondary side circuit includes a filter capacitor adapted to provide the output voltage or the output current to the at least one load; and A control circuit, wherein based on at least one of the output voltage, the output current, the input signal, and at least one external control signal, the control circuit selects one of two or more modulation methods at any given time and provides the switching control signal according to the selected modulation method to control the operation of the switching devices of the primary side circuit, Among them, The control circuit operates the switching devices based on the output voltage or the output current, and the control circuit is configured to operate in a first modulation method and a second modulation method, wherein in the first modulation method, the switching devices are operated to generate the output voltage including three different voltage levels, and wherein in the second modulation method, the switching devices are operated to generate the output voltage including two different voltage levels; Wherein the control circuit operates in a fourth modulation method between operating in the first modulation method and operating in the second modulation method. During the transition from the first modulation method to the second modulation method in the fourth modulation method, the duty cycle of every other pulse of one of the first pair of switching devices is monotonically reduced from 50% to 0%, and the duty cycle of every other pulse of one of the second pair of switching devices is reduced from 50% to 0%. Or during the transition from the second modulation method to the first modulation method, the duty cycle of every other pulse of one of the first pair of switching devices is monotonically increased from 0% to 50%, and the duty cycle of every other pulse of one of the second pair of switching devices is monotonically increased from 0% to 50%.

2. The power converter according to claim 1, wherein, One of the multiple modulation methods operates in a symmetric modulation method. The first pair of switching devices and the second pair of switching devices each have a first switching device and a second switching device. In the symmetric modulation method, the control signals of the first switching device and the second switching device in the first pair of switching devices are complementary, and the control signals of the first switching device and the second switching device in the second pair of switching devices are complementary. Among them, the switching control signals of the first switching device of the first pair of switching devices and the second switching device of the second pair of switching devices are in-phase signals and have a duty cycle of 50%.

3. The power converter according to claim 1, wherein, One of the multiple modulation methods operates in an asymmetric modulation method. The first pair of switching devices and the second pair of switching devices each have a first switching device and a second switching device. In the asymmetric modulation method, the control signals of the first switching device and the second switching device in the first pair of switching devices are complementary, and the control signals of the first switching device and the second switching device in the second pair of switching devices are complementary. Among them, the switching control signals of the first switching device of the first pair of switching devices and the second switching device of the second pair of switching devices operate at a duty cycle of 25% at the same switching frequency, and their phases are 180 degrees different from each other.

4. The power converter according to claim 1 further includes a first input capacitor and a second input capacitor, wherein, The first pair of switching devices has a first switching device and a second switching device, and the second pair of switching devices has a third switching device and a fourth switching device. The first input capacitor is connected across the first switching device and the second switching device, and the second input capacitor is connected across the third switching device and the fourth switching device. Among them, when the voltage of the first input capacitor is greater than the voltage of the second input capacitor, the control circuit delays the switching control signals of the first switching device and the second switching device for a first time, and advances the switching control signals of the third switching device and the fourth switching device by the first time; when the voltage of the second input capacitor is greater than the voltage of the first input capacitor, the control circuit advances the switching control signals of the first switching device and the second switching device for a second time, and delays the switching control signals of the third switching device and the fourth switching device by the second time.

5. The power converter according to claim 1, wherein, The control circuit uses one or a combination of frequency shift, phase shift, and duty cycle shift to cause the control circuit to switch the modulation method.

6. The power converter according to claim 1, wherein, The first pair of switching devices includes a first switching device and a second switching device, and the second pair of switching devices includes a third switching device and a fourth switching device, wherein: (1) when both the first switching device and the fourth switching device are non-conductive, the voltages at the first electrical node and the second electrical node are at a first voltage level; (2) when one of the first switching device and the fourth switching device is in a conductive state, the voltage across the first electrical node and the second electrical node changes to a second voltage level; and (3) when both the first switching device and the fourth switching device are conductive, the voltage across the first electrical node and the second electrical node reaches a third voltage level, where the first voltage level is grounded, the second voltage level is substantially half of the input signal, and the third voltage level is substantially the input signal.

7. The power converter according to claim 1, wherein, The first pair of switching devices includes a first switching device and a second switching device, and the second pair of switching devices includes a third switching device and a fourth switching device. The first switching device, the second switching device, the third switching device, and the fourth switching device are controlled by a first switch control signal, a second switch control signal, a third switch control signal, and a fourth switch control signal respectively. Among them, in a first modulation mode: (1) the first switch control signal of the first switching device and the second switch control signal of the second switching device are complementary to each other, and the third switch control signal of the third switching device and the fourth switch control signal of the fourth switching device are complementary to each other; (2) the first switch control signal of the first switching device, the second switch control signal of the second switching device, the third switch control signal of the third switching device, and the fourth switch control signal of the fourth switching device have the same switching period; (3) within the switching period, the switch control signal of each switching device has two rising edges and two falling edges, and the first rising edge of the first switch control signal of the first switching device lags behind the first rising edge of the fourth switch control signal of the fourth switching device by a first predetermined time, and the second rising edge of the first switch control signal of the first switching device leads the second rising edge of the fourth switch control signal of the fourth switching device by a second predetermined time, or the first rising edge of the first switch control signal of the first switching device leads the first rising edge of the fourth switch control signal of the fourth switching device by a first predetermined time, and the second rising edge of the first switch control signal of the first switching device lags behind the second rising edge of the fourth switch control signal of the fourth switching device by a second predetermined time; and (4) the control circuit can change the switching period, the first predetermined time, and the second predetermined time.

8. The power converter according to claim 7, wherein, The control circuit generates the first switching control signal of the first switching device, the second switching control signal of the second switching device, the third switching control signal of the third switching device, and the fourth switching control signal of the fourth switching device according to a second modulation method different from the first modulation method. The second modulation method operates as a symmetric modulation method. One of the first modulation method and the second modulation method includes variable frequency modulation, and one of the first modulation method and the second modulation method includes fixed frequency modulation control. The control circuit changes the gain of the power converter by operating the power converter in the first modulation method in a first time interval and operating the power converter in the second modulation method in a second time interval.

9. The power converter according to claim 1, wherein, The control circuit is further configured to operate in a third modulation method, in which the switching device generates a zero voltage level, and the control circuit operates in the third modulation method between operating in the first modulation method and operating in the second modulation method.

10. A power converter having a first input terminal and a second input terminal for receiving an input signal from a voltage source and providing an output voltage or an output current to at least one load, the power converter comprising: A transformer including a first winding and a second winding; A primary side circuit including: A first switching device, a second switching device, a third switching device, and a fourth switching device connected in series, wherein the first switching device, the second switching device, the third switching device, and the fourth switching device are connected across the first input terminal and the second input terminal. The first switching device and the second switching device are controlled by a first switching control signal and a second switching control signal respectively, and the third switching device and the fourth switching device are controlled by a third switching control signal and a fourth switching control signal respectively; and A resonant circuit electrically connected between a first electrical node and a second electrical node, wherein the first electrical node is a common terminal between the first switching device and the second switching device connected in series, the second electrical node is a common terminal between the third switching device and the fourth switching device connected in series, and the resonant circuit is coupled to the first winding of the transformer; A secondary side circuit connected to the second winding of the transformer, wherein the secondary side circuit includes a filter capacitor adapted to provide the output voltage or the output current to the load; and A control circuit is adapted to provide the switch control signals to control the operation of the switching devices of the primary side circuit. Among them, the first switch control signal, the second switch control signal, the third switch control signal, and the fourth switch control signal have the same switching period. During the switching period, (1) each of the switch control signals has two rising edges and two falling edges; (2) the first rising edge of the first switch control signal lags behind the first rising edge of the fourth switch control signal by a first predetermined time, and the second rising edge of the first switch control signal leads the second rising edge of the fourth switch control signal by a second predetermined time, or (3) the first rising edge of the first switch control signal leads the first rising edge of the fourth switch control signal by the first predetermined time, and the second rising edge of the first switch control signal lags behind the second rising edge of the fourth switch control signal by the second predetermined time; and (4) the control circuit can change the switching period, the first predetermined time, and the second predetermined time. Among them, the control circuit operates the switching devices according to the output voltage or the output current. The control circuit is configured to operate in a first modulation mode and a second modulation mode. In the first modulation mode, the switching devices are operated to generate the output voltage including three different voltage levels. In the second modulation mode, the switching devices are operated to generate the output voltage including two different voltage levels. Among them, the control circuit operates in a fourth modulation mode between operating in the first modulation mode and operating in the second modulation mode. In the fourth modulation mode, during the transition from the first modulation mode to the second modulation mode, the duty cycle of every other pulse of one of the first switching device and the second switching device monotonically decreases from 50% to 0%, and the duty cycle of every other pulse of one of the third switching device and the fourth switching device decreases from 50% to 0%. Or during the transition from the second modulation mode to the first modulation mode, the duty cycle of every other pulse of one of the first switching device and the second switching device monotonically increases from 0% to 50%, and the duty cycle of every other pulse of one of the third switching device and the fourth switching device increases from 0% to 50%.

11. The power converter according to claim 10, wherein, The control circuit is further configured to operate in a third modulation mode. In the third modulation mode, the switching devices generate a zero voltage level.

12. The power converter according to claim 11, wherein, The control circuit operates in the third modulation mode between operating in the first modulation mode and operating in the second modulation mode.

13. The power converter according to claim 10, wherein, One of the first modulation method, the second modulation method, and the fourth modulation method operates in an asymmetric modulation method. In the asymmetric modulation method, the control signals of the first switching device and the second switching device are complementary, and the control signals of the third switching device and the fourth switching device are complementary. Among them, one of the first switching device and the second switching device and one of the third switching device and the fourth switching device operate with a duty cycle of 25% at the same switching frequency, and their phases are 180 degrees different from each other.

14. A power converter having a first input terminal and a second input terminal for receiving an input signal from a voltage source and providing an output voltage or an output current to at least one load, the power converter comprising: A primary side circuit, comprising: A first switching device, a second switching device, a third switching device, and a fourth switching device connected in series. The first switching device, the second switching device, the third switching device, and the fourth switching device are connected across the first input terminal and the second input terminal. The first switching device and the second switching device are controlled by a first switching control signal and a second switching control signal respectively, and the third switching device and the fourth switching device are controlled by a third switching control signal and a fourth switching control signal respectively. The input signal is provided through the first switching device, the second switching device, the third switching device, and the fourth switching device. A resonant circuit electrically connected between a first electrical node and a second electrical node, where the first electrical node is a common terminal between the first switching device and the second switching device connected in series, and the second electrical node is a common terminal between the third switching device and the fourth switching device connected in series. An isolation transformer comprising a first winding and a second winding, where the first winding is connected between a third electrical node and a fourth electrical node of the resonant circuit. A secondary side circuit connected to the second winding of the transformer, where the secondary side circuit includes a filter capacitor adapted to provide the output voltage or the output current to the load; and A control circuit adapted to operate the switching devices according to at least one of the output voltage and the output current. The control circuit is configured to operate in a first modulation method and a second modulation method. In the first modulation method, the switching devices are operated to generate the output voltage including three different voltage levels. In the second modulation method, the switching devices are operated to generate the output voltage including two different voltage levels. Wherein, the control circuit operates in a fourth modulation mode between operating in the first modulation mode and operating in the second modulation mode. In the fourth modulation mode, during the transition from the first modulation mode to the second modulation mode, the duty cycle of every other pulse of one of the first pair of switching devices is monotonically reduced from 50% to 0%, and the duty cycle of every other pulse of one of the second pair of switching devices is reduced from 50% to 0%. Or during the transition from the second modulation mode to the first modulation mode, the duty cycle of every other pulse of one of the first pair of switching devices is monotonically increased from 0% to 50%, and the duty cycle of every other pulse of one of the second pair of switching devices is monotonically increased from 0% to 50%.

15. The power converter according to claim 14, wherein, The control circuit is further configured to operate in a third modulation mode. In the third modulation mode, the switching devices generate a zero voltage level, and the control circuit operates in the third modulation mode between operating in the first modulation mode and operating in the second modulation mode.

16. The power converter according to claim 14, wherein, One of the first modulation mode, the second modulation mode, and the fourth modulation mode operates in an asymmetric modulation mode. In the asymmetric modulation mode, the control signals of the first switching device and the second switching device are complementary, and the control signals of the third switching device and the fourth switching device are complementary. Wherein, one of the first switching device and the second switching device and one of the third switching device and the fourth switching device operate with a duty cycle of 25% at the same switching frequency, and their phases are 180 degrees different from each other.

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