Converters used to transfer electricity to electrical loads

Abstract

A converter (145) is described, comprising: a first input terminal (150) and a second input terminal (155), a first output terminal (165) and a second output terminal (170) that can be connected to opposite ends of an electrical load (110), a first electrical branch (201) adapted to connect the first input terminal (150) to a first intermediate electrical node (202), a second electrical branch (203) adapted to connect the first intermediate electrical node (202) to the second input terminal (155), a third electrical branch (205) adapted to connect the first output terminal (165) to a second intermediate electrical node (206), a fourth electrical branch (207) adapted to connect the second intermediate electrical node (206) to the second output terminal (170), and a first output terminal (165) and a second output terminal (170) arranged at a first input terminal (150) and a second output terminal (170) arranged at a second input terminal (150). A first active switch (180) on the second electrical branch (203) and having a first conductive terminal (185) and a resonant circuit (200) whose size is set to reduce the voltage and / or current applied to the first active switch at least when the first active switch (180) switches from the cut-off state to the saturation state; wherein the resonant circuit (200) comprises at least: a first resonant inductor (215) arranged on the first electrical branch (201), a second resonant inductor (235) arranged on the third electrical branch (205), and a first resonant capacitor (220, 285, 290); wherein the first resonant inductor (215) and the second resonant inductor (235) are inductively coupled together with a reciprocal coupling factor less than 1.

Description

Technical Field

[0001] The present invention relates to a converter for transmitting electrical power to an electrical load. The electrical load can be any electrical or electronic device that must be powered to operate and / or charge its internal battery. Typical examples of this type of electrical / electronic device include, but are not limited to, smartphones, computers, laptops, tablets, televisions, appliances, home automation systems, servers, and many other related devices. Background Art

[0002] A currently very widespread solution for transferring electrical power to an electrical load is to use a converter, ie a circuit configured to convert an input voltage to an output voltage suitable for supplying the load.

[0003] For example, AC / DC converters are known which are suitable for converting an AC voltage into a DC voltage, just as DC / AC converters are suitable for converting a DC voltage into an AC voltage, and DC / DC converters or AC / AC converters are suitable for converting a DC / AC voltage into another DC / AC voltage with different characteristics.

[0004] To ensure greater safety and robustness in use, all these converters will be made according to an isolated configuration, ie they may comprise a primary circuit adapted to receive an input voltage and a secondary circuit connected to an electrical load, both galvanically isolated from each other.

[0005] The primary circuit may comprise a wave generator, ie in the form of a switching circuit adapted to generate high frequency voltage waves, whereas the secondary circuit may comprise a rectification stage adapted to convert these voltage waves into a continuous or substantially continuous voltage to be applied to the load.

[0006] A typical implementation of this approach is represented by a flyback AC / DC converter, in which galvanic isolation is achieved via a transformer that allows electromagnetic coupling between the primary and secondary circuits. A flyback circuit is essentially a variation of a buck-boost DC / DC converter, in which the energy storage inductor is replaced by a transformer.

[0007] This type of circuit has the advantage of being particularly simple and inexpensive, but has a number of disadvantages due to the fact that it is a hard-switching converter.

[0008] A hard-switching converter is a circuit equipped with a switch that switches from on to off, and vice versa, when the current and / or voltage applied to the switch is very different from zero. As a result, large dynamic losses, also called switching losses, are generated, which in turn limit the maximum operating frequency of the system, resulting in large size and low operating efficiency.

[0009] Over the years, this circuit has been improved through quasi-resonant techniques, which are useful for switching at relatively low voltages, as well as active clamping techniques on the primary circuit and synchronous rectification on the secondary circuit, which are useful for reducing losses.

[0010] However, all these arrangements have the disadvantage of resulting in high circuit complexity, which is accompanied by high component costs and maximum frequency limitations introduced due to the real-time control of the active components.

[0011] Another strategy to galvanically isolate the primary circuit from the secondary circuit of the converter is to separate the primary circuit from the secondary circuit by means of an isolation capacitor pair, which enables capacitive coupling that enables power transfer.

[0012] Capacitively isolated converters have many advantages, mainly a significant reduction in overall size due to elimination of the transformer and the possibility of increasing the operating frequency (eg up to hundreds of kHz, MHz, tens of MHz or hundreds of MHz).

[0013] Another advantage of these capacitively isolated converters is that they are able to achieve higher operating efficiencies that remain typically stable for both light and heavy electrical loads, rather than generally low efficiencies with efficiency peaks only for certain defined load intervals as typically occurs in transformer isolated converters.

[0014] Typical examples of this type of converter are based on resonant circuit diagrams (e.g. based on resonant D, E, F, E / F, E -1 、F -1 circuit or similar circuit).

[0015] This type of circuit allows to drastically reduce dynamic losses in active components (switches, such as MOSFETs) and electromagnetic emissions (EMI), and to significantly increase the maximum operating frequency of the circuit, with benefits to overall size, weight and cost.

[0016] An example of a resonant circuit used in a converter isolated by capacitive coupling is illustrated in International Patent Application WO2013150352.

[0017] This circuit is actually a derivative of a Class E or similar amplifier, where the isolation capacitor acts as the resonant capacitor of the LC tank.

[0018] This circuit has numerous advantages, in particular high operating efficiency and low dynamic losses due to the possibility of obtaining zero voltage switching (Zero Voltage Switching) and / or zero current switching (Zero Current Switching) types of active switches.

[0019] The low dynamic losses also allow a significant increase in operating frequency compared to conventional transformer-based flyback converters, also reducing the overall size.

[0020] One of the few drawbacks of this architecture is represented by the need to place a choke inductor of large value between the supply voltage of the circuit and the active switch.

[0021] The choke inductor should have a theoretically infinite value so as to be charged by acting approximately as a current generator during the switch-on interval and discharged by powering the circuit with an approximately constant current during the switch-off interval.

[0022] Obviously, in a real circuit the choke inductor does not have an infinite value, but it is still very large and thus has a high size characteristic.

[0023] Similar dimensions characterize the output filter inductor, which is typically located downstream of the secondary rectifier stage, and which also has an ideal infinite value.

[0024] Another key component in this category of circuits is the resonant inductor in series with the isolation capacitor. This resonant inductor typically has a high inductance value and also has a small isolation capacitor necessary to compensate for low touch current (also called leakage current).

[0025] The high inductance value, high operating frequency and high current ripple (roughly sinusoidal current) make the aforementioned resonant inductor the source of almost all losses for this type of circuit, losses being realized in particular as losses due to eddy current phenomena in the core of the magnetic material (such as hysteresis and eddy currents) and losses due to eddy current phenomena in the wires wound inside it (such as skin effect, Joule effect and proximity phenomenon).

[0026] In order to address this drawback, a solution has been proposed to replace the choke inductor with a resonant inductor capable of resonating with the isolation capacitor.

[0027] An example of this solution is described in Italian patent application No. 102018000008935.

[0028] This circuit allows the use of a single inductor present on the primary circuit, acting both as a choke inductor to store energy during the on-time period and as a resonant inductor during the off-time period, thereby significantly reducing the total inductance value of the circuit and, therefore, reducing the overall size and improving efficiency.

[0029] A further improvement of this solution, also described in the aforementioned Italian patent application No. 102018000008935, envisages providing another resonant inductor on the secondary circuit, which resonant inductor can simultaneously act as an output filter reducer and resonate with the isolation capacitor and with the resonant inductor provided on the primary circuit.

[0030] Thus, the overall resonant frequency of the circuit is determined by the primary resonant inductor and the secondary resonant inductor in series with the isolation capacitor.

[0031] Besides the advantage of compactness, the circuit has the further advantage of being able to eliminate the further effect of the isolation capacitance, given the elimination of bulky inductive components, and thus having greater power delivered to the load at the same frequency and size of the isolation capacitance.

[0032] However, the circuit has the disadvantages of high peak voltages supported by the switches and high reactive currents circulating in the circuit (especially in the inductor), which are sources of losses that, while lower than in conventional circuits, represent almost all the losses in the converter.

[0033] Another disadvantage is the need to provide two resonant inductors, which are typically characterized by a lower inductance / volume ratio than the choke inductor.

[0034] In fact, the highest peak-to-peak currents discharged on resonant inductors are related to the greater presence of eddy current phenomena (such as hysteresis, skin effect and proximity effect), which are usually minimized by reducing the relative permeability of the magnetic material core (even using air gaps, materials based on nanopowders of ferromagnetic materials at high frequencies, thus similar to distributed gap inductors) and thus increasing the size of the inductor.

[0035] These disadvantages can be mitigated by similar circuits but implemented in a push-pull configuration, however, the push-pull configuration requires twice the number of components and is therefore only applicable in high output power situations where size and cost are less important for lower power circuits. Summary of the Invention

[0036] In view of the above, one object of the present invention is to provide a solution allowing to solve or at least substantially reduce the aforementioned disadvantages of the prior art.

[0037] Another aim is to achieve this within the context of a solution that is simple, compact, rational and as low-cost as possible.

[0038] These and other objects are achieved by the features of the invention as set out in independent claim 1. The dependent claims outline preferred and / or particularly advantageous aspects of the invention.

[0039] Specifically, one embodiment of the present invention provides a converter for converting electric power to an electric load, comprising:

[0040] a first input terminal and a second input terminal, between which a DC voltage or a similar voltage can be applied as a DC voltage,

[0041] a first output terminal and a second output terminal, the first output terminal and the second output terminal being connectable to opposite ends of the electrical load,

[0042] - a first electrical branch adapted to connect the first input terminal to a first intermediate electrical node,

[0043] - a second electrical branch adapted to connect the first intermediate electrical node with the second input terminal,

[0044] - a third electrical branch adapted to connect the first output terminal to a second intermediate electrical node,

[0045] - a fourth electrical branch adapted to connect the second intermediate electrical node with the second output terminal,

[0046] a first active switch (e.g., a MOSFET or other transistor), disposed on the second electrical branch and having a first conductive terminal connected to the first intermediate electrical node, a second conductive terminal connected to the second input terminal, and a control terminal adapted to receive an electrical drive signal for switching the first active switch between a saturation state and a cutoff state, wherein the saturation state allows current to flow between the first conductive terminal and the second conductive terminal, and the cutoff state prevents the current from flowing, and

[0047] a resonant circuit dimensioned to reduce the voltage and / or current applied to the first active switch at least at the instant when the first active switch switches from the cut-off state to the saturation state (preferably also at the instant when the first active switch switches from the saturation state to the cut-off state),

[0048] The resonant circuit at least comprises:

[0049] a first resonant inductor arranged on the first electrical branch and having a first terminal connected to the first input terminal and a second terminal connected to the first intermediate electrical node,

[0050] a second resonant inductor arranged on the third electrical branch and having a first terminal connected to the second intermediate electrical node and a second terminal connected to the first output terminal, and

[0051] - first resonant capacitor,

[0052] The first resonant inductor and the second resonant inductor are inductively coupled together with a reciprocal coupling factor (K) less than 1.

[0053] From a practical perspective, the inductive coupling between the first resonant inductor and the second resonant inductor can be achieved by placing the turns of the first resonant inductor close to the turns of the second resonant inductor, or when the inductors are wound on a magnetic material core, winding the turns of the first resonant inductor and the turns of the second resonant inductor on the same magnetic material core or on a magnetic path that shares a portion of the core.

[0054] In contrast to the capacitor-only implementation outlined in the background, in which there are two physically distinct resonant inductors, in the present approach the two resonant inductors are typically a single object, such as a single core of magnetic material (e.g., in the shape of a toroid, a planar, or another geometric shape) around which the turns of the first and second resonant inductors are suitably wound, to effectively halve the overall size of these two components, which typically represent the largest volume portion of the circuit.

[0055] Compared to conventional circuits with only magnetic coupling (e.g. flyback converters), the converter proposed here instead has the basic advantage of intrinsic zero voltage switching (Zero Voltage Switching) like other capacitive circuits that have been developed in the past, with all the important advantages that come with it, without the need for further quasi-resonant or active clamping circuits, and thus with much lower circuit cost.

[0056] According to one aspect of the present invention, the coupling factor (K) between the first resonant inductor and the second resonant inductor may be between 0.1 and 0.8, preferably between 0.2 and 0.6.

[0057] This low coupling factor is a further advantage over classic transformer-based converters, which require a ks close to 1 and are therefore more difficult to industrialize at low cost, and require various construction expedients in order to simultaneously ensure a high coupling factor and isolation between the primary and secondary circuits, which, on the contrary, is not necessary in a magnetic circuit with a low coupling factor, which is the object of the present invention.

[0058] Thanks to the low coupling factor between the first and second resonant inductors, the mutual inductance that determines their coupling is reduced and does not contribute to the overall resonance of the circuit, thereby allowing the converter to operate correctly. In fact, even from a theoretical perspective, the self-inductance of the first and second resonant inductors is the inductance component that determines the resonance of the circuit together with the resonant capacitance, while the mutual inductance component that is useful for transferring energy from the primary circuit to the secondary circuit via the inductance does not interfere with the circuit resonance.

[0059] According to a possible embodiment of the present invention, the first resonant capacitor may include a first terminal connected to the first intermediate electrical node and a second terminal connected to the second intermediate electrical node.

[0060] Thanks to this solution, a hybrid converter is obtained which allows power to be transferred both through the capacitance of the first resonant capacitor and through the inductance of the mutual inductive coupling between the first resonant inductor and the second resonant inductor.

[0061] This hybrid capacitive / inductive coupling scheme allows for many advantages, particularly in terms of size reduction, peak voltage reduction, and efficiency improvement.

[0062] Regarding overall size, this hybrid coupling scheme allows further reduction of the overall size of the resonant inductor thanks to its reduced magnetic flux relative to analog circuits coupled only through capacitance.

[0063] In fact, in a converter coupled only by capacitance, during the on-period of the active switch, the resonant inductor provided on the primary circuit is charged, accumulating energy in the form of magnetic flux, which is then transferred to the load only through capacitance during the subsequent off-period of the active switch.

[0064] In contrast, by means of inductive coupling between the first resonant inductor located in the primary circuit and the second resonant inductor located in the secondary circuit in the hybrid converter proposed here, a portion of the energy is already transferred to the load during the charging period of the first inductor, resulting in a reduction in the maximum magnetic flux in the resonant inductor with the same power transferred to the load.

[0065] This requires the possibility of making the size of the core of magnetic material smaller than is generally possible, or in any case the possibility of choosing a core with lower magnetic permeability, which generally corresponds to lower losses at higher frequencies, without compromising size and efficiency, as would happen using only capacitive coupling.

[0066] Considering the peak voltage, since energy is also inductively transferred during the on-period of the active switch, the first resonant inductor accumulates less magnetic energy and thus experiences a lower peak current for the same power transferred to the load.

[0067] For the same reason, during the subsequent off-phase of the first switch, the first resonant inductor has less energy to transfer to the resonant capacitor, so that both the resonant capacitor and the active switch have to withstand much lower peak voltages than in conventional solutions.

[0068] This entails the possibility of using components that can tolerate lower voltages and are therefore generally cheaper, faster and more compact.

[0069] Finally, considering the efficiency voltage, the fact that energy is transferred to the load inductively (magnetically) during the on-phase of the active switch and capacitively (electrically) during the off-period allows reducing the reactive current to be equal to the power delivered to the electrical load 110 and thus improves the efficiency of the present circuit relative to an already very efficient circuit coupled only through capacitance.

[0070] The reduction in peak voltage also allows the use of active switches (e.g., MOSFETs) that are closer to ideal and therefore characterized by lower losses (e.g., lower channel resistance allows for reduced static losses, while lower rise and fall times allow for reduced dynamic losses) compared to circuits that are only capacitively coupled.

[0071] According to one aspect of this embodiment, the resonant circuit may include a second resonant capacitor having a first terminal connected to the second output terminal and a second terminal connected to the second input terminal (or connected to a reference voltage to which the second input terminal is also connected).

[0072] Thanks to this approach, despite the galvanic coupling by mixed capacitive and inductive coupling and the advantages outlined above, the resonant capacitor also acts as an isolation capacitor which advantageously separates the converter into a primary circuit and a galvanically separated secondary circuit.

[0073] However, it is not excluded that in other embodiments, the first resonant capacitor may include a first terminal connected to the first intermediate electrical node and a second terminal connected to the second input terminal (or connected to a predetermined reference voltage to which the second input terminal is also connected), so as to be connected in parallel with the second electrical branch provided with the first active switch.

[0074] Alternatively, the first resonant capacitor may comprise a first terminal connected to the second intermediate electrical node and a second terminal connected to the second output terminal (or to a predetermined reference voltage to which the second output terminal is also connected) so as to be connected in parallel with the fourth electrical branch.

[0075] Thanks to these solutions, a purely inductive coupling system is obtained, which, however, also allows the realization of a resonator capable of efficiently transmitting electrical energy to a load with relatively small dimensions.

[0076] A different aspect of the present invention provides that the converter may further comprise a first controller configured to switch the first active switch on and off in a periodic manner, wherein the first controller is particularly configured to measure a voltage applied to the first conductive terminal of the first active switch and to switch the first active switch on when the measured voltage is eliminated (e.g. reaches a value equal to zero).

[0077] This scheme is particularly useful for ensuring high efficiency over a wide range of system operating conditions (such as changes in load or input voltage).

[0078] It should be noted that, in this specification, a controller may be any electrical / electronic device capable of determining and generating appropriate control / drive signals for corresponding active switches. Each controller may also be associated with a driver adapted to amplify the control / drive signals generated by the controller and apply them to the corresponding active switches.

[0079] A different aspect of the invention provides that the fourth electrical branch may comprise a diode having a cathode connected to the second intermediate electrical node and an anode connected to the second output terminal.

[0080] In this way, a rectifier stage can advantageously be realized which can convert the voltage wave generated by the first active switch into a DC voltage or a voltage similar to a DC voltage which is useful for applying an electrical load.

[0081] In other embodiments, instead of a diode ground, the fourth electrical branch may include a second active switch, such as a MOSFET or other transistor, the second active switch having a first conductive terminal connected to the second terminal of the first resonant capacitor, a second conductive terminal connected to the second output terminal, and a control terminal adapted to receive an electrical drive signal to switch the second active switch between a saturation state and a cutoff state, wherein the saturation state allows current to flow between the first conductive terminal and the second conductive terminal, and the cutoff state prevents the current from flowing.

[0082] Thanks to this second approach, the rectification stage can be more efficient overall, achieving for example high efficiency and low cost synchronous rectification, since it is based on a single active switch referenced to a constant potential, for example the ground terminal of the secondary circuit.

[0083] In the context of this second aspect, a preferred aspect of the invention provides that the converter may comprise a second controller configured to switch the second active switch on and off in a periodic manner, wherein the second controller is particularly configured to measure the voltage applied to the first conductive terminal of the second active switch and to switch the second active switch on when the measured voltage is eliminated (e.g. reaches a value equal to zero).

[0084] In this way, it can advantageously be ensured that the switching on of the second active switch also occurs in ZVS mode, reducing losses and thus improving the efficiency of the circuit.

[0085] Furthermore, the fact of having controlled switches based on voltage measurement allows to guarantee high efficiency in a wide range of variations in the operating conditions of the system.

[0086] According to a preferred aspect of the present solution, the second controller may be further configured to maintain the second active switch on for an on-time that is shorter than the duration of the cycle, and to turn off the second active switch at the moment when the first controller turns off the first active switch or with a certain delay (i.e., at a later moment).

[0087] Thanks to this solution, by appropriately adjusting the delay of the switching-off of the second active switch relative to the switching-off of the first active switch, the power delivered to the electric load can be effectively regulated continuously from a maximum value, which would be obtained if the second active switch were switched off at the same moment as the first active switch was switched off, to a value equal to zero, which would be obtained if the second active switch remained switched on for the entire period of time, or if the second switch was never switched off.

[0088] For example, the second controller can be configured to measure the value of a voltage applied to the electrical load (e.g., applied to the first output terminal) and vary the delay in turning off the second active switch so as to minimize (e.g., eliminate) a difference between the measured value and a desired value of the voltage.

[0089] In this way, a contracting control loop is obtained which, starting from a desired value of the voltage to be applied to the electric load, representative of the power required by the load itself, is able to automatically adjust the turn-off delay of the second active switch so that the actual value of the voltage applied to the electric load is approximately equal to the desired value of the voltage.

[0090] The electrical effect associated with this adjustment is that, by increasing the turn-off delay of the second active switch, a gradual increase in the peak value of the voltage applied to the first active switch can be observed, for example, at the first conductive terminal of the first active switch. This increase in peak voltage across the primary circuit stems from the fact that increasing the turn-off delay of the secondary switch reduces the active energy delivered to the load and increases the reactive energy accumulated in the circuit.

[0091] Taking advantage of this effect, another aspect of the present invention provides that the first controller can be configured to measure the peak value of the voltage applied to the first active switch (for example, applied to the first conductive terminal of the first active switch) and change the turn-on time of the first active switch so as to minimize (for example, eliminate) the difference between the measured value and the expected value of the voltage peak.

[0092] Thanks to this solution, it is possible to advantageously regulate the operation of the first active switch according to the needs of the electrical load to be supplied, preventing the peak voltage from reaching excessively high values, and, thanks to the reduction of the on-time of the first active switch, to always operate the second active switch with the minimum possible turn-off delay, therefore in the best efficiency conditions, roughly given by the minimum reactive energy accumulated in the system.

[0093] According to another aspect of the invention, the resonant circuit may comprise a third resonant inductor, preferably having a small inductance value, arranged along an electrical branch connecting the intermediate electrical node to the first terminal of the first resonant capacitor.

[0094] This third resonant inductor is incorporated into the calculation of the circuit's overall resonant frequency, preserving the circuit's overall inductance. In other words, for a given resonant frequency of the circuit, if the inductance of the third resonant inductor increases, the inductance of the first and / or second resonant inductors decreases accordingly, thereby maintaining a roughly constant magnitude.

[0095] The advantage given by the introduction of the third resonant inductor is that further resonant harmonics of the circuit are introduced with a frequency higher than the fundamental frequency, which involves voltage oscillations on the first terminal of the switch (e.g. the drain) during the off-time and current fluctuations in the channel of the switch during the on-time.

[0096] The voltage oscillations on the first terminal are slightly useful in at least slightly increasing the power delivered to the load because, at a higher frequency than the fundamental frequency, they more easily pass through the capacitive barrier acting as a high-pass filter.

[0097] The current fluctuations during the on-time of the switch can instead be very useful because if the period of the current oscillation is properly calculated as a sub-multiple of the on-period of the switch, the instantaneous off-current can be minimized, thus putting the circuit in a near zero-current switching condition during the off phase.

[0098] This effect is particularly useful for minimizing dynamic turn-off losses of the first switch provided on the primary circuit.

[0099] In fact, although the switch provided on the primary circuit is already zero voltage switching (and zero current switching) during the turn-on phase, the turn-off can occur under the condition of maximum current, so the non-zero dynamic loss is proportional to the current peak and the turn-off time.

[0100] On the contrary, thanks to the addition of higher harmonics generating current oscillations and a suitable choice of their frequency, the switch can be advantageously turned off in zero current switching conditions or in any case of strongly reduced current, thus significantly reducing dynamic turn-off losses.

[0101] In another aspect of the present invention, the converter may further include a tank capacitor connected in parallel with the diode or the second active switch.

[0102] Additionally or alternatively, the converter may include a tank capacitor connected in parallel with the first active switch.

[0103] A tank capacitor in parallel with the primary circuit switch or the secondary circuit diode (or switch in the case of synchronous rectification) can help achieve circuit tuning and minimize the voltage derivative at the moment of turning on the primary circuit switch, thereby ensuring lower dynamic turn-on losses due to sudden changes in current.

[0104] Furthermore, these capacitors have a positive effect, since they can make the circuit less sensitive to the construction tolerances of switches and components, in particular to the eddy current capacitances of switches and inductors.

[0105] Additionally, the tank capacitance lowers the resonant frequency of the circuit, which can be advantageous in some circumstances.

[0106] Finally, the tank capacitor can be connected in series with an additional switch only under certain voltage or power conditions (such as under conditions of high input voltage and low power delivered to the load) to simply reduce the power delivered to the load. BRIEF DESCRIPTION OF THE DRAWINGS

[0107] Other characteristics and advantages of the invention will become more apparent on reading the following description, given by way of non-limiting example with the aid of the accompanying drawings.

[0108] Figure 1FIG. 4 is an overall diagram of a power transmission system according to an embodiment of the present invention.

[0109] Figure 2 For use in Figure 1 Electrical diagram of the converter in the system.

[0110] Figure 3 for Figure 2 Variants of the converter.

[0111] Figure 4 To show the function of the drive signal Figure 2 Graph of possible shapes of voltage and current waves that can be obtained at intermediate electrical nodes of a converter.

[0112] Figure 5 According to another embodiment of the present invention, the Figure 1 Electrical diagram of the converter in the system.

[0113] Figure 6 To illustrate the situation where both active switches are turned off at the same time, Figure 5 Graphs showing trends of driving signals of the first active switch and the second active switch and waveforms of voltages applied thereto.

[0114] Figure 7 To illustrate the situation where the second active switch is turned off with a certain delay relative to the first active switch, Figure 5 Graphs showing trends of driving signals of the first active switch and the second active switch and waveforms of voltages applied thereto.

[0115] Figure 8 According to the third variant of the present invention, it can be used Figure 1 Electrical diagram of the converter in the system.

[0116] Figure 9 To show the driving signal Figure 8 Graph of possible shapes of the voltage and current waves that can be obtained at the first intermediate electrical node of the converter.

[0117] Figure 10 According to a fourth embodiment of the present invention, Figure 1 Electrical diagram of the converter in the system.

[0118] Figure 11 According to the fifth variation of the present invention, the Figure 1 Electrical diagram of the converter in the system. DETAILED DESCRIPTION

[0119] One embodiment of the present invention provides a system 100 for transmitting electrical power from a DC voltage source 105 (or at least something similar to a DC voltage) to an electrical load 110 .

[0120] Electrical load 110 (which is generally represented using a resistor symbol) may be any electrical or electronic device that must be powered to allow it to operate and / or to charge the device's own internal battery.

[0121] Typical examples of this type of electrical / electronic equipment are computers, tablets, smartphones, televisions, home appliances, home automation systems, servers, and many other devices.

[0122] In some embodiments, the DC voltage source 105 may be a DC voltage generator or a battery.

[0123] In other embodiments, the DC voltage source 105 may instead include a rectifier 115 adapted to receive an AC voltage from the AC voltage source 120 at an input, convert the AC voltage to a rectified voltage substantially similar to a DC voltage, and provide the DC voltage at an output.

[0124] The AC voltage source 120 may be, for example, a public power distribution network which may be adapted to provide an AC voltage of varying values ​​depending on country or use (e.g., industrial or domestic). By way of example only, the AC voltage source 120 may be a 50Hz-60Hz, 90V-240V AC grid.

[0125] In general, the rectifier 115 may include a first input terminal 125 and a second input terminal 130 that may be connected to the AC voltage source 120 such that the AC voltage source 120 is adapted to apply a voltage difference (AC voltage) between the two terminals that may alternate over time.

[0126] For example, the second input terminal 130 of the rectifier 115 can be connected to a reference voltage and generally referred to as a neutral terminal, and the AC voltage source 120 can be adapted to apply a voltage to the first input terminal 125 (generally referred to as a phase terminal) that varies sinusoidally with time around an average value defined by the reference voltage. It should be noted that the generator 120 can be connected to the terminal 125 and the terminal 130, and the terminals 125 and 130 can be interchanged without affecting the output of the rectifier 115.

[0127] The rectifier 115 may further include a first output terminal 135 and a second output terminal 140, between which a DC voltage difference obtained by converting the AC voltage received at the input is applied, wherein the value of the voltage applied to the first output terminal 135 is generally not lower than the value of the voltage applied to the second output terminal 140.

[0128] For example, the second output terminal 140 may be connected to a reference voltage, and a constant voltage (excluding ripples) having a value not lower than the reference voltage obtained by rectifying the AC input voltage may be applied to the first output terminal 135 .

[0129] The rectifier 115 may be manufactured in the form of a diode bridge (eg, a Graez bridge), but it is not excluded that in other embodiments the rectifier 115 may be a single diode rectifier, a combination of two diodes, a synchronous rectifier, or other devices.

[0130] Optionally, the rectifier 115 may be provided with a filtering circuit, such as a capacitor filter, whose function is to stabilize the voltage difference between the first output terminal 135 and the second output terminal 140 and reduce ripples, thereby maintaining the voltage at a substantially constant value over time.

[0131] The system 100 also comprises a converter (generally indicated at 145 ), ie a circuit adapted to receive at input the voltage supplied by the source 105 , to convert the voltage and to transmit it to the electrical load 110 .

[0132] In general, the converter 145 may include a first input terminal 150 and a second input terminal 155, between which a substantially constant voltage difference is applied, which voltage difference is obtained starting from the voltage supplied by the DC voltage source 105, wherein the value of the voltage applied to the first input terminal 150 is generally higher than the value of the voltage applied to the second input terminal 155.

[0133] For example, the second input terminal 155 may be connected to the second output terminal 140 of the rectifier 115 or to a reference voltage, while the first input terminal 150 may be connected to the first output terminal 135 of the rectifier 115 .

[0134] Optionally, an auxiliary circuit (not shown) can be inserted between the rectifier 115 and the converter 145, which auxiliary circuit is suitable for receiving the voltage supplied by the rectifier 115 at the input and converting it into another voltage, for example, into a voltage of a reduced value, which is more suitable for supplying the converter 145 and / or is useful for other purposes, such as improving the power factor and / or facilitating the control of the system 100.

[0135] Converter 145 may also include first and second output terminals 165 and 170 , which may be electrically connected to opposite ends of electrical load 110 .

[0136] In more detail, the converter 145 may comprise at least one wave generator 175 , ie a circuit supplied by the DC voltage source 105 , eg by the rectifier 115 , to generate a voltage wave, ie a series of voltage pulses following each other at a predetermined time frequency.

[0137] Preferably, the wave generator 175 is adapted to generate a high frequency voltage wave, typically a voltage wave of around hundreds of KHz, MHz, tens of MHz or hundreds of MHz.

[0138] To generate the voltage wave, the wave generator 175 includes at least one active switch 180, such as a transistor (e.g., a bipolar junction transistor (BJT), a field effect transistor (FET), a MOSFET, GaN, SiC, a MESFET, a JFET, an IGBT, etc.), which can be turned on and off (i.e., converted from a cutoff state to a saturation state and vice versa) under the command of an appropriate electrical drive signal.

[0139] More specifically, the active switch 180 may include a first conductive terminal 185 (eg, a drain of an N-type MOSFET), a second conductive terminal 190 (eg, a source of the N-type MOSFET), and a control terminal 195 (eg, a gate of the N-type MOSFET).

[0140] When the active switch 180 is off or in the off state, current cannot flow between the first conductive terminal 185 and the second conductive terminal 190 .

[0141] Conversely, when the active switch 180 is on, ie, when it is in a saturated state, current flows freely between the first conductive terminal 185 and the second conductive terminal 190 .

[0142] Switching of the active switch 180 between these two states is controlled by an electrical drive signal applied to the control terminal 195 .

[0143] In practice, when the voltage of the electrical drive signal is greater than or equal to a certain threshold, the active switch 180 is in a saturated state (switched on and able to conduct current).

[0144] On the other hand, when the voltage of the electrical drive signal is lower than the threshold value, the active switch 180 is in an OFF state (off).

[0145] In order to generate the voltage wave, the electrical drive signal may be a periodic signal varying between a minimum voltage value (possibly zero) below the threshold of the active switch 180 and a maximum value above said threshold.

[0146] For example, the electrical drive signal may be a square wave signal.

[0147] The frequency of the electric drive signal, which actually corresponds to the switching frequency of the active switch 180 and thus to the frequency of the generated voltage wave, is preferably selected to have a relatively high value, for example around hundreds of KHz, MHz, tens of MHz or hundreds of MHz.

[0148] During each cycle of the drive signal, the time period during which the electrical drive signal is greater than the turn-on threshold of the active switch 180 may be referred to as the on-time, and the time range during which the electrical drive signal is less than the turn-on threshold of the active switch 180 may be referred to as the off-time.

[0149] When the electrical drive signal, which transitions from a minimum voltage value to a maximum voltage value, exceeds a threshold value, the active switch 180 is said to be turned on or switched on.

[0150] On the other hand, when the electrical drive signal transitioning from the maximum voltage value to the minimum voltage value falls below a threshold value, the active switch 180 is said to be turned off or switched off.

[0151] The electrical drive signal for the active switch 180 may be provided by a dedicated controller (which is Figure 2 500 in the figure), the dedicated controller may be suitably connected to the control terminals 195 of the active switch 180 by any system capable of transmitting electrical signals (also wireless).

[0152] The controller 500 may be any electrical / electronic device capable of generating an electrical drive signal and possibly varying one or more of its characteristics (eg, on and off times, keeping the period constant, or varying the period).

[0153] A corresponding driver (not shown) may be associated with the controller 500 , ie, an electrical / electronic device adapted to receive the drive signal generated by the controller 500 , to appropriately amplify the drive signal and ultimately apply it to the active switch 180 .

[0154] According to a preferred aspect of the present solution, the controller 500 can be configured, for example, to measure the voltage applied to the first conductive terminal 185 of the first active switch 180 (e.g., applied to the MOSFET drain) and to turn on the active switch 180 when the measured voltage is eliminated (i.e., drops to a value equal to zero).

[0155] After the switching-on step, the controller 500 may maintain the first active switch on for a certain on-time, which may be varied to change the energy of the system.

[0156] For example, the on-time may be adjusted according to the peak voltage on the first conductive terminal 185 to ensure minimum dead energy of the system.

[0157] At the end of the on-time, the controller 500 may be configured to turn off the first active switch 180 and keep it off until the voltage is removed again.

[0158] In this manner, the off time remains substantially constant, or varies only slightly when there are small changes in the waveform of the voltage at the first conductive terminal 185 (eg, due to different levels of power).

[0159] In addition to including the active switch 180 , the wave generator 175 also includes a resonant circuit 200 , such as a fully resonant or nearly resonant reactive circuit.

[0160] The resonant circuit 200 is generally a circuit that includes one or more reactances (eg, one or more capacitors and / or inductors) that are suitably connected together and tuned to resonate at a given frequency.

[0161] The tuning of the resonant circuit 200 mainly consists in determining the magnitude of the aforementioned reactances in terms of capacitance and inductance, respectively.

[0162] In this case, the resonant circuit 200 is connected to the active switch 180 and is tuned to reduce the voltage and / or power applied to the active switch 180 during each switching phase from OFF to ON (preferably also during each switching phase from ON to OFF).

[0163] Preferably, the resonant circuit 200 is tuned so that during each switching phase of the active switch 180, the voltage and / or current applied to the active switch 180 is reduced to a value equal to zero or substantially equal to zero, thereby obtaining the wave generator 175 operating in zero voltage switching (ZVS) and / or zero current switching (ZCS) mode.

[0164] For example, the resonant circuit 200 may be tuned to resonate at a frequency that is equal to or close to the driving frequency of the active switch 180 .

[0165] In this way, the electrical losses during the switching cycles of the active switch 180 are significantly reduced, making it possible to increase the frequency of these cycles and therefore the frequency of the voltage waves generated by them, with the result that the power transmitted with the same applied voltage can be increased, or the applied voltage can be reduced with the same power transmitted.

[0166] Based on these general considerations, one possible embodiment of the converter 145 is Figure 2 As shown in the figure.

[0167] In this embodiment, converter 145 includes a primary circuit and a secondary circuit.

[0168] The primary circuit includes a first electrical branch 201 and a second electrical branch 203, wherein the first electrical branch 201 extends from the first input terminal 150 to a first intermediate electrical node 202, and the second electrical branch 203 extends from the intermediate electrical node 202 to the second input terminal 155 or to a predetermined primary circuit reference voltage 204, which can also be connected to the second input terminal 155.

[0169] The secondary circuit includes a third electrical branch 205 and a fourth electrical branch 207, wherein the third electrical branch 205 extends from the first output terminal 165 to a second intermediate electrical node 206, and the fourth electrical branch 207 extends from the intermediate electrical node 206 to the second output terminal 170 or to a predetermined secondary circuit reference voltage 208, which can also be connected to the second output terminal 170.

[0170] The active switch 180 is arranged on the second electrical branch 203, for example, such that the first conductive terminal 185 of the active switch 180 (for example, the drain of the N-type MOSFET) is connected to the intermediate electrical node 202, and the second conductive terminal 190 of the active switch 180 (for example, the source of the N-type MOSFET) is connected to the second input terminal 155 or, simply, to the reference voltage 204.

[0171] The resonant circuit 200 includes a first resonant inductor 215 arranged on the first electrical branch 201 and comprising a first terminal 216 electrically connected to the first input terminal 150 and an opposing second terminal 217 electrically connected to the intermediate electrical node 202 .

[0172] The resonant circuit 200 further comprises a first resonant capacitor 220 having a first electrical terminal 221 and a second electrical terminal 222 , each of which is connected to a corresponding armature of the first resonant capacitor 220 .

[0173] The first terminal 221 may be connected to the first intermediate electrical node 202 via an electrical branch 225 .

[0174] The second terminal 222 may be connected to the second intermediate electrical node 206 via an electrical branch 230 .

[0175] A second resonant inductor 235 may be provided on the third electrical branch 205 , the inductor including a first terminal 236 electrically connected to the second intermediate electrical node 206 and a second opposite terminal 237 electrically connected to the first output terminal 165 of the converter 145 .

[0176] The resonant circuit 200 further includes a second resonant capacitor 240 having a first electrical terminal 241 and a second electrical terminal 242 . The first electrical terminal 240 and the second electrical terminal 242 are each connected to a corresponding armature of the second resonant capacitor 240 .

[0177] The first electrical terminal 241 of the second resonant capacitor 240 may be electrically connected to the second output terminal 170 of the converter 145 via an electrical branch 245 .

[0178] The second electrical terminal 242 may be connected to the second input terminal 155 via another electrical branch, or the second electrical terminal 242 may be connected to the primary circuit reference voltage 204 .

[0179] In this way, the two resonant capacitors 220 and 240 also act as isolation capacitors galvanically separating a primary circuit, which includes at least the active switch 180 and the first resonant inductor 215 , from a secondary circuit, which includes at least the second resonant inductor 235 and the electrical load 110 .

[0180] However, if Figure 3 As shown in , it is also possible to realize a converter 145 that is not galvanically isolated, which is completely similar to the above-described converter, except that it does not have a second resonant capacitor 240 and it has an electrical branch 245 that directly connects the second output terminal 170 to the second input terminal 155 or to the primary circuit reference voltage 204.

[0181] Power may be transferred from the primary circuit to the secondary circuit thanks to the capacitive coupling achieved via the first resonant capacitor 220 and possibly via the second resonant capacitor 240 (if present).

[0182] Specifically, during each on-time of the active switch 180, the first resonant inductor 215 provided on the primary circuit is charged, accumulating energy in the form of magnetic flux, and during the subsequent off-time of the active switch 180, this energy is transferred to the electric load 110 via capacitance or capacitive electrical coupling implemented by the first resonant capacitor 220 or possibly via the second resonant capacitor 240 (if present).

[0183] As anticipated above, the first resonant inductor 215, the second resonant inductor 235, the first resonant capacitor 220 and, if present, the possible second resonant capacitor 240 are sized (tuned) such that a resonator (resonant circuit 200) is achieved that reduces (preferably eliminates) the voltage and / or current applied to the active switch 180 during each single switching phase from the cut-off state (off) to the saturation state (on) (preferably also from the saturation state (on) to the cut-off state (off)).

[0184] In this way, at the intermediate electrical node 202 , a voltage is advantageously available which, depending on the electrical drive signal of the active switch 180 , can vary over time according to a waveform capable of ensuring a ZVS and / or ZCS transition of the active switch 180 itself.

[0185] According to the electric driving signal V of the active switch 180 G ,exist Figure 4 The voltage V at the intermediate electrical node 202 is shown in FIG. D and the current I flowing through the active switch 180 D Possible trends over time.

[0186] In fact, these waveforms are, in any case, identical or similar to those obtainable with the aid of a wave generator based on a circuit structure similar to a class E or class F amplifier or on any other ZVS and / or ZCS resonant amplifier.

[0187] By obtaining the same waveform or by generating the same voltage wave, the above-mentioned wave generator 175 obtains the same advantages as the aforementioned resonant wave generator, in particular in terms of the reduction of electrical losses during the switching phases of the active switches and thus the increase of the operating frequency.

[0188] However, compared to these wave generators, the wave generator 175 has the significant advantage of not requiring a bulky choke reactor, which allows for significant reductions in size and cost.

[0189] According to an important aspect of the present invention, the first resonant inductor 215 and the second resonant inductor 235 are inductively coupled to each other with a reciprocal coupling factor K that is lower than unity (ie, less than 1).

[0190] From a practical perspective, such inductive coupling can be achieved by placing the turns of the first resonant inductor 215 closer to the turns of the second resonant inductor 235, or, in the case where the inductors are wound on a magnetic material core, winding the turns of the second resonant inductor 235 directly on the magnetic path of the magnetic material core on which the coil of the first resonant inductor 215 is also wound or on a portion of the shared core.

[0191] Thanks to this solution, the two resonant inductors 215 and 235 can be made in the form of a single component, for example a single core of magnetic material (for example in the shape of a toroid, a plane or another geometric shape) on which the turns of the first resonant inductor 215 and the second resonant inductor 235 are appropriately wound, so as to practically halve the overall size of these two components, which usually represent the largest volume portion of the global circuit.

[0192] Furthermore, by means of inductive coupling between the first resonant inductor 215 and the second resonant inductor 235 , electrical energy is transferred to the electrical load 110 not only during the OFF phase (time) but also during the ON phase (time) of the active switch 180 .

[0193] In fact, when the active switch 180 is turned on (saturation state), thanks to the inductive coupling with the second resonant inductor 235, the first resonant inductor 215, in addition to being charged and accumulating energy in the form of magnetic flux, directly and simultaneously transfers at least a portion of said energy to the secondary circuit, i.e. to the electrical load 110.

[0194] In this way, the magnetic flux is overall lower than for a similar circuit coupled only by capacitance, which allows the magnetic material core to be of smaller than normal size or in any case to choose a core with lower magnetic permeability, which generally corresponds to lower losses than a core with higher frequencies, without compromising space and efficiency.

[0195] Furthermore, since energy is also inductively transferred from the primary circuit to the secondary circuit during the on-period of the active switch 180 , the first resonant inductor 215 accumulates less magnetic energy and is thus subject to a lower peak current for the same power transferred to the electrical load 110 .

[0196] For the same reason, during the subsequent off-time of the active switch 180, the first resonant inductor 215 has less energy to transfer across the resonant / isolating capacitors 220 and 240, so that the resonant / isolating capacitors 220 and 240 and the active switch 180 must withstand much lower peak voltages than a solution coupled only through capacitance.

[0197] This entails the possibility of using components that can tolerate lower voltages and are therefore generally cheaper, faster and more compact.

[0198] Furthermore, the reduction in peak voltage allows for the use of active switches 180 (e.g., MOSFETs) that are closer to ideal and thus characterized by lower losses (e.g., lower channel resistance allows for reduced static losses, while lower rise and fall times allow for reduced dynamic losses).

[0199] Finally, the fact that energy is transferred to the electrical load 110 inductively (magnetically) during the on-period of the active switch 180 and capacitively (electrically) during the off-period allows reducing the reactive current to be equal to the power transferred to the electrical load 110 and thus improving the overall efficiency of the converter 145 relative to an already very efficient converter coupled only through capacitance.

[0200] All of these advantages are achieved while remaining within an intrinsic zero voltage switching circuit, which allows the converter 145 to be characterized by much lower losses and overall size compared to conventional circuits that are only magnetically coupled (such as a classic flyback converter).

[0201] In order for the converter 145 to operate correctly, it is preferable in any case that the mutual inductance between the first resonant inductor 215 and the second resonant inductor 235 (which is the inductive component used to inductively transfer energy from the primary circuit to the secondary circuit) does not interfere with the resonance of the circuit.

[0202] Therefore, unlike conventional transformer-based isolation converters (e.g., flyback), the coupling factor k between the first resonant inductor 215 and the second resonant inductor 235 must be less than 1, for example, between 0.1 and 0.8, and even more preferably between 0.2 and 0.6.

[0203] Reference Figure 2 and Figure 3 , the converter 145 may further include a diode 255 disposed on the fourth electrical branch 207 and generally configured to allow current to flow along the fourth electrical branch 207 in only one direction.

[0204] The diode 255 may have its own cathode connected to the second intermediate electrical node 206 and its own cathode connected to the second output terminal 170 .

[0205] When the voltage between the anode and the cathode exceeds the threshold voltage of the diode, the diode 255 is turned on, the diode enters the on-state and allows current to flow between the anode terminal and the cathode terminal.

[0206] When the current decreases to zero, the diode turns off.

[0207] In this manner, diode 255 implements a universal rectification stage that is effectively capable of converting the voltage wave from the primary circuit into a DC voltage similar to the DC voltage applied to electrical load 110 .

[0208] Possibly (although not necessarily), the rectification stage of the converter 145 may also include a capacitor 260, which is arranged on an electrical branch 265, which extends from an intermediate node 270 of the third electrical branch 205 between the second resonant inductor 235 and the first output terminal 165 to an intermediate node 275 of the electrical branch 245, so as to be connected in parallel with the electrical load 110.

[0209] like Figure 5 In other more complex but more energy efficient embodiments, at least for low current values, the diode 255 may be replaced by a second active switch 300 (eg, a MOSFET or other transistor) driven to implement, for example, synchronous rectification, as shown in FIG.

[0210] Specifically, the active switch 300 may include a first conductive terminal 256 connected to the second intermediate electrical node 206 (e.g., the drain of the N-type MOSFET), a second conductive terminal 257 connected to the second output terminal 170 (e.g., the source of the N-type MOSFET), and a control terminal 305 (e.g., the gate of the N-type MOSFET).

[0211] As an example, Figure 5 The converter 145 is shown as Figure 2 galvanically isolated as in the case of Figure 3 non-isolated situation.

[0212] When the active switch 300 is off or in the off state, current cannot flow between the first conductive terminal 256 and the second conductive terminal 257 .

[0213] Conversely, when the active switch 300 is on, ie, when it is in a saturated state, current flows freely between the first conductive terminal 256 and the second conductive terminal 257 .

[0214] Switching of the active switch 300 between these two states is controlled by an electrical drive signal applied to the control terminal 305 .

[0215] In practice, when the voltage of the electrical drive signal is greater than or equal to a certain threshold, the active switch 300 is in a saturated state (switched on and able to conduct current).

[0216] On the other hand, when the voltage of the electrical drive signal is lower than the threshold value, the active switch 300 is in the OFF state (off).

[0217] The electrical drive signal may be a periodic signal varying between a minimum voltage value (possibly zero) below the threshold of the active switch 300 and a maximum value above said threshold.

[0218] For example, the electrical drive signal may be a square wave signal.

[0219] During each cycle of the drive signal, the time period during which the electrical drive signal is greater than the turn-on threshold of the active switch 300 may be referred to as the on-time, and the time range during which the electrical drive signal is less than the turn-on threshold of the active switch 300 may be referred to as the off-time.

[0220] When the electric drive signal, which transitions from a minimum voltage value to a maximum voltage value, exceeds a threshold value, the active switch 300 is said to be turned on or switched on.

[0221] On the other hand, when the electrical drive signal transitioning from the maximum voltage value to the minimum voltage value falls below a threshold value, the active switch 300 is said to be turned off or switched off.

[0222] The electrical drive signal for the active switch 300 may be provided by a dedicated controller (which is Figure 5 The dedicated controller may be suitably connected to the control terminals 305 of the active switch 300 by any system (also wireless) capable of transmitting electrical signals.

[0223] The controller 600 may be any electrical / electronic device capable of generating an electrical drive signal and possibly varying one or more of its characteristics (eg, on and off times, keeping the period constant, or varying the period).

[0224] A corresponding driver (not shown) may be associated with the controller 600 , ie, an electrical / electronic device adapted to receive the drive signal generated by the controller 600 , to appropriately amplify the drive signal and ultimately apply it to the active switch 300 .

[0225] Active switch 300 may be driven by controller 600 to emulate an ideal diode.

[0226] In practice, the controller 600 may be configured to generate a calibrated drive signal to turn on the active switch 300 when the secondary circuit voltage (i.e., the voltage at the first terminal 256 (e.g., on the MOSFET drain)) drops below 0 V, and to turn off the active switch 300 substantially synchronously with the turning off of the active switch 180, or in any case when the current is substantially zero, to maximize the power delivered to the load 110.

[0227] Figure 6 The voltage V on the first conductive terminal 185 of the active switch 180 (eg, the drain of the MOSFET) is D The waveform and the relative drive signal V applied to the control terminal 195 G The waveform of the active switch 300 is related to the voltage V on the first conductive terminal 256 (eg, the drain of the MOSFET). D The waveform and the relative drive signal V applied to the control terminal 305G The waveforms were compared.

[0228] It can be seen that in the circuit under discussion the voltage V D The waveform of the circuit can be different from that of the classic E, F or similar circuits because of the voltage waveform V D The first part, in which the voltage V D Not null value, where the voltage V on the primary circuit D has a direct derivative, and the voltage V on the secondary circuit D At the moment of elimination, the voltage V D There is a discontinuity in the derivative of .

[0229] Generally speaking, using active rectifier switches 300 instead of diodes allows operation at higher frequencies (eg, MHz, tens of MHz, or hundreds of MHz) and generally allows for reduced static losses.

[0230] If we consider the example of using N-type or GaN-type MOSFET transistors, it is actually possible to have a low channel resistance, which allows to limit the losses compared to diodes.

[0231] Another important advantage of using the active switch 300 is the greater degree of control freedom afforded by the possibility of switching off the active switch 300 with a certain delay relative to the moment at which maximum power transmission is guaranteed (i.e. with a certain delay relative to the moment of switching off the active switch 180 provided on the primary circuit).

[0232] like Figure 7 As shown in FIG, by delaying the turn-off time of the active switch 300 relative to the active switch 180, a lower voltage V is obtained than in the previous case. D This translates into less power being delivered to the load.

[0233] In the extreme case where the active switch 300 is always on, ie, the on-time is equal to the driving signal period (eg, 100% duty cycle), no energy will be transferred to the load, and only reactive current will exist in the circuit.

[0234] As a result, between the maximum energy transfer condition to the load 110 (active switches 180 and 300 are both off simultaneously) and the condition where no power is transferred to the load (active switch 300 is always on), there are infinite possible points for turning on the active switch 300 that can be selected to continuously regulate the power transferred to the load.

[0235] In fact, if the turn-off delay of the active switch 300 is zero, both active switches 180 and 300 are turned off simultaneously, and the energy transferred to the electrical load 110 is maximum.

[0236] By increasing the turn-off delay of the active switch 300 , the energy transferred to the electrical load is gradually reduced.

[0237] The minimum value of the transferred energy is equal to zero and is reached when the delay is so high that the on-time of the active switch 300 is equal to the driving signal period (100% duty cycle), or if the active switch is always kept on.

[0238] In this way, the turn-off timing of the active switch 300 becomes a degree of freedom that can be effectively utilized for controlling the voltage, current, or power on the load 110 .

[0239] For example, the turn-off delay of active switch 300 relative to active switch 180 may advantageously be fed back to adjust the output voltage in a simple, fast and reliable manner without requiring direct feedback between the primary and secondary circuits, thus at low cost.

[0240] In practice, the controller 600 may be configured to measure the value of a voltage applied to the electrical load (e.g., applied to the first output terminal 165) and vary the delay in turning off the active switch 300 so as to minimize (e.g., eliminate) the difference between the measured value and the desired value of the voltage.

[0241] The true value of the voltage applied to the first output terminal 165 may be measured by the controller 600 through a simple electrical connection, whereas the desired value to be applied to the load may be information directly supplied to the controller 600 by the electrical load 100 itself.

[0242] In this way, a feedback control loop on the secondary circuit can be implemented that is configured such that when less power is required at the electrical load 110 , the second controller 600 delays turning off the active switch 300 , thereby reducing the voltage across the electrical load 110 and the power delivered to the electrical load 110 .

[0243] from Figure 7 A second electrical effect that can be inferred is that, with the same on-time of the active switch 180 (and therefore with the average energy in the circuit), if there is a delayed turn-on of the active switch 300, a peak voltage V D increase.

[0244] This second electrical effect can be clearly explained by the fact that, by reducing the current on the electrical load 110 , the average reactive energy in the circuit increases, which translates into a higher peak voltage on the primary circuit.

[0245] This second electrical effect can be effectively utilized to transmit information about the lower power requirements of the electrical load 110 to the primary circuit without having to use an additional communication circuit between the primary circuit and the secondary circuit (for example, an optical isolator or a digital or analog capacitive isolator suitable for transmitting information between the primary circuit and the secondary circuit).

[0246] In fact, if the controller 600 increases the delay in turning off the active switch 600 to handle the lower power demand of the electrical load 110, the primary circuit will see a peak voltage V D corresponding increase.

[0247] The peak voltage V D It can be used as a feedback signal to adjust the on-time of the active switch 180 .

[0248] In practice, the first controller 500 can be configured to measure the peak value of the voltage applied to the active switch 180 (e.g., applied to the first conductive terminal 185 of the first active switch 180 itself) and change the on-time of the first active switch 180 so as to minimize (e.g., eliminate) the difference between the measured value and the expected value of the peak value of the voltage.

[0249] The voltage V can be easily measured by the controller 500, for example using a peak detector (eg, a diode and a capacitor). D The expected value of the voltage peak can be a design parameter.

[0250] In this way, by varying the on-time of the active switch 180, it is possible to maintain the peak voltage V D Constant additional feedback control loop.

[0251] Since the voltage V D will tend to increase, so if the load 110 draws less real power, this further feedback loop will tend to reduce the on-time of the active switch 180 .

[0252] The control system eliminates the need for opto-isolators or other information transmission systems between the primary and secondary circuits, which are typically bulky (due to the minimum isolation distance required for certification purposes), costly, and unreliable, and ensures high dynamics (thanks to the feedback loop on the secondary circuit) and high efficiency (thanks to the feedback loop on the primary circuit) of the controller.

[0253] In fact, the feedback loop on the primary circuit always maintains the minimum energy required to supply the electrical load 110, so that the secondary controller 600 operates with the minimum possible delay (and thus the maximum possible ratio between active and reactive energy on the load).

[0254] Another advantage of this regulation system is that it allows light load conditions to be managed more efficiently than conventional techniques (e.g. based on pulse trains) and with fewer radiated and conducted emission problems (as conventional techniques typically introduce low frequency harmonics that are more difficult to filter with compact filters).

[0255] It should be noted that this type of control system is generally variable frequency because the off-time of the primary switch remains approximately constant (except for moderate variations caused by changes in the shape of the operating drain voltage depending on the circuit energy), while the on-time of the primary switch varies significantly depending on the energy required by the load.

[0256] As described above, the resonant circuit 200 of the above-mentioned converter 145 can ensure useful resonance for approximately ZVS and ZCS transitions during the turn-on phase, i.e., during the switching of the active switch 180 from the cut-off state (off) to the saturation state (on), and also during the turn-off phase, i.e., during the switching of the active switch 180 from the saturation state (on) to the turn-off state (on) again, to ensure useful resonance for approximately ZVS transitions.

[0257] In particular, ZVS and ZCS conditions can be ensured by simple tuning of the circuit, or more appropriately, ZVS and ZCS conditions can be implemented by measuring the voltage and / or current at the first switch-on terminal (e.g., at the MOSFET drain), which is beneficial for dynamically changing the turn-on and / or turn-off moments of switches 180 and 300.

[0258] An active control system for the turn-on timing of switches 180 and 300, particularly based on detecting the moment when the drain voltage is removed, is particularly useful for ensuring high efficiency over a wide range of operating conditions of the system (e.g., changes in load or input voltage).

[0259] In short, the off time is slightly variable depending on the operating conditions of the system, and the turn-on timing is preferably actively controlled by appropriate circuitry that detects the moment when the drain voltage of switches 180 and 300 is removed.

[0260] The on-time of the first switch 180 , counted from the switch-on moment, is preferably controlled on the primary circuit according to the drain peak voltage of the first switch 180 itself, to ensure minimum reactive energy in the system.

[0261] Conversely, the on-time of the second switch 300 is preferably calculated to ensure a delay relative to the moment the first switch 180 turns off, thereby effectively regulating the output voltage across the electrical load 110 .

[0262] The operation of the circuit can be characterized by having a current I in the active switch 180 D, these currents ID reach their maximum values ​​precisely at the instant when the same active switch 180 turns off.

[0263] In transitioning between on and off, the change in resistance of the active switch 180 occurs in a finite time.

[0264] Considering the case of a fast active switch (eg, an N-type MOSFET), a properly driven switch transitions from on to off in a few ns, tens of ns, or hundreds of ns.

[0265] During this finite transient time, the resistance of the active switch 180 gradually increases while the current decreases proportionally, causing a dissipation peak that cannot be mitigated by the ZVS transition condition.

[0266] In order to reduce this dissipation peak, the converter 145 can be modified, such as Figure 9 , namely by introducing a third resonant inductor 280 , preferably having a small inductance value, arranged along an electrical branch 225 connecting the intermediate electrical node 202 to the first terminal 221 of the first resonant capacitor 220 .

[0267] In this manner, the third resonant inductor 280 is arranged in series with the first resonant capacitor 220 in the primary circuit.

[0268] The third resonant inductor 280 is included in the calculation of the circuit's overall resonant frequency and maintains the circuit's overall inductance. In other words, for a given resonant frequency of the circuit, if the inductance of the third resonant inductor 280 increases, the inductance of the first resonant inductor 215 and / or the second resonant inductor 235 decreases accordingly, thereby maintaining a substantially constant magnitude.

[0269] The advantage given by the third resonant inductor 280 is the introduction of additional resonant harmonics of the circuit having a higher frequency than the fundamental frequency, which involves voltage oscillations V at the intermediate electrical node 202 and therefore on the first terminal 185 (e.g. MOSFET drain) of the active switch 180 during the off-time. D .

[0270] The additional harmonics also include the current I flowing through the active switch 180 during the on-time D Oscillations in .

[0271] Depends on the driving signal V G The voltage V D and current I D The possible trend is Figure 9 As shown in the figure.

[0272] Voltage V DThe oscillations of are useful for at least slightly increasing the power delivered to the electrical load 110 because, at frequencies higher than the fundamental frequency, they more easily pass through the capacitive barrier acting as a high-pass filter.

[0273] Conversely, during the on-time of the active switch 180, the current I D The oscillation is very useful to reduce dynamic losses during the turn-on phase.

[0274] In fact, if the third resonant inductor 280 is selected so that the current I D If the oscillation period of φ is a sub-multiple of the on-period of the switch 180 , the instantaneous off-current can be minimized, bringing the circuit into a condition close to zero-current switching or in any case with a strongly reduced current.

[0275] It should be observed that although Figure 8 The third resonant inductor 280 is shown for the case where the converter 145 is galvanically isolated and where the rectification stage is obtained by means of the second active switch 300, but it is not excluded that the third resonant inductor 280 can also be used for Figure 3 non-isolated case and / or case of rectification via simple diodes.

[0276] exist Figure 10 The middle figure shows a further extension of the above circuit, providing that the converter 145 may further include a first tank capacitance 285 connected in parallel with the active switch 180, the first tank capacitance 285 having, for example, a first terminal 286 connected to the first intermediate electrical node 202 and a second terminal 187 connected to the second input terminal 155 or to the primary circuit reference voltage 204.

[0277] Additionally or alternatively, the converter 145 may include a second tank capacitor 290 connected in parallel with the third electrical branch 207 or with the diode 255 or with the second active switch 300, the second tank capacitor 290 having, for example, a first terminal 291 connected to the second intermediate electrical node 206 and a second terminal 292 connected to the second output terminal 170.

[0278] Tank capacitors 285 and / or 290 may help achieve circuit tuning and minimize voltage derivatives at the moment of turning on active switch 180, thereby ensuring lower dynamic turn-on losses associated with abrupt changes in current.

[0279] Furthermore, these tank capacitors 285 and / or 290 have a positive effect, since they can make the circuit less sensitive to tolerances of switches and components, in particular to eddy current capacitances of switches and inductors.

[0280] Additionally, tank capacitors 285 and / or 290 lower the resonant frequency of the circuit.

[0281] Finally, tank capacitors 285 and / or 290 may be connected in series with an additional switch only under certain voltage or power conditions (eg, under conditions of high input voltage and low power delivered to the load) to simply reduce the power delivered to the load.

[0282] Furthermore, in this case, although Figure 10 The tank capacitors 285 and 290 are shown for the case where the converter 145 is galvanically isolated and where the rectification stage is obtained by means of the second active switch 300, but it is not excluded that the tank capacitors 285 and / or 290 can also be used for Figure 3 non-isolated case and / or case of rectification via simple diodes.

[0283] All of the above-described variations of converter 145 can be performed in a multi-phase configuration without significant conceptual changes in order to increase the power delivered to electrical load 110 .

[0284] In practice, another wave generator circuit substantially similar to wave generator 175 may be connected to the second terminal 242 of the second resonant capacitor 240 .

[0285] The active switches 180 of the two wave generators 175 can be controlled by corresponding drive signals so as to be turned on and off at the same frequency but not in phase with each other, that is, so that these active switches 180 are not always turned on or off at the same time, but there is always at least a small amount of time in each operating cycle when an active switch 180 is on while the other active switch 180 is off, and vice versa.

[0286] In this way, by appropriately adjusting the time span, ie the phase shift, between the drive signals of the two active switches 180 , it is advantageously possible to increase the power delivered to the electrical load 110 while using the same supply voltage.

[0287] If the power level to be delivered to the electrical load 110 is particularly high, the two active switches 180 can also be controlled with anti-phase drive signals therebetween, so that when one active switch 180 is turned on, the other active switch 180 is always turned off, and vice versa, thereby obtaining a push-pull operation.

[0288] Finally, it should be pointed out that in all the shown embodiments, the two resonant capacitors 220 and 240 may be discrete capacitors, ie inseparable components comprising a first terminal connected to the primary circuit and a second terminal connected to the secondary circuit.

[0289] In this manner, the system 100 may be fabricated into a unique and inseparable device, such as an isolated power supply.

[0290] However, in other embodiments, each of the resonant capacitors 220 and 240 may be formed by a pair of separable armatures, wherein one transmitting armature is connected to the primary circuit and one receiving armature is connected to the secondary circuit.

[0291] Meanwhile, the resonant inductors 215 and 235 inductively coupled together may be constituted by a coil (eg, antenna) for wireless transmission and a coil (eg, antenna) for wireless reception of power, respectively.

[0292] In this way, the primary circuit of the converter 145 can be installed in a first device, and the secondary circuit can be installed on a second device, which is physically separate and movable (removable) relative to the first device, thereby forming a wireless hybrid capacitive and inductive power transfer system.

[0293] For example, the first device may be configured as a recharging base, while the second device may be a device to be recharged or powered, such as a smartphone, a laptop computer, a television, or the like.

[0294] In this way, by appropriately bringing the second device closer to the first device, each transmitting armature can be brought closer and facing toward the corresponding receiving armature, thereby reconstructing the resonant capacitors 220 and 240, and bringing the resonant inductors 215 and 235 closer together, thereby reconstructing the inductive coupling.

[0295] Figure 11 The middle figure shows another embodiment of this solution.

[0296] This embodiment and Figure 10 The only difference from the embodiment shown in is that the resonant capacitors 220 and 240 have been eliminated, and accordingly the entire connecting branch between the first intermediate electrical node 202 and the second intermediate electrical node 206 has been eliminated, and that the first tank capacitor 285 and / or the tank capacitor 290 are also used as resonant capacitors.

[0297] In this manner, power transfer between the primary circuit and the secondary circuit is performed only inductively through mutual inductive coupling between the first resonant inductor 215 and the second resonant inductor 235 .

[0298] While this topology is similar to a conventional transformer-based converter, it actually differs from known techniques due to the low value of the coupling factor K between the first resonant inductor 215 and the second resonant inductor 235 , and due to the presence of the resonant / tank capacitors 285 and / or 290 .

[0299] It has been found that the converter 145 according to this last embodiment, although transferring less energy, operates equivalently using waveforms similar to those obtainable by the previous circuits and achieves high levels of efficiency and compact size.

[0300] Of course, this embodiment does not necessarily require the presence of both resonant / tank capacitors 285 and 290, as the presence of only one of them is sufficient.

[0301] It is obvious that several technical application modifications may be made by an expert in the field without departing from the scope of the claimed invention.

Claims

1. A converter (145) for transmitting electric power to an electric load (110), comprising: a first input terminal (150) and a second input terminal (155), between which a DC voltage can be applied or a voltage similar to the DC voltage can be applied, a first output terminal (165) and a second output terminal (170), the first output terminal (165) and the second output terminal (170) being connectable to opposite ends of the electrical load (110), - a first electrical branch (201) adapted to connect said first input terminal (150) to a first intermediate electrical node (202), - a second electrical branch (203) adapted to connect the first intermediate electrical node (202) with the second input terminal (155), - a third electrical branch (205) adapted to connect the first output terminal (165) to a second intermediate electrical node (206), - a fourth electrical branch (207) adapted to connect the second intermediate electrical node (206) with the second output terminal (170), a first active switch (180) disposed on the second electrical branch (203) and having a first conductive terminal (185) connected to the first intermediate electrical node (202), a second conductive terminal (190) connected to the second input terminal (155), and a control terminal (195) adapted to receive an electrical drive signal to switch the first active switch (180) between a saturation state and a cut-off state, wherein the saturation state allows a current to flow between the first conductive terminal (185) and the second conductive terminal (190), and the cut-off state prevents the current from flowing, and a resonant circuit (200) dimensioned to reduce a voltage and / or a current applied to the first active switch (180) at least at a time when the first active switch (180) switches from the cut-off state to the saturation state, Wherein, the resonant circuit (200) comprises at least: a first resonant inductor (215) arranged on the first electrical branch (201) and having a first terminal (216) connected to the first input terminal (150) and a second terminal (217) connected to the first intermediate electrical node (202), a second resonant inductor (235) arranged on the third electrical branch (205) and having a first terminal (236) connected to the second intermediate electrical node and a second terminal (237) connected to the first output terminal (165), and - a first resonant capacitor (220) comprising a first terminal (221) connected to the first intermediate electrical node (202) and a second terminal (222) connected to the second intermediate electrical node (206), The invention is characterized in that the first resonant inductor (215) and the second resonant inductor (235) are inductively coupled together with a reciprocal coupling factor less than 1, and the fourth electrical branch (207) includes a diode (255) or a second active switch (300), wherein the diode (255) has a cathode connected to the second intermediate electrical node (206) and an anode connected to the second output terminal (170), The second active switch (300) has a first conductive terminal (256) connected to the second intermediate electrical node (206), a second conductive terminal (257) connected to the second output terminal (170), and a control terminal (305) adapted to receive an electrical drive signal to switch the second active switch (300) between a saturation state and a cutoff state, wherein the saturation state allows current to flow between the first conductive terminal (256) and the second conductive terminal (257), and the cutoff state prevents the current from flowing.

2. The converter (145) according to claim 1, characterized in that The coupling factor between the first resonant inductor (215) and the second resonant inductor (235) is between 0.1 and 0.

8.

3. The converter (145) according to claim 2, characterized in that The coupling factor between the first resonant inductor (215) and the second resonant inductor (235) is between 0.2 and 0.

6.

4. The converter (145) according to claim 1, characterized in that The converter (145) includes a second resonant capacitor (240) having a first terminal (241) connected to the second output terminal (170) and a second terminal (242) connected to the second input terminal (155).

5. The converter (145) according to claim 1, characterized in that The converter (145) includes a first controller (500) configured to switch the first active switch (180) on and off in a periodic manner, wherein the first controller (500) is configured to measure a voltage applied to the first conductive terminal (185) of the first active switch (180) and to switch the first active switch (180) on when the measured voltage disappears.

6. The converter (145) according to claim 5, characterized in that The converter (145) includes a second controller (600) configured to turn the second active switch (300) on and off in a periodic manner, wherein the second controller (600) is configured to measure a voltage applied to the first conductive terminal (256) of the second active switch (300) and turn the second active switch (300) on when the measured voltage disappears.

7. The converter (145) according to claim 6, characterized in that The second controller (600) is configured to keep the second active switch (300) turned on for a shorter on time than the duration of the cycle, so as to turn off the second active switch (300) at the moment when the first controller (500) turns off the first active switch (180) or with a certain delay.

8. The converter (145) according to claim 7, characterized in that The second controller (600) is configured to measure a value of the voltage applied to the first output terminal (165) and to vary the delay for turning off the second active switch (300) to minimize a difference between the measured value and an expected value of the voltage.

9. The converter (145) of claim 6 or 7, wherein the first controller is configured to measure a peak value of the voltage applied to the first conductive terminal (185) of the first active switch (180), and to vary the on-time of the first active switch (180) to minimize a difference between the measured value and an expected value of the peak value of the voltage.

10. The converter (145) according to claim 1, characterized in that The converter (145) includes a third resonant inductor (280) arranged along an electrical branch (225) connecting the first intermediate electrical node (202) to the first terminal (221) of the first resonant capacitor (220).

11. The converter (145) according to claim 1, characterized in that The converter (145) includes a tank capacitor (290) connected in parallel with the diode (255) or the second active switch (300).

12. The converter (145) according to claim 1, characterized in that The converter (145) includes a tank capacitor (285) connected in parallel with the first active switch (180).

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