Full half-bridge resonant converter and voltage balance control method thereof

Abstract

The present invention discloses a full half-bridge resonant converter and a voltage balance control method thereof, wherein the full half-bridge resonant converter has a full bridge on the primary side with 4 switch tubes and a resonant inductor and capacitor, and a half bridge on the secondary side with 2 switch tubes, a total of 6 switch tubes, which reduces the number of components compared with the existing double-bridge resonant converter. The present invention adopts a fundamental wave approximation method to analyze the power model and soft switching conditions, and proposes a voltage balance control strategy, so that the converter can achieve voltage balance between the primary and secondary sides in a wide voltage range, while ensuring that all switch tubes achieve soft switching, thereby improving the converter efficiency.

Description

Technical Field

[0001] The present invention relates to the control technology of isolated high-frequency resonant converters, and specifically to a topology of a full-half bridge resonant converter and its voltage balance control method. Background Art

[0002] In the composition of new energy electric vehicles, the on-board charger OBC (On-Board-Charger) is one of the core components, which is composed of power factor correction PFC and isolated bidirectional DC-DC. The isolated bidirectional DC-DC converter has attracted more and more attention due to its advantages such as high power density, electrical isolation, and bidirectional energy transmission. However, when the voltage gain deviates from 1, problems such as loss of soft switching, increased circulating current, and transformer saturation will occur, resulting in a decrease in the efficiency of the converter.

[0003] The resonant double active bridge converter has a wide soft switching range, and some phase / shift control schemes have been proposed. Traditionally, the double active bridge converter adopts a modulation strategy of single phase shift (SPS); adopting this strategy requires both the primary and secondary bridges of the double active bridge converter to work at a 50% duty cycle, and the phase shift between the primary and secondary bridges is adjustable. In addition, the operating condition where the normalized voltage gain of the double active bridge converter is equal to 1 is defined as the voltage balance condition. Under this condition, the circulating current of the DAB converter is the lowest, and the zero voltage switching (ZVS) range is the widest. However, when the voltage conversion ratio deviates from 1, the double active bridge converter based on SPS will experience an increase in circulating current and loss of ZVS.

[0004] To solve this problem, scholars have proposed different improved modulation strategies. Among them, the extended phase shift (EPS) modulation reduces the duty cycle of the output voltage of a full bridge to less than 0.5. It is usually achieved by introducing an additional in-bridge phase shift between the drive signals of a full bridge, where the in-phase shift angle is determined by the actual voltage conversion ratio. Therefore, the output voltage of one full bridge becomes a three-level waveform, while the output voltage of the other full bridge remains a two-level square wave. Compared with SPS, the EPS scheme not only reduces the circulating current but also expands the ZVS range. However, when the converter state switches between the buck mode and the boost mode, the control signals of the two full bridges need to be interchanged.

[0005] To maintain good dynamic performance, scholars have also proposed double phase shift (DPS) modulation. Similar to EPS, DPS significantly reduces the circulating power and expands the ZVS range. In addition, different from EPS, DPS generates an identical in-bridge phase shift on both full bridges at the same time. Therefore, the output voltages of the two full bridges are both symmetric three-level waveforms. However, both EPS and DPS have only two control degrees of freedom, which prevents further improvement of performance.

[0006] Therefore, scholars have proposed triple-phase shift (TPS) modulation. Different from DPS, there are two unequal in-bridge phase shifts in TPS. However, due to the additional control degrees of freedom, it is difficult to determine the optimal modulation strategy in the form of an analytical solution. One way to solve this problem is to use Fourier analysis to derive the optimal strategy. However, since the resulting expression has an infinite number of coefficients, the analysis results are often complex and not intuitive. Another method is to use convex optimization tools to derive the results. But in many cases, the mathematical expression to be solved is a non-convex problem, so there is no global optimal solution. In addition, it highly depends on the converter parameters, making it difficult to apply in practice. Moreover, in EPS, DPS, and TPS, there is always a zero-voltage period in the output voltage of the full bridge. During this period, no active power is transferred by the converter, which weakens the power transmission ability of the DAB. Summary of the Invention

[0007] The present invention proposes a voltage balance control strategy for a full-half-bridge resonant converter to meet the requirements of a wide voltage range, for expanding the soft-switching range of the full-half-bridge resonant converter and improving the overall efficiency.

[0008] The technical solution to achieve the object of the present invention is: for a high-frequency resonant converter, a full-half-bridge resonant circuit structure and a voltage balance control strategy for expanding the zero-voltage switching (ZVS) range, reducing energy consumption, and improving the overall efficiency are proposed.

[0009] The present invention discloses a full-half-bridge resonant converter, including a full bridge on the primary side, a half bridge on the secondary side, a resonant capacitor C p , a resonant inductor L s , and a high-frequency transformer Tr with a turns ratio of 1:n;

[0010] wherein the primary-side full bridge includes switching transistors S 1 ~S 4 , body diodes d s1 ~d s4 , parasitic capacitors C s1 ~C s4 ; the secondary-side half bridge includes switching transistors S 5 ~S 6 , body diodes d s5 ~d s6 , parasitic capacitors C s5 ~C s6 , voltage-sharing capacitors C o1 and C o2 ;

[0011] V in and V out are the input voltage and the output voltage respectively, and i L and i oThey are the resonant current and the output current respectively;

[0012] Adjust the fundamental wave voltage of the primary side to be equal to the area of the positive half cycle of the fundamental wave voltage of the secondary side, achieving voltage balance.

[0013] Preferably, by setting the pulse widths of switches S 1 ~S 4 Four switches, adjust the gating signal of the pulse width controller to generate a three-level PWM voltage waveform.

[0014] Preferably, the duty cycles of switches S 1 and S 2 are 50%; the pulse width of switch S 4 is reduced to δ, the pulse width of S 3 is increased to 2π - δ, and the turn-on times of switches S 1 and S 4 are synchronized.

[0015] Preferably, by setting the pulse widths of switches S 5 ~S 6 Two switches, adjust the gating signal of the pulse width controller to generate the waveform of the secondary AC voltage v cd .

[0016] Preferably, adjust the duty cycle of S 5 -S 6 to be 50%, and a phase shift angle 1 is generated between S 5 and S such that the phase shift angle δ by which S 5 lags behind S 1 is the pulse width of switch S4.

[0017] Preferably, through steady-state analysis, obtain the waveform of the resonant current i ab based on the waveform diagrams of the midpoint primary AC voltage v cd and the secondary AC voltage v L .

[0018] The present invention proposes a voltage balance control method for a full half-bridge resonant converter. Since the converter operates in resonance, the fundamental wave approximation method is used for steady-state analysis:

[0019] From the circuit structure of the full half-bridge resonant converter, obtain the equivalent circuit diagram of the converter in the phasor domain FHA, where the two voltage sources are the normalized fundamental wave phasors of v ab and v cd / n, and obtain the normalized phasor expressions of v ab and v cd / n:

[0020]

[0021]

[0022] is the normalized vector representation form of v ab , is the normalized vector representation form of v cd , M is the voltage gain, specifically

[0023] Furthermore, the voltage gain M of the converter is obtained according to the turns ratio of the transformer; according to the normalized switching frequency F = ω s / ω N , quality factor Q = ω N L s / Z N the normalized impedance of the capacitor is obtained: QF - Q / F, where F is the normalized switching frequency and Q is the quality factor, L s is the sum of the externally connected inductor and the leakage inductance of the transformer; ω s is the switching angular frequency ω N is the standard resonant angular frequency, specifically Z N is the standard load resistance, specifically Z N = R L / n 2 , R L is the load resistance;

[0024] Using the equivalent circuit, the normalized resonant current expression is obtained:

[0025]

[0026] where and I p are the phase shift angle and peak current of the resonant current and v AB respectively;

[0027] Furthermore, the normalized output power P pu expression with respect to the pulse width δ and the phase shift angle is obtained:

[0028]

[0029] Thus, the ZVS range is further analyzed, and the ZVS conditions corresponding to each of the switches S 1 ~S 6 are obtained.

[0030] Furthermore, when the positive half-cycle areas of the fundamental voltage on the primary side and the fundamental voltage on the secondary side are equal, the voltage is considered balanced; the relational expression between δ and M can be obtained through calculation:

[0031]

[0032] Furthermore, when M = 0.5, the full-bridge on the primary side operates in the half-bridge state, and S 3 is always on while S 4 is always off. At this time, the positive half-cycle areas of v ab and v cd / n are equal; when M = 1, the primary side operates in the full-bridge state, and the voltage balance control modulation strategy is equivalent to the traditional single-phase-shift control, and the positive half-cycle areas of v ab and v cd / n are equal; when 0.5 < M < 1, the primary side operates in an intermediate state between the full-bridge and the half-bridge, and the positive half-cycle areas of v ab and v cd / n are still equal.

[0033] Compared with the prior art, the significant advantages of the present invention are as follows:

[0034] (1) The present invention proposes a full-half-bridge resonant converter topology structure, which reduces the number of switching tubes compared with the existing double-bridge resonant converter, and effectively reduces the cost.

[0035] (2) The present invention uses a resonant circuit, which has the advantages of low switching loss, easy control, and approximate sinusoidal current, and has higher efficiency than non-resonant converters.

[0036] (3) A voltage balance control strategy is proposed, enabling the converter to achieve voltage balance between the primary side and the secondary side under a wide voltage range, while ensuring that all switching tubes achieve soft switching, thereby improving the converter efficiency.

[0037] (4) The modulation strategy of the present invention does not require in-bridge phase shift, improving the power transmission capacity. Description of the Drawings

[0038] Figure 1 is the schematic diagram of the full-half-bridge resonant converter;

[0039] Figure 2 is the voltage waveform diagram generated by integrating the control methods of switches S 1 ~S 6 and the waveform diagram of the output current generated by controlling switches S 1 ~S 6 ;

[0040] Figure 3 is the equivalent circuit of the full-half-bridge resonant converter in the phasor domain FHA;

[0041] Figure 4 is the implementation diagram of the voltage balance control strategy;

[0042] Figure 5When V in = 75V, V out = 100V, M = 1, P = 200W, v ab , v cd , i L waveforms and the currents of each switching tube;

[0043] Figure 6 When V in = 125V, V out = 100V, M = 0.6, P = 200W, v ab , v cd , i L waveforms and the currents of each switching tube;

[0044] Figure 7 When V in = 150V, V out = 100V, M = 0.5, P = 200W, v ab , v cd , i L waveforms and the currents of each switching tube. Specific Embodiments

[0045] Next, in combination with the accompanying drawings in the embodiments of the present invention, the technical solutions in the embodiments of the present invention will be clearly and completely described. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative efforts fall within the protection scope of the present invention.

[0046] Next, the embodiments of the present invention will be further described in detail with reference to the accompanying drawings.

[0047] Embodiment 1

[0048] The schematic diagram of the full - half - bridge resonant converter is as shown in Figure 1 , where V in and V out are the input voltage and the output voltage respectively, i L and i o are the input and output currents respectively, C p is the resonant capacitor, C o1 , C o2 are the voltage - sharing capacitors, L s is the sum of the externally - connected inductor and the leakage inductance of the transformer, S 1 ~S 4 are the switching elements on the primary side, S 5 ~S 6 are the switching elements on the secondary side. Each of these 6 switching elements is composed of a body diode (d S1 ~d S6) and parasitic capacitance (C S1 ~C S6 ), where n is the transformer turns ratio.

[0049] First, adjust the pulse width of each switch to obtain the gating signal scheme for the high pulse width controller, thereby generating the waveform of the midpoint AC voltage V ab . As Figure 2 shown, the duty cycles of switches S 1 and S 2 are 50%. The pulse width of switch S 4 is reduced to δ, and the pulse width of S 3 is increased to 2π - δ. Thus, a three-level PWM voltage waveform is generated. Since the width of the positive pulse is δ and the width of the negative pulse is always equal to π. Adjust the duty cycle of S 5 ~S 6 to 50% to generate a symmetric square wave signal v cd . A phase shift angle 1 is generated between S 5 . By changing the width δ of the positive pulse and the phase shift angle , the normalized resonant current i and the normalized power P L,N pu pu can be controlled.

[0050] Through the steady-state analysis of the waveforms of v ab and v cd , the waveform of the resonant current i L can be obtained. To obtain the phasor expressions of the relevant quantities corresponding to changing the positive pulse width and the phase shift angle, since the resonant current is approximately a sine wave, the fundamental harmonic approximation method (FHA) is used for steady-state analysis. For convenience, all quantities are normalized by the base value, and the FHA equivalent circuit diagram of the converter in the phasor domain is obtained from the circuit structure of the full half-bridge series resonant converter. There is a phase delay, i.e., a phase angle, between switch S 1 and switch S 5 . Within one cycle, switch S 5 is closed with a pulse width of π; switch S 6 is closed with a pulse width of π as well. Thus, the waveform of the secondary AC voltage v cd is generated.

[0051] Since the resonant current is approximately a sine wave, the method of fundamental harmonic approximation (FHA) can be used for steady-state analysis. Through the steady-state analysis of the waveform diagram, the expression of the normalized resonant current i L can be obtained.

[0052] For convenience, all formulas are standardized by the base value through normalization (V Nis the standard voltage, Z N is the standard load resistance, ω N is the standard resonant angular frequency):

[0053] V N = V in

[0054] Z N = R L / n 2

[0055]

[0056] Figure 3 shows the FHA equivalent circuit of the converter in the phasor domain, where the two voltage sources are v ab and v cd / n of the normalized fundamental phasor, and we can get:

[0057]

[0058]

[0059] Next, according to the turns ratio of the transformer, the voltage gain M of the converter is obtained

[0060] M = 0.5V out / (nV in )

[0061] According to the normalized switching frequency F = ω s / ω N , the switching angular frequency ω s , and the quality factor Q = ω N L t / Z N the normalized impedance of the resonator can be obtained:

[0062] QF - Q / F

[0063] Using the equivalent circuit, the normalized resonant current expression i L,N (t) can be obtained:

[0064] i L,N (t) = I P cos(ω S t + Φ i )

[0065] where the phase angle Φ i and the peak current I p are:

[0066]

[0067]

[0068] According to the current I p rms value, v AB The normalized output power expression can be obtained through calculation of the rms value of the voltage:

[0069]

[0070] Then, according to the definition of ZVS, when the current value passing through at the switch-on moment of the switch tube is negative, it means that ZVS is achieved. According to Figure 2 , the ZVS conditions for each switch can be obtained

[0071]

[0072] Furthermore, according to Figure 4 , the voltage balance control strategy (VBT) is implemented, making the positive half-cycle of the fundamental voltage on the primary side equal to that on the secondary side. The expression is as follows:

[0073]

[0074] Furthermore, simplifying it, the obtained expression is as follows:

[0075]

[0076] According to the range of δ being in [0, π], cosδ can be obtained to be between [-1, 1], and thus the value range of M is as follows:

[0077] 1 / 2 ≤ M ≤ 1

[0078] So far, the voltage balance has been completed, and the relationship between M and δ and the range of M have been obtained.

[0079] So far, δ and can be calculated according to the normalized power magnitude and different values of M under the voltage balance control strategy.

[0080] Performing simulation according to the designed input, output, and power. At this time, soft switching can be achieved for all switch tubes within the full power range.

[0081] To verify the correctness of the theory, simulation tests were carried out in PSIM:

[0082] (1) When V in = 75V, V out = 100V, M = 1, P = 200W, v ab , v cd , i L waveforms and the currents of each switch tube are as Figure 5 shown;

[0083] (2) When V in = 125V, V out = 100V, M = 0.6, P = 200W, v ab , v cd , i L waveforms and the currents of each switching device are as Figure 6 shown;

[0084] (3) When V in = 150V, V out = 100V, M = 0.5, P = 200W, v ab , v cd , i L waveforms and the currents of each switching device Figure 7 are as shown;

[0085] After verification by combining with the simulation waveforms, it is found that the theory is consistent with the actual situation, proving that the present invention is feasible.

[0086] The above is only the preferred specific implementation manner of the present invention, but the protection scope of the present invention is not limited thereto. Any change or replacement that can be easily thought of by those skilled in the art within the technical scope disclosed by the present invention should be covered by the protection scope of the present invention. Therefore, the protection scope of the present invention should be subject to the protection scope of the claims.

Claims

1. A full half-bridge resonant converter, characterized in that: Including the full bridge on the primary side, the half bridge on the secondary side, and the resonant capacitor C p , resonant inductor L s , a high-frequency transformer Tr with a turns ratio of 1:n; The primary side full bridge includes switch tubes S1~S4, body diode d s1 ~d s4 , parasitic capacitance C s1 ~C s4 ; The secondary side half bridge includes the switch tube S5~S6 body diode d s5 ~d s6 , parasitic capacitance C s5 ~C s6 , voltage balancing capacitor C o1 and C o2 The primary side full-bridge circuit is composed of a first branch and a second branch connected in parallel; the first branch is configured with switch devices S1 and S2 in order from top to bottom; the second branch is configured with switch devices S3 and S4 in order from top to bottom; the switch devices S1 and S4 form a diagonal switch pair, and the switch devices S2 and S3 form a diagonal switch pair; the secondary side half-bridge circuit is composed of a third branch, and the third branch is configured with switch devices S5 and S6 in order from top to bottom; The primary side full bridge includes switch tubes S1-S4. S1-S2 and S3-S4 are connected in series to form left and right arms respectively. The midpoint of the left arm is connected to the resonant inductor L. S , resonant capacitor C P Then connect it to one end of the transformer primary side; the other end of the transformer primary side is connected to the midpoint of the right arm; The secondary side half bridge includes series switch tubes S5-S6 and series voltage equalizing capacitor C o1 and C o2 , the series switch tube S5-S6 and the series voltage-equalizing capacitor C o1 and C o2 In parallel, one end of the secondary side of the transformer is connected to the midpoint of the switch tube S5-S6, and the other end is connected to the capacitor C o1 -C o2 midpoint; V in and V out are the input voltage and output voltage respectively, i L and i o are the resonant current and the output current respectively; according to The positive half-cycle area of ​​the fundamental voltage on the primary side is adjusted to be equal to that of the fundamental voltage on the secondary side, and the voltage is balanced. Here, M is the voltage gain, and δ is the pulse width of the switch tube S4.

2. The full half-bridge resonant converter according to claim 1, characterized in that: By setting the pulse widths of the four switches S1 to S4 and adjusting the gate signal of the pulse width controller, a three-level PWM voltage waveform is generated.

3. The full half-bridge resonant converter according to claim 2, characterized in that: The duty cycle of switches S1 and S2 is 50%; the pulse width of switch S4 is reduced to δ, the pulse width of S3 is increased to 2π-δ, and the opening time of switches S1 and S4 is synchronized.

4. The full half-bridge resonant converter according to claim 2, characterized in that: By setting the pulse width of the two switches S5~S6 and adjusting the gate signal of the pulse width controller, a secondary AC voltage v is generated. cd waveform.

5. The full half-bridge resonant converter according to claim 4, characterized in that: Adjust the duty cycle of S5-S6 to 50%, and a phase shift angle is generated between S1 and S5. is the phase shift angle of S5 lagging behind S1.

6. The full half-bridge resonant converter according to claim 4, characterized in that: Through steady-state analysis, according to the midpoint primary AC voltage v ab and the secondary AC voltage v cd The waveform diagram of the resonant current i L waveform.

7. A voltage balance control method using the full half-bridge resonant converter according to any one of claims 1 to 6, characterized in that: Use the fundamental wave approximation method to perform steady-state analysis: The FHA equivalent circuit diagram of the converter in the phasor domain is obtained from the circuit structure of the full half-bridge resonant converter, where the two voltage sources are v ab and v cd / n, we get v ab and v cd Normalized phasor expression of / n: Yes ab The normalized vector representation of Yes cd The normalized vector representation of M is the voltage gain, specifically: φ is the phase shift angle of S5 lagging behind S1, and δ is the pulse width of switch tube S4.

8. The voltage balance control method according to claim 7, characterized in that: The voltage gain M of the converter is obtained according to the turns ratio of the transformer; according to the normalized switching frequency F = ω s / ω N , quality factor Q = ω N L s / Z N The normalized impedance of the capacitor is obtained as: QF-Q / F, where F is the normalized switching frequency, Q is the quality factor, and L s is the sum of the external inductance and the transformer leakage inductance; ω s is the switching angular frequency, ω N is the standard resonant angular frequency, specifically Z N is the standard load resistance, specifically Z N =R L / n 2 , R L is the load resistance; Using the equivalent circuit, the normalized resonant current expression is obtained: in φ and I p are the resonant current and v AB Phase shift angle and peak current; Then we get the normalized output power P pu Expressions for pulse width δ and phase shift angle φ: The range of ZVS is further analyzed, and the corresponding ZVS conditions of switches S1 to S6 are obtained.

9. The voltage balance control method according to claim 7, characterized in that: When M = 0.5, the full - bridge on the primary side operates in the half - bridge state. S3 is always on and S4 is always off. At this time, the areas of the positive half - cycles of v ab and v cd / n are equal. When M = 1, the primary side operates in the full - bridge state, and the voltage - balance control modulation strategy is equivalent to single - phase - shift control. The areas of the positive half - cycles of v ab and v cd / n are equal. When 0.5 < M < 1, the primary side operates in an intermediate state between the full - bridge and the half - bridge. At this time, the range of δ is [0, π]. Between 0 and π, the areas of the positive half - cycles of v ab and v cd / n are still equal.

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