A Method for Capturing Beidou D1 Signal with High Dynamic Range and High Sensitivity

Abstract

The present invention relates to a method for capturing Beidou D1 signals with high dynamics and high sensitivity, belonging to the field of satellite technology. After necessary preprocessing of the intermediate-frequency signal, it is stored in two caches in parallel. The data in cache A is used for the strong-star capture method with a coherent integration time of 1 ms, avoiding the NH code jump but with relatively low sensitivity. Cache B uses fixed-point discontinuous caching, which is suitable for the weak-star capture method with a coherent integration time of 4 ms, a large data time span, and high sensitivity. The strong-star and weak-star modes work alternately until the signal is successfully captured. For the successfully captured signal, on the software side, a sliding window of a certain length is used to perform secondary capture (fine capture) on the Doppler and pseudo-code of the signal in real time to adapt to relatively high dynamic stress.

Description

Technical Field

[0001] The present invention belongs to the field of satellite technology and relates to a method for capturing Beidou D1 signals with high dynamics and high sensitivity. Background Art

[0002] Capturing Beidou D1 signals with high dynamics and high sensitivity mainly involves achieving fast capture of D1 signals by GNSS receivers under high-dynamic conditions with weak received signal power while overcoming the influence of the Neumann-Hofman code (NH code) modulated on the D1 signal. It is an important link in high-dynamic and high-sensitivity navigation receivers.

[0003] The Neumann-Hofman code modulated on the Beidou D1 signal has a period of 20 ms and a chip length of 1 ms.

[0004] Its functions are as follows:

[0005] 1) Reduce the mutual interference of GNSS signals;

[0006] 2) Facilitate the bit synchronization process of the signal. However, during the signal capture stage, the receiver has no knowledge of the NH code phase of the D1 signal. If the operation of extending the coherent integration time is adopted to increase the baseband gain, the risk of energy cancellation is extremely likely to occur, further reducing the sensitivity.

[0007] To achieve high-sensitivity capture of D1 signals, the strategies generally adopted at this time are as follows:

[0008] (1) Fix the coherent integration time of the D1 signal at 1 ms and avoid the bit jump problem through non-coherent integration. However, as the integration time increases, the non-coherent loss cannot be ignored;

[0009] (2) Use the NH code phase as the fourth dimension for signal search (the other three dimensions are carrier Doppler, pseudo-code phase, and satellite PRN) for search. However, this will greatly increase resource consumption.

[0010] Continuing with the above strategy (2): To obtain sufficient coherent gain, it is necessary to strip the NH code. However, since the receiver does not know the NH code phase of each satellite's D1 signal during capture, at this time, a search work for a total of 20 code start points (corresponding to 20 different NH code phases) will be carried out. At this time, the resource consumption for completing one round of capture expands from 2*M*Ncode*Ndoppler in strategy (1) to 20*M*2*Ncode*Ndoppler, where M is the total integration time, 2*Ncode is twice the code length of the D1 signal (the number of half chips within the code period), and Ndoppler is the number of Doppler frequency bins. Summary of the Invention

[0011] In view of this, the object of the present invention is to provide a Beidou D1 signal acquisition method with high dynamic range and high sensitivity.

[0012] To achieve the above object, the present invention provides the following technical solutions:

[0013] A Beidou D1 signal acquisition method with high dynamic range and high sensitivity, the method comprising the following steps:

[0014] S1: Determine the time window T DC , within the time window, accumulate the digital intermediate frequency signal IF after analog-to-digital conversion ADC sampling, calculate the arithmetic mean at the end of the window to obtain the DC component of the intermediate frequency signal, and subtract this component from the subsequent incoming intermediate frequency signal, where T DC takes a value of 2 n / fs, fs is the 60 MHz intermediate frequency signal sampling rate;

[0015] S2: The signal is multiplied by the carrier signal output by the mixer and the local oscillator to obtain a zero intermediate frequency I / Q signal;

[0016] S3: Perform half-band filtering on the Beidou D1 signal to remove the noise components in the signal; for the B1I signal, the zero intermediate frequency signal is 2-fold decimated after passing through the first-stage half-band filter, and then passes through the second-stage half-band filter and is 2-fold decimated. After that, the intermediate frequency signal passes through the first-stage half-band filter again and is 3-fold decimated, and finally a zero intermediate frequency signal of fs / 12 = 5 MHz is obtained;

[0017] S4: After passing through several stages of half-band filters and samplers in step S3, the effective bit width of the signal gradually accumulates. To reduce the subsequent operation complexity, set a new time window T Q , calculate the probability distribution of the sampled values within the window; due to the symmetry of the normal distribution, only consider the probability distributions of +1 samples and +3 samples. If the number of +1 points is more than twice the number of +3 points, lower the threshold, otherwise raise the threshold, so that the sampled value always presents a normal distribution;

[0018] S5: Decimate the quantized signal to twice the pseudo-code rate and transfer it to the acquisition circuit;

[0019] S6: The acquisition circuit completes signal acquisition in the form of alternating strong satellite mode and weak satellite mode; in the strong satellite mode, no coherent integration is performed on the signal, that is, the coherent integration time is 1 ms; in the weak satellite mode, the coherent integration time of the signal is selected as 4 ms;

[0020] S7: Use the strong satellite mode to capture the D1 signal, and store the data entering the acquisition circuit in step S5 in the buffer A with a data capacity of V A , V A is equal to the number of samples 2*N*(T required to complete one signal acquisition in the strong satellite modenoncoh +1), where N is the number of chips in the PN code period and T noncoh is the non-coherent integration time; meanwhile, the data in step S5 is stored in parallel in the buffer B with a data capacity of V B , and the value of V B is equal to the number of samples 2*N*(T coh *T noncoh +1) required to complete signal acquisition in the weak satellite mode, where N is the number of chips in the PN code period and T coh =4 is the number of coherent integration times in the weak satellite mode, and T noncoh is the non-coherent integration time;

[0021] S8: The buffer B only caches the signal points within the time interval [T0 + K*T D1 , T0 + K*T D1 +4], that is, the above fixed-point discontinuous caching, where T D1 =20 is the navigation data period of the D1 signal, T0 is the cache start time, and K is a non-negative integer with a value range of 0 ≤ K < T noncoh , and the units are all milliseconds;

[0022] S9: Since the time for the buffer A to store all the samples is significantly less than that of the buffer B, after the buffer A finishes storing, the acquisition engine writes the PRN number of the satellite to be searched, and the acquisition circuit generates the PN code according to the PRN number; after the PN code generation is completed, the circuit extracts the data in the buffer for playback at the working frequency f acq ;

[0023] S10: First, the carrier is stripped from the signal according to the theoretical intermediate frequency value of the RF chip to obtain the I and Q zero-intermediate frequency signals;

[0024] S11: The playback signal points are passed into the N-bit shift register in the matched filtering module, and N is equal to twice the code length value of the signal; when the playback signal point count is greater than or equal to N, the signal is subjected to short-time correlation matching filtering according to the PN code sequence generated in step S10 and 2 K short-time correlation sums are obtained, where K is the radix of the FFT; since the coherent integration time in the strong satellite mode is 1 ms and the algorithm is not affected by the NH code modulation, the 2 K short-time correlation sums are directly passed to the FFT module;

[0025] S12: The 2 K short-time correlation sums in step S11 are subjected to FFT transformation to obtain 2 K signal segments in the frequency domain and cached. After the next correlation operation ends, the new signal segments will be accumulated with the cached results; when the accumulation times reach T noncoh , the average value E of the non-coherent integration energy of N code phases is obtained mean, find the maximum integral energy value E max , compare the ratio E max / E mean with a preset threshold for energy decision;

[0026] S13: Limited by hardware resources, the satellite PRN dimension must be serially searched. After the current satellite is captured, replay the samples stored in V A , and repeat steps S9 - S12 to enter the capture process of the next satellite;

[0027] S14: For the successfully captured satellite, turn on the tracking channel and read the dump timestamp T dump returned after the first integral of the channel is completed, and combine it with the timestamp T store released when the signal caching is completed to find the code start point of the signal;

[0028] After all BDS satellites on the satellite search list have completed one round of capture in the strong satellite mode, since it takes a long time for V B to store the samples required for one round of capture, it is necessary to detect whether V B has been stored to the specified capacity. If V B has not stored enough samples, repeat the strong satellite mode to capture the signal: take out the new samples in V A , and repeat the above steps S9 - S14; otherwise, turn on the weak satellite mode to capture the D1 signal;

[0029] S15: When the cache data volume of V B reaches 2 * N * (T coh * T noncoh + 1), turn on the weak satellite capture mode; the first step of the weak satellite mode is still carrier stripping and short - time correlation matching filtering. Repeat steps S9, S10, and S11. After the short - time correlation sum corresponding to 1 ms of data is generated, since the coherent integration time is greater than 1 ms, the influence of NH code modulation cannot be ignored, and NH code stripping is performed on the correlation sum;

[0030] S16: The signals used in two adjacent rounds of coherent integration are not continuous in time and are both at a fixed phase within the D1 data period. There is no need to search for the NH code phase using the method of sliding code start points. Only use the following 8 NH code combinations to perform NH code stripping on the correlation sum: {1, 1, 1, 1}, {-1, 1, 1, 1}, {1, -1, 1, 1}, {1, 1, -1, 1}, {1, 1, 1, -1}, {-1, -1, 1, 1}, {1, -1, 1, -1}, {1, -1, -1, 1};

[0031] S17: Store the coherent sum after NH code stripping into the cache V C ; V CThe accumulated data stored in [[ ]] will be coherently accumulated with the NH stripped data output by the short-time correlation filter next time until the number of coherent accumulations reaches N. coh The generated coherent sum is transmitted to the FFT module.

[0032] S18: Repeat steps S11 - S13 until signal acquisition for all satellites on the satellite search list is completed in one round under the weak satellite mode.

[0033] S19: Since under the weak satellite mode, the time span of the V B cached signal becomes significantly longer. Under high dynamics, there is a certain probability that the difference between the pseudocode phase and Doppler value searched when signal acquisition is successful and the actual code phase and Doppler at the time of enabling the tracking channel will exceed the loop tracking range. Using the three-way integral values of EPL returned by the baseband, a window with a length of 1.5 chip lengths is set to slide the signal pseudocode phase. The number of sliding times is M, and the value of M is (dT * a * f code / f carrier ) / 1.5, where a is the Doppler change rate calculated by the receiver PVT. Assume the receiver is in the positioning state, f code is the signal pseudocode frequency, f carrier is the signal carrier frequency, and dT = T dump -T store is the time span from acquisition to tracking described in step S14.

[0034] S20: After M times of pseudocode sliding are completed, read the integral energy values corresponding to 3 * M different pseudocode phases stored in the buffer, find the code phase corresponding to the maximum energy, and configure the baseband to slide the signal pseudocode to this position. Slide in the Doppler dimension of the signal and find the real-time Doppler frequency of the signal according to the returned integral energy values. After reconfiguring the code phase and Doppler, the signal transfer is successful and enters the tracking state.

[0035] Optionally, under the strong satellite mode, the signal acquisition process is as follows:

[0036] Continuous caching, carrier stripping, short-time correlation matching filtering, spectrum analysis, non-coherent integration, and energy decision.

[0037] Under the weak satellite mode, the basic signal acquisition process is as follows:

[0038] Fixed-point discontinuous caching, carrier stripping, short-time correlation matching filtering, NH code stripping, coherent integration, spectrum analysis, non-coherent integration, energy decision, Doppler and pseudocode compensation.

[0039] Optionally, for the intermediate frequency signal input to the ADC of the radio frequency chip, the DC component is removed first to prevent the signal sample distribution from being affected and interfering with the decision threshold of the subsequent quantization circuit; after passing through several stages of filters and samplers, the effective bit width of the digital signal will gradually accumulate and needs to be re-quantized to 2 bits through automatic gain control (AGC); the AGC logic accumulates and counts the MAG bits, calculates the proportional distribution within the corresponding time of the AGC, and under the characteristic that the DC-removed intermediate frequency signal follows a Gaussian distribution, if the proportional distribution exceeds 17%, it indicates that the gain is too large and a feedback signal needs to be generated to reduce the PGA gain, and conversely, if it is less than 17%, the PGA gain needs to be increased.

[0040] Optionally, the transfer function of the filter is the conjugate of the signal spectrum, and the filter must exactly match the spectrum of the input signal. Let the sampling sequence of the input signal be represented by s(n), then the output of the matched filter is expressed as:

[0041]

[0042] To ensure that the accumulated sample sequence still contains frequency information, only the sequence after the pseudo-code is stripped is segmented and accumulated, and then the FFT is used to perform spectral analysis on these partial accumulated sums; considering a Beidou intermediate frequency data sampling sequence with a length of 1 ms, denoted as:

[0043]

[0044] where T s is the sampling interval, C is the pseudo-random code, ω IF is the intermediate frequency carrier frequency, and is the initial phase; M takes values from 0 to N - 1, indicating that the number of samples within 1 ms is N, and there is no navigation message bit jump within one pseudo-random code period; the length of each short-time correlation integration is M samples, and T p represents the duration of the short-time correlation, then the output of the i-th short-time correlation accumulation is:

[0045]

[0046]

[0047] The short-time correlation result is a digital sequence {I p -jQ n with a sampling interval of T n , n = 1,..., L}, and the number of samples within all periods is L. Performing an FFT transformation on this sequence, the resulting output is:

[0048]

[0049] In the strong star mode, the data length corresponding to one FFT is 1ms. Assuming that the short-time correlation matched filter generates L=8 partial sums, and the number of signal points is expanded to 16 by the 0-filling operation, that is, the signal period is lengthened to 2ms, then the basic frequency of the FFT transformation is That is, the capture resolution of the carrier Doppler dimension is 500 Hz, and the step size is ωxL=4000 Hz; in the weak star mode, the partial sum generated by the short-time correlation matched filter is first coherently integrated, and L further accumulated partial sums are obtained every 4 ms, and then sent to the FFT module; the coherent integration only folds the samples with the same phase within the code period, and the frequency resolution and step size at this time are consistent with those in the strong star mode.

[0050] The beneficial effect of the present invention is that the present invention converts the above-mentioned search for the NH code start point into the search for several truncated 4-bit NH code sequences, which not only maintains the coherent integration strategy, but also reduces the resource consumption for completing one capture, and improves the capture sensitivity and capture time. The baseband working mode adopted by the method of the present invention is cache-playback, so in order to compensate for the inconsistency between the signal capture parameters and the real-time signal parameters caused by high dynamics, the method of the present invention uses the EPL correlator of the tracking channel to slide the pseudo-code phase of the signal, so as to achieve the purpose of successfully pulling the signal under high dynamics.

[0051] Other advantages, objectives and features of the present invention will be described in the following description to some extent, and to some extent, will be obvious to those skilled in the art based on the following examination and study, or can be taught from the practice of the present invention. The objectives and other advantages of the present invention can be realized and obtained through the following description. BRIEF DESCRIPTION OF THE DRAWINGS

[0052] In order to make the purpose, technical solutions and advantages of the present invention more clear, the present invention will be described in detail below in conjunction with the accompanying drawings, wherein:

[0053] Figure 1 It is a flowchart of the present invention. DETAILED DESCRIPTION

[0054] The following describes the embodiments of the present invention by specific examples, and those skilled in the art can easily understand other advantages and effects of the present invention from the contents disclosed in this specification. The present invention can also be implemented or applied through other different specific embodiments, and the details in this specification can also be modified or changed in various ways based on different viewpoints and applications without departing from the spirit of the present invention. It should be noted that the illustrations provided in the following embodiments only illustrate the basic concept of the present invention in a schematic manner, and the following embodiments and features in the embodiments can be combined with each other without conflict.

[0055] Among them, the attached drawings are only for illustrative purposes, showing only schematic diagrams rather than physical diagrams, and should not be construed as limiting the present invention; in order to better illustrate the embodiments of the present invention, some components in the attached drawings will be omitted, enlarged or reduced, which do not represent the dimensions of the actual product; for those skilled in the art, it is understandable that some well-known structures and their descriptions in the attached drawings may be omitted.

[0056] In the attached drawings of the embodiments of the present invention, the same or similar reference numerals correspond to the same or similar components; in the description of the present invention, it should be understood that if there are terms such as "upper", "lower", "left", "right", "front", "rear", etc. indicating the orientation or positional relationship, they are based on the orientation or positional relationship shown in the attached drawings, and are only for the convenience of describing the present invention and simplifying the description, rather than indicating or implying that the device or element referred to must have a specific orientation, be constructed and operated in a specific orientation. Therefore, the terms describing the positional relationship in the attached drawings are only for illustrative purposes and should not be construed as limiting the present invention. For those of ordinary skill in the art, the specific meanings of the above terms can be understood according to specific circumstances.

[0057] Please refer to Figure 1 , which is a high-dynamic and high-sensitivity Beidou D1 signal acquisition method.

[0058] (1) Determine a time window T DC , accumulate the digital intermediate frequency signal (IF) after analog-to-digital conversion (ADC) sampling within the time window, and calculate the arithmetic mean at the end of the window to obtain the DC component of the intermediate frequency signal, and subtract this component from the subsequent incoming intermediate frequency signal, where T DC has a value of 2 n / fs, and fs is the 60 MHz intermediate frequency signal sampling rate.

[0059] (2) The signal is multiplied by the carrier signal output by the mixer and the local oscillator to obtain a zero-intermediate frequency I / Q signal.

[0060] (3) Perform half-band (Halfband) filtering on the Beidou D1 signal to remove the noise components in the signal. Taking the B1I signal as an example, the zero-intermediate frequency signal is 2-fold decimated after passing through one-stage half-band (Halfband) filtering, and then passes through the second-stage half-band filtering and is 2-fold decimated. The subsequent intermediate frequency signal is then 3-fold decimated after passing through one-stage half-band filtering, and finally a zero-intermediate frequency signal of fs / 12 = 5 MHz is obtained.

[0061] (4) After passing through several stages of half-band filters and samplers in step (3), the effective bit width of the signal gradually accumulates. In order to reduce the subsequent operation complexity, a new time window T Q, calculate the probability distribution of the sampled values within the window. Due to the symmetry of the normal distribution, only the probability distributions of the +1 sample point and the +3 sample point need to be considered. If the number of +1 points is more than twice the number of +3 points, lower the threshold; otherwise, raise the threshold to make the sample point values always exhibit a normal distribution.

[0062] (5) Downsample the quantized signal to twice the chip rate and transfer it to the acquisition circuit.

[0063] (6) The acquisition circuit completes signal acquisition in the form of alternating strong satellite mode and weak satellite mode. In the strong satellite mode, no coherent integration is performed on the signal (i.e., the coherent integration time is 1 ms), and the influence of NH code modulation can be ignored at this time. In the weak satellite mode, the coherent integration time of the signal is selected as 4 ms. Specifically, the basic process of signal acquisition in the strong satellite mode is

[0064] Continuous caching -> Carrier stripping -> Short-time correlation matched filtering -> Spectrum analysis (FFT) -> Non-coherent integration and energy decision,

[0065] The basic process of signal acquisition in the weak satellite mode is:

[0066] Fixed-point discontinuous caching -> Carrier stripping -> Short-time correlation matched filtering -> NH code stripping -> Coherent integration -> Spectrum analysis (FFT) -> Non-coherent integration and energy decision -> Doppler and chip compensation.

[0067] (7) First, use the strong satellite mode to acquire the D1 signal. Specifically, store the data that enters the acquisition circuit in step (5) into cache A with a data capacity of V A The value of V A should be equal to the number of sample points 2 * N * (T noncoh +1) required to complete one signal acquisition in the strong satellite mode, where N is the number of chips within the chip period, and T noncoh is the non-coherent integration time. At the same time, store the data in step (5) into cache B with a data capacity of V B in parallel. The value of V B should be equal to the number of sample points 2 * N * (T coh *T noncoh +1) required to complete one signal acquisition in the weak satellite mode, where N is the number of chips within the chip period, T coh =4 is the number of coherent integration times in the weak satellite mode, and T noncoh is the non-coherent integration time.

[0068] (8) Specifically, cache B only caches the signal points within the time interval [T0 + K * T D1 , T0 + K * T D1 +4], that is, the above-mentioned fixed-point discontinuous caching, where T D1= 20 is the navigation data period of the D1 signal, T0 is the cache start time, and K is a non - negative integer with a value range of 0 ≤ K < T noncoh , and the units are all milliseconds.

[0069] (9) Since the time for cache A to store all the samples is significantly less than that of cache B, after cache A finishes storing, the acquisition engine writes the PRN numbers to be searched for satellites, and the acquisition circuit generates pseudo - codes according to the PRN numbers. After the pseudo - code generation is completed, the circuit uses the operating frequency f acq to retrieve the data in the cache for playback.

[0070] (10) First, carrier stripping is performed on the signal according to the theoretical intermediate - frequency value of the RF chip to obtain two zero - intermediate - frequency signals of I and Q.

[0071] (11) The playback signal points are passed into the N - bit shift register in the matched - filtering module, where N is equal to twice the code - length value of the signal. When the playback signal - point count is greater than or equal to N, short - time correlation matching filtering is performed on the signal according to the pseudo - code sequence generated in step (10) and 2 K short - time correlations are obtained, where K is the radix of the FFT. Since the coherent integration time is 1 ms in the strong - satellite mode and the algorithm is not affected by the NH code modulation, the 2 K short - time correlations are directly passed to the FFT module.

[0072] (12) The 2 K short - time correlations in step (11) are subjected to FFT transformation to obtain 2 K signal segments in the frequency domain, and cache operations are performed. After the next correlation operation ends, the new signal segments will be accumulated with the cache results. When the accumulation count reaches T noncoh , the average value E mean of the non - coherent integration energy of N code phases is obtained, the maximum integration energy value E max is obtained, and the ratio E max / E mean is compared with the preset threshold for energy decision.

[0073] (13) Due to hardware resource limitations, the search in the satellite PRN dimension must be serial. Therefore, after the current satellite acquisition is completed, the samples stored in V A are re - played, and steps (9) - (12) are repeated to enter the acquisition process of the next satellite.

[0074] (14) For the successfully acquired satellites, the tracking channels are enabled, and the dump timestamp T dump returned after the first integration of the channel is read, and combined with the timestamp T store released when the signal caching is completed, the code start point of the signal is obtained.

[0075] After all BDS satellites on the satellite search list have completed one round of acquisition in the strong satellite mode, since V B takes a relatively long time to store the samples required for one round of acquisition, it is necessary to detect whether V B has been stored to the specified capacity. If V B has not stored enough samples, repeat the strong satellite mode to capture the signal: take out the new samples in V A , and repeat the above steps (9)-(14). Otherwise, start the weak satellite mode to capture the D1 signal.

[0076] (15) When the amount of cached data in V B reaches 2*N*(T coh *T noncoh +1), start the weak satellite capture mode. The first step of the weak satellite mode is still carrier stripping and short-time correlation matching filtering, so repeat the above steps (9), (10), (11). After the short-time correlation sum corresponding to 1 ms of data is generated, since the coherent integration time is greater than 1 ms, the influence of NH code modulation cannot be ignored, so it is necessary to strip the NH code from the correlation sum.

[0077] (16) Specifically, since the signals used in two adjacent rounds of coherent integration are not continuous in time and are both at a fixed phase within the D1 data period, there is no need to search for the NH code phase by the method of the sliding code start point. Only the following 8 NH code combinations are needed to strip the NH code from the correlation sum: {1,1,1,1}, {-1,1,1,1}, {1,-1,1,1}, {1,1,-1,1}, {1,1,1,-1}, {-1,-1,1,1}, {1,-1,1,-1}, {1,-1,-1,1}.

[0078] (17) Store the coherent sum after NH code stripping into the cache V C . The accumulated data stored in V C will perform a coherent accumulation operation with the NH stripped data output by the next short-time correlation filter until the number of coherent times reaches N coh . Transmit the generated coherent sum to the FFT module.

[0079] (18) Repeat steps (11)~(13) until all satellites on the satellite search list have completed one round of signal capture in the weak satellite mode.

[0080] (19) Since in the weak satellite mode, V BThe time span of the cached signal becomes significantly longer. Under high dynamics, when the signal acquisition is successful, there is a certain probability that the difference between the pseudocode phase and Doppler value searched and the actual code phase and Doppler at the time of enabling the tracking channel will exceed the loop tracking range. Therefore, it is necessary to further utilize the three-channel integral values of EPL returned by the baseband and set a window with a length of 1.5 chip lengths to slide the signal pseudocode phase. The number of sliding times is M, and the value of M is (dT * a * f code / f carrier ) / 1.5, where a is the Doppler change rate calculated by the receiver PVT (assuming the receiver is in the positioning state at this time), f code is the signal pseudocode frequency, f carrier is the signal carrier frequency, and dT = T dump -T store is the time span from acquisition to tracking described in step (14).

[0081] (20) After M times of pseudocode sliding are completed, read the integral energy values corresponding to 3 * M different pseudocode phases stored in the buffer, find the code phase corresponding to the maximum energy, and configure the baseband to slide the signal pseudocode to this position. Similarly, slide in the Doppler dimension of the signal and find the real-time Doppler frequency of the signal according to the returned integral energy values. After reconfiguring the code phase and Doppler, the signal is successfully transferred to the traction state and enters the tracking state.

[0082] The present invention provides a high-dynamics and high-sensitivity Beidou D1 signal acquisition method, and the process is as Figure 1 shown. For the intermediate-frequency signal input by the ADC of the RF chip, first remove the DC component to prevent the signal sample distribution from being affected and interfering with the decision threshold of the subsequent quantization circuit. After passing through several stages of filters and samplers, the effective bit width of the digital signal will gradually accumulate and needs to be re-quantized to 2 bits through automatic gain control (AGC). The AGC logic accumulates and counts the MAG bits, calculates the proportional distribution within the corresponding time of AGC. Under the characteristic that the DC-removed intermediate-frequency signal follows a Gaussian distribution, if the proportional distribution exceeds 17%, it indicates that the gain is too large and a feedback signal needs to be generated to reduce the PGA gain. Conversely, if it is less than 17%, the PGA gain needs to be increased.

[0083] Table 1 Sample distribution of 2 bits

[0084]

[0085] The half-band filter is a special FIR filter, whose order can only be even and the length is odd. Except for the middle value being 0.5, the coefficients of the remaining even serial numbers are all 0 (thus greatly saving the multiplication and addition operations during filtering).

[0086] Taking the Beidou B1I signal as an example, the filter parameters are designed as shown in Table 2 for its effective signal bandwidth.

[0087] Table 2 Filter Parameters

[0088] Sampling rate Bandwidth Transition band Stopband attenuation 60MHz 4MHz 28MHz 65dB

[0089] The correlation-matched filtering adopted by the method of the present invention is a linear filter with the maximum ratio of the instantaneous power of the signal to the average power of the noise under white noise conditions, and it is an optimal filter. The transfer function of the matched filter is the conjugate of the signal spectrum. Therefore, the matched filter must be completely matched with the spectrum of the input signal. Assuming that the sampling sequence of the input signal is represented by s(n), the output of the matched filter can be expressed as:

[0090]

[0091] In order to ensure that the accumulated sample sequence still contains frequency information, only the sequence after pseudo-code stripping is segmented and accumulated, so that the FFT can be used for spectral analysis of these partial accumulated sums. Considering the Beidou intermediate-frequency data sampling sequence with a length of 1 ms, it can be recorded as:

[0092]

[0093] where T s is the sampling interval, C is the pseudo-random code, ω IF is the intermediate-frequency carrier frequency, and is the initial phase. M takes values from 0 to N - 1, indicating that the number of samples within 1 ms is N. Since there is no navigation message bit transition within one pseudo-random code period, the navigation message bit term is ignored here. The length of each short-time correlation integration is M samples, and here T p is used to represent the duration of short-time correlation. Then the output of the i-th short-time correlation accumulation is:

[0094]

[0095]

[0096] The short-time correlation result is a digital sequence {I p - jQ n with a sampling interval of T n , n = 1,..., L}. The number of samples within all periods is L. Performing an FFT transformation on this sequence, the obtained result is:

[0097]

[0098] In the strong satellite mode, the data length corresponding to one FFT is 1 ms. Assuming that the short-time correlation matched filtering generates a total of L = 8 partial sums, and the number of signal points is amplified to 16 through zero-padding operation (i.e., the signal period is padded to 2 ms), then the fundamental frequency of the FFT transform That is, the acquisition resolution in the carrier Doppler dimension is 500 Hz, and the step size is ω×L = 4000 Hz. In the weak satellite mode, first perform coherent integration on the partial sums generated by the short-time correlation matched filtering. Every 4 ms, L further accumulated partial sums are obtained, and then they are sent to the FFT module. Since coherent integration only folds the in-phase samples within the code period, the frequency resolution and step size at this time are the same as those in the strong satellite mode.

[0099] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention and are not intended to limit them. Although the present invention has been described in detail with reference to the preferred embodiments, those of ordinary skill in the art should understand that the technical solutions of the present invention can be modified or equivalently replaced without departing from the purpose and scope of the present technical solution, and they should all be covered within the scope of the claims of the present invention.

Claims

1. A Beidou D1 signal acquisition method with high dynamic range and high sensitivity, characterized in that: The method includes the following steps: S1: Determine the time window T DC , accumulate the digital intermediate frequency signal IF after analog-to-digital conversion ADC sampling within the time window, calculate the arithmetic mean at the end of the window to obtain the DC component of the intermediate frequency signal, and subtract this component from the subsequent incoming intermediate frequency signal, where T DC has a value of 2 n / fs, where fs is the sampling rate of the 60 MHz intermediate frequency signal; S2: The signal is multiplied by the carrier signal output by the mixer and the local oscillator to obtain a zero intermediate frequency I / Q signal; S3: Perform half-band filtering on the Beidou D1 signal to remove the noise components in the signal; for the B1I signal, the zero intermediate frequency signal is decimated by a factor of 2 after the first-stage half-band filtering, and then decimated by a factor of 2 after passing through the second-stage half-band filtering. The subsequent intermediate frequency signal is decimated by a factor of 3 after passing through the first-stage half-band filtering, and finally a zero intermediate frequency signal of fs / 12 = 5 MHz is obtained; S4: After passing through several levels of half-band filters and samplers in S3, the effective bit width of the signal gradually accumulates. To reduce the subsequent computational complexity, a new time window T is set Q , and the probability distribution of the sampled values is calculated within the window; due to the symmetry of the normal distribution, only the probability distributions of the +1 sample points and the +3 sample points are considered. If the number of +1 points is more than twice the number of +3 points, the threshold is lowered; otherwise, the threshold is raised, so that the sample point values always exhibit a normal distribution; S5: Decimate the quantized signal to twice the pseudo-code rate and transfer it to the acquisition circuit; S6: The acquisition circuit completes signal acquisition in the form of alternating strong satellite mode and weak satellite mode; in the strong satellite mode, no coherent integration is performed on the signal, that is, the coherent integration time is 1 ms; in the weak satellite mode, the coherent integration time of the signal is selected to be 4 ms; S7: Capture the D1 signal using the strong satellite mode, and store the data that enters the capture circuit in S5 into buffer A with a data capacity of V A . The value of V A is equal to the number of samples 2*N*(T noncoh + 1) required to complete one signal capture in the strong satellite mode. N is the number of chips within the pseudo-code period, and T noncoh is the non-coherent integration time. At the same time, store the data in S5 into buffer B with a data capacity of V B in parallel. The value of V B is equal to the number of samples 2*N*(T coh * T noncoh + 1) required to complete one signal capture in the weak satellite mode. N is the number of chips within the pseudo-code period, T coh = 4 is the number of coherent integration times in the weak satellite mode, and T noncoh is the non-coherent integration time. S8: Cache B only caches signal points within the time interval of [T0 + K * T D1 , T0 + K * T D1 + 4], that is, fixed-point intermittent caching, where T D1 = 20 is the navigation data period of the D1 signal, T0 is the cache start time, and K is a non-negative integer with a value range of 0 ≤ k ≤ T noncoh , and the units are all milliseconds; S9: Since the time for Cache A to store all the samples is significantly less than that of Cache B, after Cache A finishes storing, the acquisition engine writes the PRN numbers to be searched for satellites, and the acquisition circuit generates pseudo-codes based on the PRN numbers; after the pseudo-code generation is completed, the circuit retrieves the data in the cache for playback at the operating frequency f acq retrieves the data in the cache for playback; S10: First, carrier stripping is performed on the signal according to the theoretical intermediate frequency value of the RF chip to obtain two zero intermediate frequency signals of I and Q; S11: Input the playback signal points into the N-bit shift register in the matched filtering module, where N is equal to twice the code length value of the signal; when the playback signal point count is greater than or equal to N, perform short-time correlation matched filtering on the signal according to the pseudo-code sequence generated in S10 and obtain 2 K short-time correlation sums, where K is the radix of the FFT; since the coherent integration time is 1 ms in the strong star mode and the algorithm is not affected by the NH code modulation, transfer the 2 K short-time correlation sums directly to the FFT module; S12: Take the 2 in S11 K short-term correlations and perform FFT transformation to obtain 2 K signal segments in the frequency domain, and perform caching operations. After the next correlation operation ends, the new signal segments will be accumulated with the cached results; when the number of accumulations reaches T noncoh , calculate the average value E of the incoherent integration energy of N code phases mean , calculate the maximum integration energy value E max , and compare the ratio E max / E mean with a preset threshold for energy decision; S13: Limited by hardware resources, the satellite PRN dimension must be a serial search. After the current satellite is captured, replay the samples stored in V A again, repeat S9 - S12, and enter the capture process of the next satellite; S14: Turn on the tracking channel for the successfully captured satellite and read the dump timestamp T returned after the first integration of the channel is completed. dump , and combine it with the timestamp T released when the signal caching is completed. store , and calculate the code starting point of the signal. After all BDS satellites on the satellite search list have completed one round of acquisition in the strong satellite mode, due to V B The time required to store one round of sampled points for acquisition is relatively long, and it is necessary to detect whether V B has been stored to the specified capacity. If V B has not stored enough sampled points, repeat the strong satellite mode to capture the signal: take out the new sampled points in V A and repeat the above S9 - S14; otherwise, start the weak satellite mode to capture the D1 signal; S15: When V B the cache data volume reaches 2 * N * (T coh * T noncoh + 1), the weak satellite acquisition mode is enabled; the first step of the weak satellite mode is still carrier stripping and short-time correlation matched filtering. Repeat S9, S10, and S11. After the short-time correlation of the data corresponding to 1 ms is generated, since the coherent integration time is greater than 1 ms, the influence of the NH code modulation cannot be ignored, and the NH code stripping is performed on the correlation sum; S16: The signals used in two adjacent rounds of coherent integration are not continuous in time and are both at a fixed phase within the D1 data period. There is no need to search for the NH code phase using the method of the sliding code start point. Only the following 8 NH code combinations are used to perform NH code stripping on the correlation sum: {1,1,1,1}, {-1,1,1,1}, {1,-1,1,1}, {1,1,-1,1}, {1,1,1,-1}, {-1,-1,1,1}, {1,-1,1,-1}, {1,-1,-1,1}; S17: Store the coherent sum after stripping the NH code into buffer V C ; V C The accumulated data stored in V will perform a coherent accumulation operation with the NH-stripped data output by the short-time correlation filter next time until the number of coherent times reaches N coh ; Then transmit the generated coherent sum to the FFT module S18: Repeat S11 - S13 until signal acquisition in the weak satellite mode for all satellites on the satellite search list is completed; S19: Since in the weak satellite mode, the time span of the V B cache signal becomes significantly longer. Under high dynamics, there is a certain probability that the difference between the pseudo-code phase and Doppler value searched when the signal acquisition is successful and the actual code phase and Doppler at the time of turning on the tracking channel will exceed the loop tracking range; using the three-way integral values of EPL returned by the baseband, set a window with a length of 1.5 chip lengths to slide the signal pseudo-code phase; the number of sliding times is M, and the value of M is (dT * a * f code / f carrier ) / 1.5, where a is the Doppler change rate calculated by the receiver PVT. Assume that the receiver is already in the positioning state, f code is the signal pseudo-code frequency, f carrier is the signal carrier frequency, dT = T dump -T store is the time span from acquisition to tracking in S14; S20: After M times of pseudo-code sliding, read the integrated energy values corresponding to 3*M different pseudo-code phases stored in the buffer, find the code phase corresponding to the maximum energy, and configure the baseband to slide the signal pseudo-code to this position; slide in the Doppler dimension of the signal and find the real-time Doppler frequency of the signal according to the returned integrated energy values; after reconfiguring the code phase and Doppler, the signal transfer is successful and enters the tracking state.

2. A Beidou D1 signal acquisition method with high dynamic range and high sensitivity according to claim 1, characterized in that: In the strong satellite mode, the signal acquisition process is as follows: Continuous caching, carrier stripping, short-time correlation matched filtering, spectrum analysis, non-coherent integration, and energy decision; In the weak satellite mode, the basic signal acquisition process is as follows: Fixed-point discontinuous caching, carrier stripping, short-time correlation matched filtering, NH code stripping, coherent integration, spectrum analysis, non-coherent integration, energy decision, Doppler and pseudo-code compensation.

3. A Beidou D1 signal acquisition method with high dynamic range and high sensitivity according to claim 1, characterized in that: For the intermediate frequency signal input to the ADC of the RF chip, the DC component is first removed to prevent the signal sample distribution from being affected and interfering with the decision threshold of the subsequent quantization circuit. After passing through several stages of filters and samplers, the effective bit width of the digital signal will gradually accumulate and needs to be re-quantized to 2 bits through automatic gain control (AGC). The AGC logic accumulates and counts the magnitude (MAG) bits, calculates the proportional distribution within the corresponding time of the AGC. Under the characteristic that the intermediate frequency signal after DC removal follows a Gaussian distribution, if the proportional distribution exceeds 17%, it indicates that the gain is too large and a feedback signal needs to be generated to reduce the gain of the programmable gain amplifier (PGA). Conversely, if it is less than 17%, the PGA gain needs to be increased.

4. A Beidou D1 signal acquisition method with high dynamic range and high sensitivity according to claim 1, characterized in that: The transfer function of the filter is the conjugate of the signal spectrum. The filter must be completely matched with the spectrum of the input signal. Assuming the sampling sequence of the input signal is represented by s(n), the output of the matched filter is expressed as: To ensure that the accumulated sample sequence still contains frequency information, only the sequence after pseudo-code stripping is segmented and accumulated, and then the fast Fourier transform (FFT) is used to perform spectral analysis on these partial sums. Considering the Beidou intermediate frequency data sampling sequence with a length of 1 ms, denoted as: where T s is the sampling interval, C is the pseudo-random code, ω IF is the intermediate frequency carrier frequency, and is the initial phase; m takes values from 0 to N - 1, indicating that the number of samples within 1 ms is N, and there is no navigation message bit transition within one pseudo-random code period; The length of each short-time correlation integral is m samples, and T p represents the duration of the short-time correlation. Then the output of the accumulation of the i-th short-time correlation is as follows: The short-time correlation result is a digital sequence {I p - jQ n , n = 1, ..., L} with a sampling interval of T. The number of samples in all periods of the sequence is L. If the FFT transform is performed on this sequence, the resulting value is: n ​ In the strong satellite mode, the data length corresponding to one FFT is 1 ms. Assuming that the short-time correlation matched filtering generates a total of L = 8 partial sums, and the number of signal points is amplified to 16 through zero-padding operation, that is, the signal period is padded to 2 ms, then the basic frequency of the FFT transform is the acquisition resolution of the carrier Doppler dimension is 500 Hz, and the step size is ω × L = 4000 Hz; in the weak satellite mode, first perform coherent integration on the partial sums generated by the short-time correlation matched filtering, and obtain L further accumulated partial sums every 4 ms, and then send them to the FFT module; Coherent integration only folds the samples with the same phase within the code period. At this time, both the frequency resolution and the step size are the same as those in the strong star mode.

Images