Current source high frequency link matrix inverter topology with soft switching

Abstract

The present invention discloses a current-source high-frequency link matrix inverter topology with soft switching, comprising a sequentially connected power supply network, a current-source H-bridge, a series resonant tank, a high-frequency transformer, a matrix converter, a CL filter, and a load. The present invention uses series resonance to match the leakage inductance current of the high-frequency transformer with the DC-side input current, thereby avoiding large voltage spikes on the primary side of the high-frequency transformer and achieving zero-current switching of the current-source H-bridge. The present invention is also well-suited for high-frequency transformers with large leakage inductance. Furthermore, the proposed decoupling-based SPWM and SVM modulation methods for current-source matrix inverters feature mixed high- and low-frequency modulation, enabling safe commutation, effectively reducing the switching frequency of the matrix converter, and providing soft switching conditions for all switches in the matrix converter.

Description

Technical Field

[0001] The present invention relates to the field of power electronic power converter topology and modulation thereof, in particular to a current source type high frequency link matrix inverter topology structure with soft switching and a modulation method thereof. Background Art

[0002] An inverter is a power conversion device that uses power semiconductors to convert direct current (DC) into alternating current (AC). Traditional inverters use power-frequency transformers for electrical isolation, while high-frequency link inverters replace them with high-frequency transformers, overcoming the drawbacks of traditional transformers, such as bulk, noise, and cost. To improve voltage utilization between the DC and AC sides, a two-stage power conversion system is typically employed: a DAB-based DC / DC converter at the first stage and a DC / AC inverter at the second. This type of converter requires electrolytic capacitors as an intermediate energy storage element. Compared to these traditional converters, matrix converters eliminate the intermediate energy storage element and utilize bidirectional switches, enabling bidirectional energy flow. They offer a compact structure, small size, high efficiency, and independent control of the output voltage amplitude and frequency.

[0003] A current source inverter uses a large inductor in series on the DC side. Because the DC side current is essentially pulsating, the large inductor can be approximated as a DC current source. Compared to voltage source inverters, current source inverters offer boost characteristics. Furthermore, the energy storage element in current source inverters is an inductor, resulting in a longer system lifespan than voltage source inverters, which use electrolytic capacitors as the energy storage element. To reduce the size of the AC filter and DC side inductor, the switching frequency of the current source converter can be increased. Higher switching frequencies also help improve line current quality. However, increasing the switching frequency increases switching losses and electromagnetic interference in hard-switching converters. Soft switching technology offers a solution to these problems.

[0004] Due to the leakage inductance of the high-frequency transformer, when the transformer leakage current does not match the DC input current, a large voltage spike is generated on the primary side of the high-frequency transformer. The paper "Current-Source Three-Phase Three-Bridge-Leg High-Frequency Link Matrix Inverter Topology, Modulation, and Parallel Control" describes a current-source high-frequency link matrix inverter and SPWM modulation based on decoupling principles, but this paper does not consider this issue. Furthermore, when the high-frequency link matrix inverter commutates, a large voltage overshoot is generated on the power transistors of the matrix converter on the secondary side of the transformer. Therefore, safe commutation of the transformer secondary side matrix converter has always been a technical difficulty that has hindered the widespread adoption of high-frequency link inverters. Currently, there are mainly the following safe commutation strategies: ① By adding active clamps to suppress voltage overshoot, soft switching can be achieved, but the introduced clamping circuit increases the cost, and the added controllable power tube also makes the control more complicated; ② The unipolar and bipolar phase-shift control strategies use the commutation overlap of the matrix converter to achieve natural commutation of the inductor current and realize ZVS of the power tube, but there are problems such as the difficulty in controlling the commutation overlap time; ③ Introducing a series resonant circuit in the front-stage inverter to achieve soft commutation of the power tube. At this time, the power tube switching is required to occur at the zero current moment, and controlling the output energy requires judging the resonant working state of the resonant circuit, which makes the control method complicated. Summary of the Invention

[0005] The technical problem to be solved by the present invention is to provide a current source type high frequency link matrix inverter topology with soft switching to achieve soft switching of the inverter and safe commutation of the matrix converter.

[0006] In order to solve the above technical problems, the technical solution adopted by the present invention is: a current source type high frequency link matrix inverter topology with soft switching, including a power supply network, a current type H bridge, a series resonant tank, a high frequency transformer, a matrix converter, a CL type filter and a load connected in sequence; the power supply network includes a DC input voltage source U i and energy storage inductor L dc The series resonant tank includes a resonant capacitor C0 and a high-frequency transformer leakage inductance L lk The current-type H-bridge includes a first inverter bridge arm composed of a controllable switch tube S1 and a diode D1 connected in series, a second inverter bridge arm composed of a controllable switch tube S2 and a diode D2 connected in series, a third inverter bridge arm composed of a controllable switch tube S3 and a diode D3 connected in series, and a fourth inverter bridge arm composed of a controllable switch tube S4 and a diode D4 connected in series; the matrix converter includes a controllable switch tube S 1a , controllable switch tube S 4b , controllable switch tube S 4a , controllable switch tube S 1b , controllable switch tube S 3a , controllable switch tube S 6b , controllable switch tube S 6a , controllable switch tube S3b , controllable switch tube S 5a , controllable switch tube S 2b , controllable switch tube S 2a , controllable switch tube S 5b , controllable switch tube S 1a , controllable switch tube S 4b , controllable switch tube S 4a , controllable switch tube S 1b Forming the A phase bridge arm, the controllable switch tube S 3a , controllable switch tube S 6b , controllable switch tube S 6a , controllable switch tube S 3b , controllable switch tube S 5a Forming the B phase bridge arm, the controllable switch tube S 5a , controllable switch tube S 2b , controllable switch tube S 2a , controllable switch tube S 5b The C-phase bridge arm is formed, and the matrix converter is decomposed into two groups of positive and negative current-type converters. The controllable switch tube S 1a ~Controllable switch tube S 6a For the positive group switch tube, the controllable switch tube S 1b ~Controllable switch tube S 6b It is a negative group switch tube; the CL type filter includes a first capacitor C1, a second capacitor C2, a third capacitor C3, a first inductor L1, a second inductor L2 and a third inductor L3; the load includes a first load R1, a second load R2 and a third load R3.

[0007] A further improvement of the technical solution of the present invention is that: the DC input voltage source U i The positive electrode and the energy storage inductor L dc One end of the energy storage inductor L dc The other end is connected to the collector of the controllable switch tube S1 and the collector of the controllable switch tube S3 respectively; the DC input voltage source U i The negative electrode of the resonant capacitor C0 is connected to the negative electrode of the diode D2 and the negative electrode of the diode D4 respectively; one end of the resonant capacitor C0 is connected to the emitter of the controllable switch tube S2 and the negative electrode of the diode D1; the other end of the resonant capacitor C0 is connected to the leakage inductance L of the high-frequency transformer lk One end of the high-frequency transformer primary side is connected to the high-frequency transformer leakage inductance L lk The other end of the high-frequency transformer primary side is connected to the emitter of the controllable switch tube S4 and the negative electrode of the diode D3; one end of the high-frequency transformer secondary side is respectively connected to the controllable switch tube S 1a Collector, controllable switch tube S 3a Collector, controllable switch tube S 5aThe other end of the secondary side of the high frequency transformer is connected to the collector of the controllable switch tube S 1b Collector, controllable switch tube S 3b Collector, controllable switch tube S 5b The collector of the controllable switch tube S 1a The emitter and the controllable switch tube S 4b The emitter of the controllable switch tube S 3a The emitter and the controllable switch tube S 6b The emitter of the controllable switch tube S 5a The emitter and the controllable switch tube S 2b The emitter of the controllable switch tube S 1b The emitter and the controllable switch tube S 4a The emitter of the controllable switch tube S 3b The emitter and the controllable switch tube S 6a The emitter of the controllable switch tube S 5b The emitter and the controllable switch tube S 2a The emitter of the controllable switch tube S 4a The collector and the controllable switch tube S 4b The collector of the controllable switch tube S 6a The collector and the controllable switch tube S 6b The collector of the controllable switch tube S 2a The collector and the controllable switch tube S 2b The collector of the controllable switch tube S 4a The collector and the controllable switch tube S 4b The collector of the controllable switch S is connected to one end of the first capacitor C1 and one end of the first inductor L1 respectively; 6a The collector and the controllable switch tube S 6b The collector of the controllable switch S is connected to one end of the second capacitor C2 and one end of the second inductor L2 respectively; 2a The collector and the controllable switch tube S 2b The collector of the first capacitor C1 is connected to one end of the third capacitor C3 and one end of the third inductor L3 respectively; the other end of the first capacitor C1 is connected to the other end of the second capacitor C2 and the other end of the third capacitor C3 respectively; the other end of the first inductor L1 is connected to one end of the first load R1; the other end of the second inductor L2 is connected to one end of the second load R2; the other end of the third inductor L3 is connected to one end of the third load R3; the other end of the first load R1 is connected to the other end of the second load R2 and the other end of the third load R3 respectively.

[0008] The technical solution of the present invention is further improved in that: the topological structure adopts a current-type SPWM modulation method based on the decoupling idea, in which the modulation wave is a three-phase line voltage sine wave, the carrier is a high-frequency sawtooth wave, the three-phase modulation wave is compared with the carrier to generate a high-frequency signal, and the high-frequency signal is synthesized into a high-frequency square wave through logical processing; the current-type H-bridge on the primary side of the high-frequency transformer converts the DC current into a high-frequency square wave current under the high-frequency square wave drive, and its driving signal is two high-frequency square wave coupling signals V n and V p , V n and V p is a pair of high-frequency square waves with overlapping on-time and duty cycle greater than 50%, V n and V p As the driving signal of the controllable switch tubes S1~S4 and the coupling signal of the matrix converter; the matrix converter is decomposed into two groups of ordinary three-phase current type inverters, namely the positive group inverter and the negative group inverter. The positive group inverter consists of the controllable switch tube S 1a ~Controllable switch tube S 6a The negative group inverter is composed of controllable switch tube S 1b ~Controllable switch tube S 6b The driving signal of each group of the current-source three-phase inverter is obtained by logically combining six SPWM signals with coupling signals.

[0009] The further improvement of the technical solution of the present invention is that: the topological structure adopts a current-type SVM modulation method based on the decoupling idea, in which the modulation wave is a three-phase phase voltage sine wave, and the current-type SVM modulation method based on the decoupling idea includes sector judgment, vector action time calculation and switch signal generation. The sector judgment is specifically as follows: within one power frequency cycle, the three-phase modulation wave is divided into 12 sectors according to the size of the instantaneous value; the calculation of the vector action time is divided into the calculation of the effective vector and the zero vector action time; the switch signal generation includes six basic switch signals and one zero vector signal, and the zero vector signal is logically transformed to generate two coupling signals as the driving signal of the four switch tubes of the current-type H bridge; the post-stage matrix converter is decomposed into two groups of ordinary three-phase current-type converters, namely the positive group converter and the negative group converter, and the positive group converter consists of a controllable switch tube S 1a ~Controllable switch tube S 6a The negative group converter consists of a controllable switch tube S 1b ~Controllable switch tube S 6b The driving signals of all the switch tubes of the matrix converter are synthesized by coupling logic of the coupling signal and the six basic switch signals.

[0010] Due to the adoption of the above technical solution, the technical advancements achieved by the present invention are:

[0011] 1. The present invention uses series resonance to match the leakage inductance current of the high-frequency transformer with the DC-side input current, avoiding large voltage spikes on the primary side of the high-frequency transformer and achieving zero-current switching of the current-type H-bridge. The present invention is also well-suited for high-frequency transformers with large leakage inductance; at the same time, the transformer current is continuous, avoiding discontinuous operation mode.

[0012] 2. The SPWM and SVM modulation methods based on the decoupling concept proposed in the present invention, which are suitable for current source matrix inverters, have the characteristics of high- and low-frequency mixed modulation, can achieve safe commutation, effectively reduce the switching times of the matrix converter, and provide soft switching conditions for all switching tubes of the matrix converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] Figure 1 The present invention provides a current source type high frequency link matrix inverter topology with soft switching;

[0014] Figure 2 This is a topological decoupling diagram of the current source type high frequency link matrix inverter of the present invention;

[0015] Figure 3 The present invention provides a front-stage H-bridge modulation of the SPWM modulation method based on the decoupling idea;

[0016] Figure 4 Schematic diagram of the SVM modulation method based on the decoupling concept provided by the present invention;

[0017] Figure 5 A waveform diagram of the inverter working process within a high-frequency switching cycle provided by the present invention;

[0018] FIG6( a ) is a circuit diagram of mode 1 of a positive group converter operating within a high-frequency switching cycle provided by the present invention;

[0019] FIG6( b ) is a circuit diagram of mode 2 of the positive group converter operating within a high-frequency switching cycle provided by the present invention;

[0020] FIG6( c ) is a circuit diagram of mode 3 of the positive group converter operating within a high-frequency switching cycle provided by the present invention;

[0021] FIG6( d ) is a circuit diagram of mode 4 of the positive group converter operating within a high-frequency switching cycle provided by the present invention;

[0022] FIG6( e ) is a circuit diagram of mode 5 of the positive group converter operating within a high-frequency switching cycle provided by the present invention; DETAILED DESCRIPTION

[0023] The present invention is described in further detail below in conjunction with the embodiments:

[0024] like Figure 1 As shown, the current source type high frequency link matrix inverter topology with soft switching includes a power supply network, a current type H bridge, a series resonant tank, a high frequency transformer, a matrix converter, a CL type filter and a load connected in sequence; the power supply network includes a DC input voltage source U i and energy storage inductor L dc The series resonant tank includes a resonant capacitor C0 and a high-frequency transformer leakage inductance L lk The current-type H-bridge includes a first inverter bridge arm composed of a controllable switch tube S1 and a diode D1 connected in series, a second inverter bridge arm composed of a controllable switch tube S2 and a diode D2 connected in series, a third inverter bridge arm composed of a controllable switch tube S3 and a diode D3 connected in series, and a fourth inverter bridge arm composed of a controllable switch tube S4 and a diode D4 connected in series; the matrix converter includes a controllable switch tube S 1a , controllable switch tube S 4b , controllable switch tube S 4a , controllable switch tube S 1b , controllable switch tube S 3a , controllable switch tube S 6b , controllable switch tube S 6a , controllable switch tube S 3b , controllable switch tube S 5a , controllable switch tube S 2b , controllable switch tube S 2a , controllable switch tube S 5b , controllable switch tube S 1a , controllable switch tube S 4b , controllable switch tube S 4a , controllable switch tube S 1b Forming the A phase bridge arm, the controllable switch tube S 3a , controllable switch tube S 6b , controllable switch tube S 6a , controllable switch tube S 3b , controllable switch tube S 5a Forming the B phase bridge arm, the controllable switch tube S 5a , controllable switch tube S 2b , controllable switch tube S 2a , controllable switch tube S 5b The C-phase bridge arm is formed; the CL filter includes a first capacitor C1, a second capacitor C2, a third capacitor C3, a first inductor L1, a second inductor L2 and a third inductor L3; the load includes a first load R1, a second load R2 and a third load R3.

[0025] The DC input voltage source U i The positive electrode and the energy storage inductor L dc One end of the energy storage inductor Ldc The other end is connected to the collector of the controllable switch tube S1 and the collector of the controllable switch tube S3 respectively; the DC input voltage source U i The negative electrode of the resonant capacitor C0 is connected to the negative electrode of the diode D2 and the negative electrode of the diode D4 respectively; one end of the resonant capacitor C0 is connected to the emitter of the controllable switch tube S2 and the negative electrode of the diode D1; the other end of the resonant capacitor C0 is connected to the leakage inductance L of the high-frequency transformer lk One end of the high-frequency transformer primary side is connected to the high-frequency transformer leakage inductance L lk The other end of the high-frequency transformer primary side is connected to the emitter of the controllable switch tube S4 and the negative electrode of the diode D3; one end of the high-frequency transformer secondary side is respectively connected to the emitter of the controllable switch tube S 1a Collector, controllable switch tube S 3a Collector, controllable switch tube S 5a The other end of the secondary side of the high frequency transformer is connected to the collector of the controllable switch tube S 1b Collector, controllable switch tube S 3b Collector, controllable switch tube S 5b The collector of the controllable switch tube S 1a The emitter and the controllable switch tube S 4b The emitter of the controllable switch tube S 3a The emitter and the controllable switch tube S 6b The emitter of the controllable switch tube S 5a The emitter and the controllable switch tube S 2b The emitter of the controllable switch tube S 1b The emitter and the controllable switch tube S 4a The emitter of the controllable switch tube S 3b The emitter and the controllable switch tube S 6a The emitter of the controllable switch tube S 5b The emitter and the controllable switch tube S 2a The emitter of the controllable switch tube S 4a The collector and the controllable switch tube S 4b The collector of the controllable switch tube S 6a The collector and the controllable switch tube S 6b The collector of the controllable switch tube S 2a The collector and the controllable switch tube S 2b The collector of the controllable switch tube S 4a The collector and the controllable switch tube S 4b The collector of the controllable switch S is connected to one end of the first capacitor C1 and one end of the first inductor L1 respectively; 6a The collector and the controllable switch tube S 6bThe collector of the controllable switch S is connected to one end of the second capacitor C2 and one end of the second inductor L2 respectively; 2a The collector and the controllable switch tube S 2b The collector of the first capacitor C1 is connected to one end of the third capacitor C3 and one end of the third inductor L3 respectively; the other end of the first capacitor C1 is connected to the other end of the second capacitor C2 and the other end of the third capacitor C3 respectively; the other end of the first inductor L1 is connected to one end of the first load R1; the other end of the second inductor L2 is connected to one end of the second load R2; the other end of the third inductor L3 is connected to one end of the third load R3; the other end of the first load R1 is connected to the other end of the second load R2 and the other end of the third load R3 respectively.

[0026] The present invention adopts the following two modulation methods for the above topology:

[0027] 1. A current-mode SPWM modulation method based on the decoupling concept is adopted. In the current-mode SPWM modulation method based on the decoupling concept, the modulation wave is a three-phase line voltage sine wave, and the carrier is a high-frequency sawtooth wave. The three-phase modulation wave is compared with the carrier to generate a high-frequency signal, and the high-frequency signal is synthesized into a high-frequency square wave through logical processing; Figure 3 The principle of the pre-stage current-type H-bridge modulation of the current-type SPWM modulation method based on the decoupling idea is given, which gives two high-frequency square wave coupling signals V n and V p The current type H bridge on the primary side of the high-frequency transformer converts the DC current into a high-frequency square wave current under the high-frequency square wave drive. Its driving signal is two high-frequency square wave coupled signals V n and V p , V n and V p It is a pair of high-frequency square waves with overlapping conduction time and duty cycle greater than 50%. The overlapping time is to prevent the DC side inductor current from being open-circuited while maintaining the DC side input current continuity. The two high-frequency square waves are coupled to the signal V n and V p As the driving signal of the controllable switch tubes S1~S4 and the coupling signal of the matrix converter; Figure 2 As shown, the matrix converter is decomposed into two groups of ordinary three-phase current source inverters, namely the positive group inverter and the negative group inverter. The positive group inverter consists of controllable switch tube S 1a ~Controllable switch tube S 6a The negative group inverter is composed of controllable switch tube S 1b ~Controllable switch tube S 6b The driving signal of each group of the current-source three-phase inverter is obtained by logically combining six SPWM signals with coupling signals.

[0028] Second, the current-type SVM modulation method based on the decoupling idea is adopted. Figure 4 This is the schematic diagram of the SVM modulation method, which shows the correspondence between the effective vectors and the matrix converter switches in 12 sectors. The 12 sectors are further divided based on the six sectors, according to the magnitude of the modulation waveform line voltage. After this, the vector action times must be calculated, including the action times of the effective vector and the zero vector. The action times of the vectors vary in different sectors. After determining the vector action times, the action times of the vectors in each sector must also be determined. The basic principle is to first calculate the zero vector, then the two effective vectors, and finally the zero vector. Finally, based on the corresponding vector action times in each sector, the corresponding drive signals, namely the six basic switch signals, are generated. While synthesizing the six basic switch signals, the zero vector switch signal must also be synthesized. The zero vector signal is logically synthesized from the six basic switch signals. The zero vector switch signal is then logically processed to generate two coupling signals. These two coupling signals are then logically ANDed with the six basic switch signals to generate the drive signals for the bidirectional switches in the matrix converter. Considering that the polarity of the high-frequency transformer's output voltage and the direction of the current depend on the switching mode of the H-bridge, the two coupling signals serve as the drive signals for the preceding current-mode H-bridge.

[0029] Figure 5 This is a waveform diagram of the inverter working state principle within a high-frequency switching cycle of the SPWM modulation based on the decoupling idea provided by the present invention. pah 、S pal 、S nah 、S nal 、S pbh 、S pbl 、S nbh 、S nbl 、S pch 、S pcl 、S nch 、S ncl They are the controllable switch tubes S in the matrix converter. 1a , controllable switch tube S 4a , controllable switch tube S 1b , controllable switch tube S 4b , controllable switch tube S 3a , controllable switch tube S 6a , controllable switch tube S 3b , controllable switch tube S 6b , controllable switch tube S 5a , controllable switch tube S 2a , controllable switch tube S 5b and controllable switch tube S 2bThe corresponding drive signals, Q1~Q4 are the drive signals corresponding to the controllable switch tubes S1, S2, S3 and S4 in the current type H bridge, u p 、u s and i0 are the primary voltage, secondary voltage and primary current of the high-frequency transformer respectively, and u c0 is the voltage at the resonant capacitor C0.

[0030] Figure 6(a)-Figure 6(e) To correspond Figure 5 The modal circuit diagram of the positive group converter working in a high-frequency cycle of SPWM modulation based on the decoupling concept. Assuming that all components in the topology are ideal components, according to the working principle, there are five positive group working states in a high-frequency cycle. The specific modal analysis is as follows:

[0031] Mode 1 [t0~t1]:

[0032] As shown in Figure 6(a), the current-mode H-bridge and the A-phase bridge arm of the matrix converter are in the full-on state, and the primary and secondary voltages of the high-frequency transformer are zero, providing a zero-voltage switching condition for the next stage. The primary and secondary currents of the high-frequency transformer are equal to the DC-side inductor current i dc At this time, the resonant capacitor C0 is charged with a constant current, u c0 The primary current of the high-frequency transformer flows through the controllable switch tubes S1 and S4 of the current-type H-bridge, and no current flows through S2 and S3, providing a zero-current shutdown condition. 1a 、S 4b 、S 4a 、S 1b , the output filter capacitor provides a current path for the output filter inductor and supplies power to the load. No power is transferred in this stage.

[0033] Mode 2 [t1~t2]:

[0034] As shown in Figure 6(b), at time t1, the controllable switches S2 and S4 of the current-mode H-bridge achieve zero current shutdown, and the controllable switches S 4b 、S 4a 、S 1b Achieved zero voltage shutdown, controllable switch tube S 6a Zero voltage switching is achieved, and the voltage between the primary and secondary sides of the high-frequency transformer is equal to u ab The DC side inductor current flows through controllable switches S1 and S4, and the resonant capacitor C0 continues to charge, causing the voltage to increase further. During this stage, the DC power supply transfers power to the load through the high-frequency transformer.

[0035] Mode 3 [t2~t3]:

[0036] As shown in Figure 6(c), the process is similar to mode 2. The DC side inductor current passes through the controllable switch tubes S1 and S4, the resonant capacitor C0 continues to charge, and the voltage further increases. At time t2, due to the controllable switch tube S 3b The reverse blocking effect of the body diode in the controllable switch tube S 6a Zero voltage shutdown, while the high frequency transformer leakage inductance limits the rate of rise of the secondary current, making the controllable switch tube S 2a It can be turned on with zero current. At this time, the voltage between the primary and secondary sides of the high-frequency transformer is equal to u ac , the secondary current of the high-frequency transformer flows through the controllable switch tube S 1a , controllable switch tube S 4b The body diode and controllable switch tube S 2a and controllable switch tube S 5b The body diode in the MOSFET transmits DC power to the load.

[0037] Mode 4 [t3~t4]:

[0038] As shown in Figure 6(d), at time t3, all current-type H-bridge switches are in the on state. Since the DC side inductance limits the current rise rate, the controllable switches S2 and S3 achieve zero current turn-on. At the same time, the switch tubes of the A-phase bridge arm of the matrix converter are also in the full on state, with the controllable switch tube S 4b Zero voltage turn-on, and the high-frequency transformer leakage inductance limits the rate of increase of the secondary current, the controllable switch tube S 1b 、S 4a Able to turn on at zero current, controllable switch tube S 5b The reverse blocking characteristics of the body diode in the controllable switch tube S 2a It is capable of zero-voltage shutdown. At the beginning of this phase, the resonant capacitor resonates with the primary leakage inductance of the high-frequency transformer, gradually reducing the current flowing through switches S1 and S4. The DC-side inductor current gradually shifts from these controllable switches S1 and S4 to controllable switches S2 and S3. By the end of this phase, the current flowing through these controllable switches S1 and S4 drops to zero, the high-frequency transformer leakage current equals the DC-side inductor current, and the resonant capacitor voltage is approximately equal to its value at time t3.

[0039] Mode 5 [t4~t5]:

[0040] As shown in Figure 6(e), the process is similar to that in Mode 1. In this stage, the DC-side inductor current flowing through controllable switches S2 and S3 remains essentially unchanged. Compared to Mode 1, the direction of the high-frequency transformer leakage current changes, causing the resonant capacitor to begin a constant current discharge, and the voltage gradually decreases. By time t5, the resonant capacitor voltage has dropped to near zero.

[0041] The operating principle of decoupling-based SVM modulation is essentially the same as that of decoupling-based SPWM modulation within a high-frequency switching cycle. The effective vector of decoupling-based SVM modulation corresponds to the operating phase t1 to t3 in the diagram, while the zero vector corresponds to the operating phases t0 to t1 and t3 to t5 in the diagram. The difference lies in the SVM effective vector's ability to more flexibly adjust the order of vector action, thereby changing the order in which the controllable switches operate.

[0042] Compared with the prior art, the current source high frequency link matrix inverter topology and pulse width modulation method provided by the present invention have the following advantages:

[0043] By resonating an external resonant capacitor with the inherent leakage inductance of the high-frequency transformer, the voltage spike problem caused by the mismatch between the high-frequency transformer leakage inductance current and the DC input current can be avoided, thus achieving zero-current switching of the current-type H-bridge. This method is also well-suited for high-frequency transformers with large leakage inductance; at the same time, the transformer current is continuous, avoiding discontinuous operation. Existing modulation methods for current-source high-frequency link matrix inverters mostly use SVPWM. The SVPWM method mentioned in the document "Current-Fed Isolated Three-Phase Matrix-Type Grid Inverter With Soft-Switching Capability" increases the harmonic content of the output current while eliminating the mismatch between the DC inductance current and the transformer leakage inductance current, and also places high demands on the transformer leakage inductance. The decoupling-based SPWM and SVM modulation methods proposed in this patent are suitable for current-source matrix inverters and feature mixed high- and low-frequency modulation, effectively reducing the switching frequency of the matrix converter.

[0044] The above description is merely a preferred embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any changes or substitutions that can be easily conceived by a person skilled in the art within the technical scope disclosed in the present invention should be included in the scope of protection of the present invention. Therefore, the scope of protection of the present invention should be based on the scope of protection of the claims.

Claims

1. A current source high frequency link matrix inverter topology with soft switching, characterized by: It includes a power supply network, a current type H bridge, a series resonant tank, a high frequency transformer, a matrix converter, a CL type filter and a load connected in sequence; the power supply network includes a DC input voltage source U i and energy storage inductor L dc The series resonant tank includes a resonant capacitor C0 and a high-frequency transformer leakage inductance L lk The current-type H-bridge includes a first inverter bridge arm composed of a controllable switch tube S1 and a diode D1 connected in series, a second inverter bridge arm composed of a controllable switch tube S2 and a diode D2 connected in series, a third inverter bridge arm composed of a controllable switch tube S3 and a diode D3 connected in series, and a fourth inverter bridge arm composed of a controllable switch tube S4 and a diode D4 connected in series; the matrix converter includes a controllable switch tube S 1a , controllable switch tube S 4b , controllable switch tube S 4a , controllable switch tube S 1b , controllable switch tube S 3a , controllable switch tube S 6b , controllable switch tube S 6a , controllable switch tube S 3b , controllable switch tube S 5a , controllable switch tube S 2b , controllable switch tube S 2a , controllable switch tube S 5b , controllable switch tube S 1a , controllable switch tube S 4b , controllable switch tube S 4a , controllable switch tube S 1b Forming the A-phase bridge arm, the controllable switch tube S 3a , controllable switch tube S 6b , controllable switch tube S 6a , controllable switch tube S 3b , controllable switch tube S 5a Forming the B phase bridge arm, the controllable switch tube S 5a , controllable switch tube S 2b , controllable switch tube S 2a , controllable switch tube S 5b The CL filter comprises a first capacitor C1, a second capacitor C2, a third capacitor C3, a first inductor L1, a second inductor L2 and a third inductor L3; the load comprises a first load R1, a second load R2 and a third load R3; the DC input voltage source U i The positive electrode and the energy storage inductor L dc One end of the energy storage inductor L dc The other end is connected to the collector of the controllable switch tube S1 and the collector of the controllable switch tube S3 respectively; the DC input voltage source U i The negative electrode of the resonant capacitor C0 is connected to the negative electrode of the diode D2 and the negative electrode of the diode D4 respectively; one end of the resonant capacitor C0 is connected to the emitter of the controllable switch tube S2 and the negative electrode of the diode D1; the other end of the resonant capacitor C0 is connected to the leakage inductance L of the high-frequency transformer lk One end of the high-frequency transformer primary side is connected to the high-frequency transformer leakage inductance L lk The other end of the high-frequency transformer primary side is connected to the collector of the controllable switch tube S4 and the negative electrode of the diode D3; one end of the high-frequency transformer secondary side is respectively connected to the collector of the controllable switch tube S 1a Collector, controllable switch tube S 3a Collector, controllable switch tube S 5a The other end of the secondary side of the high frequency transformer is connected to the collector of the controllable switch tube S 1b Collector, controllable switch tube S 3b Collector, controllable switch tube S 5b The collector of the controllable switch tube S 1a The emitter and the controllable switch tube S 4b The emitter of the controllable switch tube S 3a The emitter and the controllable switch tube S 6b The emitter of the controllable switch tube S 5a The emitter and the controllable switch tube S 2b The emitter of the controllable switch tube S 1b The emitter and the controllable switch tube S 4a The emitter of the controllable switch tube S 3b The emitter and the controllable switch tube S 6a The emitter of the controllable switch tube S 5b The emitter and the controllable switch tube S 2a The emitter of the controllable switch tube S 4a The collector and the controllable switch tube S 4b The collector of the controllable switch tube S 6a The collector and the controllable switch tube S 6b The collector of the controllable switch tube S 2a The collector and the controllable switch tube S 2b The collector of the controllable switch tube S 4a The collector and the controllable switch tube S 4b The collector of the controllable switch S is connected to one end of the first capacitor C1 and one end of the first inductor L1 respectively; 6a The collector and the controllable switch tube S 6b The collector of the controllable switch S is connected to one end of the second capacitor C2 and one end of the second inductor L2 respectively; 2a The collector and the controllable switch tube S 2b The collector of the first capacitor C1 is connected to one end of the third capacitor C3 and one end of the third inductor L3 respectively; the other end of the first capacitor C1 is connected to the other end of the second capacitor C2 and the other end of the third capacitor C3 respectively; the other end of the first inductor L1 is connected to one end of the first load R1; the other end of the second inductor L2 is connected to one end of the second load R2; the other end of the third inductor L3 is connected to one end of the third load R3; the other end of the first load R1 is connected to the other end of the second load R2 and the other end of the third load R3 respectively; the topology adopts a current-type SPWM modulation method based on the decoupling idea, in which the modulation wave is a three-phase line voltage sine wave, the carrier is a high-frequency sawtooth wave, the three-phase modulation wave is compared with the carrier to generate a high-frequency signal, and the high-frequency signal is synthesized into a high-frequency square wave through logical processing; the current-type H-bridge on the primary side of the high-frequency transformer converts the DC current into a high-frequency square wave current under the high-frequency square wave drive, and its driving signal is two high-frequency square wave coupling signals V n and V p , V n and V p is a pair of high-frequency square waves with overlapping on-time and duty cycle greater than 50%, V n and V p As the driving signal of the controllable switch tubes S1~S4 and the coupling signal of the matrix converter; the matrix converter is decomposed into two groups of three-phase current type inverters, namely the positive group inverter and the negative group inverter. The positive group inverter is composed of the controllable switch tube S 1a ~Controllable switch tube S 6a The negative group inverter is composed of controllable switch tube S 1b ~Controllable switch tube S 6b The driving signal of each group of three-phase current source inverters is obtained by logically combining six SPWM and coupling signals.

2. The current source type high frequency link matrix inverter topology with soft switching according to claim 1, characterized in that: The topology adopts a current-type SVM modulation method based on the decoupling idea. In the current-type SVM modulation method based on the decoupling idea, the modulation wave is a three-phase phase voltage sine wave. The current-type SVM modulation method based on the decoupling idea includes sector judgment, vector action time calculation and switch signal generation. The sector judgment is specifically as follows: within one power frequency cycle, the three-phase modulation wave is divided into 12 sectors according to the size of the instantaneous value; the calculation of the vector action time is divided into the calculation of the effective vector and the zero vector action time; the switch signal generation includes six basic switch signals and one zero vector signal. The zero vector signal generates two coupling signals through logical transformation as the driving signal of the four switch tubes of the current-type H-bridge; the post-stage matrix converter is decomposed into two groups of ordinary three-phase current-type converters, namely the positive group converter and the negative group converter. The positive group converter consists of a controllable switch tube S 1a ~Controllable switch tube S 6a The negative group converter consists of a controllable switch tube S 1b ~Controllable switch tube S 6b The driving signals of all the switch tubes of the matrix converter are synthesized by coupling logic of the coupling signal and the six basic switch signals.

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