Pulse radar signal processing method based on integration of measurement and communication functions

Abstract

The present invention discloses a pulse radar signal processing method based on the integration of measurement and communication functions, which mainly solves the problems in the prior art that the radar has no communication function, and the distance measurement is inaccurate and the speed measurement takes a long time due to the Doppler frequency. The scheme is: set the radar transmission signal, receive the echo signal of the transmission signal at the receiving end and pre-process it to obtain the double frequency of the frequency difference, sine, cosine signal and 4 intermediate signals; according to the quadrant where the phase of the frequency difference sine and cosine signal is located, select the corresponding intermediate signal as the baseband signal; demodulate the baseband signal by different edge jumps of the double frequency difference signal to realize data communication; 2D The two adjacent and identical transition edges are counted to calculate the target speed υ; the baseband signal is pulse compressed to obtain the pulse compression peak, and the target distance is calculated. The present invention avoids the blind speed of the pulse radar, improves the ranging accuracy, and can be used for radar detection and communication navigation.

Description

Technical Field

[0001] The invention belongs to the technical field of signal processing, and in particular relates to a pulse radar signal processing method, which can be used for radar detection, communication navigation, etc. Background Art

[0002] Radar uses pulse compression signals to achieve long range and high range resolution at the same time. Pulse compression signals mainly include linear frequency modulation signals, nonlinear frequency modulation signals, and phase-coded pulse signals. Among them, the advantages of linear and nonlinear frequency modulation signals are that they have a wide adaptability range for Doppler frequency shift, and the disadvantage is that the movement of the target increases the Doppler frequency shift of the echo signal. The larger the Doppler frequency shift, the greater the delay of the pulse compression peak, which affects the ranging accuracy. The advantages of phase-coded signals are high range resolution, strong anti-interference ability, and the ability to distinguish the address identities of multiple targets. The disadvantage is that they are sensitive to Doppler frequency shift. When the Doppler frequency shift increases, the amplitude of its pulse compression peak will decrease or even disappear.

[0003] Radar measurement includes distance measurement and speed measurement. When the radar is measuring speed, the pulse radar transmits a radio frequency signal in the form of a pulse according to a repetitive cycle, and transmits a pulse signal in each cycle to receive the echo signal reflected by the target. The radar mixes the echo signal with the local oscillator signal generated by itself at the receiving end, and down-converts it from the radio frequency signal to an intermediate frequency signal. The intermediate frequency signal is then A / D sampled, and then digitally down-converted to convert the intermediate frequency signal into a baseband signal. Finally, the baseband signal is processed to obtain the pulse compression peak. Due to the relative motion between the radar and the target, the received echo signal is added with a Doppler frequency shift, and the radar measures the Doppler frequency shift value f. Δ , and then according to the Doppler frequency shift f Δ The relationship formula with the target speed υ is υ=(f Δ ×c)÷(2×f0), f0 is the frequency of the RF signal at the transmitter, and c is the speed of light. It can be seen that speed measurement is the same as frequency measurement.

[0004] When the radar is measuring distance, it uses large transmission energy to transmit wide pulse signals to achieve a long effective distance, ensuring that the radar has sufficient detection power; the transmission signal adopts a pulse compression signal in the form of phase coding. At the transmitting end, a single pulse of the transmission signal is phase-encoded with a pseudo-random sequence code, and at the receiving end, the received echo wide pulse signal is pulse compressed to obtain a narrow pulse. This narrow pulse is called a pulse compression peak; the radar measures distance based on the time difference τ between the received signal and the transmitted signal. This time difference τ is obtained by measuring the time difference between the pulse compression peak and the transmitted signal by the radar, minus the fixed time overhead such as the signal processing time, and then the target distance R is obtained according to the distance formula R = (c×τ) ÷ 2.

[0005] Since pulse radar has a speed measurement ambiguity problem, which is called blind speed, the existing speed measurement method adopts "staggered" measures to solve it. The radar uses more than two different repetition frequencies to work alternately, which is called staggered repetition frequency. The negative effect is that it cannot eliminate the secondary echo of far-field clutter, and the response time within the speed range is prolonged, the number of calculation steps is increased, and the processing difficulty is increased. Therefore, in engineering implementation, "group staggered" measures are usually taken to deal with the blind speed problem. In order to obtain one effective speed information, the radar needs to transmit three pulse groups continuously. Each pulse group has m continuous pulses. The repetition frequency of the pulses in the same pulse group is the same, and the repetition frequencies of different pulse groups are staggered. For example, in the MTD processing of moving target detection, m adjacent and partially overlapping Doppler filters constitute a Doppler filter group to process m continuous pulses. The Doppler filter group covers the entire range of the target Doppler frequency shift, can detect moving targets hidden in meteorological clutter and eliminate the secondary echo of clutter, but the speed response time is more time-consuming.

[0006] The existing pulse radar measures target distance, speed and other data information, which is used by itself or transmitted to the subsequent system through the connected data interface. The pulse radar itself does not have a data link communication function.

[0007] Existing pulse radars usually use linear frequency modulation signals or phase coded signals for ranging, and take the "group difference" measure to solve the blind speed problem in speed measurement. However, there are two problems in its implementation:

[0008] 1. It requires a lot of hardware resources, many calculation steps, and the speed test takes a long time.

[0009] Pulse radar needs three pulse groups to transmit signals at staggered periods in each repetition cycle to calculate one valid speed information. There are m continuous pulses in each pulse group. During the MTD processing of moving target detection, m continuous pulses are processed by m Doppler filters. The larger the range of the target Doppler frequency shift, the more Doppler filters are required. Therefore, more hardware resources are required, more calculation steps are required, and the speed response takes a long time. In addition, if the target is accidentally lost, one of the three pulse groups will not obtain an echo signal, and the calculated speed information cannot be trusted.

[0010] 2. Due to the influence of Doppler frequency shift, pulse radar has technical limitations in using linear frequency modulation signals or phase coded signals for ranging.

[0011] The adaptability range of linear frequency modulation signal to Doppler frequency shift is wider. When the Doppler frequency shift does not exceed the signal bandwidth, the filter can still obtain the pulse compression peak, but the time delay of the pulse compression peak is proportional to the Doppler frequency shift, resulting in delay error and affecting the ranging accuracy. In order to obtain accurate distance information, two linear frequency modulation signals need to be transmitted successively. These two signals are a pair of pulse signals with positive slope frequency modulation and negative slope frequency modulation. The time delay of their pulse compression peak is proportional to and inversely proportional to the Doppler frequency shift respectively. The delay error can be eliminated by averaging the time delays of the two pulse compression peaks. The phase-coded signal is required for ranging. The phase-coded signal has a narrow adaptability range to the Doppler frequency shift. The radar needs to continuously adjust the local signal frequency to keep it carrier synchronized with the received signal.

[0012] The above two deficiencies and the lack of communication function of the pulse radar itself result in the inability of the pulse radar to perform signal processing for target speed measurement, distance measurement and data communication in each pulse repetition period. Summary of the invention

[0013] The purpose of the present invention is to address the deficiencies of the above-mentioned prior art and to propose a pulse radar signal processing method based on the integration of measurement and communication functions, so that when there is a Doppler frequency shift, the pulse radar can achieve speed measurement, distance measurement, and data communication for the target in each repetition cycle.

[0014] To achieve the above object, the technical solution of the present invention includes the following:

[0015] (1) The radar transmits a radio frequency pulse signal RF(t) in each cycle, which is a pulse compression signal, a data start mark, and N-bit data in sequence. The radar receives the echo signal RF(t)' at the receiving end and down-converts it into an intermediate frequency signal IF(t). The pulse compression signal uses a phase coded signal, the data start mark uses binary 01, and the data start mark and N-bit data both use BPSK phase modulation.

[0016] (2) At the receiving end, the intermediate frequency signal IF(t) is subjected to a frequency of f S A / D sampling is used to obtain the digital signal of the intermediate frequency signal: Among them, f S =4×f I , f I is the intermediate frequency signal frequency, f Δ is the Doppler frequency shift; θ represents the two-phase modulation and takes the value of 0 or π; is the phase difference between the received signal and the transmitted signal, ranging from 0 to 2π; the sampling interval n = t × f S ;

[0017] (3) By output frequency f I -f PThe numerically controlled oscillator NCO generates a cosine signal of cos2π(f I -f P )n, the sinusoidal signal is sin2π(f I -f P )n a pair of initial orthogonal signals, f P is the intermediate frequency offset;

[0018] (4) The digital signal IF(n) in step (2) is respectively combined with the cosine signal cos2π(f I -f P )n、sine signal sin2π(f I -f P )n performs orthogonal transformation and low-pass filtering respectively to generate I and Q signals:

[0019]

[0020] Where f Δ +f P is the frequency difference;

[0021] (5) performing four-way parallel addition and subtraction operations on the I-way signal and the Q-way signal to obtain four-way operation results I+Q, IQ, (-I)+(-Q), and QI respectively, taking the sign bits of the four-way operation results respectively, to obtain four intermediate signals, namely, a first intermediate signal FLAG1, a second intermediate signal FLAG2, a third intermediate signal FLAG3, and a fourth intermediate signal FLAG4;

[0022] (6) Take the sign bits of the I-path signal and the Q-path signal in step (4) respectively, and obtain a pair of initial intermediate signals I FG , Q FG , the initial intermediate signal I FG and Q FG Multiply to get the frequency difference f Δ +f P The double frequency signal f 2D :

[0023] (7) Use a digital frequency divider to double the frequency difference signal f 2D Perform frequency division to obtain the sine signal AS2 of the frequency difference and the cosine signal AS1 of the frequency difference:

[0024]

[0025]

[0026] (8) Construct a lookup table. This lookup table has two address input terminals, two value output terminals, and stores the 2-bit values ​​of the high bit SE1 and the low bit SE0 internally.

[0027] (9) AS2 and AS1 in step (7) are used as address inputs of the lookup table, the high bit of the address input is AS2, and the low bit is AS1. The value output of the lookup table is connected to the address input of the four-to-one selector, and the first intermediate signal FLAG1, the second intermediate signal FLAG2, the third intermediate signal FLAG3, and the fourth intermediate signal FLAG4 obtained in step (5) are used as data inputs of the four-to-one selector;

[0028] (10) The phase of the frequency difference signal in step (7) The quadrant determines the value of the lookup table input address AS2 AS1 in step (8), and then selects different internal preset values ​​SE1 SE0;

[0029] (11) According to the internal preset values ​​SE1 SE0 output from the lookup table, the four-to-one selector outputs the corresponding intermediate signal to obtain the baseband signal JD;

[0030] (12) using a matched filter to perform matched filtering on the baseband signal JD obtained in step (11) to obtain a pulse compression peak;

[0031] (13) Check whether the pulse compression peak is effective:

[0032] If the pulse compression peak is less than the set threshold, the pulse compression peak is invalid and no operation is performed;

[0033] If the pulse compression peak value is greater than the set threshold value, the pulse compression peak value is valid, and (14) is executed;

[0034] (14) Use pulse compression peak to complete the pulse radar's ranging, speed measurement and data communication:

[0035] (14a) Obtain the time difference between the pulse compression peak and the emission of the transmission signal, calculate the distance between the target and the radar, and complete the ranging of the pulse radar;

[0036] (14b) finding the data start mark, demodulating the baseband signal JD obtained in step (11) according to the data start mark, and completing the signal processing of the pulse radar data communication;

[0037] (14c) Detection frequency difference f Δ +f P The double frequency signal f 2DThe pulse radar detects the jumping edge and counts two adjacent and identical jumping edges through the sampling clock to calculate the target speed υ and complete the speed measurement of the pulse radar.

[0038] Compared with the prior art, the present invention has the following advantages:

[0039] 1) The time consumption of pulse radar speed measurement is reduced.

[0040] The existing pulse radar is affected by blind speed and needs to transmit signals at a "staggered" period.

[0041] The present invention is not affected by blind speed, does not require a "staggered" cycle, uses a fixed repetition frequency for each repetition cycle, can extract the Doppler frequency shift in each repetition cycle to obtain the target speed, greatly reduces the time consumption of speed measurement, and does not require the use of m Doppler filters, saving hardware resources. The present invention can measure the speed of low-speed targets and high-speed moving targets as low as zero frequency, and has a wide range of applications.

[0042] 2) Added the communication function of pulse radar data link.

[0043] Prior art pulse radars do not have data link communication capabilities.

[0044] The pulse radar of the present invention has a data link communication function. In each repetition period, the transmitted signal carries N bits of data information. The data information adopts a binary phase shift keying BPSK modulation method and does not require carrier synchronization. The modulation imposed on the signal by the Doppler frequency shift can be eliminated in each repetition period to obtain the data information.

[0045] 3) When pulse radar uses phase-coded signals for ranging, it is not affected by Doppler frequency shift.

[0046] When the pulse radar of the prior art uses linear frequency modulation signals for ranging, the existence of Doppler frequency shift will cause the delay of the pulse compression peak to increase, affecting the ranging accuracy; when using phase-coded signals for ranging, because the signal has a narrow adaptability range to Doppler frequency shift, the radar needs to continuously adjust the local signal frequency to keep it synchronized with the carrier of the received signal.

[0047] The phase coding signal adopted in the present invention can eliminate the modulation imposed on the signal by the Doppler frequency shift in each repetition period, so that the signal is not affected by the Doppler frequency shift when performing distance measurement. BRIEF DESCRIPTION OF THE DRAWINGS

[0048] Figure 1 It is a flow chart of the implementation of the present invention;

[0049] Figure 2 is the signal format of each transmitted signal in the present invention;

[0050] Figure 3 It is a principle block diagram of generating four intermediate signals and three frequency difference signals in the present invention;

[0051] Figure 4 It is a principle block diagram of radar ranging in the present invention;

[0052] Figure 5 It is a principle block diagram of data demodulation in the present invention;

[0053] Figure 6 It is a principle block diagram of radar speed measurement in the present invention;

[0054] Figure 7 It is a pulse compression function simulation of an example of the present invention;

[0055] Figure 8 It is a communication function simulation of an example of the present invention;

[0056] Fig. 9 It is a simulation of the speed measurement function of an example of the present invention. DETAILED DESCRIPTION

[0057] The embodiments of the present invention are described in detail below with reference to the accompanying drawings.

[0058] Reference Figure 1 , the implementation steps of this embodiment include the following:

[0059] Step 1: Set the format of the radar transmission signal.

[0060] The transmission signal used by the pulse radar in the prior art is easily affected by Doppler frequency shift, resulting in inaccurate distance and speed measurement data, and the pulse radar itself does not have a communication function. In order to prevent the pulse radar from being affected by Doppler frequency shift and to achieve accurate speed measurement, distance measurement and data communication for the target, the transmission signal of the pulse radar is set.

[0061] Reference Figure 2 The signal format of each radar transmission signal is set as follows: pulse compression signal, data start mark, N-bit data. The pulse compression signal uses a phase coded signal, the data start mark uses binary 01, and the modulation form of the data start mark and the N-bit data both use BPSK phase modulation.

[0062] Step 2: pre-process the echo signal of the radar receiving and transmitting signal to generate 4 intermediate signals and 3 frequency difference signals.

[0063] Reference Figure 3 , the specific implementation of this step is as follows:

[0064] 2.1) Down-convert the received echo signal RF(t)' into an intermediate frequency signal IF(t), where the echo signal IF signal

[0065] 2.2) Perform A / D sampling and orthogonal transformation on the intermediate frequency signal IF(t) to generate I and Q signals:

[0066] 2.2.1) The intermediate frequency signal is subjected to S A / D sampling is used to obtain the digital signal of the intermediate frequency signal Where f0 is the RF signal frequency, f S =4×f I , f I is the intermediate frequency signal frequency, f Δ is the Doppler frequency, θ represents the two-phase modulation and takes the value of 0 or π; It is the phase difference between the received signal and the transmitted signal, ranging from 0 to 2π; the sampling interval n = t × f S , t is the time;

[0067] 2.2.2) By outputting the frequency f I -f P The numerically controlled oscillator NCO generates a cosine signal of cos2π(f I -f P )n, the sinusoidal signal is sin2π(f I -f P )n a pair of initial orthogonal signals, f P is the intermediate frequency offset;

[0068] 2.2.3) The digital signal IF(n) is respectively combined with the cosine signal cos2π(f I -f P )n、sine signal sin2π(f I -f P )n to complete the orthogonal transformation, and then perform low-pass filtering on the two signals after the orthogonal transformation to generate I and Q signals:

[0069]

[0070] In the formula, f Δ +f P is the frequency difference;

[0071] 2.3) Perform four-way parallel addition and subtraction operations on the I-way signal and the Q-way signal, and take the sign bits of the four-way operation results respectively to obtain four intermediate signals:

[0072] 2.3.1) Add the I-channel signal and the Q-channel signal to obtain the added signal A1:

[0073]

[0074] Taking the sign bit of A1 to obtain the first intermediate signal FLAG1;

[0075] 2.3.2) Subtract the Q signal from the I signal to get the subtraction signal R1:

[0076]

[0077] Taking the sign bit of R1 to obtain the second intermediate signal FLAG2;

[0078] 2.2.3) Invert the I and Q signals at the same time and add them together to obtain the inverted signal A2:

[0079]

[0080] Taking the sign bit of A2 to obtain the third intermediate signal FLAG3;

[0081] 2.3.4) Subtract the I-channel signal from the Q-channel signal to obtain the inverse subtraction signal R2:

[0082]

[0083] The sign bit of R2 is taken to obtain the fourth intermediate signal FLAG4.

[0084] 2.4) Take the sign bits of the I-path signal and the Q-path signal to obtain a pair of initial intermediate signals I FG , Q FG .

[0085] 2.5) For the initial intermediate signal I FG , Q FG After processing, three frequency difference signals are obtained:

[0086] 2.5.1) The initial intermediate signal I FG With Q FG Multiply and filter the narrow pulse to obtain the frequency difference f Δ +f P The double frequency signal f 2D :

[0087]

[0088] Among them, 2(f Δ +f P ) is the frequency of the double frequency difference signal;

[0089] 2.5.2) Use a digital frequency divider to double the frequency difference signal f 2D Perform frequency division to obtain the sine signal AS2 of the frequency difference and the cosine signal AS1 of the frequency difference:

[0090]

[0091]

[0092] Among them, f Δ +f P are the frequencies of the two signals.

[0093] Step 3: construct a lookup table, use the lookup table and a four-to-one selector to output a baseband signal JD, and use the signal for ranging.

[0094] Reference Figure 4 , the specific implementation of this step is as follows:

[0095] 3.1) Construct a lookup table with 2 external address input terminals, 2 value output terminals, and an internal storage of the 2-bit value of high bit SE1 and low bit SE0;

[0096] 3.2) Use AS2 and AS1 in 2.5.2) as the address input of the lookup table, the high bit of the address input is AS2, and the low bit is AS1, the value output of the lookup table is connected to the address input of the four-to-one selector, and use the first intermediate signal FLAG1, the second intermediate signal FLAG2, the third intermediate signal FLAG3, and the fourth intermediate signal FLAG4 obtained in step 2 as the data input of the four-to-one selector;

[0097] 3.3) According to 2.5.2), the phase of the frequency difference signal The quadrant determines the value of the lookup table input address AS2 AS1, and then selects different internal preset values ​​SE1 SE0:

[0098] When the phase When in the first quadrant, the input address of the lookup table AS2AS1=11, and the internal preset value SE1 SE0 is selected as 00;

[0099] When the phase When in the second quadrant, the input address of the lookup table AS2AS1=10, and the internal preset values ​​SE1 SE0 are selected as 01;

[0100] When the phase When in the third quadrant, the input address of the lookup table AS2AS1 = 00, and the internal preset value SE1 SE0 is selected as 10;

[0101] When the phase When in the fourth quadrant, the input address AS2AS1 of the lookup table is 01, and the internal preset value SE1 SE0 is selected to be 11.

[0102] 3.4) According to the internal preset values ​​SE1 SE0 output by the lookup table, the four-to-one selector outputs the corresponding intermediate signal, which is the baseband signal JD:

[0103] If the internal preset values ​​SE1 SE0 output by the lookup table are 00, the first intermediate signal FLAG1 output by the four-to-one selector is the baseband signal JD;

[0104] If the internal preset values ​​SE1 SE0 output by the lookup table are 01, the four-to-one selector outputs the second intermediate signal FLAG2, which is the baseband signal JD;

[0105] If the internal preset values ​​SE1 SE0 output by the lookup table are 10, the four-to-one selector outputs the third intermediate signal FLAG3, which is the baseband signal JD;

[0106] If the internal preset values ​​SE1 SE0 output by the lookup table are 11, the four-to-one selector outputs the fourth intermediate signal FLAG4, which is the baseband signal JD.

[0107] 3.5) Perform matched filtering on the obtained baseband signal to obtain the pulse compression peak, and use this pulse compression peak to calculate the distance between the target and the radar to complete the ranging of the pulse radar:

[0108] 3.5.1) Use a matched filter to perform matched filtering on the baseband signal JD to obtain the pulse compression peak;

[0109] 3.5.2) Check whether the pulse compression peak is valid:

[0110] If the pulse compression peak is less than the set threshold, the pulse compression peak is invalid and no operation is performed;

[0111] If the pulse compression peak is greater than the set threshold, the pulse compression peak is valid and continue to execute 3.5.3);

[0112] 3.5.3) First, use the time difference between the peak of the pulse compression and the time of the transmitted signal to obtain the time difference τ between the received signal and the transmitted signal, and then use the distance formula to calculate the distance between the target and the radar R = (c × τ) ÷ 2 to complete the pulse radar ranging. Where c is the speed of light, c = 3 × 10 8 m / s.

[0113] Step 4, phase demodulate the baseband signal JD obtained in step 3 according to the data start mark to obtain N-bit data, completing the signal processing of the pulse radar communication.

[0114] Reference Figure 5, the specific implementation of this step is as follows:

[0115] 4.1) Find the data start mark, that is, detect the transition edge of the baseband signal JD;

[0116] 4.2) Demodulate the baseband signal JD according to the different transition edges of the baseband signal:

[0117] 4.2.1) Use a two-to-one selector, take the baseband signal JD as its first input, take the inverted baseband signal RJD as its second input, and take the transition edge of the baseband signal as its address input SE;

[0118] 4.2.2) According to the different transition edges of the baseband signal JD, set the address input SE of the two-to-one selector:

[0119] If the edge of the baseband signal changes to a rising edge 01, it means that the baseband signal is in phase with the data in the transmit signal, and the address input SE of the two-to-one selector is set to 0;

[0120] If the edge of the baseband signal changes to a falling edge 10, it means that the baseband signal is opposite to the data in the transmission signal, and the address input SE of the two-to-one selector is set to 1;

[0121] 4.2.3) According to different address inputs SE, the two-choice selector selects the signal output of the input end to obtain the demodulated N-bit data:

[0122] When the address input SE is 0, the two-to-one selector directly outputs the baseband signal JD of the first input terminal to obtain the demodulated N-bit data;

[0123] When the address input SE is 1, the two-to-one selector directly outputs the inverted baseband signal RJD of the second input terminal to obtain the demodulated N-bit data;

[0124] Step 5: Detect the double frequency signal f of the intermediate frequency difference in step 2 2D The jump edge is detected, and two adjacent and identical jump edges are counted to calculate the target speed υ, thus completing the speed measurement of the pulse radar.

[0125] Reference Figure 6 , the specific implementation of this step is as follows:

[0126] 5.1) Detect the frequency difference double frequency signal f 2D The time between two adjacent rising edges is twice the frequency signal f 2D In one cycle, the time between two adjacent falling edges is also the double frequency signal f 2D a cycle of

[0127] 5.2) According to the different transition edges of the double frequency difference signal, four counters are used to count the sampling clock, rising edge, and falling edge, respectively, to obtain the average number a of the count values ​​between two adjacent rising edges and the average number b of the count values ​​between two adjacent falling edges:

[0128] When the frequency difference is twice the frequency signal f 2D When it is at the rising edge, the first counter counts the sampling clock, and the second counter counts the rising edge. When L rising edges are counted, the lower 4 bits of the count value of the first counter are discarded, and the high bits are retained for output to obtain the average number a of the count values ​​between two adjacent rising edges;

[0129] When the frequency deviation is twice the frequency signal f 2D When the falling edge is at the falling edge, the third counter counts the sampling clock, and the fourth counter counts the falling edge. When L falling edges are counted, the lower 4 bits of the count value of the third counter are discarded, and the higher bits are retained for output, so as to obtain the average number b of the count values ​​between two adjacent falling edges;

[0130] 5.3) Add the rising edge average a and the falling edge average b to get the total average e, and calculate the Doppler frequency f based on the total average e Δ =f S ÷ef P ;

[0131] 5.4) According to the relationship between Doppler frequency shift and target speed, the target speed υ=(f Δ ×c)÷(2×f0), that is, the speed measurement of the pulse radar is completed, where f0 is the frequency of the RF signal, c is the speed of light, c=3×10 8 m / s.

[0132] In this example, due to 2 4 =16 and L=16, let f0=12GHz, f I =20MHz, f S =80MHz, f P =30kHz, when the supersonic aircraft flies toward or away from the radar at the maximum speed of υ = 340m / s, the Doppler frequency shift is f Δ =27.2kHz, f Δ =-27.2kHz, so the range of Doppler frequency shift, that is, the required frequency measurement range, is not narrower than -27.2kHz to +27.2kHz. The present invention simulates the pulse compression function, the communication function, and the speed measurement function according to the above input values, respectively. Figure 7 , Figure 8 , Fig. 9 .

[0133] The above description is only a specific example of the present invention and does not constitute any limitation to the present invention. It is obvious that for professionals in this field, after understanding the content and principles of the present invention, it is possible to make various modifications and changes in form and details without departing from the principles and structures of the present invention. However, these modifications and changes based on the ideas of the present invention are still within the scope of protection of the claims of the present invention.

Claims

1. A signal processing method based on a pulse radar with integrated measurement and communication functions, characterized in that: The steps include: (1) The radar transmits a radio frequency pulse signal RF(t) in each cycle, which is a pulse compression signal, a data start mark, and N-bit data in sequence. The radar receives the echo signal RF(t)' at the receiving end and down-converts it into an intermediate frequency signal IF(t). The pulse compression signal uses a phase coded signal, the data start mark uses binary 01, and the data start mark and N-bit data both use BPSK phase modulation. (2) At the receiving end, the intermediate frequency signal IF(t) is subjected to a frequency of f S A / D sampling is used to obtain the digital signal of the intermediate frequency signal: Among them, f S =4×f I , fI is the intermediate frequency signal frequency, f Δ is the Doppler frequency shift; θ represents the two-phase modulation and takes the value of 0 or π; is the phase difference between the received signal and the transmitted signal, ranging from 0 to 2π; the sampling interval n = t × f S ; (3) By output frequency f I -f P The numerically controlled oscillator NCO generates a cosine signal of cos2π(f I -f P )n, the sinusoidal signal is sin2π(f I -f P )n a pair of initial orthogonal signals, f P is the intermediate frequency offset; (4) The digital signal IF(n) in step (2) is respectively combined with the cosine signal cos2π(f I -f P )n、sine signal sin2π(f I -f P )n performs orthogonal transformation and low-pass filtering respectively to generate I and Q signals: Where f Δ +f P is the frequency difference; (5) performing four-way parallel addition and subtraction operations on the I-way signal and the Q-way signal to obtain four-way operation results I+Q, IQ, (-I)+(-Q), and QI respectively, taking the sign bits of the four-way operation results respectively, to obtain four intermediate signals, namely, a first intermediate signal FLAG1, a second intermediate signal FLAG2, a third intermediate signal FLAG3, and a fourth intermediate signal FLAG4; (6) Take the sign bits of the I-path signal and the Q-path signal in step (4) respectively, and obtain a pair of initial intermediate signals I FG , Q FG , the initial intermediate signal I FG and Q FG Multiply to get the frequency difference f Δ +f P The double frequency signal f 2D : (7) Use a digital frequency divider to double the frequency difference signal f 2D Perform frequency division to obtain the sine signal AS2 of the frequency difference and the cosine signal AS1 of the frequency difference: (8) Construct a lookup table. This lookup table has two address input terminals, two value output terminals, and stores the 2-bit values ​​of the high bit SE1 and the low bit SE0 internally. (9) AS2 and AS1 in step (7) are used as address inputs of the lookup table, the high bit of the address input is AS2, and the low bit is AS1. The value output of the lookup table is connected to the address input of the four-to-one selector, and the first intermediate signal FLAG1, the second intermediate signal FLAG2, the third intermediate signal FLAG3, and the fourth intermediate signal FLAG4 obtained in step (5) are used as data inputs of the four-to-one selector; (10) The phase of the frequency difference signal in step (7) The quadrant determines the value of the lookup table input address AS2 AS1 in step (8), and then selects different internal preset values ​​SE1 SE0; (11) According to the internal preset values ​​SE1 SE0 output from the lookup table, the four-to-one selector outputs the corresponding intermediate signal to obtain the baseband signal JD; (12) using a matched filter to perform matched filtering on the baseband signal JD obtained in step (11) to obtain a pulse compression peak; (13) Check whether the pulse compression peak is effective: If the pulse compression peak is less than the set threshold, the pulse compression peak is invalid and no operation is performed; If the pulse compression peak value is greater than the set threshold value, the pulse compression peak value is valid, and step (14) is executed; (14) Use pulse compression peak to complete the pulse radar's ranging, speed measurement and data communication: (14a) Obtain the time difference between the pulse compression peak and the emission of the transmission signal, calculate the distance between the target and the radar, and complete the ranging of the pulse radar; (14b) finding the data start mark, demodulating the baseband signal JD obtained in step (11) according to the data start mark, and completing the signal processing of the pulse radar data communication; (14c) Detection frequency difference f Δ +f P The double frequency signal f 2D The pulse radar detects the jumping edge and counts two adjacent and identical jumping edges through the sampling clock to calculate the target speed υ and complete the speed measurement of the pulse radar.

2. The method according to claim 1, characterized in that: The four-way operation results of I+Q, IQ, (-I)+(-Q), and QI obtained in step (5) are expressed as follows:

3. The method according to claim 1, characterized in that: In step (10), different internal preset values ​​SE1SE0 are selected, which are implemented as follows: When the phase When in the first quadrant, according to the input address AS2AS1=11 of the lookup table, the internal preset values ​​SE1 SE0 are selected as 00; When the phase When in the second quadrant, according to the input address AS2AS1=10 of the lookup table, the internal preset values ​​SE1 SE0 are selected as 01; When the phase When in the third quadrant, according to the input address AS2AS1=00 of the lookup table, the internal preset value SE1 SE0 is selected as 10; When the phase When in the fourth quadrant, according to the input address AS2AS1=01 of the lookup table, the internal preset value SE1 SE0 is selected to be 11.

4. The method according to claim 1, characterized in that: In step (11), according to the internal preset values ​​SE1 SE0 output from the lookup table, the four-to-one selector outputs the corresponding intermediate signal to obtain the baseband signal JD, which is implemented as follows: If the internal preset values ​​SE1 SE0 output by the lookup table are 00, the four-to-one selector outputs the first intermediate signal FLAG1 to obtain the baseband signal JD; If the internal preset values ​​SE1 SE0 output by the lookup table are 01, the four-to-one selector outputs the second intermediate signal FLAG2 to obtain the baseband signal JD; If the internal preset values ​​SE1 SE0 output by the lookup table are 10, the four-to-one selector outputs the third intermediate signal FLAG3 to obtain the baseband signal JD; If the internal preset values ​​SE1 SE0 output by the lookup table are 11, the four-to-one selector outputs the fourth intermediate signal FLAG4 to obtain the baseband signal JD.

5. The method according to claim 1, characterized in that: In step (14a), the distance between the target and the radar is calculated by first using the time difference between the peak of the pulse compression and the time when the transmitted signal is sent, minus the fixed time overhead of the signal processing time, to obtain the time difference τ between the received signal and the transmitted signal; then the distance formula is used to calculate the distance between the target and the radar R = (c × τ) ÷ 2, where c is the speed of light, c = 3 × 10 8 m / s.

6. The method according to claim 1, characterized in that: Step (14b) searches for the data start mark and demodulates the baseband signal JD obtained in step (11) according to the data start mark, which is implemented as follows: (14b1) detecting the transition edge of the baseband signal JD; (14b2) According to the different transition edges of the baseband signal, the baseband signal JD is obtained in step (11): If the edge of the baseband signal changes to a rising edge 01, it means that the baseband signal is in phase with the data in the transmitted signal, and the baseband signal is directly output to obtain N-bit data after phase demodulation; If the edge of the baseband signal changes to a falling edge 10, it means that the baseband signal is inverted to the data in the transmission signal, and the baseband signal is then inverted and output to obtain N-bit data after phase demodulation.

7. The method according to claim 1, characterized in that: In step (14c), the sampling clock counts two adjacent and identical transition edges to calculate the target speed υ, which is specifically implemented as follows: (14c1) According to the different transition edges of the double frequency difference signal, the sampling clock is used to perform the following operations on two adjacent rising edges and two adjacent falling edges: When the frequency difference is twice the frequency signal f 2D When it is on the rising edge, the time between two adjacent rising edges is recorded as the double frequency signal f 2D One cycle of the sampling clock and the double frequency signal f 2D After L cycles are counted, the average number of count values ​​between two adjacent rising edges is calculated according to the first total count value sum1 to obtain a=sum1 / L; When the frequency difference is twice the frequency signal f 2D When it is on the falling edge, the time between two adjacent falling edges is recorded as the double frequency signal f 2D The same sampling clock is used to sample the double frequency signal f 2D After L cycles are counted, the average number of count values ​​between two adjacent falling edges is calculated according to the second total count value sum2 to obtain b=sum2 / L; (14c2) The total average e is obtained by adding the rising edge average a and the falling edge average b. The Doppler frequency f is calculated based on the total average e. Δ =f S ÷ef P ; (14c3) According to the relationship between Doppler frequency shift and target speed, the target speed υ is calculated as follows: Δ ×c)÷(2×f0), where f0 is the frequency of the RF signal, c is the speed of light, and c=3×10 8 m / s.

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