Integrated circuit comprising a matching and filtering network with a direct current supply stage and corresponding matching and filtering method

Abstract

An integrated circuit comprises a power amplifier (PA), an antenna (ANT), and a matching and filtering network (MFN) comprising a direct current feed stage (DCFD), a first section (SCT1), and a second section (SCT2) on an output node of the power amplifier (PA). The direct current feed stage (DCFD) and the two sections comprise an inductance-capacitance "LC" arrangement configured to have an impedance matched to an output of the power amplifier (PA) in a base frequency band. The LC arrangements of the direct current feed stage (DCFD) and the first section (SCT1) are further configured to have a resonance frequency adapted to attenuate a harmonic frequency band of the base frequency band, respectively.

Description

Technical Field

[0001] Examples and implementations involve integrated circuits including matching and filtering networks, typically located between the output of a power amplifier and an antenna. Summary of the Invention

[0002] On the one hand, a power amplifier operates at maximum efficiency when the load on its output is at its optimal impedance. The optimal impedance at the output of a power amplifier is typically different from the impedance of the antenna to which it is connected.

[0003] Therefore, impedance matching circuitry is typically provided between the output of a power amplifier and the antenna to transform the antenna's impedance to the amplifier's ideal impedance. Impedance matching circuitry is usually produced using passive components such as inductors and capacitors. Impedance matching circuitry can occupy a potentially large surface area, which can be particularly detrimental when it is fabricated within a small integrated circuit.

[0004] On the other hand, the emission spectrum of a power amplifier often contains interference signals, such as harmonic frequencies of the fundamental frequency or noise generated due to the internal nonlinearity of the power amplifier.

[0005] Therefore, a filtering circuit is typically provided between the output of the power amplifier and the antenna to specifically filter the harmonic bands, such as up to the fifth order. The filtering circuit is usually produced using passive components such as resistors, inductors, and capacitors, and also occupies a potentially large surface area, which is particularly detrimental when impedance matching circuits are produced in an integrated manner.

[0006] Impedance matching and filtering circuits have specific constraints and are typically designed separately.

[0007] While techniques that separate these two functions (impedance matching and filtering) are satisfactory in terms of performance, they require many passive components, making them expensive and cumbersome.

[0008] However, reducing the size of integrated circuits reduces the surface area available for passive devices, making it more difficult to integrate both functions at a satisfactory performance level.

[0009] The use of surface acoustic wave (SAW) filters reduces the space required, but increases the cost.

[0010] Therefore, there is a need for compact and inexpensive solutions that can achieve impedance matching and filtering while obtaining good performance levels.

[0011] According to embodiments and implementations, the present invention proposes providing impedance matching and filtering in a single network, wherein impedance matching is provided by a network with the minimum quality factor to achieve optimal response over the widest possible frequency range while keeping the number of passive components as low as possible; filtering is provided by replacing passive components with a rationally selected inductor-capacitor arrangement to resonate at the frequency to be filtered, without altering the impedance transformation in the transmission frequency. Specifically, the resonant frequency of the inductor-capacitor arrangement is selected to optimize the network size and response.

[0012] In this regard, one aspect proposes an integrated circuit comprising a power amplifier designed to provide a signal in a baseband frequency band, an antenna, and a matching and filtering network comprising:

[0013] - DC power supply stage between the supply voltage node and the output node of the power amplifier;

[0014] -The first segment between the output node and the intermediate node of the power amplifier; and

[0015] -The second segment between the intermediate node and the antenna input node,

[0016] The DC power supply stage and the two sections include inductor-capacitor (LC) arrangements configured to have impedances that match the output of the power amplifier in the baseband.

[0017] The LC arrangement of the DC power supply stage and the LC arrangement of the first section are also configured to have resonant frequencies that are suitable for attenuating the harmonic frequency bands of the base spectrum.

[0018] For convenience and in accordance with conventional use in electronics, the symbol "LC" is used to denote the term "inductor-capacitor".

[0019] Therefore, a matching and filtering network with two LC arrangement sections is proposed, wherein all LC arrangements in the DC power supply stage and the first section's LC arrangement are configured for impedance matching and harmonic frequency filtering.

[0020] In particular, it should be noted that the LC arrangement of the DC power supply stage and the LC arrangement of the first section are configured for simultaneous matching and filtering functions, which differs from the conventional technique where one arrangement is dedicated to matching and the other, at least partially separate, arrangement is dedicated to filtering.

[0021] Therefore, based on this design, it proposes the complete integration of matching and filtering functions in the LC arrangement of the DC power supply stage and the LC arrangement of the first segment, allowing for the production of particularly compact integrated circuits without sacrificing performance or increasing cost.

[0022] According to one embodiment, the LC arrangement of the second segment includes: an LC arrangement configured to have a resonant frequency suitable for attenuating the harmonic bands of the base spectrum; and an inductor or capacitor element configured not to introduce resonance suitable for attenuating the harmonic bands of the base spectrum.

[0023] In the second section, inductors or capacitors arranged in an LC configuration are not integrated and are dedicated to impedance matching, which in particular allows for the preservation of high-frequency attenuation.

[0024] According to one embodiment, the second segment includes a series LC arrangement coupled between the input node and the ground node of the antenna, and an inductive element coupled between the input node and the intermediate node of the antenna, or includes a parallel LC arrangement coupled between the input node and the intermediate node of the antenna, and a capacitive element coupled between the intermediate node and the ground node.

[0025] In other words, the second section may be provided with series-coupled inductor elements (dedicated to impedance matching and allowing high-frequency filtering) and a series LC arrangement (shunted to ground to remove signals around its resonant frequency); or alternatively, it may be provided with shunted-coupled capacitor elements (dedicated to impedance matching and allowing high-frequency filtering) and a series-coupled parallel LC arrangement to block signals around its resonant frequency.

[0026] According to one embodiment, the first segment includes a series LC arrangement coupled between the output node and the ground node of the power amplifier, and the second segment includes a series LC arrangement coupled between the input node and the ground node of the antenna, the series LC arrangement of the second segment being configured to have a resonant frequency greater than that of the series LC arrangement of the first segment.

[0027] More specifically, for impedance matching, the capacitor element is typically larger on the power amplifier side than on the antenna side. On the other hand, the resonant frequency of an LC arrangement is inversely proportional to the size of the inductor and capacitor elements in the LC arrangement. Therefore, this embodiment proposes to optimize the resonant frequency by increasing the size of the inductor element to ensure resonance of the capacitor element provided for impedance matching. The total space requirement (particularly the space required for the inductor element in a series LC arrangement) is thus optimized to be as small as possible.

[0028] According to one embodiment, the DC power supply stage includes a low-pass T-shaped LC arrangement, and the first section includes a parallel LC arrangement coupled between an intermediate node and the output node of the power amplifier, and a series LC arrangement coupled between an intermediate node and a ground node.

[0029] The term “low-pass T-shaped LC arrangement” should be understood to refer to a typical inductor-capacitor arrangement, which includes two inductor elements coupled in series and connected at the center node, and a capacitor element coupled between the center node and ground.

[0030] Therefore, a parallel LC arrangement blocks the signal at its resonant frequency, while a series LC arrangement routes the signal at its resonant frequency to ground. A low-pass T-shaped LC arrangement includes additional filtering consisting of an inductor coupled to the amplifier's output node and a capacitor coupled to ground in series.

[0031] According to one embodiment, the parallel LC arrangement of the first segment is configured to have a highest resonant frequency among the resonant frequencies of the LC arrangement that is suitable for attenuating the harmonic band of the base spectrum.

[0032] This embodiment is particularly advantageous for additional constraints that arise when the baseband is used at high frequencies, i.e., above which the interfering inductive elements associated with the metal connection (especially the metal connection to ground) have an effect that is no longer negligible at these frequencies (especially in the harmonic band of the baseband).

[0033] More specifically, the interference inductive element generated by the physical connection to ground can cause the capacitive element coupled to ground (through an interference arrangement equivalent to a series LC arrangement) to resonate at frequencies that may be below the harmonic band (e.g., the fourth and fifth harmonics).

[0034] Therefore, this embodiment provides attenuation of the fourth and fifth harmonic frequency bands through the resonant frequency of the parallel LC arrangement of the first segment, which does not suffer from the problem of interference inductance on the ground connection.

[0035] Furthermore, the resonant frequencies of one or more series LC arrangements in each segment (first and second segments) can be selected to be distributed in the harmonic frequency bands (e.g., the second and third harmonics) that are still to be attenuated, so as to include inductor elements of the minimum size combined with capacitor elements intended for impedance matching.

[0036] More specifically, since the value of the inductor required to make the capacitor resonate is inversely proportional to the capacitance value and inversely proportional to the square of the resonant frequency, the highest resonant frequency is advantageously associated with the lowest capacitance value in order to minimize the value of the inductor to be increased.

[0037] This again allows for optimization of overall space requirements by locating the resonant frequency in a way that optimizes the size of the inductor that performs the filtering function in each segment.

[0038] In this regard, according to one embodiment, the LC arrangement is configured according to at least one of the following criteria:

[0039] - The low-pass T-shaped LC arrangement of the DC power supply stage is configured to have a resonant frequency in half of the second harmonic frequency band.

[0040] - The series LC arrangement of the first section is configured to have a resonant frequency in the other half of the second harmonic frequency band.

[0041] - The series LC arrangement of the second section is configured to have a resonant frequency in the third harmonic band;

[0042] - The parallel LC arrangement of the first section is configured to have a resonant frequency between the fourth harmonic band and the fifth harmonic band or in the common part of the fourth harmonic band and the fifth harmonic band.

[0043] This embodiment provides the possibility of locating the resonant frequency to optimize total space requirements and performance.

[0044] In particular, note the resonant frequency of the parallel LC arrangement in the first section (rather than the series LC arrangement in the second section) in the fourth and fifth harmonic bands. This positioning advantageously prevents potential problems caused by interfering inductive elements connected to metallic connections (especially to ground), the effects of which become non-negligible as the frequency increases, particularly in the fourth and fifth harmonic bands.

[0045] According to another aspect, the present invention proposes a method for impedance matching and filtering between the output of a power amplifier providing a signal in the baseband and an antenna, the method comprising dimensioning a virtual matching network comprising:

[0046] - DC power supply stage between the supply voltage node and the output node of the power amplifier;

[0047] -The first segment between the intermediate node and the output node of the power amplifier; and

[0048] -The second segment between the antenna's input node and intermediate node,

[0049] The DC power supply stage includes inductor elements, and a first segment and a second segment include inductor and capacitor elements. The dimensions are determined to have an impedance that matches the output of the power amplifier in the baseband. The method includes generating a true matching and filtering network that replaces the inductor elements of the DC power supply stage and the inductor and capacitor elements of the first segment of the virtual matching network with a corresponding “LC” inductor-capacitor arrangement. The “LC” inductor-capacitor arrangement is configured to have an equivalent impedance that matches the output of the power amplifier in the baseband and also has resonant frequencies that are suitable for attenuating the harmonic bands of the baseband.

[0050] Based on this approach, a method for determining the size of a virtual impedance matching network is proposed, without providing filtering functionality, so as to first determine the size required for impedance matching.

[0051] The term “virtual” is understood to mean, for example, “existing in a state of pure possibility or chance rather than in a state of material realization,” such as, in particular, intermediate calculations.

[0052] Secondly, filtering is then introduced into the real matching and filtering network (i.e., a matching and filtering network that "truly exists for matching and filtering the transmitted signal," as opposed to "virtual"). The real matching and filtering network is obtained by replacing the virtual components with real components that are equivalent for impedance matching requirements and also have filtering capabilities.

[0053] In practice, for any embodiment of the power amplifier and antenna, this method allows matching and filtering functions to be fully incorporated in the LC arrangement of the first segment and the LC arrangement of the second segment in a compact, efficient and inexpensive manner.

[0054] According to one implementation, generating the real matching and filtering network includes replacing the inductor or capacitor elements of the second segment of the virtual matching network with an “LC” inductor-capacitor arrangement, which is configured to have an equivalent impedance that matches the output of the power amplifier in the baseband and also has a resonant frequency suitable for attenuating the harmonic bands of the baseband.

[0055] According to one implementation, the second segment of the virtual matching network includes an inductive element coupled between the input node and the intermediate node of the antenna, and a capacitive element coupled between the input node and the ground node of the antenna, wherein generating the real matching and filtering network includes replacing the capacitive element with a series LC arrangement or replacing the inductive element with a parallel LC arrangement.

[0056] According to one implementation, generating the real matching and filtering network includes replacing the capacitive elements of the virtual matching network with a series LC arrangement in the first and second segments, wherein the resonant frequency of the series LC arrangement in the second segment is selected such that it is higher than the resonant frequency of the series LC arrangement in the first segment.

[0057] According to one implementation, the first segment of the virtual matching network includes an inductor coupled between the output node and the intermediate node of the power amplifier, and a capacitor coupled between the intermediate node and the ground node. The generation of the real matching and filtering network includes: replacing the inductor of the DC power supply stage of the virtual matching network with a low-pass T-shaped LC arrangement, replacing the inductor of the first segment with a parallel LC arrangement, and replacing the capacitor of the first segment with a series LC arrangement.

[0058] According to one implementation, the resonant frequency of the parallel LC arrangement in the first segment is selected such that it is the highest among the resonant frequencies of LC arrangements suitable for attenuating the harmonic frequency band of the fundamental frequency band.

[0059] According to one implementation, the true matching and filtering network is generated based on at least one of the following criteria:

[0060] - The T-shaped LC arrangement of the DC power supply stage has a resonant frequency in half of the second harmonic frequency band;

[0061] -The series LC arrangement of the first section has a resonant frequency in the other half of the second harmonic frequency band;

[0062] - The series LC arrangement in the second section has a resonant frequency in the third harmonic band;

[0063] - The parallel LC arrangement in the first section has a resonant frequency between the fourth harmonic band and the fifth harmonic band, or in the common part of the fourth harmonic band and the fifth harmonic band. Attached Figure Description

[0064] Other advantages and features of the invention will become apparent upon review of the detailed description of the non-limiting embodiments and implementations and from the accompanying drawings, in which: Detailed Implementation

[0065] Figure 1 The matching and filtering network MFN (e.g., integrated into an integrated circuit) between the output node of the power amplifier PA and the input node of the antenna ANT is shown.

[0066] The power amplifier (PA) is configured to provide a transmission signal in the baseband, particularly a radio frequency suitable for wireless communication such as 4G, 5G or LTE, Wi-Fi or Bluetooth telecommunications.

[0067] The matching and filtering network MFN includes two matching and filtering sections, SCT1 and SCT2, and a DC power supply stage, DCFD.

[0068] The DC-powered stage DCFD includes two inductor elements L11 and L12 connected in series between the supply voltage terminal VCC and the output node of the power amplifier PA, and includes a decoupling capacitor element between the supply voltage terminal VCC and the ground reference voltage terminal GND.

[0069] On the one hand, the DC-powered stage DCFD allows the necessary voltage level and current for the matching and filtering network MFN to be supplied from the output node of the power amplifier PA.

[0070] The series-connected inductors L31 and L32 are configured to present the imaginary part of the ideal impedance of the power amplifier PA at the output node of the power amplifier PA.

[0071] The DC-powered stage DCFD also includes another capacitor C3 coupled to the center node between the two inductor elements L11, L12 and ground GND to form a low-pass T-arrangement 30. The low-pass T-arrangement is a conventional LC filter structure, so named because of the shape of its wiring diagram, which resembles the letter T.

[0072] The low-pass T-shaped inductor-capacitor arrangement of 30 does not affect the power supply function of the stage DCFD, but it adds a short-circuit effect to ground for signals at its resonant frequency (commonly referred to as a shunt).

[0073] More specifically, the first inductor L31 and capacitor C30 can be considered as a series LC arrangement mounted between the output node of the power amplifier PA and ground GND, thus suitable for attenuating the transmitted signal at the resonant frequency.

[0074] The T-shaped LC arrangement 30 (particularly elements L31 and C30) is configured to have a resonant frequency suitable for attenuating the harmonic bands of the base spectrum.

[0075] Harmonics are frequencies that are integer multiples of the fundamental frequency of the transmitted signal.

[0076] The first section SCT1 is located between the output node of the power amplifier PA and the intermediate node N1, and the second section SCT2 is located between the intermediate node N1 and the input node of the antenna ANT.

[0077] Each of the two sections SCT1 and SCT2 includes an inductor and a capacitor, which is, for convenience, is traditionally referred to as an "LC" inductor-capacitor arrangement.

[0078] The LC arrangement of the two sections SCT1 and SCT2 is configured to have an impedance that matches the output of the power amplifier PA in the baseband.

[0079] This impedance is matched to the output of the power amplifier PA, because at this impedance, the amplifier delivers the desired power to the antenna with the best possible efficiency.

[0080] Furthermore, the LC arrangement of the first segment SCT1 and the second segment SCT2 is configured to have resonant frequencies that are respectively suitable for attenuating the harmonic bands of the base spectrum.

[0081] Specifically, the first section SCT1 includes a parallel LC arrangement 11 coupled between the output node of the power amplifier PA and the intermediate node N1, and a series LC arrangement 12 coupled between the intermediate node N1 and the ground node GND.

[0082] The second section SCT2 includes an inductor L2 coupled between the intermediate node N1 and the input node of the antenna ANT, and a series LC arrangement 22 coupled between the input node of the antenna ANT and the ground node GND.

[0083] Inductor L2 is not integrated into the resonant LC arrangement that provides filtering at the resonant frequency, but attenuation is still allowed at high and very high frequencies. In particular, the inductor L2 in the second section is dedicated to impedance matching.

[0084] Alternatively (not shown), the second segment SCT2 may include a capacitive element coupled between the input node of the antenna ANT and the ground node GND, and a parallel LC arrangement coupled between the intermediate node N1 and the input node of the antenna ANT.

[0085] The capacitor element in this alternative embodiment is not integrated into the resonant LC arrangement that provides filtering at the resonant frequency, but attenuation is still allowed at high and very high frequencies. The capacitor element in the second segment of this alternative embodiment is specifically designed for impedance matching.

[0086] It should be noted that, traditionally, a coupling capacitor CC is provided between the input node of the antenna and the antenna ANT to block the DC component of the voltage, and its capacitance value is chosen to be large enough to have a negligible effect on impedance matching.

[0087] Therefore, the parallel LC arrangement 11 of the first segment SCT1 blocks the transmission of the signal at its resonant frequency from the output of the power amplifier PA to the antenna ANT via the intermediate node N1 along the serial channel. In the alternative embodiment described above, this also applies to the parallel LC arrangement of the second segment SCT2.

[0088] In addition, the series LC arrangement 12, 22 routes the signal flowing from the output of the power amplifier PA to the antenna ANT at its resonant frequency to ground GND via intermediate node N1 (commonly referred to as the shunt).

[0089] According to an advantageous example embodiment, the parallel LC arrangement 11 is configured in a dual manner: on the one hand, it has an impedance in the base frequency band that matches the inductor element used for impedance matching (…). Figure 2 The corresponding equivalent impedance, on the other hand, has the ability to operate in the harmonic frequency band ( Figure 5 The resonant frequency f11 selected from one of the following is... Figure 5 ).

[0090] Similarly, each series LC arrangement 12, 22 is advantageously configured in a dual manner: on the one hand, it has an impedance in the baseband of the capacitor element used for impedance matching ( Figure 2 The corresponding equivalent impedance, and has harmonic frequency band ( Figure 5 The resonant frequencies f12 and f22 selected from one of them are... Figure 5 This allows the inductance of a series LC arrangement to be minimized to the equivalent impedance.

[0091] In particular, it should be noted that each LC arrangement 11, 12, 30 of the DC-powered stage DCFD and the first section SCT1 is configured as a whole for both impedance matching and filtering functions. That is, there are no components in the LC arrangements of the DC-powered stage DCFD and the first section SCT1 that are dedicated solely to impedance matching or filtering functions.

[0092] More specifically, the decoupling capacitor element between the supply voltage VCC and ground GND (and the capacitor element CC between the antenna input node and the antenna ANT) is usually always provided and can be considered separately from the circuitry of the matching and filtering network MFN.

[0093] In addition, the second section SCT2 includes an LC arrangement 22 that is configured simultaneously for both matching and filtering functions, and separately includes an inductor L2 dedicated solely to impedance matching, but allowing for high-frequency filtering.

[0094] In the above alternative embodiments, the inductor L2 can be replaced by a parallel LC arrangement that is configured simultaneously for both matching and filtering functions, and the series LC arrangement 22 can be replaced by a capacitor element dedicated solely to impedance matching, but allowing for high-frequency filtering.

[0095] In another alternative embodiment (not shown), the second segment SCT2 may include an inductor L2 coupled between the intermediate node N1 and the input node of the antenna ANT, and a capacitor coupled between the input node of the antenna ANT and ground GND, both dedicated solely to impedance matching. This additional alternative embodiment can improve high-frequency filtering, but at the cost of less peak attenuation at the resonant frequency (i.e., poorer selectivity in harmonic frequency attenuation).

[0096] refer to Figures 2 to 4 This demonstrates an advantageous method for determining the above reference. Figure 1 The dimensions of the inductor and capacitor elements in the described impedance matching and filtering network MFN.

[0097] Figure 2 A virtual matching network MN0 is shown, which will be used as a reference benchmark to determine the reference. Figure 1 The dimensions of the inductor and capacitor elements of the described matching and filtering network MFN.

[0098] The matching network MN0 is described as "virtual" because it is used solely for computational purposes to determine the magnitude of the impedance matching requirement. The results of this dimensional determination will be used as the basis for calculations to evaluate the actual generated components, in order to compare them with a reference network. Figure 1 The described network MFN implements matching and filtering.

[0099] The virtual matching network MN0 includes two virtual matching segments SCT01 and SCT02 and a virtual DC power supply stage DCFD0.

[0100] Each of the two virtual segments SCT01 and SCT02 includes an inductive element on a serial channel from the output node of the power amplifier PA through the intermediate node N1 to the input node of the antenna ANT, and a capacitive element that is shunt-coupled to ground GND on the intermediate node N1 and the input node of the antenna ANT.

[0101] The virtual matching network MN0 thus corresponds to a low-pass network with the minimum quality factor in the two segments SCT01 and SCT02, which are provided to match the impedance between the output of the power amplifier PA and the antenna ANT.

[0102] Now for reference Figure 3 .

[0103] Figure 3 The Smith chart is shown with the impedance normalized by the antenna ANT, such that the impedance of the antenna RANT is located at the center of the Smith chart.

[0104] The sizing is performed by converting the impedance of the antenna RANT to the ideal impedance of the power amplifier RPA.

[0105] Therefore, the intermediate impedance R1 is calculated by the geometric mean between the antenna impedance RANT and the ideal impedance RPA presented at the output node of the power amplifier PA.

[0106] In other words: R1 = (RA x RL) 1 / 2

[0107] Strictly speaking, the calculation is performed using the following: RPA is the reciprocal of the real part of the admittance presented at the output node of the power amplifier PA, and RANT is the reciprocal of the real part of the admittance presented by the antenna ANT.

[0108] The intermediate impedance R1 corresponds to the impedance that will be presented at the intermediate node N1 of the virtual matching network MN0 (strictly speaking, the reciprocal of the real part of the admittance).

[0109] Capacitor C0 and inductor L0 Figure 4 The value of ) is obtained by reading the Smith chart and by referencing Figure 4 Equations 1 and 2 are defined for each segment SCT01, SCT02 (or SCTk- Figure 4 It is exported by ).

[0110] Equation 1:

[0111] Equation 2:

[0112] Where ω is the angular frequency at a frequency (f0) selected from the fundamental frequency band, R L The impedance presented to the left of each segment SCTk (strictly speaking, the reciprocal of the real part of the admittance), k∈[01; 02]; and R G It is the impedance presented to the right of each SCTk segment (strictly speaking, the reciprocal of the real part of the admittance), such as Figure 4 As shown.

[0113] The imaginary part of the admittance of the load at the output node of the power amplifier PA is generated by the inductor L3 of the virtual DC power supply stage DCFD0, through which the power amplifier PA is powered.

[0114] The method then includes generating a real matching and filtering network MFN (as referenced above) from inductor L0 and capacitor C0 elements of thus determined size in each segment SCT01, SCT02 of the virtual matching network MN0. Figure 1 (As described).

[0115] In this regard, refer to Figure 1 and Figure 2 .

[0116] The generation of the real matching and filtering network MFN involves replacing the inductor L3 of the virtual DC power supply stage DCFD0 with LC arrangement 30, and replacing the inductor and capacitor elements of the virtual first segment SCTO1 with corresponding LC arrangements 11 and 12. LC arrangements 30, 11, and 12 are configured to have impedances equivalent to the matching impedance of the virtual matching network MN0 in the baseband, and also to have resonant frequencies f30, f11, and f12, respectively, suitable for attenuating the harmonic bands of the baseband. Figure 5 ).

[0117] Furthermore, the generation of the real matching and filtering network MFN advantageously includes replacing the capacitive elements (or inductor elements) of the virtual second segment SCT02 with an LC arrangement, which is also configured to have an impedance equivalent to the matching impedance of the virtual matching network MN0 in the baseband, and also to have a resonant frequency f22 matched to attenuate the harmonic band of the baseband. Figure 5 ).

[0118] Specifically, one or more inductor elements L0 of the virtual matching network MN0 are replaced by one or more parallel LC arrangements 11, which are selected at a frequency f0 and a corresponding resonant frequency f within the fundamental frequency band. r They have the same impedance at the same location.

[0119] On the one hand, the inductance value L31 and the capacitance value C30 are selected together so that the resonant frequency of the LC filter (series shunt L31, C30) is located at the desired value seen from the output node of the power amplifier PA.

[0120] On the other hand, the inductance values ​​L31 and L32 of the inductor elements of the T-shaped LC arrangement 30 and the capacitance value C30 of the T-shaped LC arrangement 30, together with the resonance in the selected frequency band, are selected to have an imaginary part of the equivalent admittance that is equal to that required for impedance matching at the output node of PA, that is, such that the entire T-shaped LC arrangement 30 has an impedance equivalent to that of the inductor element L3 of the virtual network MN0 in the base frequency band.

[0121] For each segment SCT1 (SCT2), the inductance L and capacitance C of one or more components in the parallel LC arrangement 11 are given by equations 3 and 4:

[0122] Equation 3:

[0123] Equation 4:

[0124] Where L0 is the inductance value of the inductor element replaced in each segment SCT01 (SCT02), f0 is the frequency in the baseband, and f rIt is the resonant frequency of the corresponding parallel LC arrangement.

[0125] Furthermore, specifically, one or more capacitive elements C0 of the virtual matching network MN0 are replaced by series LC arrangements 12, 22, which have the same impedance at a selected frequency f0 within the fundamental frequency band and at the corresponding resonant frequency f r There is resonance at that point.

[0126] For each segment SCT1, SCT2, the capacitance C and inductance L of the components arranged in series LC 12, 22 are given by Equations 5 and 6.

[0127] Equation 5:

[0128] Equation 6:

[0129] Where C0 is the capacitance value of the capacitor element that is replaced in each segment SCT01 and SCT02, f0 is the frequency in the baseband, and f r It is the resonant frequency of the corresponding parallel LC arrangement.

[0130] Choose the resonant frequency f r This is to attenuate the harmonic frequency band of the base spectrum.

[0131] Now for reference Figure 5 .

[0132] Figure 5 As shown in the reference Figure 1 The description and references Figures 2 to 4 The obtained result of the matched and filtered network MFN has a resonant frequency (f r The advantageous positioning of f11, f12, f22, and f30.

[0133] Curve 51 shows the transmission gain of the matched and filtered network MFN, curve 52 shows the transmission gain of the network MFN in the baseband FB, curve 53 shows the real part of the impedance of the network MFN in the baseband FB, and curve 54 shows the imaginary part of the impedance of the network MFN in the baseband FB.

[0134] The resonant frequency of the parallel LC arrangement 11 of the first section SCT1 is denoted as f11, the resonant frequency of the series LC arrangement 12 of the first section SCT1 is denoted as f12, the resonant frequency of the series LC arrangement 22 of the second section SCT2 is denoted as f22, and the resonant frequency of the T-shaped LC arrangement 30 of the DC-DC power supply stage DCFD is denoted as f30 (reference). Figure 1 Different resonant frequencies will be directly represented by their corresponding references.

[0135] In an advantageous example, the resonant frequency f30 is located in half (e.g., the upper half) of the second harmonic band HB2, and the resonant frequency f12 is located in the other half (e.g., the lower half) of the second harmonic band HB2. The resonant frequency f22 is located in the third harmonic band HB3. The resonant frequency f11 is located between the fourth harmonic band HB4 and the fifth harmonic band HB5, that is, in the case where the fourth harmonic band HB4 and the fifth harmonic band HB5 overlap, it is located in the common part of said bands HB4 and HB5.

[0136] On the one hand, this provides attenuation greater than 35 dB up to the fourth harmonic band HB4, and attenuation greater than 25 dB at the fifth harmonic band HB5.

[0137] Furthermore, this example corresponds to a spatially optimized embodiment of a matched and filtered network (MFN) that exhibits optimized performance at high transmission frequencies.

[0138] On the one hand, when the baseband is so high that the interfering inductive elements associated with the grounded metal connection have a non-negligible effect in certain harmonic bands (usually the fourth and fifth harmonics), filtering of these harmonic bands cannot be performed using a series LC arrangement with ground coupling.

[0139] Therefore, the attenuation of the fourth and fifth harmonic frequency bands is primarily provided by the resonant frequency of the parallel LC arrangement 11 of the first segment SCT1, without considering the increased overall size, since the parallel LC arrangement is the only option unaffected by interference inductance issues on the ground connection.

[0140] On the other hand, it is assumed that in this type of embodiment, the inductor has a much larger size and occupies much more space than the capacitor; it is assumed that the capacitor C0 coupled to ground in the virtual matching network MN0 has a lower value near the antenna ANT and a higher value near the power amplifier PA; it is assumed that the value of the inductor required to make the capacitor resonate is inversely proportional to the square of the capacitance value and the resonant frequency; and the number of inductors added to make the series capacitor resonate should be minimized.

[0141] Therefore, the minimum capacitance value C0 is first associated with the maximum residual resonant frequency (in this example, the third harmonic band HB3). This minimizes the value of the inductor L in the series LC arrangement 22 of the second segment SCT2, as determined by Equation 6.

[0142] Then, the capacitive element of the first segment SCT01 of the virtual network MN0 is selected to resonate in the highest harmonic frequency band (i.e., the second harmonic frequency band HB2) that has not yet been filtered. This minimizes the value of the inductor L of the series LC arrangement 12 of the first segment SCT1, as determined by Equation 6, while also covering the second harmonic frequency band HB2.

[0143] Therefore, the two inductor elements of the series LC arrangement 12 and 22 in the two sections SCT1 and SCT2 require a minimum total space, while having resonant frequencies distributed in a manner suitable for attenuating harmonic frequency bands that are not attenuated by the parallel LC arrangement of the first section HB3 and HB2.

[0144] The resonant frequency of the T-shaped LC arrangement 30 can then be more freely positioned in the second harmonic band HB2 because the additional space required for the added capacitor element is small compared to the space required for the inductor element, especially compared to the space savings achieved by optimizing the inductor elements of the series LC arrangements 12, 22.

[0145] The inductor of the second section SCT2 is not replaced by a resonant LC arrangement in order to maintain a certain level of attenuation at high frequencies, which yields the best results in this configuration.

[0146] Curve 52 shows that the maximum loss in the baseband FB in this example is on the order of 1.6 dB.

[0147] Curves 53 and 54 show that, in this example, the real part of the impedance is contained as approximately 10% of, for example, 3.5 ohms, and the imaginary part is also contained as approximately 10 pF (picofarads).

[0148] Alternatively, for this example, if the base spectrum is not high enough to encounter interference inductance in the metallic ground connection, the resonant frequencies f12, f22 can be selected first for each series LC arrangement so that they are distributed in different harmonic bands of the base spectrum and such that the series LC arrangement that replaces the capacitor element of the virtual matching network with the minimum capacitance value has the maximum resonant frequency.

[0149] In other words, the minimum capacitance value C0 can first be associated with the maximum desired resonant frequency, i.e., in this alternative embodiment, with the fourth and fifth harmonic bands HB4 and HB5; then, another capacitor element (i.e., the capacitor element of the first segment SCT01 of the virtual network MN0) is selected to resonate in the highest harmonic band that has not yet been filtered (i.e., the third harmonic band HB3 in this alternative embodiment).

[0150] The resonant frequency is then selected for each parallel LC arrangement and low-pass T-shaped LC arrangement so that it is distributed in different harmonic bands of the fundamental frequency band, along with the resonant frequency of the series LC arrangement. The latter choice has no particular limitations in terms of space requirements, because the additional space required by the added capacitor element is smaller compared to the space required by the inductor element.

[0151] Therefore, a matching and filtering technique has been described that is advantageous in terms of size and performance, with a very inexpensive passive component arrangement.

[0152] In summary, matching is performed by a low-pass filter with two segments, SCT01 and SCT02, with the minimum quality factor in order to provide the real part of the impedance transformation.

[0153] The imaginary part of the optimal impedance of the power amplifier is generated by the inductor L3 of the DC power supply stage DCFD0.

[0154] A capacitive coupling element CC is added before the antenna to block DC voltage. Its value is chosen to be large enough to have a small effect on impedance transformation.

[0155] Harmonic suppression (filtering) is achieved by replacing the series inductor with a parallel LC arrangement 11 and by replacing the shunt capacitors with series LC arrangements 12 and 22. In the baseband, the equivalent reactance of the LC arrangements is maintained equal to the reactance of the elements they replace.

[0156] The inductor L3 of the DC-powered stage DCFD is also replaced by a low-pass T-shaped LC arrangement 30 to provide additional filtering at the resonant frequency of the T-shaped LC arrangement 30.

[0157] The resonant frequency of the LC arrangement can be advantageously selected as follows:

[0158] - A T-shaped LC arrangement introduces a short circuit to the top (or possibly the bottom) of the second harmonic frequency band at the output of the power amplifier;

[0159] - The first parallel LC arrangement blocks the fourth and fifth harmonic frequency bands;

[0160] - The first shunt is arranged in series with an LC circuit on the output side of the power amplifier, releasing frequencies in the bottom (or possibly top) of the second harmonic band;

[0161] The second branch is arranged in series with LC circuits to release frequencies in the third harmonic band.

[0162] - The inductor element in the last section, on the antenna side, is not replaced by a resonant circuit in order to provide attenuation for higher-order harmonics (i.e., greater than 5).

Claims

1. An integrated circuit comprising a power amplifier (PA) designed to provide a signal in a baseband frequency band, an antenna (ANT), and a matching and filtering network (MFN), said matching and filtering network comprising: - A DC power supply stage (DCFD) between the supply voltage node (VCC) and the output node of the power amplifier (PA). - The first segment (SCT1) between the output node and the intermediate node (N1) of the power amplifier (PA); and - The second segment (SCT2) between the intermediate node (N1) and the input node of the antenna (ANT). The DC-DC power supply stage (DCFD) and the two sections include an LC arrangement configured to have an impedance that matches the output of the power amplifier (PA) in the base frequency band. The LC arrangement of the DC power supply stage (DCFD) and the first segment (SCT1) is further configured to have resonant frequencies suitable for attenuating the harmonic bands of the base frequency band. The DC power supply stage (DCFD) includes a low-pass T-shaped LC arrangement, and the first section (SCT1) includes a parallel LC arrangement (11) coupled between the output node of the power amplifier (PA) and the intermediate node (N1), and a series LC arrangement (12) coupled between the intermediate node (N1) and the ground node (GND). The parallel LC arrangement (11) of the first segment (SCT1) is configured to have a highest resonant frequency (f11) suitable for attenuating the harmonic frequency band of the base spectrum band among the resonant frequencies of the DC power supply stage and the LC arrangement of the first segment.

2. The integrated circuit of claim 1, wherein the LC arrangement of the second segment (SCT2) comprises a series LC arrangement (22) and an inductor (L2) or a capacitor, the series LC arrangement (22) of the second segment being configured to have a resonant frequency suitable for attenuating the harmonic band of the base spectrum, and the inductor (L2) or the capacitor being configured not to introduce resonance within the harmonic band of the base spectrum.

3. The integrated circuit according to claim 2, wherein the second segment (SCT2) includes a series LC arrangement (22) coupled between the input node and the ground node (GND) of the antenna (ANT), and an inductor (L2) coupled between the intermediate node (N1) and the input node of the antenna (ANT), or includes a parallel LC arrangement coupled between the intermediate node (N1) and the input node of the antenna (ANT), and a capacitor coupled between the intermediate node (N1) and the ground node (GND).

4. The integrated circuit of claim 3, wherein the first segment (SCT1) includes a series LC arrangement (12) coupled between the output node of the power amplifier (PA) and the ground node (GND), and the second segment (SCT2) includes a series LC arrangement (22) coupled between the input node of the antenna (ANT) and the ground node (GND), the series LC arrangement (22) of the second segment (SCT2) being configured to have a resonant frequency (f22) greater than the resonant frequency (f12) of the series LC arrangement (12) of the first segment (SCT1).

5. The integrated circuit according to claim 3 or 4, wherein the LC arrangement is configured according to at least one of the following criteria: - The low-pass T-shaped LC arrangement (30) of the DC power supply stage (DCFD) is configured to have a resonant frequency (f30) in half of the second harmonic frequency band (HB2). - The series LC arrangement (12) of the first segment (SCT1) is configured to have a resonant frequency (f12) in the other half of the second harmonic frequency band (HB2). - The series LC arrangement (22) of the second section (SCT2) is configured to have a resonant frequency (f22) in the third harmonic band (HB3). - The parallel LC arrangement (11) of the first segment (SCT1) is configured to have a resonant frequency (f11) between the fourth harmonic band (HB4) and the fifth harmonic band (HB5), or in the common part of the fourth harmonic band (HB4) and the fifth harmonic band (HB5).

6. A method for impedance matching and filtering between the output of a power amplifier (PA) providing a signal in a baseband and an antenna (ANT), comprising determining the dimensions of a virtual matching network (MN0), the virtual matching network (MN0) comprising: - A DC power supply stage (DCFD0) between the supply voltage node (VCC) and the output node of the power amplifier (PA). - The first segment (SCT01) between the output node and the intermediate node (N1) of the power amplifier (PA); and The second segment (SCT02) between the intermediate node (N1) and the input node of the antenna (ANT). The DC power supply stage (DCFD0) includes an inductor (L3), and the first section (SCT01) and the second section (SCT02) include inductors and capacitors. The dimensions are designed to have an impedance that matches the output of the power amplifier (PA) in the baseband. The method includes generating a true matching and filtering network (MFN) by replacing the inductor (L3) of the DC power supply stage (DCFD0) and the inductor and capacitor of the first segment (SCT01) of the virtual matching network (MN0) with corresponding LC arrangements (30, 11, 12), wherein the corresponding LC arrangements (30, 11, 12) are configured to have an equivalent impedance matching the output of the power amplifier (PA) in the base frequency band and resonant frequencies (f30, f11, f12) respectively suitable for attenuating the harmonic bands of the base frequency band. The first segment (SCT01) of the virtual matching network (MN0) includes an inductive element coupled between the output node of the power amplifier (PA) and the intermediate node (N1), and a capacitive element coupled between the intermediate node (N1) and the ground node (GND). The generation of the real matching and filtering network (MFN) includes: replacing the inductor (L3) of the DC power supply stage (DCFD0) of the virtual matching network (MN0) with a low-pass T-shaped LC arrangement; replacing the inductor of the first section (SCT01) with a parallel LC arrangement (11); and replacing the capacitor of the first section (SCT01) with a series LC arrangement (12). The resonant frequency (f11) of the parallel LC arrangement (11) of the first segment (SCT1) of the real matched and filtered network is selected such that it is the highest among the resonant frequencies of the LC arrangement suitable for attenuating the harmonic band of the base spectrum.

7. The method of claim 6, wherein generating the True Matching and Filtering Network (MFN) comprises: The inductor or capacitor element of the second segment (SCTO2) of the virtual matching network (MN0) is replaced by a series LC arrangement (22), which is configured to have an equivalent impedance that matches the output of the power amplifier (PA) in the base frequency band and also has a resonant frequency (f22) suitable for attenuating the harmonic band of the base frequency band.

8. The method of claim 7, wherein the second segment (SCTO2) of the virtual matching network (MN0) includes an inductive element coupled between the intermediate node (N1) and the input node of the antenna (ANT), and a capacitive element coupled between the input node of the antenna (ANT) and the ground node (GND), and wherein the generation of the true matching and filtering network (MFN) includes: The capacitor element can be replaced by a series LC arrangement (22), or the inductor element can be replaced by a parallel LC arrangement.

9. The method of claim 8, wherein generating the True Matching and Filtering Network (MFN) comprises: The capacitor element of the virtual matching network (MN0) is replaced by a series LC arrangement (12, 22) in the first segment (SCT1) and the second segment (SCT2), and the resonant frequency (f22) of the series LC arrangement (22) in the second segment (SCT2) is selected such that it is higher than the resonant frequency (f12) of the series LC arrangement (12) in the first segment (SCT1).

10. The method of claim 8 or 9, wherein the true matched and filtered network (MFN) is generated according to at least one of the following criteria: - The T-shaped LC arrangement (30) of the DC power supply stage (DCFD) of the real matching and filtering network has a resonant frequency (f30) in half of the second harmonic frequency band (HB2). - The series LC arrangement (12) of the first segment (SCT1) has a resonant frequency (f12) in the other half of the second harmonic frequency band (HB2). - The series LC arrangement (12) of the second section (SCT2) has a resonant frequency (f22) in the third harmonic band (HB3). - The parallel LC arrangement (11) of the first segment (SCT1) has a resonant frequency (f11) between the fourth harmonic band (HB4) and the fifth harmonic band (HB5) or a resonant frequency (f11) in the common part of the fourth harmonic band (HB4) and the fifth harmonic band (HB5).

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