Use of electrochemical impedance spectroscopy (EIS) in continuous glucose monitoring

Abstract

Electrochemical impedance spectroscopy (EIS) is used in conjunction with continuous glucose monitors and continuous glucose monitoring (CGM) to enable in-vivo sensor calibration, total (sensor) failure analysis, and intelligent sensor diagnostics and failure detection. An equivalent circuit model is defined and each circuit element is used to characterize sensor behavior.

Description

[0001] This application is a divisional application of the invention patent application with application number 201480075588.5, application date November 26, 2014, entitled "Use of Electrochemical Impedance Spectroscopy (EIS) in Continuous Glucose Monitoring".

[0002] Relevant application information

[0003] This application claims the benefit of U.S. Provisional Application No. 61 / 916,637, filed December 16, 2013, the entire contents of which are incorporated herein by reference. Technical Field

[0004] Embodiments of the present invention generally relate to methods and systems for continuous glucose monitoring (CGM) using a continuous glucose monitor and electrochemical impedance spectroscopy (EIS), and more specifically, to the use of EIS in in vivo sensor calibration, total (sensor) fault analysis, and intelligent sensor diagnostics and fault detection. They also relate to application-specific integrated circuits (ASICs) for implementing the EIS applications for single-electrode and multi-electrode (redundant) sensors. Background Technology

[0005] Patients and healthcare professionals wish to monitor readings of physiological conditions within the patient's body. For example, patients may wish to continuously monitor their blood glucose levels. Currently, patients can measure their blood glucose (BG) using blood glucose measuring devices (i.e., glucose meters), such as test strips, continuous glucose measurement systems (or continuous blood glucose monitors), or hospital hemacues. BG measuring devices use various methods to measure a patient's BG levels, such as using a blood sample, a sensor in contact with bodily fluids, an optical sensor, an enzyme sensor, or a fluorescence sensor. When a BG measuring device produces a BG measurement result, the result is displayed on the device.

[0006] Current continuous glucose monitoring systems include subcutaneous (or short-term) sensors and implantable (or long-term) sensors. For each of these sensors, patients must wait a certain amount of time for the continuous glucose sensor to stabilize and provide accurate readings. In most continuous glucose sensors, the patient must wait up to three hours for the sensor to stabilize before any glucose measurement results can be used. This is inconvenient for patients and, in some cases, may prevent them from using the continuous glucose monitoring system.

[0007] Furthermore, when a glucose sensor is first inserted into a patient's skin or subcutaneous layer, the sensor is unstable. The electronic readings from the sensor, representing the patient's glucose level, vary over a wide range. In the past, sensor stabilization often took several hours. Techniques for sensor stabilization are detailed in U.S. Patent No. 6,809,653 (“653 Patent”), filed December 19, 1999, and granted to Mann et al. on October 26, 2004 (assigned to Medtronic Minimized, Inc.), the contents of which are incorporated herein by reference. In said 653 Patent, the initialization process for sensor stabilization can be reduced to approximately one hour. A high voltage (e.g., 1.0–1.2 volts) can be applied for one to two minutes to stabilize the sensor, and then a low voltage (e.g., between 0.5 and 0.6 volts) is applied for the remaining time of the initialization process (e.g., approximately 58 minutes). Even with this process, sensor stabilization still requires a significant amount of time.

[0008] Before using the sensor electrodes, it is desirable to ensure they are adequately "wetted" or hydrated. Insufficient hydration can result in inaccurate readings of the patient's physiological condition. Users of existing blood glucose sensors are instructed not to power them on immediately. Using them too early will prevent existing blood glucose sensors from operating optimally or effectively. There is no automated process or measurement technology to determine when to power on the sensor. This manual process is inconvenient and places too much responsibility on patients who may forget to use or turn on the power.

[0009] Besides stability and wetting issues during the initial phase of sensor lifespan, other problems may arise during the sensor's lifespan. For example, all sensors are preset with a defined operating lifespan. In existing short-term sensors on the market today, for instance, sensors are typically effective for 3 to 5 days. Although a sensor can continue to operate and transmit signals after its preset operating lifespan, sensor readings eventually become inconsistent and unreliable after the preset operating lifespan ends. The exact sensor lifespan varies for each individual sensor, but all sensors are certified to have at least a preset operating lifespan. Therefore, manufacturers have instructed users to replace sensors after their preset operating lifespan ends. Although continuous glucose monitoring systems can monitor the time since sensor insertion and indicate the end of the sensor's operating life to notify the user to replace the sensor, they do not have sufficient safeguards to prevent sensor use beyond its operating lifespan. While the monitor can simply stop operating once the sensor's operating lifespan is reached, patients can circumvent these safeguards by simply disconnecting and reconnecting the same sensor. Therefore, there is a vulnerability in the system where users can keep the sensor active for longer than recommended, thus compromising the accuracy of the blood glucose values ​​returned by the glucose monitor.

[0010] Furthermore, during the sensor's lifespan, it frequently absorbs contaminants such as peptides and small protein molecules. These contaminants reduce electrode surface area or diffusion channels for analytes and / or reaction byproducts, thereby decreasing sensor accuracy. During sensor operation, it is crucial to determine when these contaminants affect the sensor signal and how to remedy this situation.

[0011] The current state of continuous glucose monitoring (CGM) is largely adjunctive, meaning that readings from CGM devices (including, for example, implantable or subcutaneous sensors) cannot be used for clinical decision-making without a reference value. This reference value must be obtained via finger puncture using, for example, a BG meter. This reference value is necessary because the amount of information obtainable from the sensor / sensing component is limited. Specifically, currently, the only partially available information from the sensing component for processing is the raw sensor value (i.e., sensor current or Isig) and the counter voltage. Therefore, during analysis, if the raw sensor signal is abnormal (e.g., if the signal is decreasing), the only way to distinguish between sensor malfunction and physiological changes in the user / patient (i.e., changes in blood glucose levels) is to obtain a reference glucose value via finger puncture. This reference finger puncture is also known to be used to calibrate the sensor.

[0012] Various methods have been explored in this field to eliminate, or at least minimize, the finger punctures required for calibrating and evaluating sensor health. However, given the sheer number and complexity of various sensor failure modes, a satisfactory solution has yet to be found. At most, diagnostic methods based on direct evaluation of the Isig or comparison of two Isigs have been developed. In either case, because the Isig follows the body's glucose levels, it is by definition not independent of the analyte. Therefore, by itself, the Isig is not a reliable source of information for sensor diagnostics, nor a reliable predictor of continuous sensor performance.

[0013] Another limitation in this field to date is the lack of sensor electronics that, while managing sensor power, can not only operate the sensor but also perform real-time sensor and electrode diagnostics, including for redundant electrodes. Admittedly, the concept of electrode redundancy has existed for quite some time. However, until now, there has been little success in using electrode redundancy not only for acquiring more than one reading at a time but also for evaluating the relative health of redundant electrodes, the overall reliability of the sensor, and (if any) the frequency required to calibrate reference values.

[0014] Furthermore, even when redundant detection electrodes are used, their number is typically limited to two. This is partly due to a lack of advanced electronics for real-time operation, evaluation, and management of a large number of independent working electrodes (e.g., up to five or more). However, another limiting argument is that redundant electrodes are used to obtain “independent” sensor signals, and for this purpose, two redundant electrodes are sufficient. As noted, while this is one function of using redundant electrodes, it is not the only one.

[0015] The art is also exploring various circuit models to develop more accurate and reliable means for providing self-calibrating sensors and for performing sensor diagnostics. These models typically attempt to correlate circuit elements with parameters that can be used for intelligent diagnostics, total fault analysis, and real-time self-calibration. However, most such models have so far achieved only limited success. Summary of the Invention

[0016] According to one embodiment of the present invention, a method for real-time self-calibration of a sensor (the sensor having sensing electronics, a microcontroller, and at least one working electrode) includes: for the at least one working electrode, performing an electrochemical impedance spectroscopy (EIS) procedure via the microcontroller to obtain a value for at least one impedance-based parameter of the at least one working electrode; for the working electrode, periodically repeating the EIS procedure via the microcontroller to obtain additional values ​​for the at least one impedance-based parameter; calculating a value for at least one EIS-based parameter via the microcontroller based on the obtained and additional values ​​for the at least one impedance-based parameter; monitoring changes in the calculated values ​​of the at least one EIS-based parameter; and adjusting a calibration factor of the sensor via the microcontroller based on the changes in the calculated values.

[0017] According to another embodiment of the present invention, a method for real-time self-calibration of a sensor (the sensor having sensing electronics, a microcontroller, and at least one working electrode and a counter electrode) includes: for the at least one working electrode, executing a plurality of electrochemical impedance spectroscopy (EIS) programs via the microcontroller; generating a plurality of Nyquist curves via the microcontroller based on the respective outputs of the plurality of EIS programs; setting a baseline Nyquist curve length; setting a baseline higher frequency Nyquist curve slope; monitoring the Nyquist curve length and higher frequency Nyquist slope of the plurality of Nyquist curves via the microcontroller to detect changes in the Nyquist curve length and the higher frequency Nyquist slope; and adjusting a calibration factor of the sensor based on changes in the Nyquist curve length and the Nyquist slope.

[0018] In another embodiment of the invention, a method for performing a diagnosis subcutaneously or on an implanted sensor (the sensor having at least one working electrode) includes: defining a vector containing values ​​associated with one or more electrochemical impedance spectroscopy (EIS) parameters and values ​​associated with one or more non-EIS parameters; defining a corresponding threshold for each of the EIS-based parameters and each of the non-EIS-based parameters; performing a first EIS procedure to generate a first set of data for the values ​​associated with the one or more EIS-based parameters; performing a second EIS procedure after a predetermined (calculated) time interval to generate a second set of data for the values ​​associated with the one or more EIS-based parameters; updating the vector with the first set of data and the second set of data; and monitoring the vector values ​​to determine whether the sensor has lost sensitivity.

[0019] According to another embodiment of the present invention, a method for determining the age of a sensor includes: performing an electrochemical impedance spectroscopy (EIS) procedure before sensor initialization; generating a Nyquist curve based on the output of the EIS procedure; and determining whether the sensor is new or reused based on a lower frequency Nyquist slope.

[0020] According to another embodiment of the present invention, a method for distinguishing different glucose sensor types includes: performing an electrochemical impedance spectroscopy (EIS) procedure on the sensor; generating a Nyquist curve based on the output of the EIS procedure; and identifying one or more different sensor types based on the Nyquist curve. Attached Figure Description

[0021] Embodiments of the invention will be described in detail with reference to the accompanying drawings, in which the same reference numerals denote the same parts.

[0022] Figure 1 This is a perspective view of a subcutaneous sensor insertion assembly and a block diagram of a sensing electronic device according to an embodiment of the present invention.

[0023] Figure 2A A substrate with two sides is shown, the first side of which contains an electrode structure and the second side of which contains electronic circuitry.

[0024] Figure 2B The diagram shows the overall block diagram of the circuit used to detect the sensor output.

[0025] Figure 3 The illustration shows a block diagram of a sensing electronic device and a sensor including multiple electrodes according to an embodiment of the present invention.

[0026] Figure 4 The illustration shows alternative embodiments of the present invention, including sensors and sensing electronics according to embodiments of the present invention.

[0027] Figure 5 An electronic block diagram of a sensor electrode and a voltage applied to the sensor electrode according to an embodiment of the present invention is shown.

[0028] Figure 6A The illustration depicts a method for applying a pulse during a settling time range to reduce the settling time range according to an embodiment of the present invention.

[0029] Figure 6B The illustration shows a method for stabilizing a sensor according to an embodiment of the present invention.

[0030] Figure 6CThe illustration shows the utilization of feedback information during the stabilization of a sensor according to an embodiment of the present invention.

[0031] Figure 7 The illustration shows the function of a stabilizing sensor according to an embodiment of the present invention.

[0032] Figure 8A The illustration shows a block diagram of a sensing electronic device and a sensor according to an embodiment of the present invention. The sensing electronic device includes a voltage generating device.

[0033] Figure 8B The illustration shows a voltage generating device that implements this embodiment of the present invention.

[0034] Figure 8C The illustration shows a voltage generating device according to an embodiment of the present invention generating two voltage values.

[0035] Figure 8D The illustration shows a voltage generating device having three voltage generating systems according to an embodiment of the present invention.

[0036] Figure 9A The illustration shows a sensing electronics device including a microcontroller for generating voltage pulses according to an embodiment of the present invention.

[0037] Figure 9B The illustration shows a sensing electronic device including an analysis module according to an embodiment of the present invention.

[0038] Figure 10 The diagram illustrates a block diagram of a sensor system including hydrated electronic components according to an embodiment of the present invention.

[0039] Figure 11 The illustration shows an embodiment of the present invention including a mechanical switch that helps determine hydration time.

[0040] Figure 12 The illustration shows a method for detecting hydration according to an embodiment of the present invention.

[0041] Figure 13A The illustration shows a method for hydrating a sensor according to an embodiment of the present invention.

[0042] Figure 13B Other methods for verifying sensor hydration according to embodiments of the present invention are illustrated.

[0043] Figure 14A , Figure 14B and Figure 14C The illustration shows a method for combining sensor hydration and sensor stabilization according to an embodiment of the present invention.

[0044] Figure 15AThe illustration shows an EIS-based analysis of a system that responds to the application of a periodic AC signal according to an embodiment of the present invention.

[0045] Figure 15B The diagram illustrates a known circuit model used for electrochemical impedance spectroscopy.

[0046] Figure 16A An example of a Nyquist curve according to an embodiment of the present invention is illustrated, wherein an AC voltage plus a DC voltage (DC bias) is applied to the working electrode for a selected spectrum from 0.1 Hz to 1000 MHz.

[0047] Figure 16B Another example is shown with a Nyquist curve featuring a linear fit for relatively low frequencies and an intercept close to the real impedance value at relatively high frequencies.

[0048] Figure 16C and Figure 16D An infinite glucose sensor and a finite glucose sensor, respectively, responding to a sinusoidal operating potential, are shown.

[0049] Figure 16E Bode plots of amplitude according to an embodiment of the present invention are shown.

[0050] Figure 16F Bode plots of the phase according to an embodiment of the present invention are shown.

[0051] Figure 17 The figure illustrates the Nyquist curve of sensor impedance as a function of sensor age according to an embodiment of the present invention.

[0052] Figure 18 The illustration shows a method for applying EIS technology in sensor stabilization and sensor age detection according to an embodiment of the present invention.

[0053] Figure 19 The diagram illustrates a timeline for executing the EIS procedure according to an embodiment of the present invention.

[0054] Figure 20 The illustration shows a method for detecting and repairing sensors using an EIS procedure combined with remedial measures according to an embodiment of the present invention.

[0055] Figure 21A and Figure 21B An example of a sensor remedial measure according to an embodiment of the present invention is illustrated.

[0056] Figure 22The Nyquist curve of a normally functioning sensor is shown, where the Nyquist slope gradually increases and the intercept gradually decreases as the sensor is worn for a period of time.

[0057] Figure 23A The original current signals (Isig) from two redundant working electrodes according to an embodiment of the present invention are shown, as well as the actual impedances of the electrodes at 1 kHz.

[0058] Figure 23B It shows Figure 23A Nyquist curve of the first working electrode (WE1).

[0059] Figure 23C It shows Figure 23A Nyquist curve of the second working electrode (WE2).

[0060] Figure 24 An example of signal dip for two redundant working electrodes according to an embodiment of the present invention is illustrated, as well as the actual impedance of each electrode at 1 kHz.

[0061] Figure 25A The illustration shows the real impedance, virtual impedance, and substantial glucose independence of the phase of a normally functioning glucose sensor according to an embodiment of the present invention at relatively high frequencies.

[0062] Figure 25B An exemplary example of different glucose-dependent levels of real impedance at relatively low frequencies is shown according to an embodiment of the present invention.

[0063] Figure 25C An exemplary example of different glucose dependence levels at relatively low frequencies is shown according to an embodiment of the present invention.

[0064] Figure 26 The diagram illustrates the changing trends of the 1kHz real impedance, 1kHz virtual impedance, and relatively high frequency phase as the sensitivity of the glucose sensor decreases due to oxygen deficiency at the sensor insertion site, according to an embodiment of the present invention.

[0065] Figure 27 The isig and phases of hypoxia under different glucose concentration conditions in vitro are shown according to an embodiment of the present invention.

[0066] Figures 28A to 28C An example of sensitivity loss due to hypoxia according to an embodiment of the present invention is shown, wherein redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes are included.

[0067] Figure 28D It shows Figures 28A to 28CIn the example, the EIS-induced spike signal in the original Isig.

[0068] Figure 29 An example of sensitivity loss caused by hypoxia due to occlusion according to an embodiment of the present invention is shown.

[0069] Figures 30A to 30C An example of sensitivity loss due to biological contamination according to an embodiment of the present invention is shown, wherein redundant working electrodes WE1 and WE2 and EIS-based parameters of the electrodes are provided.

[0070] Figure 30D It shows Figures 30A to 30C The spike signal induced by EIS in the original Isig instance.

[0071] Figure 31 A diagnostic procedure for sensor fault detection according to an embodiment of the present invention is shown.

[0072] Figure 32A and Figure 32B Another diagnostic procedure for sensor fault detection according to an embodiment of the present invention is shown.

[0073] Figure 33A A top-level flowchart of a current-based (Isig) fusion algorithm according to an embodiment of the present invention is shown.

[0074] Figure 33B A top-level flowchart of a fusion algorithm based on sensor glucose (SG) according to an embodiment of the present invention is shown.

[0075] Figure 34 An embodiment of the present invention is shown. Figure 33B Details of the fusion algorithm based on sensor glucose (SG).

[0076] Figure 35 An embodiment of the present invention is shown. Figure 33A Details of the current-based (Isig) fusion algorithm.

[0077] Figure 36 This is an example of a sensor calibrating a stable state according to an embodiment of the present invention.

[0078] Figure 37 This is an example of a sensor used in calibration conversion according to an embodiment of the present invention.

[0079] Figure 38A This is an example of an EIS-based dynamic slope (with slope adjustment) for sensor calibration according to an embodiment of the present invention.

[0080] Figure 38BA flowchart illustrating an EIS-assisted sensor calibration process including low-start detection according to an embodiment of the present invention is shown.

[0081] Figure 39 The sensor current (Isig) and impedance amplitude at 1 kHz are shown when an external simulated interference object is adjacent to the sensor according to an embodiment of the present invention.

[0082] Figure 40A and Figure 40B They are shown respectively Figure 39 The Bode plots of phase and impedance in the simulation are shown.

[0083] Figure 40C It shows Figure 39 The simulated Nyquist curve is shown.

[0084] Figure 41 Another in vitro simulation with interfering agents is shown according to an embodiment of the present invention.

[0085] Figure 42A and Figure 42B An ASIC block diagram according to an embodiment of the present invention is illustrated.

[0086] Figure 43 A potentiostat configuration for a sensor with redundant working electrodes according to an embodiment of the present invention is shown.

[0087] Figure 44 It shows having Figure 43 The equivalent AC circuit between the electrodes of the sensor is shown in the potentiometer configuration.

[0088] Figure 45 A main block diagram of the EIS circuit in the analog front-end IC of a glucose sensor according to an embodiment of the present invention is shown.

[0089] Figures 46A to 46F It shows that Figure 45 The signal of the EIS circuit shown is simulated as a current with a 0-degree phase and 0-degree phase multiply.

[0090] Figures 47A to 47F It shows that Figure 45 The signal of the EIS circuit shown is simulated as a current with a 0-degree phase and a 90-degree phase multiply.

[0091] Figure 48 A circuit model according to an embodiment of the present invention is shown.

[0092] Figures 49A to 49C An example of a circuit model according to an alternative embodiment of the present invention is shown.

[0093] Figure 50A It is the Nyquist curve simulated by the overlaying equivalent circuit according to an embodiment of the present invention.

[0094] Figure 50B yes Figure 50A A magnified view of the high-frequency portion.

[0095] Figure 51 The Nyquist curve showing Cdl increasing along the direction of arrow A according to an embodiment of the present invention is shown.

[0096] Figure 52 The Nyquist curve showing α increasing along the direction of arrow A according to an embodiment of the present invention is shown.

[0097] Figure 53 R is shown according to an embodiment of the present invention. p The Nyquist curve increases in the direction of arrow A.

[0098] Figure 54 The Nyquist curve showing the increase of Warburg admittance along the direction of arrow A according to an embodiment of the present invention is shown.

[0099] Figure 55 The Nyquist curve showing λ increasing along the direction of arrow A according to an embodiment of the present invention is shown.

[0100] Figure 56 The effect of membrane capacitance on the Nyquist curve according to an embodiment of the present invention is shown.

[0101] Figure 57 The Nyquist curve showing the increase of membrane resistance along the direction of arrow A according to an embodiment of the present invention is shown.

[0102] Figure 58 The Nyquist curve, in which Rsol increases along the direction of arrow A according to an embodiment of the present invention, is shown.

[0103] Figures 59A to 59C The changes in EIS parameters associated with circuit elements during startup and calibration are shown according to an embodiment of the present invention.

[0104] Figures 60A to 60C The variations of a different set of EIS parameters associated with circuit elements during startup and initialization, according to an embodiment of the present invention, are illustrated.

[0105] Figures 61A to 61C This illustrates another set of different EIS parameters associated with circuit elements during startup and initialization, according to an embodiment of the present invention.

[0106] Figure 62 The EIS response of a plurality of electrodes according to an embodiment of the present invention is shown.

[0107] Figure 63 The diagram illustrates the effect of increasing glucose on the Isig-calibrated Nyquist curve according to an embodiment of the present invention.

[0108] Figure 64 The effect of the oxygen (Vcntr) response on the Nyquist curve according to an embodiment of the present invention is shown.

[0109] Figure 65 The displacement of the Nyquist curve due to temperature change is shown according to an embodiment of the present invention.

[0110] Figure 66 The relationship between Isig and blood glucose is illustrated according to an embodiment of the present invention.

[0111] Figures 67A to 67B Sensor drift according to an embodiment of the present invention is illustrated.

[0112] Figure 68 An increase in membrane resistance during sensitivity loss is shown according to an embodiment of the present invention.

[0113] Figure 69 The decrease in Warburg admittance during sensitivity loss is shown according to an embodiment of the present invention.

[0114] Figure 70 A calibration curve according to an embodiment of the present invention is shown.

[0115] Figure 71 The invention illustrates that, according to an embodiment of the invention, higher frequency semicircles become more pronounced on the Nyquist curve.

[0116] Figure 72A and Figure 72B The reduction of Vcntr rail and Cdl according to an embodiment of the present invention is shown.

[0117] Figure 73 The variation in the slope of the calibration curve according to an embodiment of the present invention is shown.

[0118] Figure 74 The variation in the length of the Nyquist curve according to an embodiment of the present invention is shown.

[0119] Figure 75 It shows Figure 74 Enlarged views of the lower and higher frequency regions of the Nyquist curve.

[0120] Figure 76A and Figure 76B The combined effect of increased membrane resistance, decreased Cdl, and Vcntr rails according to an embodiment of the present invention is illustrated.

[0121] Figure 77 The Cdl values ​​of two working electrodes according to an embodiment of the present invention are shown.

[0122] Figure 78 The Rp values ​​of two working electrodes according to an embodiment of the present invention are shown.

[0123] Figure 79 The combined effect of changing EIS parameters on the calibration curve according to an embodiment of the present invention is illustrated.

[0124] Figure 80 This illustrates that, according to an embodiment of the invention, the Nyquist curve is longer in the lower frequency region, even with sensitivity loss.

[0125] Figure 81 This is a flowchart of sensor self-calibration based on the detection of sensitivity changes according to an embodiment of the present invention.

[0126] Figure 82 The illustration shows the horizontal displacement of the Nyquist curve due to sensitivity loss according to an embodiment of the present invention.

[0127] Figure 83 A method for developing a heuristic EIS metric based on a Nyquist curve, according to an embodiment of the present invention, is shown.

[0128] Figure 84 The relationship between Rm and calibration factor according to an embodiment of the present invention is shown.

[0129] Figure 85 The relationship between Rm and normalized Isig according to an embodiment of the present invention is shown.

[0130] Figure 86 The Isig curves as a function of time are shown under various glucose level conditions according to embodiments of the present invention.

[0131] Figure 87 Cdl curves as a function of time for various glucose levels according to embodiments of the present invention are shown.

[0132] Figure 88 An embodiment of the present invention is shown. Figure 86 The second inflection point of the curve.

[0133] Figure 89 The invention illustrates the relationship with the present invention. Figure 88 The peak value corresponds to the second inflection point of Rm.

[0134] Figure 90 An example of the relationship between the calibration factor (CF) and Rmem+Rsol according to an embodiment of the present invention is shown.

[0135] Figure 91A This is a graph illustrating in vivo results of MARD based on all effective BGs during approximately the first 8 hours of the sensor's lifespan, according to an embodiment of the present invention.

[0136] Figure 91B This is a graph illustrating the median ARD number based on all valid BGs during approximately the first 8 hours of the sensor's lifespan, according to an embodiment of the present invention.

[0137] Figures 92A to 92C The calibration factor adjustment according to an embodiment of the present invention is shown.

[0138] Figures 93A to 93C The calibration factor adjustment according to an embodiment of the present invention is shown.

[0139] Figures 94A to 94C The calibration factor adjustment according to an embodiment of the present invention is shown.

[0140] Figure 95 An exemplary example of the initial decay in Cdl according to an embodiment of the present invention is shown.

[0141] Figure 96 The effect of removing illegal Radius current on Isig is shown according to an embodiment of the present invention.

[0142] Figure 97A The calibration factors for the two working electrodes before the removal of the illegal Radius current are shown according to an embodiment of the present invention.

[0143] Figure 97B The calibration factors for the two working electrodes after the removal of the illegal Radius current are shown according to an embodiment of the present invention.

[0144] Figure 98A and Figure 98B The effect of removing illegal Radius current on MARD according to an embodiment of the present invention is shown.

[0145] Figure 99 This is a graph showing the change of double-layer capacitance over time according to an embodiment of the present invention.

[0146] Figure 100The diagram illustrates the displacement of Rmem+Rsol during sensitivity loss and the appearance of a higher frequency semicircle according to an embodiment of the present invention.

[0147] Figure 101A A flowchart illustrating the use of combinational logic to detect sensitivity loss according to an embodiment of the present invention is shown.

[0148] Figure 101B A flowchart illustrating the use of combinational logic to detect sensitivity loss according to an embodiment of the present invention is shown.

[0149] Figure 102 An exemplary method for using the Nyquist slope as a marker to distinguish between new and old sensors, according to an embodiment of the present invention, is shown.

[0150] Figures 103A to 103C Exemplary examples of Nyquist curves with different lengths for different sensor configurations according to embodiments of the present invention are shown.

[0151] Figure 104 It shows Figures 103A to 103C The length of the Nyquist curve of the sensor as a function of time. Detailed Implementation

[0152] The following description takes into account the accompanying drawings, which form part of the invention, illustrating several embodiments of the invention. It should be understood that other embodiments may be used and changes in structure and operation may be made without departing from the scope of the invention.

[0153] The invention is described below with reference to example flowcharts of methods, systems, devices, apparatuses, programming, and computer program products. It should be understood that each block in the example flowcharts, and combinations of blocks in the example flowcharts, can be implemented by programming instructions, including computer program instructions (as can be depicted in the figures as any menu screen). These computer program instructions can be loaded onto a computer or other programmable data processing device (e.g., a controller, microcontroller, or processor in a sensing electronic device) to produce a machine, such that the instructions, which execute on the computer or other programmable data processing device, produce instructions for implementing the functions specified in the flowcharts or multiple flowcharts. These computer program instructions can also be stored in a computer-readable storage medium to instruct the computer or other programmable data processing device to operate in a particular manner, such that the instructions stored in the computer-readable storage medium produce an article of manufacture including instructions for implementing the functions specified in the flowcharts or multiple flowcharts. The computer program instructions can also be loaded onto a computer or other programmable data processing device to produce a series of operational steps that execute in the computer or other programmable device to produce a computer-executable program, such that the instructions, which execute on the computer or other programmable device, provide steps for implementing the functions specified in the flowcharts or multiple flowcharts and / or in the menus present herein. Programming instructions may also be stored in and / or implemented by electronic circuitry, including integrated circuits (ICs) and application-specific integrated circuits (ASICs) used with sensor devices, apparatuses, and systems.

[0154] Figure 1 This is a perspective view of a subcutaneous sensor insertion assembly according to an embodiment of the present invention, and a block diagram of a sensing electronics device. Figure 1 As shown, the subcutaneous sensor assembly 10 is configured for subcutaneously placing the active portion or similar portion of a flexible sensor 12 (see, for example, FIG. 2) at a selected site within the user's body. The subcutaneous or percutaneous portion of the sensor assembly 10 includes a hollow, grooved insertion needle 14 and a cannula 16. The needle 14 facilitates quick and easy subcutaneous placement of the cannula 16 at the subcutaneous insertion site. The interior of the cannula 16 contains the detection portion 18 of the sensor 12, and one or more sensor electrodes 20 are exposed to the user's bodily fluids through windows 22 formed on the cannula 16. In embodiments of the invention, one or more sensor electrodes 20 may include a counter electrode, a reference electrode, and one or more working electrodes. After insertion, the insertion needle 14 is withdrawn to properly leave the cannula 16 with the detection portion 18 and sensor electrodes 20 at the selected insertion site.

[0155] In certain embodiments, the subcutaneous sensor assembly 10 facilitates the precise placement of such a flexible thin-film electrochemical sensor 12 for monitoring specific blood parameters indicative of the user's condition. The sensor 12 monitors glucose levels in the body and can be used in conjunction with external or implantable automated or semi-automated drug infusion pumps to control insulin delivery to diabetic patients, as described in U.S. Patents 4,562,751, 4,678,408, 4,685,903, or 4,573,994.

[0156] In a particular embodiment, the flexible electrochemical sensor 12 may be constructed using thin-film masking technology as an elongated thin-film conductor embedded or wrapped between selected layers and films of insulating material, such as polyimide films or sheets. When the detection portion 18 (or active portion) of the sensor 12 is placed subcutaneously at the insertion site, the sensor electrode 20 at the end of the detection portion 18 is exposed through one of the insulating layers, directly contacting the patient's blood or other bodily fluids. The detection portion 18 is connected to a connection portion 24 terminating at a conductive contact or similar element, which is also exposed through one of the insulating layers. In alternative embodiments, other types of implantable sensors may also be used, such as chemically-based implantable sensors, optically-based implantable sensors, etc.

[0157] As is known in the art, the connection portion 24 and the contact pad are generally adapted to be directly electrically connected via wires to a suitable monitor or sensing electronics 100 for monitoring the user's condition in response to signals from the sensor electrodes 20. Further description of this general type of flexible thin-film sensor can be found in U.S. Patent No. 5,391,250, entitled "METHOD OFFABRICATING THIN FILM SENSORS," which is incorporated herein by reference. The connection portion 24 can be conveniently electrically connected to the monitor or sensing electronics 100, or conveniently electrically connected to the monitor or sensing electronics 100 via a connector block 28 (or the like), which is shown and described in U.S. Patent No. 5,482,473, entitled "FLEX CIRCUIT CONNECTOR," which is incorporated herein by reference. Therefore, according to embodiments of the invention, the subcutaneous sensor assembly 10 can be configured or formed to operate with a wired or wireless feature monitoring system.

[0158] The sensor electrode 20 can be used in a variety of detection applications and can be configured in various ways. For example, the sensor electrode 20 can be used in physiological parameter detection applications where certain types of biomolecules act as catalysts. For example, the sensor electrode 20 can be used in glucose and oxygen sensors having glucose oxidase (GOx) that catalyzes the reaction with the sensor electrode 20. The sensor electrode 20, together with biomolecules or some other catalysts, can be placed in vascular or non-vascular environments within the human body. For example, the sensor electrode 20 and biomolecules can be placed in a vein and be exposed to blood flow, or they can be placed in the subcutaneous region or abdominal region of the human body.

[0159] The monitor 100 may also be referred to as a sensing electronics device 100. The monitor 100 may include a power supply 110, a sensor interface 122, processing electronics 124, and data formatting electronics 128. The monitor 100 can be connected to the sensor assembly 10 via a connector and cable 102, the connector being electrically connected to the connection block 28 of the connection portion 24. In an alternative embodiment, the cable may be omitted. In this embodiment of the invention, the monitor 100 may include a suitable connector that is directly connected to the connection portion 104 of the sensor assembly 10. The sensor assembly 10 may be modified to place the connector portion 104 in a different location, such as on top of the sensor assembly, to facilitate the placement of the monitor 100 on top of the sensor assembly.

[0160] In embodiments of the invention, the sensor interface 122, processing electronics 124, and data formatting electronics 128 are formed as multiple separate semiconductor chips. However, alternative embodiments may combine various different semiconductor chips into a single custom semiconductor chip or multiple custom semiconductor chips. The sensor interface 122 is connected to a cable 102 connected to the sensor assembly 10.

[0161] The power source 110 may be a battery. The battery may include three sets of silver oxide 357 battery cells. In alternative embodiments, different battery chemistry, such as lithium-based chemistry, alkaline batteries, nickel metal hydride, etc., may be used, and different numbers of batteries may be used. The monitor 100 supplies power to the sensor assembly via the power source 110 through cable 102 and cable connector 104. In embodiments of the invention, the power is the voltage supplied to the sensor assembly 10. In embodiments of the invention, the power is the current supplied to the sensor assembly 10. In embodiments of the invention, the power is the voltage supplied to the sensor assembly 10 in a specific number of volts.

[0162] Figure 2A and Figure 2B An example is given of an implantable sensor and electronic components for driving the implantable sensor according to an embodiment of the present invention. Figure 2AA substrate 220 with two sides is shown, wherein a first side 222 of the two sides contains an electrode structure and a second side 224 of the two sides contains electronic circuitry. Figure 2A As can be seen, the first side 222 of the substrate includes two pairs of working electrodes 240, 242, 244, and 246 located opposite each other to the reference electrode 248. The second side 224 of the substrate includes circuitry. As shown, the electronic circuitry can be housed within a sealed enclosure 226, which provides a protective shell for the circuitry. This allows the sensor substrate 220 to be inserted into a vascular environment or other environment that would expose the electronic circuitry to fluids. By sealing the electronic circuitry within the sealed enclosure 226, the electronic circuitry can operate without the risk of short circuits caused by surrounding fluids. Figure 2A The diagram also shows a pad 228 for connecting the input and output lines of the electronic circuit. The electronic circuit itself can be manufactured in various ways. According to an embodiment of the invention, the electronic circuit can be manufactured into an integrated circuit using industry-standard techniques.

[0163] Figure 2B An example block diagram of a circuit for detecting sensor output according to an embodiment of the present invention is provided. At least one pair of sensor electrodes 310 can be connected to a data converter 312, the output of which can be connected to a counter 314. The counter 314 can be controlled by control logic 316. The output of the counter 314 can be connected to a line interface 318. The line interface 318 can be connected in parallel with input / output lines 320 and can also be connected to control logic 316. The input / output lines 320 can also be connected to a power rectifier 322.

[0164] The sensor electrode 310 can be used in a variety of detection applications and can be configured in various ways. For example, the sensor electrode 310 can be used in physiological parameter detection applications where certain types of biomolecules act as catalysts. For example, the sensor electrode 310 can be used in glucose and oxygen sensors having glucose oxidase (GOx) that catalyzes the reaction with the sensor electrode 310. The sensor electrode 310, together with biomolecules or other types of catalysts, can be placed in vascular or non-vascular environments within the human body. For example, the sensor electrode 310 and biomolecules can be placed in a vein and affected by blood flow.

[0165] Figure 3A block diagram illustrating a sensing electronics device and a sensor including multiple electrodes according to an embodiment of the present invention is provided. The sensor assembly or system 350 includes a sensor 355 and sensing electronics 360. The sensor 355 includes a counter electrode 365, a reference electrode 370, and a working electrode 375. The sensing electronics 360 includes a power supply 380, a regulator 385, a signal processor 390, a measurement processor 395, and a display / transmission module 397. The power supply 380 provides power (in the form of voltage, current, or voltage including current) to the regulator 385. The regulator 385 transmits a regulated voltage to the sensor 355. In an embodiment of the invention, the regulator 385 transmits voltage to the counter electrode 365 of the sensor 355.

[0166] Sensor 355 generates a sensor signal representing the concentration of the physiological characteristic being measured. For example, the sensor signal may represent a blood glucose reading. In embodiments of the invention using a subcutaneous sensor, the sensor signal may represent the hydrogen peroxide level in the subject's body. In embodiments of the invention using a blood sensor or a cranial sensor, the sensor signal represents the amount of oxygen being measured by the sensor. In embodiments of the invention using an implantable or long-term sensor, the sensor signal may represent the oxygen level in the subject's body. The sensor signal is measured at the working electrode 375. In embodiments of the invention, the sensor signal may be a current measured at the working electrode. In embodiments of the invention, the sensor signal may be a voltage measured at the working electrode.

[0167] After a sensor signal is measured at sensor 355 (e.g., a working electrode), signal processor 390 receives the sensor signal (e.g., a measured current or voltage). Signal processor 390 processes the sensor signal and generates a processed sensor signal. Measurement processor 395 receives the processed sensor signal and calibrates the processed sensor signal using a reference value. In embodiments of the invention, the reference value is stored in a reference memory and provided to measurement processor 395. Measurement processor 395 generates sensor measurement values. Sensor measurement values ​​may be stored in measurement memory (not shown). Sensor measurement values ​​may be sent to a display / transmission device for display on a display within a housing with sensing electronics or transmitted to an external device.

[0168] The sensor electronics 360 can be a monitor, which includes a display showing physiological characteristic readings. The sensor electronics 360 can also be installed in a desktop computer, pager, television with communication capabilities, laptop computer, server, network computer, personal digital assistant (PDA), mobile phone with computer functionality, infusion pump with display, glucose sensor with display, and / or a combination of infusion pump / glucose sensor. The sensor electronics 360 can be installed in a Blackberry device, network device, home network device, or device connected to a home network.

[0169] Figure 4 Examples of alternative embodiments of the invention, including sensors and sensing electronics according to embodiments of the invention, are illustrated. Sensor assembly or sensor system 400 includes sensing electronics 360 and sensor 355. The sensor includes a counter electrode 365, a reference electrode 370, and a working electrode 375. Sensing electronics 360 includes a microcontroller 410 and a digital-to-analog converter (DAC) 420. Sensing electronics 360 may also include a current-to-frequency converter (I / F converter) 430.

[0170] Microcontroller 410 includes software program code or programmable logic that, when executed, causes microcontroller 410 to transmit a signal to DAC 420, or the programmable logic causes microcontroller 410 to transmit a signal to DAC 420, wherein the signal represents a voltage level or voltage value to be applied to sensor 355. DAC 420 receives the signal and generates a voltage value at the level indicated by microcontroller 410. In embodiments of the invention, microcontroller 410 may frequently or infrequently change the representation of the voltage level in the signal. For example, a signal from microcontroller 410 may instruct DAC 420 to apply a first voltage value for one second and a second voltage value for two seconds.

[0171] Sensor 355 may receive a voltage level or voltage value. In an embodiment of the invention, counter electrode 365 may receive the output of an operational amplifier having a reference voltage and voltage value from DAC 420 as inputs. The application of the voltage level causes sensor 355 to generate a sensor signal representing the concentration of the physiological characteristic being measured. In an embodiment of the invention, microcontroller 410 may measure the sensor signal (e.g., current value) from the working electrode. For example, sensor signal measurement circuit 431 may measure the sensor signal. In an embodiment of the invention, sensor signal measurement circuit 431 may include a resistor and current may flow through the resistor to measure the value of the sensor signal. In an embodiment of the invention, the sensor signal may be a current level signal and sensor signal measurement circuit 431 may be a current-to-frequency (I / F) converter 430. Current-to-frequency converter 430 may measure the sensor signal based on a current reading, convert it into a frequency-based sensor signal, and transmit the frequency-based sensor signal to microcontroller 410. In an embodiment of the invention, microcontroller 410 is more likely to receive frequency-based sensor signals than non-frequency-based sensor signals. Microcontroller 410 receives sensor signals, whether frequency-based or non-frequency-based, and determines physiological characteristic values ​​of the subject, such as blood glucose levels. Microcontroller 410 may include program code that, when executed or run, is capable of receiving sensor signals and converting them into physiological characteristic values. In embodiments of the invention, microcontroller 410 may convert sensor signals into blood glucose levels. In embodiments of the invention, microcontroller 410 may use measurements stored in internal memory to determine the subject's blood glucose levels. In embodiments of the invention, microcontroller 410 may use measurements stored in external memory to help determine the subject's blood glucose levels.

[0172] After determining the physiological characteristic value via microcontroller 410, microcontroller 410 can store the measured value of the physiological characteristic value for a period of time. For example, blood glucose values ​​can be sent from the sensor to microcontroller 410 every one second or every five seconds, and the microcontroller can store the sensor measurement value of BG readings for 5 minutes or 10 minutes. Microcontroller 410 can transmit the measured value of the physiological characteristic value to a display on sensor electronics 360. For example, sensor electronics 360 can be a monitor including a display that provides blood glucose readings to the patient. In embodiments of the invention, microcontroller 410 can transmit the measured value of the physiological characteristic value to an output interface of microcontroller 410. The output interface of microcontroller 410 can transmit the measured value of the physiological characteristic value (e.g., blood glucose value) to an external device, such as an infusion pump, a combination of an infusion pump / blood glucose meter, a computer, a personal digital assistant, a pager, a network device, a server, a mobile phone, or any computing device.

[0173] Figure 5 An electronic block diagram illustrating a sensor electrode and a voltage applied to the sensor electrode according to an embodiment of the present invention is provided. Figure 5 In the illustrated embodiment of the invention, operational amplifier 530 or other servo control device can be connected to sensor electrode 510 via circuit / electrode interface 538. Operational amplifier 530, using feedback information from the sensor electrode, attempts to maintain a predetermined voltage (which is the expected applied voltage of the DAC) between reference electrode 532 and working electrode 534 by adjusting the voltage at counter electrode 536. Current can then flow from counter electrode 536 to working electrode 534. This current can be measured to detect the electrochemical reaction between sensor electrode 510 and the sensor's biomolecules, which are placed near sensor electrode 510 and act as catalysts. Figure 5 The disclosed circuitry can be used in long-term or implantable sensors, or in short-term or subcutaneous sensors.

[0174] In a long-term sensor implementation, when glucose oxidase (GOx) is used as the catalyst in the sensor, current can flow from the counter electrode 536 to the working electrode 534 as long as oxygen is present in the vicinity of the enzyme and sensor electrode 510. For example, if the voltage setpoint at the reference electrode 532 is maintained at approximately 0.5 volts, the amount of current flowing from the counter electrode 536 to the working electrode 534 has an appropriate linear relationship with a uniform slope to the amount of oxygen present in the region surrounding the enzyme and electrode. Therefore, the accuracy of determining the oxygen content in the blood can be improved by maintaining the reference electrode 532 at approximately 0.5 volts and using current-voltage curves of different blood oxygen levels in this region. Different embodiments of the invention may use different sensors with biomolecules different from glucose oxidase, and thus a voltage different from the 0.5 volt setpoint may be present at the reference electrode.

[0175] As discussed above, during the initial implantation or insertion of sensor 510, sensor 510 may produce inaccurate readings due to the patient's adjustment of the sensor and electrochemical byproducts generated by the catalyst used in the sensor. A settling time is required for many sensors to enable sensor 510 to provide accurate readings of the patient's physiological parameters. During the settling time, sensor 510 does not provide accurate blood glucose measurements. Sensor users and manufacturers would like to improve the range of sensor settling times so that the sensor can be used quickly after insertion into the patient's body or subcutaneous layer.

[0176] In previous sensor electrode systems, the settling time or time range was one to three hours. To shorten the settling time or time range and improve the timeliness of sensor accuracy, the sensor (or the sensor's electrodes) can be subjected to multiple pulses, rather than applying one pulse followed by another voltage. Figure 6AAn example is provided of a method according to an embodiment of the invention for applying multiple pulses during a settling time range to shorten the settling time range. In an embodiment of the invention, a voltage applying device applies a first voltage to an electrode for a first time or a first time period (600). In an embodiment of the invention, the first voltage may be a constant DC voltage. This results in the generation of an anode current. In an alternative embodiment of the invention, a digital-to-analog converter or another voltage source may apply a voltage to the electrode for a first time period. An anode current means driving electrons toward the electrode to which the voltage has been applied. In an embodiment of the invention, instead of a voltage, the applying device may apply a current. In an embodiment of the invention that applies voltage to a sensor, after the first voltage is applied to the electrode, a voltage regulator may wait (i.e., not apply voltage) for a second time, time range, or time period (605). In other words, the voltage applying device waits until the second time period expires. Not applying voltage generates a cathode current, allowing the electrode to acquire electrons. After applying the first voltage to the electrode for a first time period, not applying voltage for a second time period is repeated several times (610). This may be referred to as an anode and cathode cycle. In an embodiment of the invention, the stabilization method is repeated a total of three times, that is, a voltage is applied three times for a first time period, and after each voltage application, no voltage is applied for a second time period. In an embodiment of the invention, the first voltage may be 1.07 volts. In an embodiment of the invention, the first voltage may be 0.535 volts. In an embodiment of the invention, the first voltage may be approximately 0.7 volts.

[0177] Repeatedly applying and not applying voltage causes the sensor (and electrodes) to undergo an anode-cathode cycle. This anode-cathode cycle reduces electrochemical byproducts generated by the patient's body reacting to the insertion or implantation of the sensor. In embodiments of the invention, electrochemical byproducts lead to background current generation, which results in inaccurate measurements of the subject's physiological parameters. In embodiments of the invention, electrochemical byproducts can be eliminated. Under other operating conditions, electrochemical byproducts can be reduced or significantly reduced. A successful stabilization method achieves equilibrium in the anode-cathode cycle, significantly reduces electrochemical byproducts, and minimizes background current.

[0178] In an embodiment of the present invention, the first voltage applied to the electrode of the sensor can be a positive voltage. In an embodiment of the present invention, the applied first voltage can be a negative voltage. In an embodiment of the present invention, the first voltage can be applied to the working electrode. In an embodiment of the present invention, the first voltage can be applied to the counter electrode or the reference electrode.

[0179] In embodiments of the invention, the duration of the voltage pulse can be equal to the duration of no voltage application, for example, both being three minutes. In embodiments of the invention, the duration of voltage application or the duration of the voltage pulse can be different values; for example, the first time and the second time can be different. In embodiments of the invention, the first time period can be five minutes and the waiting time can be two minutes. In embodiments of the invention, the first time period can be two minutes and the waiting time (or the second time range) can be five minutes. In other words, the duration of applying the first voltage can be two minutes and the period of no voltage application can be five minutes. This time range is merely illustrative and not intended to limit the invention. For example, the first time range can be two minutes, three minutes, five minutes, or ten minutes, and the second time range can be five minutes, ten minutes, twenty minutes, etc. The time ranges (e.g., the first time and the second time) can depend on the unique characteristics of different electrodes, sensors, and / or the patient's physiological characteristics.

[0180] In embodiments of the invention, more or fewer pulses than three can be used to stabilize the glucose sensor. In other words, the number of repetitions can be greater than three or less than three. For example, four voltage pulses (e.g., a high voltage followed by no voltage) or six voltage pulses can be applied to one of the electrodes.

[0181] For example, three consecutive pulses of 1.07 volts (followed by their respective waiting periods) may be sufficient for a sensor implanted subcutaneously. In embodiments of the invention, three consecutive voltage pulses of 0.7 volts may be used. For sensors implanted in blood or cranial fluid (e.g., long-term or permanent sensors), the three consecutive pulses may have higher or lower voltage values, and may be negative or positive voltages. Furthermore, more than three (e.g., five, eight, twelve) pulses may be used to create an anodic-cathode cycle between the anodic and cathodic currents in any of the sensors, whether subcutaneous, blood, or cranial fluid.

[0182] Figure 6B An example is given of a method for stabilizing a sensor according to an embodiment of the present invention. Figure 6B In the illustrated embodiment of the invention, a voltage applying device may apply a first voltage to the sensor for a first time to initiate an anodic cycle (630) at the sensor electrode. The voltage applying device may be a DC power supply, a digital-to-analog converter, or a voltage regulator. After the first time period has elapsed, a second voltage is applied to the sensor for a second time to initiate a cathode cycle (635) at the sensor electrode. For example, unlike not applying a voltage, such as... Figure 6AAs illustrated in the method, different voltages (different from the first voltage) are applied to the sensor over a second time period. In embodiments of the invention, the application of the first voltage for a first time and the application of the second voltage for a second time are repeated several times (640). In embodiments of the invention, the application of the first voltage for a first time and the application of the second voltage for a second time may each be applied for a stable time period, for example, 10 minutes, 15 minutes, or 20 minutes, rather than repeated several times. This stable time period is the entire time period of the stabilization process, for example, until the sensor is stabilized (and thus the electrodes are stabilized). The advantages of this stabilization method are faster sensor operation, lower background current (in other words, suppression of some background current), and better glucose response.

[0183] In an embodiment of the invention, the first voltage can be 0.535 volts, applied for five minutes, and the second voltage can be 1.070 volts, applied for two minutes. In other words, in this embodiment, the voltage pulse scheme is repeated three times. The pulse method can be changed, extending the second time range, for example, the application time of the second voltage, from two minutes to five minutes, ten minutes, fifteen minutes, or twenty minutes. Furthermore, in this embodiment, after three repetitions, the rated operating voltage of 0.535 volts can be applied.

[0184] 1.070 and 0.535 volts are exemplary voltage values. Other voltage values ​​may be selected based on a variety of factors. These factors may include the type of enzyme used in the sensor, the membrane used in the sensor, the sensor's runtime, pulse length, and / or pulse amplitude. Under some operating conditions, the first voltage may range from 1.00 to 1.09 volts, and the second voltage may range from 0.510 to 0.565 volts. In other operating embodiments, the range encompassing the first and second voltages may have a higher range depending on the voltage sensitivity of the sensor electrodes, for example, 0.3 volts, 0.6 volts, or 0.9 volts. Under other operating conditions, the voltage range may be from 0.8 volts to 1.34 volts, and other voltage ranges may be from 0.335 to 0.735 volts. Under other operating conditions, the higher voltage range may be narrower than the lower voltage range. For example, the higher voltage range may be from 0.9 volts to 1.09 volts, while the lower voltage range may be from 0.235 volts to 0.835 volts.

[0185] In embodiments of the present invention, the first voltage and the second voltage can be positive voltages, or optionally, in other embodiments of the present invention, they can be negative voltages. In embodiments of the present invention, the first voltage can be positive and the second voltage can be negative, or optionally, the first voltage can be negative and the second voltage can be positive. The first voltage can have different voltage levels in each repetition. In embodiments of the present invention, the first voltage can be a constant DC voltage. In other embodiments of the present invention, the first voltage can be a ramp power supply, a sinusoidal voltage, a step voltage, or other commonly used voltage waveforms. In embodiments of the present invention, the second voltage can be a constant DC voltage, a ramp voltage, a sinusoidal voltage, a step voltage, or other commonly used voltage waveforms. In embodiments of the present invention, the first voltage or the second voltage can be an AC signal superimposed on a DC waveform. In embodiments of the present invention, the first voltage can be one type of voltage, such as a ramp voltage, and the second voltage can be another type of voltage, such as a sinusoidal voltage. In embodiments of the present invention, the first voltage (or the second voltage) can have different waveforms in each repetition. For example, if there are three cycles in the stabilization method, then in the first cycle, the first voltage can be a ramp voltage, in the second cycle, the first voltage can be a constant voltage, and in the third cycle, the first voltage can be a sinusoidal voltage.

[0186] In embodiments of the invention, the duration of the first time range and the duration of the second time range may have the same value, or alternatively, the duration of the first time range and the duration of the second time range may have different values. For example, the duration of the first time range may be two minutes, and the duration of the second time range may be five minutes, and the number of repetitions may be three. As discussed above, the stabilization method may include multiple repetitions. In embodiments of the invention, the duration of the first time range and the duration of the second time range may change in different repetitions of the stabilization method. For example, in the first repetition of the anode-cathode cycle, the first time range may be two minutes and the second time range may be five minutes. In the second repetition, the first time range may be one minute and the second time range may be three minutes. In the third repetition, the first time range may be three minutes and the second time range may be ten minutes.

[0187] In an embodiment of the invention, a first voltage of 0.535 volts is applied to the electrodes of the sensor for two minutes to initiate an anodic cycle, followed by a second voltage of 1.07 volts applied to the electrodes for five minutes to initiate a cathode cycle. Then, the first voltage of 0.535 volts is again applied for two minutes to initiate an anodic cycle, and the second voltage of 1.07 volts is applied to the sensor for five minutes. In a third repetition, 0.535 volts is applied for two minutes to initiate an anodic cycle, followed by 1.07 volts for five minutes. Then, within the actual operating time of the sensor, for example, when the sensor provides readings of the patient's physiological characteristics, a voltage of 0.535 volts is applied to the sensor.

[0188] Shorter duration voltage pulses can be used Figure 6A and Figure 6B In this embodiment, a shorter duration voltage pulse can be used to apply a first voltage, a second voltage, or both. In this embodiment, the amplitude of the shorter duration voltage pulse for the first voltage is -1.07 volts, and the amplitude of the shorter duration voltage pulse for the second voltage is approximately half the amplitude of the higher amplitude, for example, -0.535 volts. Alternatively, the amplitude of the shorter duration pulse for the first voltage may be 0.535 volts, and the amplitude of the shorter duration pulse for the second voltage may be 1.07 volts.

[0189] In embodiments of the invention using short-duration pulses, the voltage may not be applied continuously throughout the entire first time period. Instead, the voltage applying device may transmit multiple short-duration pulses within the first time period. In other words, multiple small-width or short-duration voltage pulses may be applied to the sensor electrodes within the first time period. The width of each small-width or short-duration pulse may be multiple milliseconds. For example, the pulse width may be 30 milliseconds, 50 milliseconds, 70 milliseconds, or 200 milliseconds. These values ​​are intended to be illustrative and not limiting of the invention. In embodiments of the invention, for example... Figure 6A In the illustrative embodiment, these short-duration pulses are applied to the sensor (electrode) for a first time period, and then no voltage is applied for a second time period.

[0190] In embodiments of the invention, each short-duration voltage pulse may have the same duration within a first time period. For example, each short-duration voltage pulse may have a time width of 50 milliseconds and the pulse delay between each pulse may be 950 milliseconds. In this embodiment, if the measurement time for the first time range is two minutes, then 120 short voltage pulses may be applied to the sensor. In embodiments of the invention, each short voltage pulse may have a different duration. In embodiments of the invention, each short-duration voltage pulse may have the same amplitude. In embodiments of the invention, each short-duration voltage pulse may have a different amplitude. By using short-duration voltage pulses, rather than continuously applying voltage to the sensor, the same anode and cathode cycle can be generated and the sensor (e.g., electrodes) consumes less total energy or charge over time. Compared to applying a continuous voltage to the electrodes, the application of short-duration voltage pulses uses less power because less energy is applied to the sensor (and thus the electrodes).

[0191] Figure 6C An example illustrates the use of feedback information during sensor stabilization according to an embodiment of the invention. The sensor system may include a feedback mechanism to determine whether additional pulses are needed to stabilize the sensor. In an embodiment of the invention, the sensor signal generated by the electrode (e.g., the working electrode) may be analyzed to determine whether the sensor signal is stable. A first voltage is applied to the electrode for a first time period to initiate an anode cycle (630). A second voltage is applied to the electrode for a second time period to initiate a cathode cycle (635). In an embodiment of the invention, an analysis module may analyze the sensor signal (e.g., the current emitted by the sensor signal, the resistance at a specific point on the sensor, the impedance at a specific node on the sensor) and determine whether a threshold measurement has been reached (e.g., by comparing with a threshold measurement to determine whether the sensor is providing an accurate reading) (637). If the sensor reading is determined to be accurate, this indicates that the electrode is stable (and thus the sensor is stable) (642), and no additional first and / or second voltages need to be applied. In an embodiment of the invention, if stabilization is not achieved, then additional anode / cathode cycles are initiated by applying the first voltage to the electrode for a first time period (630) followed by applying the second voltage to the electrode for a second time period (635).

[0192] In embodiments of the present invention, the analysis module can be used after three anode / cathode cycles of applying a first voltage and a second voltage to the electrodes of the sensor. In embodiments of the present invention, as... Figure 6C As illustrated by the example, the analysis module can be used after applying a first voltage and a second voltage.

[0193] In embodiments of the invention, the analysis module can be used to measure the voltage generated after current has flowed through the electrodes or through both electrodes. The analysis module can monitor the voltage level at the electrodes or the voltage level at the receiving level. In embodiments of the invention, if the voltage level is above a certain threshold, this indicates that the sensor is stable. In embodiments of the invention, if the voltage level drops below the threshold level, this indicates that the sensor is stable and ready to provide readings. In embodiments of the invention, current can be introduced into the electrodes or flow through both electrodes. The analysis module can monitor the current level transmitted by the electrodes. In this embodiment of the invention, if the current differs from the sensor signal current by an order of magnitude, the analysis module can monitor the current. If the current is above or below a current threshold, this indicates that the sensor is stable.

[0194] In an embodiment of the invention, the analysis module measures the impedance between the two electrodes of the sensor. The analysis module compares the impedance to a threshold impedance value or a target impedance value, and if the measured impedance is lower than the target impedance or threshold impedance, then the sensor is stable (and thus the sensor signal is stable). In another embodiment of the invention, the analysis module measures the resistance between the two electrodes of the sensor. In this embodiment, if the analysis module compares the resistance to a threshold resistance value or a target resistance value, and the measured resistance value is less than the threshold resistance value or the target resistance value, then the analysis module determines that the sensor is stable and the sensor signal is usable.

[0195] Figure 7 The function of a stabilized sensor according to an embodiment of the present invention is illustrated by example. Line 705 represents the glucose sensor reading of the glucose sensor when using a conventional single-pulse stabilization method. Line 710 represents the glucose sensor reading when three voltage pulses are applied (e.g., the three voltage pulses last for two minutes, and there is no voltage applied for five minutes after each voltage pulse). The x-axis 715 represents the amount of time. Points 720, 725, 730, and 735 represent glucose readings obtained using a finger-prick method and subsequently entered into the blood glucose meter. As shown, the conventional single-pulse stabilization method takes approximately 1 hour and 30 minutes to stabilize to a desired glucose reading, such as 100 units. In contrast, the three-pulse stabilization method takes only approximately 15 minutes to stabilize the glucose sensor and produces a significantly improved stabilization time range.

[0196] Figure 8AA block diagram illustrating a sensing electronics device and sensor according to an embodiment of the present invention is provided. The sensing electronics device includes a voltage generating device. The voltage generating or applying device 810 includes electronic components, logic, or circuitry for generating voltage pulses. The sensing electronics device 360 ​​may also include an input device 820 for receiving reference values ​​and other useful data. In an embodiment of the invention, the sensing electronics device may include a measurement memory 830 for storing sensor measurements. In this embodiment, a power supply 380 provides power to the sensing electronics device. The power supply 380 may also supply power to a regulator 385, which provides a regulated voltage to the voltage generating or applying device 810. In the illustrated embodiment of the invention, a connection terminal 811 represents a connection terminal that couples or connects the sensor 355 to the sensing electronics device 360.

[0197] exist Figure 8A In the illustrated embodiment of the invention, voltage generating or applying device 810 provides a voltage (e.g., a first voltage or a second voltage) to an input of operational amplifier 840. Voltage generating or applying device 810 may also provide a voltage to the working electrode 375 of sensor 355. Another input of operational amplifier 840 is connected to a reference electrode 370 of the sensor. The voltage applied to operational amplifier 840 by voltage generating or applying device 810 causes the voltage measured at counter electrode 365 to be close to or equal to the voltage applied to working electrode 375. In an embodiment of the invention, voltage generating or applying device 810 can be used to apply a desired voltage between counter electrode and working electrode. This can occur by applying a fixed voltage directly to counter electrode.

[0198] exist Figure 6A and Figure 6B In the illustrated embodiment of the invention, voltage generating device 810 generates a first voltage to be applied to the sensor within a first time range. Voltage generating device 810 sends this first voltage to operational amplifier 840, which drives the voltage at the counter electrode 365 of sensor 355 to reach the first voltage. In another embodiment of the invention, voltage generating device 810 may also directly transmit the first voltage to the counter electrode 365 of sensor 355. Figure 6A In the illustrated embodiment of the invention, the voltage generating device 810 subsequently does not transmit the first voltage to the sensor 355 for a second time period. In other words, the voltage generating device 810 is turned off or disconnected. The voltage generating device 810 can be programmed to continuously cycle between applying the first voltage and not applying the voltage several times or continuously cycle for a stable time period (e.g., for twenty minutes). Figure 8BAn example of a voltage generating device implementing this embodiment of the invention is provided. A voltage regulator 385 transmits a regulated voltage to the voltage generating device 810. A control circuit 860 controls the opening and closing of a switch 850. If the switch 850 is closed, then voltage is applied. If the switch 850 is open, then no voltage is applied. A timer 865 provides a signal to the control circuit 860 to instruct the control circuit 860 to turn the switch 850 on or off. The control circuit 860 includes logic that can instruct the circuit to open and close the switch 850 multiple times (to match the desired number of repetitions). In an embodiment of the invention, the timer 865 may also send a stabilization signal to identify the end of a stable sequence, i.e., the stable time range has elapsed.

[0199] In an embodiment of the present invention, the voltage generating device generates a first voltage lasting for a first time period and generates a second voltage lasting for a second time period. Figure 8C An example is provided to illustrate how a voltage generating device produces two voltage values ​​to implement an embodiment of the invention. In this embodiment, a two-position switch 870 is used. For example, if the first switch position 871 is turned on or off by a timer 865 commanding the control circuit 860, the voltage generating device 810 generates a first voltage for a first time range. After the first voltage has been applied for the first time range, the timer sends a signal indicating that the first time range has elapsed to the control circuit 860, and the control circuit 860 instructs the switch 870 to move to a second position 872. When the switch 870 is in the second position 872, the regulated voltage enters a voltage buck converter or buck converter 880, thereby reducing the regulated voltage to a lower value. This lower value is then delivered to the operational amplifier 840 for a second time range. After the timer 865 sends a signal indicating that the second time range has elapsed to the control circuit 860, the control circuit 860 moves the switch 870 back to the first position. This continues until the desired number of repetitions or a stable time range has elapsed. In an embodiment of the present invention, after the sensor has stabilized for a certain period of time, the sensor sends a sensor signal 350 to a signal processor 390.

[0200] Figure 8D Example of a voltage application device 810 for applying voltage to a sensor in a more complex manner. Voltage application device 810 may include a control device 860, a switch 890, a sinusoidal voltage generator 891, a ramp voltage generator 892, and a constant voltage generator 893. In other embodiments of the invention, voltage application may generate an AC wave based on a DC signal or other various voltage pulse waveforms. Figure 8DIn the illustrated embodiment of the invention, the control device 860 can move a switch to one of three voltage generation systems 891 (sine wave), 892 (ramp), and 893 (constant DC). This causes each of the voltage generation systems to produce a defined voltage waveform. Under certain operating conditions, for example, when a sine wave pulse is applied for all three pulses, the control device 860 can cause the switch 890 to connect the voltage from the voltage regulator 385 to the sine wave voltage generator 891, thereby causing the voltage application device 810 to generate a sine wave voltage. Under other operating conditions, for example, when a ramp voltage is applied to the sensor as the first voltage of the first pulse of three pulses, a sinusoidal voltage is applied to the sensor as the first voltage of the second pulse of three pulses, and a constant DC voltage is applied to the sensor as the first voltage of the third pulse of three pulses, the control device 860 can move the switch 890 between connecting the voltage from the voltage generating or applying device 810 to the ramp voltage generating system 892, then to the sinusoidal voltage generating system 891, and then to the constant DC voltage generating system 893 during a first time range in the anode / cathode cycle. In this embodiment of the invention, the control device 860 can also instruct or control the switch to connect one of the voltage generating subsystems to the voltage from the regulator 385 during a second time range (e.g., during the application of the second voltage).

[0201] Figure 9A An example is illustrated of a sensing electronics device according to an embodiment of the present invention, which includes a microcontroller for generating voltage pulses. Advanced sensing electronics devices may include a microcontroller 410 (see...). Figure 4 The system includes a digital-to-analog converter (DAC) 420, an operational amplifier 840, and a sensor signal measurement circuit 431. In an embodiment of the invention, the sensor signal measurement circuit may be a current-to-frequency (I / F) converter 430. Figure 9A In the illustrated embodiment of the invention, the software or programmable logic in the microcontroller 410 provides instructions to send a signal to the DAC 420, thereby instructing the DAC 420 to output a specific voltage to the operational amplifier 840. For example... Figure 9AAs shown by line 911, the microcontroller 410 can also be instructed to output a specific voltage to the working electrode 375. As discussed above, applying a specific voltage to the operational amplifier 840 and the working electrode 375 can drive the voltage measured at the counter electrode to reach a specific voltage amplitude. In other words, the microcontroller 410 outputs a signal representing the voltage or voltage waveform to be applied to the sensor 355 (e.g., the transport amplifier 840 connected to the sensor 355). In an alternative embodiment of the invention, a fixed voltage can be set by applying a voltage directly from the DAC 420 between the reference electrode and the working electrode 375. A similar result can also be obtained by applying a voltage to each electrode such that the difference between the voltages of the respective electrodes is equal to the fixed voltage applied between the reference electrode and the working electrode. Furthermore, the fixed voltage can be set by applying a voltage between the reference electrode and the counter electrode. Under certain operating conditions, the microcontroller 410 can generate pulses of a specific amplitude, which the DAC 420 understands as representing the application of a voltage of a specific amplitude to the sensor. After the first time interval, the microcontroller 410 (via program or programmable logic) outputs a second signal that instructs the DAC 420 not to output a voltage (for a given time interval). Figure 6A The method described is applicable to the sensing electronics 360 that operates, or outputs a second voltage (for the sensor electronics 360 that operates according to the method described above). Figure 6B The method described herein applies to the sensing electronics 360. After the second time period has elapsed, the microcontroller 410 repeats the following cycle: sending a signal indicating that a first voltage to be applied (lasting for a first time period) and subsequently sending a signal indicating that no voltage is applied or a second voltage is applied (lasting for a second time period).

[0202] Under other operating conditions, microcontroller 410 may generate a signal sent to DAC 420 instructing the DAC to output a ramp voltage. Under other operating conditions, microcontroller 410 may generate a signal sent to DAC 420 instructing the DAC 420 to output a voltage that is an analog sinusoidal voltage. These signals may be incorporated into any pulse modulation method discussed in the foregoing paragraphs of this application or earlier. In embodiments of the invention, microcontroller 410 may generate a series of instructions and / or pulses, which DAC 420 receives and understands to represent a pulse sequence to be applied. For example, microcontroller 410 may send a series of instructions (by signals and / or pulses) instructing DAC 420 to generate a constant voltage for a first repetition over a first time range, a ramp voltage for a first repetition over a second time range, a sinusoidal voltage for a second repetition over the first time range, and a square wave with two values ​​for a second repetition over the second time range.

[0203] Microcontroller 410 may include programmable logic or a program to continue the loop for a sustained, stable time range or repeat it multiple times. For example, microcontroller 410 may include counting logic to identify when a first time range or a second time range has passed. Furthermore, microcontroller 410 may include counting logic to identify when a stable time range has passed. If any of the aforementioned time ranges have passed, the counting logic may instruct the microcontroller to send a new signal or stop sending signals to DAC 420.

[0204] The use of microcontroller 410 allows multiple voltage amplitudes to be applied in multiple sequences and for multiple durations. In an embodiment of the invention, microcontroller 410 may include control logic or a program to instruct digital-to-analog converter 420 to send a voltage pulse of approximately 1.0 volt for a first time period of one minute, followed by a voltage pulse of approximately 0.5 volt for a second time period of four minutes, and repeat this cycle four times. In an embodiment of the invention, microcontroller 420 may be programmed to send a signal causing DAC 420 to apply voltage pulses of the same amplitude to each first voltage in each repetition. In an embodiment of the invention, microcontroller 410 may be programmed to send a signal causing DAC 420 to apply voltage pulses of different amplitudes to each first voltage in each repetition. In this embodiment of the invention, microcontroller 410 may also be programmed to send a signal causing DAC 420 to apply voltage pulses of different amplitudes to each second voltage in each repetition. For example, the microcontroller 410 can be programmed to send signals to cause the DAC 420 to apply a first voltage pulse of approximately 1.0 volt in the first repetition, a second voltage pulse of approximately 0.5 volt in the first repetition, a first voltage of 0.7 volt and a second voltage of 0.4 volt in the second repetition, and a first voltage of 1.2 volt and a second voltage of 0.8 volt in the third repetition.

[0205] The microcontroller 410 can also be programmed to instruct the DAC 420 to provide multiple short-duration voltage pulses for a first time period. In this embodiment of the invention, instead of applying a single voltage over the entire first time period (e.g., two minutes), multiple shorter-duration pulses can be applied to the sensor. In this embodiment, the microcontroller 410 can also be programmed to instruct the DAC 420 to provide multiple short-duration voltage pulses to the sensor for a second time period. For example, the microcontroller 410 can signal the DAC to apply multiple short-duration voltage pulses, where the short duration is 50 milliseconds or 100 milliseconds. Between these short-duration pulses, the DAC may not apply voltage or may apply a minimum voltage. The microcontroller can cause the DAC 420 to apply short-duration voltage pulses for the first time period, for example, two minutes. The microcontroller 410 can then signal the DAC to not apply any voltage or to apply short voltage pulses to the sensor at the magnitude of a second voltage for a second time period, for example, the second voltage may be 0.75 volts and the second time period may be five minutes. In an embodiment of the invention, the microcontroller 410 may send a signal to the DAC 420 to cause the DAC 420 to apply voltages of different amplitudes for each short-duration pulse within a first time range and / or a second time range. In another embodiment, the microcontroller 410 may send a signal to the DAC 420 to cause the DAC 420 to use a voltage amplitude with a fluctuating pattern for the short-duration voltage pulses for a duration within the first or second time range. For example, the microcontroller may send a signal or pulse instructing the DAC 420 to apply thirty 20-millisecond pulses to the sensor within the first time range. Each of the thirty 20-millisecond pulses may have the same amplitude or different amplitudes. In this embodiment of the invention, the microcontroller 410 may instruct the DAC 420 to apply short-duration pulses within the second time range or may instruct the DAC 420 to apply another voltage waveform within the second time range.

[0206] While Figures 6 through 8 disclose the application of voltage, current can also be applied to the sensor to initiate the stabilization process. For example, in Figure 6BIn the embodiments of the invention shown, a first current can be applied within a first time range to initiate an anode or cathode response, and a second current can be applied within a second time range to initiate a corresponding anode or cathode response. The application of the first and second currents can be repeated continuously multiple times or can be sustained for a stable time range. In embodiments of the invention, a first current can be applied within the first time range and a first voltage can be applied within the second time range. In other words, one of the anode or cathode cycles can be triggered by a current applied to the sensor, and the other of the anode or cathode cycle can be triggered by a voltage applied to the sensor. As described above, the applied current can be a constant current, a ramp current, a step-pulse current, or a sinusoidal current. In some operating conditions, the current is applied within the first time range as a short-duration pulse sequence.

[0207] According to an embodiment of the present invention, Figure 9B Examples are given of sensors and sensing electronics that use an analysis module to process feedback information during stable periods. Figure 9B An analysis module 950 is introduced into the sensing electronics 360. The analysis module 950 uses feedback information from the sensor to determine whether the sensor is stable. In embodiments of the invention, the microcontroller 410 may include instructions or commands to control the DAC 420 to apply voltage or current to a portion of the sensor 355. Figure 9B An example is given illustrating that a voltage or current can be applied between the reference electrode 370 and the working electrode 375. However, a voltage or current can be applied between the electrodes or directly to one of the electrodes, and the invention is not limited to this. Figure 9B An illustrative implementation is shown. The applied voltage or current is indicated by the dashed line 955. The analysis module 950 can measure the voltage, current, resistance, or impedance in the sensor 355. Figure 9BThe example illustrates that the measurement occurs at the working electrode 375, but this should not limit the invention, as other embodiments of the invention may measure voltage, current, resistance, or impedance between the electrodes of the sensor or directly at the reference electrode 370 or the counter electrode 365. The analysis module 950 may receive the measured voltage, current, resistance, or impedance and may compare the measured values ​​with stored values ​​(e.g., thresholds). The dashed line 956 indicates that the analysis module 950 reads or acquires the measured values ​​of voltage, current, resistance, or impedance. Under some operating conditions, if the measured voltage, current, resistance, or impedance is above the threshold, the sensor is stable, and the sensor signal provides an accurate reading of the patient's physiological condition. Under other operating conditions, if the measured voltage, current, resistance, or impedance is below the threshold, the sensor is stable. Under other operating conditions, the analysis module 950 may verify that the measured voltage, current, resistance, or impedance is stable over a specific time range (e.g., one minute or two minutes). This may indicate that the sensor 355 is stable and that the sensor signal is transmitting accurate measurements of the patient's physiological parameters (e.g., blood glucose levels). After the analysis module 950 has determined that the sensor is stable and that the sensor signal provides accurate measurements, the analysis module 950 can transmit a signal (e.g., a sensor stability signal) to the microcontroller 410, indicating that the sensor is stable and that the microcontroller 410 can begin using or receiving signals from the sensor 355. This is indicated by the dashed line 957.

[0208] Figure 10 A block diagram illustrating a sensor system including hydration electronic components according to an embodiment of the present invention is provided. The sensor system includes a connector 1010, a sensor 1012, and a monitor or sensing electronics device 1025. The sensor 1012 includes electrodes 1020 and a connection portion 1024. In embodiments of the invention, the sensor 1012 can be connected to the sensing electronics device 1025 via the connector 1010 and a cable. In other embodiments of the invention, the sensor 1012 can be directly connected to the sensing electronics device 1025. In other embodiments of the invention, the sensor 1012 can be incorporated into the same physical device as the sensing electronics device 1025. The monitor or sensing electronics device 1025 may include a power supply 1030, a regulator 1035, a signal processor 1040, a measurement processor 1045, and a processor 1050. The monitor or sensing electronics device 1025 may also include a hydration detection circuit 1060. The hydration detection circuit 1060 is connected to the sensor 1012 to determine whether the electrodes 1020 of the sensor 1012 are sufficiently hydrated. If electrode 1020 is not adequately hydrated, it will not provide accurate glucose readings. Therefore, it is crucial to know when electrode 1020 is adequately hydrated. Once electrode 1020 is adequately hydrated, accurate glucose readings can be obtained.

[0209] exist Figure 10 In the illustrated embodiment of the invention, the hydration detection circuit 1060 may include a delay or timer module 1065 and a connection detection module 1070. In embodiments of the invention using a short-term sensor or a subcutaneous sensor, after the sensor 1012 has been inserted into the subcutaneous tissue, the sensing electronics or monitor 1025 is connected to the sensor 1012. The connection detection module 1070 identifies that the sensing electronics 1025 has been connected to the sensor 1012 and sends a signal to the timer module 1065. This is in Figure 10 Arrow 1084 indicates that detector 1083 detects a connection and sends a signal to connection detection module 1070 indicating that sensor 1012 and sensing electronics 1025 are connected. In embodiments of the invention using implantable or long-term sensors, connection detection module 1070 identifies that the implantable sensor has been inserted into the body. Timer module 1065 receives the connection signal and waits for a set or predetermined hydration time. For example, the hydration time may be two minutes, five minutes, ten minutes, or twenty minutes. These embodiments are intended to be illustrative and not limiting of the invention. The time range is not necessarily a set of minutes and may include any number of seconds. In embodiments of the invention, after timer module 1065 has waited for the set hydration time, timer module 1065 can notify processor 1050 by sending a hydration signal: sensor 1012 has hydrated, which is indicated by line 1086.

[0210] In this embodiment of the invention, processor 1050 may receive a hydration signal and only begin using sensor signals (e.g., sensor measurements) after receiving the hydration signal. In another embodiment of the invention, hydration detection circuit 1060 may be connected between the sensor (sensor electrode 1020) and signal processor 1040. In this embodiment of the invention, hydration detection circuit 1060 may prevent sensor signals from being sent to signal processor 1040 until timer module 1065 notifies hydration detection circuit 1060 that the set hydration time has elapsed. This is indicated by the dashed lines marked by reference numerals 1080 and 1081. For example, timer module 1065 may send a connection signal to a switch (or transistor) to turn on the switch and allow sensor signals to be transmitted to signal processor 1040. In an alternative embodiment of the invention, in hydration detection circuit 1060, timer module 1065 may send a connection signal after the hydration time has elapsed to turn on switch 1088 (or close switch 1088), thereby applying voltage from regulator 1035 to sensor 1012. In other words, in this embodiment of the invention, the voltage from the regulator 1035 may not be applied to the sensor 1012 before the hydration time has elapsed.

[0211] Figure 11An example of the present invention includes a mechanical switch that helps determine hydration time. In an embodiment of the invention, a single housing may include a sensor assembly 1120 and sensing electronics 1125. In an embodiment of the invention, the sensor assembly 1120 may be located within one housing and the sensing electronics 1125 may be located within another housing, but the sensor assembly 1120 and the sensing electronics 1125 may be connected together. In this embodiment of the invention, the connection detection mechanism 1160 may be a mechanical switch. The mechanical switch can detect that the sensor 1120 is physically connected to the sensing electronics 1125. In an embodiment of the invention, a timer circuit 1135 may also be activated when the mechanical switch 1160 detects that the sensor 1120 is connected to the sensing electronics 1125. In other words, the mechanical switch may be closed and a signal may be transmitted to the timer circuit 1135. Once the hydration time has elapsed, the timer circuit 1135 sends a signal to the switch 1140 to cause the regulator 1035 to apply voltage to the sensor 1120. In other words, no voltage is applied before the hydration time has elapsed. In embodiments of the invention, once the hydration time has elapsed, current may be applied to the sensor instead of voltage. In an alternative embodiment of the invention, power may initially be applied to the sensor 1120 when the mechanical switch 1160 detects that the sensor 1120 is physically connected to the sensing electronics 1125. The power sent to the sensor 1120 causes a sensor signal to be output from the working electrode of the sensor 1120. The sensor signal may be measured and sent to the processor 1175. The processor 1175 may include a counter input. Under certain operating conditions, after a set hydration time, starting from the time the sensor signal is input to the processor 1175, has elapsed, the processor 1175 may begin processing the sensor signal as an accurate measurement of glucose in the subject. In other words, the processor 1170 has received the sensor signal from the constant potential circuit 1170 for a period of time but has not processed the signal until it receives an instruction from the counter input of the processor, recognizing that the hydration time has elapsed. In embodiments of the invention, the constant potential circuit 1170 may include a current-to-frequency converter 1180. In this embodiment of the invention, the current-to-frequency converter 1180 can receive sensor signals as current values ​​and convert the current values ​​into frequency values, which is easier for the processor 1175 to process.

[0212] In an embodiment of the invention, when sensor 1120 is disconnected from sensing electronics 1125, mechanical switch 1160 can also notify processor 1175. This is in Figure 11The dotted line 1176 indicates this. This allows the processor 1170 to shut off or reduce power to several components, chips, and / or circuits of the sensing electronics 1125. If the sensor 1120 is not connected, and if components or circuits of the sensing electronics 1125 are in the ON state, the battery or power supply may be depleted. Therefore, if the mechanical switch 1160 detects that the sensor 1120 has been disconnected from the sensing electronics 1125, the mechanical switch can indicate this to the processor 1175, which can then shut off or reduce power to one or more of the circuits, chips, or components of the sensing electronics 1125.

[0213] Figure 12 An example of an electrical method for detecting hydration according to an embodiment of the present invention is illustrated. In this embodiment, an electronic detection mechanism can be used to detect sensor connections. In this embodiment, the hydration detection electronics 1250 may include an AC power source 1255 and a detection circuit 1260. The hydration detection electronics 1250 may be located in a sensing electronics 1225. The sensor 1220 may include a counter electrode 1221, a reference electrode 1222, and a working electrode 1223. Figure 12 As shown, the AC source 1255 is connected to the voltage setting device 1275, the reference electrode 1222, and the detection circuit 1260. In this embodiment of the invention, an AC signal from the AC source is applied to the reference electrode connection, as shown... Figure 12 As shown by the dashed line 1291 in the diagram. In an embodiment of the invention, the AC signal is coupled to the sensor 1220 through an impedance, and if the sensor 1220 is connected to the sensing electronics 1225, the coupled signal is significantly weakened. Therefore, a low-level AC signal appears at the input of the detection circuit 1260. This is also referred to as a high-attenuation signal or a signal with a high attenuation level. Under some operating conditions, the voltage level of the AC signal can be V. 施加 *(C 耦合 ) / (C 耦合 +C 传感器 If the detection circuit 1260 detects a high-level AC signal (low-attenuation signal) at its input, no interrupt signal is sent to the microcontroller 410 because the sensor 1220 is not yet fully hydrated or activated. For example, the input of the detection circuit 1260 could be a comparator. If the sensor 1220 is fully hydrated (or wetted), an effective capacitance is formed between the counter electrode and the reference electrode (e.g., ...). Figure 12 The capacitor C in r-c ), and form an effective capacitance between the reference electrode and the working electrode (e.g., Figure 12 The capacitor C in w-rIn other words, effective capacitance refers to the capacitance formed between two nodes, and does not imply the placement of an actual capacitor between two electrodes in the circuit. In an embodiment of the present invention, the AC signal from AC source 1255 is controlled by capacitor C. r-c and C w-r Sufficient attenuation is achieved, and the detection circuit 1260 detects the presence of a low-level or highly attenuated AC signal from the AC source 1255 at its input. This embodiment of the invention is crucial because the use of existing connections between the sensor 1120 and the sensing electronics 1125 reduces the number of sensor connections. In other words, Figure 11 The disclosed mechanical switch requires associated connections between the switch and sensor 1120 and sensing electronics 1125. Eliminating the mechanical switch is advantageous because the continuous reduction in size of sensor 1120 and the elimination of components contribute to this size reduction. In an alternative embodiment of the invention, the AC signal may be applied to different electrodes (e.g., counter electrode or working electrode), and the invention may operate in a similar manner.

[0214] As described above, after the detection circuit 1260 detects a low-level AC signal at its input, it can subsequently detect a low-attenuation high-level AC signal at its input. This indicates that the sensor 1220 has been disconnected from the sensing electronics 1225 or that the sensor is not functioning properly. If the sensor has been disconnected from the sensing electronics 1225, the AC source can couple to the input of the detection circuit 1260 with little or no attenuation. As described above, the detection circuit 1260 can generate an interrupt signal sent to the microcontroller. This interrupt signal can be received by the microcontroller, and the microcontroller can reduce or remove power to one or more components or circuits in the sensing electronics 1225. This can be referred to as a second interrupt signal. Moreover, this helps to reduce the power consumption of the sensing electronics 1225, especially when the sensor 1220 is not connected to the sensing electronics 1225.

[0215] exist Figure 12 In an optional embodiment of the invention shown, an AC signal may be applied to a reference electrode 1222, as indicated by reference numeral 1291, and an impedance measuring device 1277 may measure the impedance of a region within the sensor 1220. For example, this region may be the area between the reference electrode and the working electrode, such as... Figure 12As shown by dashed line 1292. Under certain operating conditions, if the measured impedance has decreased below an impedance threshold or other set standard, the impedance measuring device 1277 may send a signal to the detection circuit 1260. This indicates that the sensor is fully hydrated. Under other operating conditions, once the impedance exceeds the impedance threshold, the impedance measuring device 1277 may send a signal to the detection circuit 1260. The detection circuit 1260 then sends an interrupt signal to the microcontroller 410. In another embodiment of the invention, the impedance measuring device 1277 may directly send an interrupt signal or other signal to the microcontroller.

[0216] In an alternative embodiment of the invention, the AC source 1255 can be replaced by a DC source. If a DC source is used, a resistance measuring element can be used instead of the impedance measuring element 1277. In an embodiment of the invention using a resistance measuring element, once the resistance drops below a resistance threshold or a set standard, the resistance measuring element can send a signal to the detection circuit 1260 (indicated by the dashed line 1293) or directly to the microcontroller, indicating that the sensor is fully hydrated and that power can be applied to the sensor.

[0217] exist Figure 12 In the illustrated embodiment of the invention, if the detection circuit 1260 detects a low-level or high-attenuation AC signal from an AC source, it generates an interrupt signal to the microcontroller 410. This interrupt signal indicates that the sensor is fully hydrated. In this embodiment of the invention, in response to the interrupt signal, the microcontroller 410 generates a signal that is transmitted to the digital-to-analog converter 420 to instruct or cause the digital-to-analog converter 420 to apply voltage or current to the sensor 1220. Figure 6A , Figure 6B or Figure 6CAny different pulse or short pulse sequence described in the relevant text describing pulse application may be applied to sensor 1220. For example, a voltage from DAC 420 may be applied to operational amplifier 1275, the output of which is applied to counter electrode 1221 of sensor 1220. This causes the sensor (e.g., working electrode 1223 of the sensor) to generate a sensor signal. Because the sensor is fully hydrated, as identified by the interrupt signal, the sensor signal generated at working electrode 1223 is a precisely measured glucose. The sensor signal is measured by sensor signal measuring device 431 and transmitted to microcontroller 410, where physiological parameters of the patient are measured. The generation of the interrupt signal indicates that the sensor is fully hydrated and that sensor 1220 is providing a precise glucose measurement. In this embodiment of the invention, the hydration time may depend on the sensor type and / or manufacturer and the sensor's response to insertion or implantation in the patient. For example, a sensor 1220 may have a hydration time of five minutes, and a sensor 1220 may have a hydration time of one minute, two minutes, three minutes, six minutes, or 20 minutes. Moreover, any amount of time can be an acceptable amount of hydration time for the sensor, but a smaller amount of time is preferred.

[0218] If sensor 1220 has been connected but is not adequately hydrated or wetted, then the effective capacitance C... r-c and C w-c The AC signal from AC source 1255 may not be attenuated. The electrodes in sensor 1120 are dry before insertion, and because the electrodes are dry, there is no good electrical path (or conductive path) between the two electrodes. Therefore, high-level AC signals or low-attenuation AC signals can still be detected by detection circuit 1260 without generating an interrupt signal. Once the sensor is inserted, the electrodes are immersed in a conductive fluid. This creates a leakage path with low DC resistance. Moreover, a boundary layer capacitor is formed at the metal / fluid interface. In other words, a considerable capacitance is formed between the metal / fluid interface, and this large capacitance appears as two capacitors connected in series between the sensor electrodes. This can be referred to as the effective capacitance. In practice, the conductivity of the electrolyte on the electrodes is measured. In some embodiments of the invention, a glucose-limiting membrane (GLM) also exemplifies impedance that blocks electrical efficiency. Non-hydrated GLMs produce high impedance, while highly hydrated GLMs produce low impedance. Low impedance is desirable for accurate sensor measurements.

[0219] Figure 13AAn example of a method for hydrating a sensor according to an embodiment of the present invention is illustrated. In an embodiment of the invention, the sensor may be physically connected to a sensing electronics device (1310). After connection, in one embodiment of the invention, a timer or counter is started to calculate the hydration time (1320). After the hydration time has elapsed, a signal may be transmitted to a subsystem in the sensing electronics device to begin applying voltage to the sensor (1330). As discussed above, in an embodiment of the invention, a microcontroller may receive the signal and instruct a DAC to apply voltage to the sensor, or in another embodiment of the invention, a switch may receive the signal, which causes a regulator to apply voltage to the sensor. The hydration time may be five minutes, two minutes, ten minutes, and may vary depending on the patient and the type of sensor.

[0220] In an optional embodiment of the invention, after the sensor is connected to the sensing electronics, an AC signal (e.g., a low-voltage AC signal) can be applied to the sensor, for example, to the sensor's reference electrode (1340). Since the connection between the sensor and the sensing electronics allows the application of an AC signal to the sensor, an AC signal can be applied. After the AC signal is applied, an effective capacitance (1350) is formed between the voltage-applied electrode of the sensor and two other electrodes. The detection circuit determines the level of the AC signal present at the input of the detection circuit (1360). If a low-level AC signal (or a highly attenuated AC signal) is present at the input of the detection circuit, an interrupt signal is generated by the detection circuit and sent to the microcontroller (1370) due to the effective capacitance forming a good conductive path between the electrodes and the resulting AC signal attenuation.

[0221] The microcontroller receives an interrupt signal generated by the detection circuit and sends a signal to the digital-to-analog converter, instructing or causing the digital-to-analog converter to apply voltage to the sensor's electrodes (e.g., counter electrodes) (1380). Applying voltage to the sensor's electrodes causes the sensor to generate or produce a sensor signal (1390). The sensor signal measuring device 431 measures the generated sensor signal and transmits the sensor signal to the microcontroller. The microcontroller receives the sensor signal from the sensor signal measuring device connected to the working electrode and processes the sensor signal to extract measurements of the physiological characteristics of the subject or patient (1395).

[0222] Figure 13B Other methods for verifying sensor hydration according to embodiments of the present invention are illustrated. Figure 13BIn the illustrated embodiment of the invention, the sensor is physically connected to a sensing electronics device (1310). In this embodiment, an AC signal (1341) is applied to an electrode (e.g., a reference electrode) in the sensor. Optionally, in this embodiment, a DC signal (1341) is applied to the electrode in the sensor. If an AC signal is applied, an impedance measuring element measures the impedance at a point in the sensor (1351). Optionally, if a DC signal is applied, a resistance measuring element measures the resistance at a point in the sensor (1351). If the resistance or impedance is below a resistance threshold or impedance threshold (or other set standard), the impedance (or resistance) measuring element sends a signal (or causes a signal to be transmitted) to a detection circuit (1361), and the detection circuit sends an interrupt signal to the microcontroller, recognizing that the sensor is hydrated. Reference numerals 1380, 1390, and 1395 are shown in the figures. Figure 13A and Figure 13B They are the same because they represent the same operation.

[0223] The microcontroller receives an interrupt signal and sends a signal to a digital-to-analog converter to apply voltage to the sensor (1380). In an alternative embodiment of the invention, as discussed above, the digital-to-analog converter may apply current to the sensor. The sensor, such as a working electrode, generates a sensor signal (1390) representing a patient's physiological parameters. The microcontroller receives a sensor signal (1395) from a sensor signal measuring device that measures the sensor signal at an electrode (e.g., the working electrode) in the sensor. The microcontroller processes the sensor signal to extract measurements of the patient's or subject's physiological characteristics, such as the patient's blood glucose level.

[0224] Figure 14A and Figure 14B An example is provided illustrating a combined method for sensor hydration and sensor stabilization according to an embodiment of the present invention. Figure 14AIn the illustrated embodiment of the invention, the sensor is connected to a sensing electronics device (1405). An AC signal is applied to the sensor electrodes (1410). A detection circuit determines the AC signal level present at the input of the detection circuit (1420). If the detection circuit determines that a low-level AC signal is present at the input (meaning the AC signal is attenuating at a high level), then an interrupt signal is sent to the microcontroller (1430). Once the interrupt signal is sent to the microcontroller, the microcontroller knows to start or initiate a stabilization sequence (1440), i.e., applying multiple voltage pulses to the sensor electrodes as described above. For example, the microcontroller may cause a digital-to-analog converter to apply three voltage pulses (with an amplitude of +0.535 volts) to the sensor, each of which is followed by a duration of three voltage pulses (with an amplitude of 1.07 volts to be applied). This may be referred to as a stabilization sequence of the transmitted voltage. The microcontroller may induce the above-mentioned voltage stabilization sequence by executing a software program in a read-only memory (ROM) or random access memory. After the stabilization sequence is completed, the sensor may generate a sensor signal, which is measured and transmitted to the microcontroller (1450).

[0225] In an embodiment of the invention, the detection circuit can determine that a high-level AC signal has been continuously present at the input of the detection circuit (e.g., the input of a comparator), even after a hydration time threshold has elapsed (1432). For example, the hydration time threshold could be 10 minutes. After 10 minutes have elapsed, the detection circuit can still detect the presence of a high-level AC signal. At this point, the detection circuit can send a hydration assist signal to the microcontroller (1434). If the microcontroller receives the hydration assist signal, it can send a signal to cause the DAC to apply a voltage pulse or a series of voltage pulses to assist in sensor hydration (1436). In an embodiment of the invention, the microcontroller can send a signal to cause the DAC to apply a portion of a stabilization sequence or other voltage pulses to assist in sensor hydration. In this embodiment of the invention, the application of the voltage pulse can generate a low-level AC signal (or a highly attenuated signal) that is detected in the detection circuit (1438). At this point, the detection circuit can send an interrupt signal, as disclosed in step 1430, and the microcontroller can begin stabilization.

[0226] Figure 14BA second embodiment combining the hydration and stabilization methods is illustrated, wherein feedback information is used during the stabilization process. The sensor is connected to sensing electronics (1405). An AC signal (or DC signal) is applied to the sensor (1411). In this embodiment, the AC signal (or DC signal) is applied to the sensor's electrodes, such as a reference electrode. An impedance measuring device (or resistance measuring device) measures the impedance (or resistance) within a specified area of ​​the sensor (1416). In this embodiment, the impedance (or resistance) between the reference electrode and the working electrode can be measured. The measured impedance (or resistance) can be compared to a certain impedance or resistance value (1421) to observe whether the impedance (or resistance) within the sensor is sufficiently low, indicating that the sensor is hydrated. If the impedance (or resistance) is lower than the impedance value (resistance value) or other set standard (which may be a threshold), an interrupt signal is transmitted to the microcontroller (1431). Upon receiving the interrupt signal, the microcontroller sends a signal to the DAC, instructing the DAC to apply a stabilization sequence of voltage (or current) to the sensor (1440). After a stabilizing sequence is applied to the sensor, a sensor signal is generated in the sensor (e.g., at the working electrode). This sensor signal is measured by a sensor signal measuring device, transmitted through the sensor signal measuring device, and received by the microcontroller (1450). Because the sensor is hydrated and a voltage stabilizing sequence has been applied to the sensor, the sensor signal is a precisely measured physiological parameter (i.e., blood glucose).

[0227] Figure 14CA third embodiment of the invention combining the stabilization and hydration methods is illustrated. In this embodiment, the sensor is connected to a sensing electronics device (1500). After the sensor is physically connected to the sensing electronics device, an AC signal (or DC signal) is applied to the electrodes of the sensor (e.g., a reference electrode) (1510). Simultaneously or substantially simultaneously, the microcontroller sends a signal to cause the DAC to apply a stabilizing voltage sequence to the sensor (1520). In an alternative embodiment of the invention, instead of a stabilizing voltage sequence, a stabilizing current sequence may be applied to the sensor. A detection circuit determines the level of the AC signal (or DC signal) present at the input of the detection circuit (1530). If a low-level AC signal (or DC signal) representing a highly attenuated AC signal (or DC signal) is present at the input of the detection circuit, an interrupt signal is sent to the microcontroller (1540). Because the microcontroller has initiated the stabilization sequence, the microcontroller receives the interrupt signal and sets a first indicator indicating that the sensor is fully hydrated (1550). After the stabilization sequence ends, the microcontroller sets a second indicator indicating that the stabilization sequence is complete (1555). The application of a stabilizing voltage sequence causes a sensor (e.g., a working electrode) to generate a sensor signal (1560), which is measured by a sensor signal measurement circuit and sent to a microcontroller. If a second indicator indicating completion of the stabilizing sequence is set and a first indicator indicating completion of hydration is set, the microcontroller can use the sensor signal (1570). If one or both of these indicators are not set, the microcontroller cannot use the sensor signal because the sensor signal cannot represent a precise measurement of the patient's physiological values.

[0228] In general, the hydration and stabilization processes described above can be used as part of a larger continuous glucose monitoring (CGM) approach. The current state of continuous glucose monitoring in the art is primarily an adjunct, meaning that readings provided by CGM devices (including, for example, implantable or subcutaneous sensors) cannot be used to make clinical decisions without a reference value. The reference value must be obtained via finger puncture, for example, using a BG meter. Because the available information from the sensor / detection element is limited, a reference value is necessary. Specifically, currently only a few pieces of information provided by the detection element for processing are the raw sensor value (i.e., sensor current or Isig) and the reverse voltage, which is the voltage between the counter electrode and the reference electrode (see, for example...). Figure 5 Therefore, during analysis, if the original sensor signal appears abnormal (e.g., if the signal is reduced), the only way for someone skilled in the art to distinguish between sensor malfunction and physiological changes in the user / patient's body (i.e., changes in blood glucose levels) is by obtaining a reference glucose value through a finger prick. As is known in the art, finger prick reference values ​​are also used to calibrate the sensor.

[0229] The embodiments of the invention described herein relate to advances and improvements in continuous glucose detection, resulting in more independent systems, related devices, and methods in which the need for finger-puncture reference values ​​can be minimized or eliminated, and thereby clinical decisions can be made based solely on information from sensor signals with a high degree of reliability. From a sensor design perspective, according to embodiments of the invention, this independence can be achieved through electrode redundancy, sensor diagnostics, and Isig and / or sensor glucose (SG) fusion.

[0230] As will be further explored below, redundancy can be achieved by using multiple working electrodes (e.g., in addition to the counter electrode and the reference electrode) to generate multiple signals representing the patient's blood glucose (BG) level. These multiple signals can then be used to assess the relative health of the electrodes (working electrodes), the overall reliability of the sensor, and the frequency at which the reference value needs to be calibrated (if necessary).

[0231] Sensor diagnostics involves using additional (diagnostic) information that provides real-time observation of the sensor's health status. In this regard, electrochemical impedance spectroscopy (EIS) has been found to provide this additional information in the form of sensor impedance and impedance-related parameters at different frequencies. Moreover, advantageously, it has been further discovered that, for some frequency ranges, the impedance and / or impedance-related data are substantially independent of glucose. This glucose independence enables the use of various EIS-based labels or markers not only to produce robust, highly reliable sensor glucose values ​​(through fusion methods) but also to assess the condition, health status, age, and effectiveness of individual electrodes and the entire sensor substantially independently of glucose-dependent ISIg methods.

[0232] For example, analysis of glucose-independent impedance data provides information about how quickly the sensor hydrates and how quickly it becomes ready for data acquisition, such as using 1 kHz real impedance values, 1 kHz virtual impedance values, and Nyquist slope values ​​(described in detail below). Furthermore, glucose-independent impedance data provides information about potential occlusion (e.g., using 1 kHz real impedance values) that may exist on the sensor membrane surface, temporarily blocking glucose access to the sensor and thus causing signal degradation. Additionally, glucose-independent impedance data, using phase angle and / or virtual impedance values ​​at, for example, 1 kHz and higher frequencies, provides information about sensor sensitivity loss during extended wear, potentially due to localized hypoxia at the insertion site.

[0233] In the description of electrode redundancy and EIS in this paper, the fusion algorithm can be used to obtain diagnostic information about each redundant electrode provided by EIS and to independently assess the reliability of each electrode. Weights as a measure of reliability can then be added to each individual signal, and a single fused signal can be calculated to generate the sensor glucose value observed by the patient / treatment recipient.

[0234] As can be seen from the above, the combined use of redundancy, EIS-based sensor diagnostics, and EIS-based fusion algorithms makes the entire CGM system more reliable than currently available CGM systems. Redundancy has advantages in at least two aspects. First, redundancy eliminates the risk of single-point failure by providing multiple signals. Second, providing multiple electrodes (working electrodes) where a single electrode may be sufficient allows the output of redundant electrodes to be used as a check on the main electrode, thereby reducing and potentially eliminating the need for frequent calibration. Furthermore, EIS diagnostics independently and thoroughly examines the health status of each electrode without requiring a reference glucose value (finger puncture), thus reducing the number of reference values ​​required. However, the use of EIS technology and EIS diagnostics is not limited to redundant systems (i.e., those with more than one working electrode). In particular, as discussed below in conjunction with embodiments of the invention, EIS can be advantageously used in conjunction with single-electrode and / or multi-electrode sensors.

[0235] The EIS or AC impedance method was used to study systems that respond to the application of periodic small-amplitude AC signals. This is in Figure 15A The diagram illustrates, where E is the applied potential, I is the current, and the impedance (Z) is defined as ΔE / ΔI. However, while impedance itself can be simply defined mathematically as ΔE / ΔI, commercial success has yet to be achieved in applying EIS technology to continuous glucose monitoring. This is partly because glucose sensors are highly complex systems, and, to date, no mathematical model has been developed in the art that fully explains the complexity of the EIS output of a glucose sensor.

[0236] Figure 15B A simplified circuit model for describing electrochemical impedance spectroscopy is shown. In this example, IHP represents the inner Helmholtz surface, OHP represents the outer Helmholtz surface, CE is the counter electrode, WE is the working electrode, and C... d It is a double-layer capacitor, R p It is a polarization resistor, Z w It is the Warburg impedance, and R s It is the solution resistance. The last four components—double-layer capacitance (C) d ), Warburg impedance (Z) w ), polarization resistance (R) p) and solution resistance (R) s Each of these components can play a significant role in sensor performance and can be measured individually by applying a low-frequency or high-frequency alternating operating potential. For example, the Warburg impedance is closely related to the diffusion impedance of the electrochemical system—which is primarily a low-frequency impedance—and thus, the Warburg impedance is present in all diffusion-limited electrochemical sensors. Therefore, by associating one or more of these components with one or more components and / or layers of a glucose sensor, those skilled in the art can use EIS technology as a sensor diagnostic tool.

[0237] As is known in the art, impedance can be defined in terms of magnitude and phase, where the magnitude (|Z|) is the ratio of the voltage difference amplitude to the current amplitude, and the phase (θ) is the phase shift of the current exceeding the voltage. When a circuit is driven solely by direct current (DC), impedance is the same as resistance; that is, resistance is a special case of impedance under zero-phase-angle conditions. However, as a complex number, impedance can also be expressed by its real and imaginary parts. In this respect, real impedance and imaginary impedance can be obtained from impedance magnitude and phase using the following equations:

[0238] Real Impedance(ω)=Magnitude(ω)x cos(Phase(ω) / 180xπ)

[0239] Imaginary Impedance(ω)=Magnitude(ω)x sin(Phase(ω) / 180xπ)

[0240] Wherein, ω represents the input frequency at which amplitude (ohms) and phase (degrees) are measured. The following section will examine in detail sensing electronics including application-specific integrated circuits (ASICs)—on the one hand, the relationship between impedance and current, and on the other hand, the relationship between impedance and voltage—including how the former is calculated based on measurements of the latter, the ASICs being developed for embodiments of the present invention.

[0241] continue Figure 15B The circuit model shown can be simplified to the following:

[0242]

[0243] Among them, Z w (ω) is the Warburg impedance, ω is the angular velocity, j is the imaginary unit (used instead of the traditional "i" to avoid confusion with current), and C d Rp and R s These are the double-layer capacitance, polarization resistance, and solution resistance (as defined above). The Warburg impedance can be calculated as follows:

[0244]

[0245]

[0246]

[0247] Where D is the diffusivity, L is the sensor film thickness, C is the peroxide concentration, and m: 1 / 2 corresponds to the 45° Nyquist slope.

[0248] The Nyquist curve is a graphical representation in which the real part (Real Z) of the impedance is plotted relative to its imaginary part (Img Z) over the entire frequency spectrum. Figure 16A This shows a general example of a Nyquist curve, where the X value is the real part of the impedance and the Y value is the imaginary part of the impedance. The phase angle is the angle between the impedance point (X,Y)—defined as a vector with amplitude |Z|—and the X-axis.

[0249] Figure 16A The Nyquist curve is generated by applying an AC voltage plus a DC voltage (DC bias) between the working electrode and the counter electrode at a selected frequency from 0.1 Hz to 1000 MHz (i.e., a frequency sweep). Starting from the right, the frequency increases from 0.1 Hz. The real and imaginary impedances can be calculated and plotted for each frequency condition. As shown, a typical Nyquist curve for an electrochemical system might look like a semicircle connected to a straight line at the inflection point, where the semicircle and the straight line represent the plotted impedance. In some implementations, the impedance at the inflection point is very important because it is the easiest to identify in the Nyquist curve and the intercept can be defined. Typically, the inflection point is close to the X-axis, and the X-value at the inflection point is close to the sum of the polarization resistance and the solution resistance (R0). p +R s ).

[0250] refer to Figure 16BThe Nyquist curve is typically described using a lower frequency region 1610 and a higher frequency region 1620, where the labels “higher frequency” and “lower frequency” are used in a relative sense and are not intended to limit the invention. Thus, for example, the lower frequency region 1610 may exemplarily include data points obtained in a frequency range of approximately 0.1 Hz to approximately 100 Hz (or higher), and the higher frequency region 1620 may exemplarily include data points obtained in a frequency range of approximately 1 kHz (or lower) to approximately 8 kHz (and higher). In the lower frequency region 1610, the Nyquist slope represents the slope of a linear fit 1630 of the lower frequency data points in the Nyquist curve. As shown, in the higher frequency region 1620, the virtual impedance value is minimal and negligible. Therefore, the intercept 1600 is essentially the value of the real impedance at higher frequencies (e.g., in this case, approximately in the range of 1 kHz to 8 kHz). Figure 16B In the middle, the intercept of 1600 is approximately 25 k ohms.

[0251] Figure 16C and Figure 16D This illustrates how the glucose sensor responds to a sinusoidal (i.e., alternating) operating potential. In these diagrams, GLM is the glucose restriction membrane of the sensor. AP is the adhesion promoter, HSA is human serum albumin, GOX is glucose oxidase (layer), and E... dc It is a DC potential, E ac It is the AC potential, and C' 过氧化物 This refers to the concentration of peroxides during AC application. For example... Figure 16C As shown, if the sensor diffusion length is smaller than the membrane (GOX) length, the system provides a relatively linear response (i.e., infinite) with a constant phase angle, where the sensor diffusion length is a function of the AC potential frequency, molecular diffusivity, and membrane thickness. Conversely, if the diffusion length is equal to the membrane (GOX) length, the system response becomes finite, producing a response such as... Figure 16D The semicircular Nyquist curve is shown. The latter is generally applicable to low-frequency EIS, where non-Faraday processes are negligible.

[0252] During EIS analysis, various AC voltages and DC biases of different frequencies can be applied, for example, between the working electrode and the reference electrode. In this respect, EIS is an improvement over previous methods, which may have been limited to applying a simple DC current or a single-frequency AC voltage. Although, in general, EIS can be performed at frequencies ranging from μHz to MHz, in embodiments of the present invention, a narrower frequency range (e.g., from about 0.1Hz to about 8kHz) is sufficient to perform EIS. Therefore, in embodiments of the present invention, an AC potential in the frequency range of about 0.1Hz to about 8kHz can be applied, with a programmable amplitude of at least 100mV, and preferably about 50mV.

[0253] Within the aforementioned frequency range, relatively higher frequencies—that is, those typically in the range of approximately 1 kHz to approximately 8 kHz—are used to examine the capacitive properties of the sensor in detail. Based on the film thickness and permeability, the typical impedance range at these relatively higher frequencies can be, for example, approximately 500 ohms to 25 k ohms, and the typical phase range can be, for example, 0 degrees to -40 degrees. On the other hand, relatively lower frequencies—that is, those typically in the range of approximately 0.1 Hz to approximately 100 Hz—are used to examine the resistive properties of the sensor in detail. Therefore, based on the electrode design and degree of metallization, the typical functionalized range of the output real impedance can be, for example, approximately 50 k ohms to 300 k ohms, and the typical phase range can be approximately -50 degrees to approximately -90 degrees. The above exemplary ranges are, for example... Figure 16E and Figure 16F It is shown in the Bode plots.

[0254] As stated above, the phrases "higher frequency" and "lower frequency" are intended to be used relative to each other, not in an absolute sense, and "higher frequency" and "lower frequency," along with the typical impedance and phase ranges described above, are exemplary and not limiting of the invention. Nevertheless, the basic principle remains the same: the capacitive and resistive behavior of a sensor can be examined in detail by analyzing impedance data across the entire frequency spectrum, where lower frequencies typically provide information about components with higher resistance (e.g., electrodes), while higher frequencies provide information about capacitive components (e.g., membranes). However, the actual frequency range in each case depends on the overall design, including, for example, the type of electrode, the surface area of ​​the electrode, the membrane thickness, the membrane permeability, etc. See also Figure 15B Regarding the general correspondence between high-frequency circuit elements and sensor films, and the general correspondence between low-frequency circuit elements and Faraday processes (including, for example, electrodes).

[0255] EIS can be used in sensor systems where the sensor includes a single working electrode, as well as in sensor systems where the sensor includes multiple (redundant) working electrodes. In one implementation, EIS provides useful information about the sensor's age (or aging). Specifically, the amplitude and phase angle of the impedance change under different frequency conditions. Figure 17 As shown, the sensor impedance—specifically, the sum of Rp and Rs—reflects the sensor's age and operating condition. Therefore, from... Figure 17 The different curves show that new sensors typically have higher impedance than used sensors. Therefore, considering the sum of Rp and Rs (X value), a threshold can be used to determine when the sensor's age exceeds its specified operating life. It should be noted that although... Figure 17 Figure 21 illustrates an exemplary embodiment and is discussed below. However, the real impedance value at the inflection point (i.e., Rp+Rs) is used to determine the aging, condition, stability, and hydration of the sensor. In addition to the real impedance, alternative implementations may use other EIS-based parameters, such as virtual impedance, phase angle, Nyquist slope, etc. Alternatively, alternative implementations may use other EIS-based parameters, such as virtual impedance, phase angle, Nyquist slope, etc., instead of the real impedance.

[0256] Figure 17 An example of a Nyquist curve relative to sensor lifetime is illustrated. The points indicated by the arrows are the inflection points for each sweep within the spectral range. For example, before initialization (time t = 0), Rs+Rp is above 8.5 k ohms, and after initialization (time t = 0.5 hours), the value of Rs+Rp drops below 8 k ohms. Over the next six days, Rs+Rp continues to decrease, thus reaching below 6.5 k ohms at the end of the specified sensor lifetime. Based on these examples, a threshold can be set to specify when the Rs+Rp value represents the end of the specified sensor operating lifetime. Therefore, EIS technology eliminates the possibility of reusing the sensor after its specified operating time. In other words, if a patient attempts to reuse the sensor after it has reached its specified operating time by disassembling and reconnecting it, EIS will measure an abnormally low impedance, causing the system to reject the sensor and prompt the patient to use a new one.

[0257] Furthermore, EIS can detect sensor malfunctions by detecting when the sensor impedance drops below a low impedance threshold level, indicating that the sensor may have been overused and is no longer functioning properly. The system can then stop the sensor before its specified operating life. As discussed in more detail below, sensor impedance can also be used to detect other sensor malfunctions (modes). For example, when a sensor enters a low-current state for any reason (i.e., a sensor malfunction), the sensor impedance may also increase above a certain high impedance threshold. If the impedance becomes abnormally high during sensor operation due to factors such as protein or peptide contamination, macrophage attachment, or any other factor, the system can also stop the sensor before its specified operating life.

[0258] Figure 18 Examples illustrate how EIS technology can be applied in the sensor stabilization process and in detecting sensor age according to embodiments of the present invention. Figure 18 The logic begins after the hydration process, and the aforementioned sensor initialization process has been completed (1800). In other words, the sensor is considered sufficiently hydrated, and the first initialization process has been used to initialize the sensor. Preferably, the initialization process can be in the form of a voltage pulse as described in detail above. However, in alternative embodiments, different waveforms can be used for the initialization process. For example, a sine wave can be used instead of a pulse to accelerate the wetting or conditioning of the sensor. Furthermore, some portions of the waveform may need to be greater than the sensor's normal operating voltage, i.e., 0.535 volts.

[0259] In block 1810, an EIS procedure is applied and the impedance is compared to a first high threshold and a first low threshold. Examples of the first high threshold and the first low threshold could be 7 k ohms and 8.5 k ohms, respectively, although these values ​​can be set higher or lower as needed. If the impedance, for example, Rp + Rs, is higher than the first high threshold, then in block 1820, the sensor undergoes an additional initialization process (e.g., applying one or more additional pulses). Ideally, the total number of initialization processes used to initialize the sensor can be optimized to limit the impact on sensor battery life and the total time required to stabilize the sensor. Therefore, by using EIS, fewer initialization processes can be performed initially, and the number of initializations can be gradually increased to obtain just the right number of initializations to prepare the sensor for use. Similarly, in an alternative implementation, EIS can be used during the hydration process to minimize the number of initializations required for the auxiliary hydration process shown in Figures 13 and 14.

[0260] On the other hand, if the impedance, for example, Rp+Rs, is below a first low threshold, then at block 1860, the sensor can be identified as faulty and immediately stopped. A message will be sent to the user to replace the sensor and restart the hydration process. If the impedance is within the high and low threshold range, then at block 1830, the sensor can begin normal operation. Subsequently, the logic proceeds to block 1840, where an additional EIS is performed to check the sensor's age. Upon first reaching block 1840, the microcontroller performs an EIS to assess the sensor's age, thus preventing the user from plugging and unplugging the same sensor. During subsequent repetitions of the EIS procedure, when the logic returns to block 1840, the microcontroller can perform an EIS at fixed intervals throughout the sensor's specified lifespan. In a preferred embodiment, the fixed interval is set to every two hours; however, longer or shorter time periods can also be used.

[0261] In block 1850, the impedance is compared to a second set of high and low thresholds. Examples of the second high and low thresholds can be 5.5k ohms and 8.5k ohms, respectively, although these values ​​can be set higher or lower as needed. As long as the impedance value is within the range of the second high and low thresholds, the logic proceeds to block 1830, whereby the sensor operates normally until the specified sensor lifespan is reached, for example, five days. Of course, as described with reference to block 1840, EIS can be performed at normally set intervals throughout the specified sensor lifespan. However, if, after performing EIS, in block 1850, the impedance is determined to have fallen below the second lower threshold or risen above the second higher threshold, then in block 1860, the sensor is stopped. In a further optional implementation, a secondary check can be performed when a sensor failure reading occurs. For example, if the EIS shows that the impedance exceeds the range of the second high and low thresholds, then, before determining sensor termination (block 1860), the logic can perform a second EIS to confirm that it does not meet the second set of thresholds (and confirm that the first EIS was performed correctly).

[0262] Figure 19 Based on the above detailed description, a possible schedule for performing diagnostic EIS procedures according to a preferred embodiment of the invention is described in detail. Each diagnostic EIS procedure is optional, and it is possible to have no diagnostic EIS procedures or any combination of one or more diagnostic EIS procedures. Figure 19The schedule begins with sensor insertion at time 1900. After insertion, the sensor undergoes a hydration period of 1910. This hydration period is crucial because, as mentioned above, an insufficiently hydrated sensor may provide inaccurate readings to the user. During the hydration period 1910, at time 1920, a first optional diagnostic EIS procedure is performed to ensure adequate hydration of the sensor. The first diagnostic EIS procedure 1920 measures the sensor impedance to determine if the sensor is adequately hydrated. If the first diagnostic EIS procedure 1920 determines that the impedance is within the set high and low threshold ranges, indicating adequate hydration, the sensor controller may allow the sensor to be powered on at time 1930. Conversely, if the first diagnostic EIS procedure 1920 determines that the impedance is outside the set high and low threshold ranges, indicating insufficient hydration, the sensor hydration period 1910 may be extended. After extended hydration, once a certain capacitance is reached between the sensor electrodes, signifying adequate hydration, power-on at time 1930 can occur.

[0263] The second optional diagnostic EIS procedure 1940 is arranged after the sensor is powered on at time 1930, but before the sensor initialization begins at time 1950. Arranged here, the second diagnostic EIS procedure 1940 can detect whether the sensor is to be reused before the initialization begins at time 1950. The test to determine whether the sensor is to be reused is performed in… Figure 18 It is described in detail. However, unlike before... Figure 18 As described, an aging test is performed after initialization is complete. Figure 19 In the test, the aging test was conducted before initialization. It is important to understand that... Figure 19 The timeline of the EIS procedure can be rearranged without affecting the overall teachings of this application, and the order of some steps can be interchanged. As previously explained, the second diagnostic EIS procedure 1940 detects a reused sensor by determining the sensor impedance value and subsequently comparing that value with set high and low thresholds. If the impedance is outside the set threshold range, this indicates that the sensor has been reused, and the sensor can then be rejected, prompting the user to replace it with a new sensor. This prevents complications that may arise from the reuse of old sensors. Conversely, if the impedance falls within the set threshold range, sensor initialization 1950 can begin, confirming that a new sensor is being used.

[0264] The third optional diagnostic EIS procedure 1960 can be scheduled after the initialization begins at time point 1950. This third diagnostic EIS procedure 1960 tests the sensor's impedance value to determine if the sensor is adequately initialized. The third diagnostic EIS procedure 1960 should be performed for the minimum amount of time required for any sensor to be adequately initialized. When the third diagnostic EIS procedure is performed at this time, sensor lifespan is maximized by limiting the time a adequately initialized sensor is not used, and over-initialization is avoided by ensuring adequate initialization before it occurs. Preventing over-initialization is crucial because it generates suppressed current, which can cause inaccurate readings. However, insufficient initialization is also problematic; therefore, if the third diagnostic EIS procedure 1960 indicates insufficient sensor initialization, optional initialization can be performed at time point 1970 to adequately initialize the sensor. Insufficient initialization is detrimental because excessive current results in readings unrelated to the actual glucose concentration. Due to the dangers of both insufficient and excessive initialization, the third diagnostic EIS procedure plays a vital role in ensuring proper sensor operation during use.

[0265] In addition, the optional periodic diagnostic EIS procedure 1980 can be scheduled according to time after the sensor is fully initialized. The EIS procedure 1980 can be scheduled at any set interval. As discussed in detail below, the EIS procedure 1980 can also be triggered by other sensor signals, such as abnormal current or abnormal counter electrode voltage. Furthermore, as needed, as few or as many EIS procedures can be scheduled. In a preferred embodiment, the EIS procedure used during hydration, the EIS procedure used during sensor life checks, the EIS procedure used during initialization, or the EIS procedure used during periodic diagnostic testing is the same procedure. In alternative embodiments, the EIS procedure can be shortened or lengthened (i.e., checking fewer or more frequency ranges) for different EIS procedures, depending on the need for a specific impedance range. The periodic diagnostic EIS procedure 1980 monitors impedance values ​​to ensure the sensor operates continuously at an optimal level.

[0266] If the sensor current decreases due to contaminants, sensor age, or a combination of both, the sensor may not be operating optimally. Sensors older than a certain length are no longer useful, but sensors impaired by contaminants may be repairable. Contaminants reduce the surface area of ​​the electrodes or the diffusion pathways of analytes and reaction byproducts, leading to a decrease in sensor current. These contaminants carry a charge and gradually accumulate on the electrodes or membrane surface under certain voltage conditions. Previously, contaminants would impair sensor effectiveness. Now, if the periodic diagnostic EIS procedure 1980 detects an impedance value indicating the presence of contaminants, remedial measures can be taken. (Reference) Figure 20 This describes when remedial measures should be taken. Therefore, regular diagnostic EIS procedures (1980) become crucial, as these procedures can trigger sensor remedial measures that restore sensor current to normal levels and extend sensor life. Two possible implementations of sensor remedial measures are described below. Figure 21A and Figure 21B As described in the text.

[0267] Furthermore, any scheduled diagnostic EIS procedure 1980 may be suspended or rescheduled when certain events are identified as imminent. These events may include any situation requiring the patient to check sensor readings, including, for example, when the patient uses a test strip to measure his or her BG level to calibrate the sensor, when the patient is warned of a calibration error and needs to measure his or her BG level again using the test strip, or when a warning of hyperglycemia or hypoglycemia has been issued but not received.

[0268] Figure 20 An example illustrates a method for combining diagnostic EIS procedures with sensor remediation measures according to an embodiment of the present invention. The diagnostic procedure in block 2000 may be... Figure 19 A detailed description of any periodic diagnostic EIS procedure 1980. The logic of this method begins as follows: In block 2000, a diagnostic EIS procedure is performed to detect the impedance value of the sensor. As described herein, in a particular embodiment, the EIS procedure uses a combination of DC bias and AC voltages of different frequencies, wherein the impedance detected by performing the EIS procedure is plotted on a Nyquist curve, and the inflection point of the Nyquist curve is approximately the sum of the polarization resistance and the solution resistance (i.e., the real impedance value). After the diagnostic EIS procedure in block 2000 detects the impedance value of the sensor, the logic moves to block 2010.

[0269] In box 2010, the impedance value is compared with the set high and low thresholds to determine if the impedance value is normal. If the impedance is within the set high and low threshold range in box 2010, then in box 2020, normal sensor operation resumes, and... Figure 20The logic will halt until another diagnostic EIS procedure is scheduled. Conversely, if the impedance is determined to be abnormal in block 2010 (i.e., exceeding the set high and low threshold ranges), then remedial action is triggered in block 2030. Examples of acceptable high and low thresholds during the sensor's lifetime are 5.5 k ohms and 8.5 k ohms, respectively, although these values ​​can be set higher or lower as needed.

[0270] The remedial measures in block 2030 are performed to remove any contaminants that could cause abnormal impedance values. In a preferred embodiment, the remedial measures are performed by applying a reverse current or reverse voltage between the working electrode and the reference electrode. A detailed description of the remedial measures will be provided with reference to FIG21. After the remedial measures are performed in block 2030, the impedance value is re-detected in block 2040 using a diagnostic EIS procedure. Subsequently, when the impedance value from the diagnostic EIS procedure in block 2040 is compared with a set high or low threshold, the success of the remedial measures is determined in block 2050. As in block 2010, if the impedance is within the set threshold range, the impedance is considered normal; if the impedance is outside the set threshold range, the impedance is considered abnormal.

[0271] If the sensor impedance value is determined to return to normal at box 2050, then normal sensor operation is performed at box 2020. If the impedance remains abnormal, indicating that sensor age is the cause of the abnormal impedance or that remedial measures to remove contaminants have been unsuccessful, then the sensor is subsequently stopped at box 2060. In an alternative implementation, instead of immediately stopping the sensor, the sensor may initially generate a sensor message requiring the user to wait and then perform further remedial measures after a set time period has elapsed. This optional step may be combined with separate logic to determine whether the impedance value is close to or within a high and low threshold after the initial remedial measures are performed. For example, if the sensor impedance value does not change, then the sensor may then decide to stop. However, if the sensor impedance value is close to a preset range but still outside the range after the initial remedial measures, then additional remedial measures may be performed. In yet another alternative implementation, the sensor may generate a message requiring the user to calibrate the sensor using a finger puncture device measurement to further confirm whether the sensor has truly failed. All of the above implementations are designed to prevent users from using a faulty sensor that produces inaccurate readings.

[0272] Figure 21AAn example of one implementation of the sensor remediation method described above is given. In this implementation, the blockage caused by contaminants is removed by reversing the voltage applied between the working electrode and the reference electrode of the sensor. The reversed DC voltage dissipates the charged contaminants on the electrode or membrane surface, clearing the diffusion path. Through the cleared path, the sensor current returns to normal levels, and the sensor can generate accurate readings. Therefore, the remediation method saves the user the time and cost associated with replacing another working sensor.

[0273] Figure 21B An example of an alternative implementation of the sensor remediation method described above is illustrated. In this implementation, a reversed DC voltage applied between the working electrode and the reference electrode is combined with an AC voltage. By adding the AC voltage, some substances tightly adsorbed onto the surface layer or on the surface layer can be removed because the AC voltage allows its driving force to extend further from the electrodes and penetrate all layers of the sensor. The AC voltage can appear in any number of different waveforms. Some examples of waveforms that can be used include square waves, triangular waves, sine waves, or pulses. As with the previous implementation, once the contaminant is removed, the sensor can return to normal operation, and both sensor lifespan and accuracy are improved.

[0274] While the above embodiments primarily illustrate the use of real impedance data in sensor diagnostics, embodiments of the present invention also relate to the use of other EIS-based and substantially analyte-independent parameters (besides real impedance) in sensor diagnostic procedures. For example, as described above, analysis of impedance data (essentially) independent of glucose (e.g., 1 kHz real impedance and 1 kHz virtual impedance, and the Nyquist slope) can provide information about how quickly the sensor hydrates and how quickly it is ready to acquire data, reflecting sensor efficiency. Furthermore, impedance data (essentially) independent of glucose, such as 1 kHz real impedance, provides information about potential blockages that may exist on the sensor membrane surface, which could temporarily block the pathway of glucose into the sensor and thus cause signal degradation.

[0275] Furthermore, impedance data (essentially) independent of glucose, such as higher frequency phase angle values ​​and / or virtual impedance values ​​at 1 kHz and higher frequencies, provide information about sensor sensitivity loss during extended wear, which may potentially be due to local hypoxia at the insertion site. In this regard, the basic mechanism of hypoxia-induced sensitivity loss can be described as follows: when local oxygen is lacking, the sensor output (i.e., Isig and SG) becomes oxygen-dependent rather than glucose-dependent, thus the sensor loses sensitivity to glucose. Other indicators, including the 0.1 Hz real impedance, counter electrode voltage (Vcntr), and EIS-induced Isig spike signal, can also be used to detect hypoxia-induced sensitivity loss. Moreover, in redundant sensor systems, the relative difference between the 1 kHz real impedance, 1 kHz virtual impedance, and 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biocontamination.

[0276] According to embodiments of the invention, EIS-based sensor diagnostics require consideration and analysis of EIS data relating to one or more of at least three main factors, namely, potential sensor failure modes: (1) signal start-up; (2) signal degradation; and (3) sensitivity loss. It is evident that this paper has found that most impedance-related parameters used for these diagnostic analyses and procedures can be studied under a specific frequency condition or over multiple frequency ranges, wherein these parameters are substantially independent of the analyte, allowing sensor-diagnostic procedures to be performed independently of analyte levels in the patient. Therefore, while EIS-based sensor diagnostics can be triggered, for example, by large fluctuations in the analyte-dependent isig, the impedance-related parameters used for these sensor diagnostic procedures are themselves substantially independent of analyte levels. As discussed in more detail below, it has been found that in most cases where glucose is observed to affect the amplitude (or other characteristics) of EIS-based parameters, this effect is typically small enough—for example, the difference between the EIS-based measurement and the glucose affecting it is at least an order of magnitude—that this effect can be filtered out from the measurement by, for example, software in an IC.

[0277] By definition, "start-up" refers to the integrity of the sensor signal during the first few hours after insertion (e.g., t = 0-6 hours). For example, in current devices, the sensor signal is considered unreliable during the first two hours after insertion; similarly, the sensor glucose value is masked to the patient / user. In cases where the sensor takes a longer time to hydrate, the sensor signal is lower in the hours following insertion. By using EIS, additional impedance information can be provided (by performing the EIS procedure) after sensor insertion. In this regard, the total impedance equation can be used to explain the principle behind low-start-up detection using 1kHz real impedance. At relatively high frequencies—1kHz and above—the virtual impedance is very small (as confirmed by in vivo data), thus the total impedance is summarized as:

[0278]

[0279] As the sensor gradually completes wetting, the double-layer capacitor (C) d The total impedance will decrease because, as shown in the equation above, the total impedance is related to C. d Inversely proportional. This is in, for example... Figure 16B The example shown illustrates the form of a 1600 intercept on the real impedance axis. Importantly, the 1kHz virtual impedance can also be used for the same purpose, as it also includes a capacitive component and is inversely proportional to it.

[0280] Another indicator for low-start detection is the Nyquist slope, which depends entirely on the impedance at a relatively low frequency, which in turn corresponds to the Warburg impedance component of the total impedance (see, for example...). Figure 15B ). Figure 22The Nyquist curve of a normally functioning sensor is shown, where arrow A represents the time progression, i.e., the time the sensor has been worn, starting from t=0. Therefore, an EIS at a relatively low frequency is performed immediately after sensor insertion (t=0), producing real and virtual impedance data plotted using a first linear fit 2200 with a first (Nyquist) slope. At time intervals after t=0, a second (lower frequency) sweep is run, producing a second linear fit 2210 with a second (Nyquist) slope larger than the first Nyquist slope, and so on. As the sensor becomes more hydrated, the Nyquist slope increases, and the intercept decreases, becoming steeper and closer to the Y-axis, as reflected in lines 2200, 2210, etc. Regarding low-start detection, clinical data show that the Nyquist slope typically increases sharply after sensor insertion and initialization, then stabilizes to a certain level. One explanation for this is that as the sensor gradually wets, the diffusion rate and concentration of the substance undergo a sharp change, which is reflected in the Warburg impedance.

[0281] exist Figure 23A In this embodiment, the Isig 2230 of the first working electrode WE1 starts below expectations (approximately 10 nA) and takes some time to catch up with the Isig 2240 of the second working electrode WE2. Therefore, in this particular embodiment, WE1 is designated as having a low start-up. EIS data reflects this low start-up in two ways. First, as... Figure 23A As shown, the real impedance of WE1 at 1kHz (2235) is much higher than the real impedance of WE2 at 1kHz (2245). Second, when compared with the Nyquist slope of WE2 ( Figure 23C When comparing, the Nyquist slope of WE1 ( Figure 23B The impedance starts low, has a large intercept of 2237, and takes longer to stabilize. As discussed later, these two indicators—1kHz real impedance and Nyquist slope—can be used as diagnostic inputs in the fusion algorithm to determine which of the two electrodes can carry a higher weight when calculating the fused signal. Furthermore, one or both of these indicators can be used in diagnostic procedures to determine whether the sensor is acceptable overall or whether it should be stopped and replaced.

[0282] By definition, a signal (or Isig) drop refers to a condition where the sensor signal is low, and this drop is largely temporary in nature, for example, lasting only a few hours. This low signal may be caused by, for example, some form of biological occlusion on the sensor surface, or by pressure applied at the insertion site (e.g., when sleeping on this side). During this period, the sensor data is considered unreliable; however, the signal does eventually recover. In EIS data, this type of signal drop—as opposed to a signal drop caused by changes in blood glucose levels within the patient—is reflected in 1 kHz real impedance data, such as... Figure 24 As shown.

[0283] Specifically, in Figure 24 In the data, the Isig 2250 of the first working electrode WE1 and the Isig 2260 of the second working electrode WE2 both start at approximately 25 nA at the far left (i.e., 6 pm). Over time, these two Isig values ​​fluctuate, reflecting fluctuations in glucose levels near the sensor. For approximately the first 12 hours (i.e., until approximately 6 am), both Isig values ​​are fairly stable, as are their respective 1 kHz real impedances of 2255 and 2265. However, between approximately 12 and 18 hours—i.e., from 6 am to noon—the Isig 2260 of WE2 begins to decrease and continues to decline over the next few hours until approximately 9 pm. During this period, the Isig 2250 of WE1 also shows some decrease, but it is much more stable and decreases less than the Isig 2260 of WE2. The changes in the Isig values ​​of WE1 and WE2 are also reflected in their respective 1 kHz real impedance data. Therefore, as... Figure 24 As shown, during the aforementioned time period, although the 1kHz real impedance (2255) of WE1 remained relatively stable, the 1kHz real impedance (2265) of WE2 increased significantly.

[0284] By definition, sensitivity loss refers to a decrease in the sensor signal (Isig) and a lack of response over an extended period of time, which is typically irrecoverable. Sensitivity loss can occur for a variety of reasons. For example, electrode poisoning significantly reduces the active surface area of ​​the working electrode, severely limiting the current amplitude. Sensitivity loss can also occur due to hypoxia or lack of oxygen at the insertion site. Furthermore, sensitivity loss can occur due to some form of extreme surface occlusion (i.e., a more prolonged form of signal decline caused by biological or other factors), which restricts the pathway of glucose and oxygen across the sensor membrane, thereby reducing the number / frequency of chemical reactions that generate current in the electrode and ultimately produce the sensor signal (Isig). It should be noted that the various causes of sensitivity loss described above apply to both short-term (7 to 10 days of wear) and long-term (6 months of wear) sensors.

[0285] In EIS data, the absolute values ​​of phase (|phase|) and virtual impedance (|virtual impedance|) typically increase over a relatively high frequency range (e.g., 128 Hz and above, and 1 kHz and above) before sensitivity loss. Figure 25A An example of a normally functioning glucose sensor is shown, in which the sensor current 2500 responds to glucose—that is, Isig 2500 follows glucose fluctuations—but all relevant impedance outputs, such as the 1kHz real impedance 2510, the 1kHz virtual impedance 2530, and the phase (2520) at frequencies of approximately 128Hz or greater, remain constant because they are essentially independent of glucose.

[0286] Specifically, Figure 25A The top figure shows that after the first few hours, the 1kHz real impedance 2510 remained fairly stable at approximately 5k ohms (and the 1kHz virtual impedance 2530 remained fairly stable at approximately -400 ohms). In other words, at 1kHz, the real impedance data 2510 and the virtual impedance data 2530 are essentially independent of glucose, thus, according to analysis, they can be used as indicators or independent indicators of sensor health and ultimately reliability. However, as mentioned earlier, different impedance-related parameters can show glucose independence at different frequency ranges, and in each case, this range can depend on the overall sensor design, such as electrode type, electrode surface area, membrane thickness, membrane permeability, etc.

[0287] Therefore, in the embodiments Figure 25B In a 90% short tubeless electrode design, the top figure again shows the sensor current 2501 responding to glucose, and after the first few hours, the 1kHz real impedance 2511 remains fairly stable at about 7.5k ohms. Figure 25B The bottom figure shows the real impedance data at frequencies between 0.1 Hz (2518) and 1 kHz (2511). As can be seen from the figure, the real impedance data at 0.1 Hz (2518) is largely dependent on glucose. However, as indicated by reference numerals 2516, 2514, and 2512, the real impedance becomes increasingly independent of glucose as the frequency increases from 0.1 Hz to 1 kHz (i.e., the impedance data measured closer to 1 kHz).

[0288] Back Figure 25AThe middle graph shows that phase 2520 at relatively high frequencies is largely independent of glucose. However, it should be noted that "relatively high frequencies" related to this sensor parameter (phase) during the analysis mean frequencies of 128 Hz and above. In this respect, the graph shows that the phase at all frequencies between 128 Hz and 8 kHz is stable throughout the entire time period shown. On the other hand, as... Figure 25C As shown in the bottom figure, while phase 2522 is stable at 128 Hz (and above), phase 2524 fluctuates at frequencies gradually lower than 128 Hz—that is, the phase becomes increasingly dependent on glucose and the phase degree changes. It should be noted that... Figure 25C Electrode design of the embodiment and Figure 25B The same electrode design is used, and Figure 25C The top image and Figure 25B top Figure 1 To.

[0289] Figure 26 An example of sensitivity loss due to hypoxia at the insertion site is shown. In this case, the insertion site showed sensitivity loss on the fourth day (due to...). Figure 26 (Indicated by the black vertical line in the image) Oxygen deprivation occurred immediately afterward, causing the sensor current 2600 to drop and no response to occur. The 1kHz real impedance 2610 remained stable, indicating no physical blockage at the sensor. However, as shown by the various downward arrows, the changes in the relatively high frequency phase 2622 and the 1kHz virtual impedance 2632 are consistent with the sensitivity loss, indicating that this type of loss is due to oxygen deprivation at the insertion site. Specifically, Figure 26 The diagram shows that at higher frequencies (2620) and at 1 kHz, the phase and virtual impedance (2630) become more negative—represented by the black vertical line—before the sensor loses sensitivity—and continue to trend downwards as the sensor sensitivity continues to decrease. Therefore, as described above, the absolute values ​​of the phase (|phase|) and virtual impedance (|virtual impedance|) increase in a relatively high frequency range (e.g., 128 Hz and above, and 1 kHz and above) before this sensitivity loss occurs; in other words, this sensitivity loss is predicted by the increase in the absolute values ​​of the phase (|phase|) and virtual impedance (|virtual impedance|) in a relatively high frequency range (e.g., 128 Hz and above, and 1 kHz and above).

[0290] The above characteristics can be confirmed through in vitro testing. Figure 27 An example of in vitro testing is shown. Figure 27Results of in vitro sensor testing are shown, simulating hypoxia under different glucose concentrations. In the top plot, Isig fluctuates with glucose concentrations increasing from 100 mg / dL (2710) to 200 mg / dL (2720), 300 mg / dL (2730), 400 mg / dL (2740) and then decreasing back to 200 mg / dL (2750). In the bottom plot, the phase is generally stable at relatively high frequencies, indicating that the phase is independent of glucose. However, under very low oxygen concentrations, e.g., 0.1% O2, as indicated by the circled areas and arrows 2760 and 2770, the phase fluctuates at relatively high frequencies. It should be noted that the amplitude and / or direction (i.e., positive or negative) of the fluctuations depend on various factors. For example, the higher the glucose-to-oxygen ratio, the greater the amplitude of the phase fluctuations. Furthermore, the specific sensor design and sensor age (i.e., measured over time after implantation) affect these fluctuations. Therefore, for example, the older the sensor, the more prone it is to fluctuations.

[0291] Figures 28A to 28D This illustrates another example of sensitivity loss due to hypoxia when using redundant working electrodes WE1 and WE2. Figure 28A As shown, even when the sensor current 2800 fluctuates and eventually becomes unresponsive, the 1kHz real impedance 2810 remains stable. Moreover, as previously mentioned, the change in the 1kHz virtual impedance 2820 corresponds to the sensor sensitivity loss. However, furthermore, Figure 28B The data shows the real impedance and virtual impedance at 0.105 Hz (denoted as 2830 and 2840, respectively). The latter, more commonly referred to as the "0.1 Hz data," shows that although the virtual impedance appears fairly stable at 0.1 Hz, the real impedance 2830 at 0.1 Hz increases significantly when the sensor loses sensitivity. Furthermore, as... Figure 28C As shown, under conditions where sensitivity is lost due to hypoxia, V cntr 2850 reaches the voltage rail of 1.2 volts.

[0292] In short, the charts illustrate the following findings: Sensitivity loss due to hypoxia is accompanied by a lower 1kHz virtual impedance (i.e., the latter becomes more negative), a higher 0.105Hz real impedance (i.e., the latter becomes more positive), and V... cntr Furthermore, the hypoxia process and the Vcntr-rail (counter electrode voltage rail: V) cntr -rail) is generally associated with an increase in the capacitive component in an electrochemical circuit. It should be noted that in some diagnostic procedures described later, the 0.105 Hz real impedance may not be used, as this relatively low-frequency real impedance data may appear to be analyte-dependent.

[0293] Finally, regarding Figures 28A to 28D In some instances, it should be noted that impedance measurements at 1 kHz or higher frequencies typically cause EIS-induced spikes in the Isig signal. This is in Figure 28D The diagram shows the original Isig of WE2 plotted relative to time. The significant increase in Isig at the start of the spike signal is a non-Radaic process due to the charge of the double-layer capacitor. Therefore, in addition to the lower 1kHz virtual impedance, higher 0.105Hz real impedance, and V discussed above, cntr Outside of orbit, the loss of sensitivity caused by hypoxia can also be accompanied by a high EIS-induced spike signal.

[0294] Figure 29 This example illustrates another instance of sensitivity loss. This instance can be considered as... Figure 24 Regarding the extreme form of the aforementioned decrease in Isig. Here, the sensor current 2910 was observed to be low from the start of insertion, indicating a problem causing electrode blockage during insertion. Compared to Figure 25A For the same parameter values ​​of a normally functioning sensor, the 1kHz real impedance 2920 is significantly higher, while the relatively higher frequency phase 2930 and 1kHz virtual impedance 2940 both become more negative. The changes in the relatively higher frequency phase 2930 and 1kHz virtual impedance 2940 suggest that the sensitivity loss may be due to oxygen deficiency, which in turn may be due to occlusion of the sensor surface.

[0295] Figures 30A to 30D Data from another redundant sensor is shown, where the relative differences between the 1 kHz real impedance and 1 kHz virtual impedance, and the 0.1 Hz real impedance between two or more working electrodes can be used to detect sensitivity loss due to biocontamination. In this example, WE1 exhibits more sensitivity loss than WE2, which is evident from WE2's higher 1 kHz real impedance 3010, WE2's lower 1 kHz virtual impedance 3020, and WE2's much higher real impedance at 0.105 Hz (3030). However, furthermore, in this example, V cntr The 3050 did not reach the voltage rail. Furthermore, as... Figure 30D As shown, the height of the spike signal in the original Isig data did not change significantly over time. This indicates that, for sensitivity loss caused by biological contamination, V cntr The increase in the height of the spike signal is correlated with the increase in the height of the spike signal. Furthermore, the fact that the height of the spike signal in the original Isig data did not change significantly over time indicates that the capacitive component of the circuit did not change significantly over time. Thus, the sensitivity loss caused by biocontamination is related to the resistive component of the circuit (i.e., diffusion).

[0296] The aforementioned impedance-related parameters can be used individually or in combination, and input to: (1) an EIS-based sensor diagnostic program; and / or (2) a fusion algorithm for generating more reliable sensor glucose values. For the former, Figure 31 Examples illustrate how EIS-based data—that is, impedance-related parameters or characteristics—can be used in diagnostic procedures to determine in real time whether a sensor is functioning correctly or whether it should be replaced.

[0297] Figure 31 The diagnostic procedure shown in the flowchart is based on the periodic acquisition of EIS data, for example, hourly, every half hour, every ten minutes, or at any other interval—including continuous acquisition—which may be adapted to a specific sensor depending on the analysis. Within each such interval, EIS may operate across the entire spectrum (i.e., “full sweep”), or within a selected frequency range, or even at a single frequency. Thus, for example, for a scheme involving hourly data acquisition, as described above, EIS may be performed in a frequency range from μHz to MHz, or in a narrower frequency range such as from about 0.1Hz to about 8kHz. In embodiments of the invention, EIS data acquisition may alternate between full sweep and narrower frequency ranges, or may be performed according to other schemes.

[0298] The timing and frequency of EIS implementation and data acquisition can be influenced by various factors. For example, each EIS implementation consumes a certain amount of power, typically supplied by the sensor's battery, which powers the sensor electronics, including the ASIC described later. Thus, battery capacity and remaining sensor lifespan help determine the number of EIS runs and the frequency width of sampling per run. Furthermore, embodiments of the invention envision scenarios where monitoring of EIS parameters (e.g., real impedance at 1 kHz) at a specific frequency may be required based on a first schedule (e.g., every few seconds or minutes), while the same parameters may be monitored at other frequencies based on a second schedule (e.g., at lower frequencies). In these scenarios, the diagnostic procedure can be tailored to the specific sensor and requirements, thus conserving battery power and avoiding unnecessary and / or redundant EIS data acquisition.

[0299] It should be noted that, in embodiments of the present invention, diagnostic procedures, such as Figure 31The diagnostic procedure shown involves performing a series of individual "tests" to monitor the sensor in real time. Multiple tests or flags—also known as "multiple flags"—are used because data on multiple impedance-based parameters or characteristics can be collected each time the EIS is run (i.e., each time an EIS procedure is executed). This data can be used to detect sensor condition or quality, including whether the sensor is malfunctioning or faulty. In sensor diagnostics, sometimes a diagnostic test may indicate a sensor fault, while other diagnoses may indicate no fault. Therefore, the validity of multiple impedance-related parameters and the implementation of multiple test procedures are advantageous, as some of the multiple tests can act as checks on the validity of others. Thus, real-time monitoring using a multi-flag approach can include a degree of built-in redundancy.

[0300] From the above content, we can see that... Figure 31 The diagnostic procedure illustrated begins at step 3100, after the sensor has been inserted / implanted and the EIS has been run to provide EIS data as input. In step 3100, using the EIS data as input, it is first determined whether the sensor is still in the proper position. Therefore, if the |Z| slope is found to be constant throughout the entire test band (or range), and / or the phase angle is approximately -90°, then the sensor is determined to no longer be in the proper position, and an alarm is sent to, for example, the patient / user, indicating that the sensor has been removed. The specific parameters (and their respective values) described herein for detecting sensor removal are based on the finding that once the sensor is outside the body and the membrane is no longer hydrated, the impedance spectrum response behaves like that of a capacitor.

[0301] If it is determined that the sensor is still in the proper position, the logic moves to step 3110 to determine whether the sensor has been properly initialized. As shown, “Init.Check” is performed by determining the following: (1) whether |(Z) is true at 1kHz. n -Z1) / Z1|>30%, where, as discussed above, Z1 is the actual impedance measured at the first moment, Z n (1) The impedance measured at the next interval, and (2) Whether the phase angle change is greater than 10° at 0.1Hz. If the answer to either of the above two questions is "yes", then the test is satisfactory, that is, test 1 passes. Otherwise, test 1 is marked as failed.

[0302] In step 3120, Test 2 asks whether the frequency difference (f2-f1) between two consecutive EIS runs is greater than 10Hz under a phase angle of -45°. Again, a "no" answer is marked as failing; otherwise, Test 2 is satisfactory.

[0303] Test 3 in step 3130 is a hydration test. In this step, the current impedance Z is queried at 1 kHz. n Is it less than the initial impedance Z? pi If so, the test is satisfactory; otherwise, test 3 is marked as failed. Test 4 in step 3140 is also a hydration test, but this time it is performed at a lower frequency. Therefore, this test queries the Z-axis at 0.1Hz during sensor operation after initialization. n Is it less than 300k ohms? Similarly, a "no" answer indicates the sensor failed test 4.

[0304] In step 3150, test 5 queries whether the low-frequency Nyquist slope increases from 0.1 Hz to 1 Hz over the entire range. As discussed above, for a properly functioning sensor, the relatively low-frequency Nyquist slope should increase over time. Therefore, if the answer to this query is "yes," then the test is satisfactory; otherwise, the test is marked as failed.

[0305] Step 3160 is the final test of this diagnostic procedure implementation. In this step, it is asked whether the real impedance decreases throughout the entire range. Again, as discussed above, in a normally functioning sensor, it is foreseeable that the real impedance should decrease over time. Therefore, the answer "yes" in this step indicates that the sensor is functioning normally; otherwise, the sensor fails test 6.

[0306] Once all six tests have been performed, a decision is made at 3170 whether the sensor is operating normally or malfunctioning. In this embodiment, the sensor is determined to be operating normally (3172) if it passes at least three of the six tests. In other words, to determine that the sensor has malfunctioned (3174), the sensor must fail at least four of the six tests. In alternative embodiments, different rules can be used to evaluate normal operation and sensor malfunction. Furthermore, in embodiments of the invention, each test can be weighted such that the assigned weight reflects, for example, the importance of the test or the importance of the specific parameter challenged by the test in determining overall sensor operation (normal vs. malfunction). For example, the weight of one test could be twice that of another test, but only half the weight of a third test, and so on.

[0307] In other alternative implementations, different numbers of tests and / or different groups of EIS-based parameters for each test may be used. Figure 32A and Figure 32B An example of a diagnostic procedure for real-time monitoring, including seven tests, is shown. See also Figure 32AAfter the sensor has been inserted / implanted, the logic begins at step 3200, and the EIS procedure has been executed, providing EIS data as input. In 3200, using the EIS data as input, it is first determined whether the sensor is still in the proper position. Therefore, if the |Z| slope is found to be constant within the detection frequency bandwidth (or range), and / or the phase angle is approximately -90°, then the sensor is determined to no longer be in the proper position, and an alarm is sent to, for example, the patient / user, indicating that the sensor has been removed. On the other hand, if the sensor is determined to be in the proper position, then the logic moves to begin the diagnostic examination (3202).

[0308] In step 3205, test 1 is similar to the above regarding... Figure 31 Test 1 of the diagnostic procedure discussed, except that Test 1 in this embodiment specifies that Z is measured two hours after the first measurement. n Thus, in this embodiment, Z n =Z 2hr More specifically, Test 1 compares the actual impedance two hours after (sensor implantation and) initialization with the value before initialization. Similarly, another part of Test 1 queries whether the phase difference between two hours after initialization and the phase before initialization is greater than 10° at 0.1Hz. As mentioned earlier, if the answer to either query is yes, then the sensor is determined to be normal and initialized, and Test 1 is satisfactory; otherwise, the sensor fails the test. It should be noted that even though the test in this embodiment queries about impedance and phase changes two hours after initialization, the time interval between any two consecutive EIS runs can be shorter or longer, depending on various factors, including, for example, sensor design, electrode redundancy level, the extent to which diagnostic procedures include redundancy testing, battery power, etc.

[0309] Moving to step 3210, the logic then performs a sensitivity loss check by querying whether the percentage change in impedance amplitude and the percentage change in Isig at 1 kHz after a two-hour interval (n+2) are greater than 30%. If the answer to both queries is "yes," then the sensor is determined to have lost sensitivity, and thus, Test 2 is determined to have failed. It should be noted that although Test 2 illustrated herein is based on a preferred 30% difference percentage, in other embodiments, for this test to be performed, the percentage difference in impedance amplitude and the percentage difference in Isig at 1 kHz may fall within the range of 10% to 50%.

[0310] Test 3 (in 3220) is similar to Figure 31The test algorithm 5 is illustrated as an example. In this example, as previously stated, the problem is whether the low-frequency Nyquist slope increases from 0.1 Hz to 1 Hz over the entire range. If so, the test passes; otherwise, it fails. As shown in 3220, the test can also be modified to set a threshold or acceptable range for the percentage change in the low-frequency Nyquist slope. Exceeding this threshold or range, the sensor is considered to have failed the test, at least triggering further diagnostic tests. In embodiments of the invention, this threshold / acceptable range for the percentage change in the low-frequency Nyquist slope may fall within the range of about 2% to about 20%. In some preferred embodiments, the threshold may be about 5%.

[0311] The logic then moves to step 3230, another low-frequency test, this time involving phase and impedance amplitude. More specifically, the phase test queries whether the phase at 0.1 Hz continuously increases over time. If so, the test fails. As with other tests involving monitoring parameter trends, the low-frequency phase test of test 4 can also be modified to set a threshold or acceptable range for the percentage change in low-frequency phase; exceeding this threshold or range may cause the sensor to be considered a failure, or at least raise concerns. In embodiments of the invention, this threshold / acceptable range for the percentage change in low-frequency phase may fall within the range of approximately 5% to approximately 30%. In some preferred embodiments, the threshold may be approximately 10%.

[0312] As described above, Test 4 also includes a low-frequency impedance amplitude test, in which the impedance amplitude at 0.1 Hz is questioned to determine whether it continuously increases over time. If so, the test fails. It should be noted that Test 4 is considered "failed" if either the phase test or the impedance amplitude test fails. The low-frequency impedance amplitude test of Test 4 can also be modified to set a threshold or acceptable range for the percentage change in low-frequency impedance amplitude. If this threshold or range is exceeded, the sensor may be considered to have failed the test, or at least, the sensor should raise concerns. In embodiments of the invention, this threshold / acceptable range for the percentage change in low-frequency impedance amplitude may fall within the range of approximately 5% to approximately 30%. In some preferred embodiments, the threshold may be approximately 10%, where the impedance amplitude of a normal sensor typically ranges from approximately 100 kΩ to approximately 200 kΩ.

[0313] Test 5 (in step 3240) is another sensitivity loss check that can be considered a supplement to Test 2. In this test, if both the percentage change in Isig and the percentage change in impedance amplitude are greater than 30% at 1 kHz, then the sensor is determined to be recovering from sensitivity loss. In other words, the sensor is determined to have previously experienced some sensitivity loss, even if the sensitivity loss was not detected by Test 2 for some reason. As with Test 2, although Test 5 is illustrated based on a preferred 30% difference percentage, in other embodiments, for the purpose of performing this test, the percentage difference in Isig and the percentage difference in impedance amplitude at 1 kHz may be in the range of 10% to 50%.

[0314] Moving to step 3250, Test 6 provides a sensor functional test using specific failure criteria, determined based on observed data and a specific sensor design. Specifically, in one implementation, the sensor may be determined to fail the test if at least two of the following three criteria are met: (1) Isig is less than 10 nA; (2) the virtual impedance at 1 kHz is less than -1500 ohms; and (3) the phase at 1 kHz is less than -15°, thus the sensor is unlikely to respond to glucose. Therefore, if any two of (1) to (3) are not met, then Test 6 is determined to have passed. It should be noted that in other implementations, the Isig test in this test may fail if Isig is less than about 5 nA to about 20 nA. Similarly, the second test may fail if the virtual impedance at 1 kHz is less than about -1000 ohms to about -2000 ohms. Finally, the phase test may fail if the phase at 1 kHz is less than about -10° to about -20°.

[0315] Finally, step 3260 provides another sensitivity check, wherein the parameters are evaluated under low-frequency conditions. Therefore, test 7 queries whether the difference between the ratio of the virtual impedance to Isig(n+2) at 0.1Hz and the previous ratio is greater than 30% of the previous ratio. If so, the test fails; otherwise, the test passes. In this embodiment, while test 7 is illustrated based on a preferred 30% difference percentage, in other embodiments, the difference percentage may range from 10% to 50% for this test to be performed.

[0316] Once all seven tests have been performed, a decision is made at 3270 regarding whether the sensor is functioning correctly or whether an alarm should be issued to indicate a sensor malfunction (or potential malfunction). As shown, in this embodiment, if the sensor passes at least four of the seven tests, then the sensor is determined to be functioning correctly (3272). In other words, to determine that the sensor has malfunctioned or at least warrants attention (3274), the sensor must fail at least four of the seven tests. If the sensor is determined to be "bad" (3274), then an alarm indicating this condition can be sent to, for example, a patient / user. As previously mentioned, in alternative embodiments, different rules can be used to evaluate normal operation and sensor malfunction / concern. Furthermore, in embodiments of the invention, each test can be weighted so that the assigned weight reflects, for example, the importance of the test, or the importance of the specific parameter inquired by the test, in determining the overall operation of the sensor (normal vs. malfunction).

[0317] As described above, in embodiments of the present invention, the various impedance-related parameters described above can be used individually or in combination as input to one or more fusion algorithms for generating more reliable sensor glucose values. Specifically, it is known that, unlike a single sensor (i.e., a single working electrode) system, multiple detection electrodes provide more reliable glucose readings. Instead, glucose readings obtained from two or more working electrodes are fused as multiple signals to provide a single sensor glucose value. This signal fusion uses quantitative input provided by the EIS to calculate the most reliable output sensor glucose value from the redundant working electrodes. It should be noted that while the following discussion may describe various different fusion algorithms based on the first working electrode (WE1) and the second working electrode (WE2) as redundant electrodes, this is illustrative and not limiting, as the algorithms and their basic principles described herein are applicable to and can be used in redundant sensor systems with more than two working electrodes.

[0318] Figure 33A and Figure 33B The top-level flowchart for two alternative methods is shown, each of which includes a fusion algorithm. Specifically, Figure 33A This is a flowchart involving a current-based (Isig) fusion algorithm. Figure 33B This is a flowchart involving glucose (SG) fusion with a sensor. As can be seen from the diagram, the main difference between the two methods lies in the calibration time. Therefore, Figure 33AAs shown, for Isig fusion, calibration 3590 is performed after fusion 3540. That is, redundant Isigs from WE1 to WEn are fused into a single Isig 3589, which is then calibrated to produce a single sensor glucose value 3598. On the other hand, for SG fusion, calibration 3435 is performed on each individual Isig from WE1 to WEn to produce a calibrated SG value (e.g., 3436, 3438) for each working electrode. Therefore, the SG fusion algorithm provides independent calibration for each of the multiple Isigs, which may be preferred in embodiments of the invention. Once calibrated, the multiple calibrated SG values ​​are fused into a single SG value 3498.

[0319] It should be noted that, importantly, Figure 33A and Figure 33B Each of the flowcharts shown includes a spike signal filtering process (3520, 3420). As discussed above regarding sensitivity loss, impedance measurements at frequencies of 1 kHz or higher typically induce EIS-induced spike signals in the Isig. Therefore, once the EIS procedure is performed on each of the electrodes WE1 to WEN, it is preferable to first filter Isig 3410, 3412, etc., and 3510, 3512, etc., to obtain their respective filtered Isig 3422, 3424, etc., and 3522, 3524, etc., for both SG fusion and Isig fusion. As discussed in detail below, the filtered Isig is then used for Isig fusion, or first calibrated and then used for SG fusion. As will become apparent in the following discussion, the two fusion algorithms calculate and assign weights based on various different factors.

[0320] Figure 34 Details of the fusion algorithm 3440 used for SG fusion are shown. Essentially, four factors need to be checked before determining the fusion weights. First, the integrity check 3450 includes determining whether each of the following parameters falls within a specified range for normal sensor operation (e.g., predetermined lower and upper thresholds): (i) Isig; (ii) 1 kHz real and virtual impedance; (iii) 0.105 Hz real and virtual impedance; and (iv) Nyquist slope. As shown, the integrity check 3450 includes a bound check 3452 and a noise check 3456, where the above parameters are used as input parameters for each check. It should be noted that, in short, real and / or virtual impedances appear under conditions of one or more frequencies. Figures 33A to 35 Impedance is abbreviated as "Imp" in Chinese. Furthermore, both real and virtual impedances can use impedance magnitude and phase (in... Figure 33A and Figure 33BThe input is also shown in the diagram for calculation.

[0321] The outputs of each of the boundary check 3452 and the noise check 3458 are the respective reliability index (RI) for each of the redundant working electrodes. Therefore, the outputs of the boundary check include, for example, RI_bound_We1 (3543) and RI_bound_We2 (3454). Similarly, for the noise check, the outputs include, for example, RI_noise_We1 (3457) and RI_noise_We2 (3458). The boundary and noise reliability indices for each working electrode can be calculated based on conformity to the aforementioned ranges for normal sensor operation. Therefore, if any of the parameters is outside the specified range for a particular electrode, the reliability index of that particular electrode decreases.

[0322] It should be noted that the thresholds or ranges of the above parameters can depend on various factors, including specific sensor and / or electrode designs. Nevertheless, in a preferred embodiment, typical ranges for some of the above parameters may be as follows: threshold for 1 kHz real impedance = [0.3e+4 - 2e+4]; threshold for 1 kHz virtual impedance = [-2e+3, 0]; threshold for 0.105 Hz real impedance = [2e+4 - 7e+4]; threshold for 0.105 Hz virtual impedance = [-2e+5 - 0.25e+5]; threshold for Nyquist slope = [2, 5]. Noise can be calculated, for example, using a second-order central difference method, where noise is considered to exceed noise limits if it exceeds a certain percentage (e.g., 30%) of the median value of each variable buffer.

[0323] Second, the sensor drop can be detected using the sensor current (Isig) and the 1kHz real impedance. Therefore, as... Figure 34 As shown, Isig and "Imp" are used as inputs for drop detection 3460. The first step here is to determine if there is any divergence between Isig values, and whether any such divergence is reflected in the 1kHz real impedance data. This is achieved using a mapping 3465 between the Isig similarity index (RI_sim_isig12) 3463 and the 1kHz real impedance similarity index (RI_sim_imp12) 3464. This mapping is crucial because it helps avoid false positives in cases where the drop is not genuine. If the Isig deviation is genuine, the algorithm selects the sensor with the higher Isig value.

[0324] According to an embodiment of the present invention, the divergence / convergence of two signals (e.g., two Isig signals, or two 1kHz real impedance data points) can be calculated as follows:

[0325] diff_va1=abs(va1-(va1+va2) / 2);

[0326] diff_va2=abs(va2-(va1+va2) / 2);

[0327] RI_sim=1-(diff_va1+diff_va2) / (mean(abs(va1+va2)) / 4)

[0328] Here, va1 and va2 are two variables, and RI_sim (similarity index) is an index that measures the convergence or divergence of the signal. In this implementation, RI_sim must be between 0 and 1. Therefore, if the RI_sim calculated above is less than 0, then RI_sim is set to 0; if it is greater than 1, then RI_sim is set to 1.

[0329] Mapping is performed using ordinary linear regression (OLR) 3465. However, when OLR does not work well, robust median slope linear regression (RMSLR) can be used. For example, for the Isig similarity index and the 1kHz real impedance index, the following two mapping processes are required: (i) mapping the Isig similarity index to the 1kHz real impedance similarity index; and (ii) mapping the 1kHz real impedance similarity index to the Isig similarity index. These two mapping processes produce two residuals: res12 and res21. Each of the decremental reliability indices 3467 and 3468 can then be calculated as follows:

[0330] RI_dip=1–(res12+res21) / (RI_sim_isig+RI_sim_1K_real_impedance).

[0331] The third factor is sensitivity loss 3470, which can be detected using, for example, a 1kHz virtual impedance trend over the past 8 hours. If the trend of one sensor becomes negative, the algorithm will rely on the other sensor. If both sensors suffer sensitivity loss, a simple average is taken. The trend can be calculated by using a strong low-pass filter to eliminate the 1kHz virtual impedance, which becomes noise, and the trend can be determined by using a correlation coefficient or linear regression over time (e.g., the past eight hours) to determine whether the correlation coefficient or the slope is negative. Each of the sensitivity loss reliability indices 3473 and 3474 is then assigned a binary value of 1 or 0.

[0332] The overall reliability index (RI) for each of we1, we2, ..., wen is calculated as follows:

[0333] RI_we1=RI_dip_we1×RI_sensitivity_loss_we1×RI_bound_we1×RI_noise_we1

[0334] RI_we2=RI_dip_we2×RI_sensitivity_loss_we2×RI_bound_we2×RI_noise_we2

[0335] RI_we3=RI_dip_we3×RI_sensitivity_loss_we3×RI_bound_we3×RI_noise_we3

[0336] RI_we4=RI_dip_we4×RI_sensitivity_loss_we4×RI_bound_we4×RI_noise_we4

[0337]

[0338] RI_we n =RI_dip_we n ×RI_sensitivity_loss_we n ×RI_bound_we n ×RI_noise_we n

[0339] After calculating the reliability index of each individual working electrode, the weight of each electrode is calculated as follows:

[0340] weight_we1=RI_we1 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0341] weight_we2=RI_we2 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0342] weight_we3=RI_we3 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0343] weight_we4=RI_we4 / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0344]

[0345] weight_we n =RI_we n / (RI_we1+RI_we2+RI_we3+RI_we4+…+RI_we n )

[0346] Based on the above, the fused SG 3498 is then calculated as follows:

[0347] SG=weight_we1×SG_we1+weight_we2×SG_we2+weight_we3×SG_we3+weight_we4×SG_we4+...+weight_we n ×SG_we n

[0348] The final factor involves spurious information in the final sensor readout, which may be caused by transient weight changes during sensor fusion. This can be avoided by using a low-pass filter 3480 to smooth the RI of each electrode, or by applying a low-pass filter to the final SG. When the former is used, the reliability exponents of the filter—e.g., RI_We1* and RI_We2* (3482, 3484)—are used to calculate the weight of each electrode, and thus to calculate the fused SG 3498.

[0349] Figure 35 The details of the Isig fusion algorithm 3540 are shown. As can be seen from the figure, this algorithm is similar to... Figure 34 The fusion algorithms shown for SG fusion are essentially similar, with only two differences. First, as mentioned earlier, for Isig fusion, calibration constitutes the final step of the method, where the Isig3589 of a single fusion is calibrated to produce a single sensor glucose value of 3598. See also... Figure 33B Second, although SG fusion uses the SG values ​​of multiple electrodes to calculate the final SG value 3498, the fused Isig value 3589 is calculated by using the filtered Isig (3522, 3524, etc.) of multiple electrodes.

[0350] In a closed-loop study involving a non-diabetic population, the aforementioned fusion algorithm provided a significant improvement in mean absolute relative difference (MARD) on day 1 and throughout the entire lifespan (i.e., the seven-day lifespan of the sensor). On day 1, the low start-up problem was most pronounced and thus had a significant impact on the accuracy and reliability of the sensor. The study evaluated an 88% distributed layout design with a high current density (rated) coating using three different methods: (1) calculating a sensor glucose value (SG) using Medtronic Minimed's Ferrari Algorithm 1.0 (the SG fusion algorithm discussed earlier); (2) calculating an SG by identifying a better ISIG value using 1kHz EIS data (via the Isig fusion algorithm described above); and (3) calculating an SG using a higher ISIG value (i.e., without using EIS). Details of the data used in the study are as follows:

[0351] (1) For a distributed arrangement with 88% of the coatings having high current density (rated), based on Ferrari 1.0Alg's SG

[0352]

[0353]

[0354]

[0355]

[0356]

[0357]

[0358] (2) For a distributed arrangement with 88% of the coatings having high current density (rated), based on the SG of ISIG with better 1kHz EIS.

[0359]

[0360]

[0361]

[0362]

[0363]

[0364] (3) For a distributed arrangement with 88% of the coatings having a high current density (rated), based on a higher ISIG SG

[0365]

[0366]

[0367]

[0368]

[0369]

[0370] The data above reveals that, using the first method, the MARD (%) on the first day was 19.52%, and the overall MARD was 12.28%. For the second method, the MARD on the first day was 15.96%, and the overall MARD was 11.83%. Finally, for the third method, the MARD on the first day was 17.44%, and the overall MARD was 12.26%. Therefore, for this design using redundant electrodes, the better ISIG calculation SG based on 1kHz EIS (i.e., the second method) offers the greatest advantage. Specifically, the lower first-day MARD can be attributed, for example, to better detection of low startup using EIS. Furthermore, the overall MARD percentage in this study is greater than 1% lower than the average overall MARD of 13.5% for WE1 and WE2. It should be noted that in the above methods, data transformation can be controlled, for example, by filtering methods to minimize the degree of transformation, such as by using the aforementioned... Figures 33A to 35 The low-pass filter discussed is 3480.

[0371] It is worth reiterating that sensor diagnostics, including, for example, assessments of low-start-up, sensitivity loss, and signal degradation events, depend on a wide variety of factors, including sensor design, the number of electrodes (i.e., redundancy), electrode distribution / configuration, and so on. Therefore, the actual frequency or frequency range based on ESI parameters can be substantially independent of glucose, and thus, independent indicators or indications for one or more of the aforementioned failure modes can also depend on the specific sensor design. For example, as mentioned above, although it has been found that sensitivity loss can be predicted using virtual impedance at relatively high frequencies—where virtual impedance is substantially independent of glucose—glucose-dependent levels, the specific frequency range of virtual impedance as an indicator of sensitivity loss can vary (become higher or lower) depending on the actual sensor design.

[0372] More specifically, as sensor designs increasingly utilize redundant working electrodes, the size of these electrodes must become smaller to maintain the overall sensor size. The size of the electrodes, in turn, affects the frequency required for specific diagnostics. In this regard, it is important to note the points discussed herein and... Figures 33A to 35The fusion algorithms shown are intended to be exemplary and not limiting. Each algorithm may be modified as needed based on the type of sensor in the analysis to use EIS-based parameters at frequencies exhibiting minimal glucose dependence.

[0373] Furthermore, experimental data indicate that human tissue structure can also affect glucose dependence at different frequencies. For example, in children, the real impedance at 0.105 Hz was found to be a low-start detection indicator that is largely independent of glucose. This is thought to stem from changes in children's tissue structure, such as Warburg impedance, which primarily involves the resistive component. See also the following discussion on interference detection.

[0374] Embodiments of the invention described herein also relate to the use of EIS in optimizing sensor calibration. As background, in current methods, the slope of the BG-Isig curve used to calibrate subsequent Isig values ​​is calculated as follows:

[0375]

[0376] Here, α is an exponential function of the time constant, β is a function of the blood glucose variable, and the offset (compensation) is a constant. For sensors in steady-state conditions, this method provides fairly accurate results. For example... Figure 36 As shown, BG and Isig follow a fairly linear relationship, and offset can be obtained as a constant.

[0377] However, there are situations where the above linear relationship does not apply, such as during sensor switching processes. From Figure 37 It is clear that, based on the relationship between Isig and BG, Isig-BG differs significantly for cases 1 and 2 from that for cases 3 and 4. For these types of cases, the use of the constant offset is prone to producing inaccurate results.

[0378] To address this problem, one embodiment of the present invention involves using EIS-based dynamic compensation, wherein EIS measurements are used to define a sensor state vector, as follows:

[0379] V={real_imp_1K,img_imp_1K,Nyquist_slope,Nyquist_R_square}

[0380] In this vector, all elements are essentially independent of BG. It should be noted that Nyquist_R_square is the R-squared of the linear regression used to calculate the Nyquist slope, i.e., the square of the correlation coefficient between the real and imaginary impedances at relatively low frequencies, and a low R-squared indicates anomalies in sensor performance. A state vector is assigned to each Isig-BG pair. If a significant difference is detected in the state vector—for example… Figure 37 The example shown, |V2-V3|, assigns different offset values ​​to 3 and 4 when comparing them with 1 and 2. Therefore, by using this dynamic compensation method, it is possible to maintain the linear relationship between Isig and BG.

[0381] In the second embodiment, an EIS-based segmentation method can be used for calibration. Figure 37 Based on the embodiments and vector V, it can be determined that the sensor state in cases 1 and 2 is significantly different from the sensor state in cases 3 and 4. Therefore, the calibration buffer can be divided into the following two parts:

[0382] Isig_buffer1=[Isig1,Isig2]; BG_buffer1=[BG1,BG2]

[0383] Isig_buffer2=[Isig3,Isig3]; BG_buffer2=[BG3,BG3]

[0384] Therefore, when the sensor is operating in scenarios 1 and 2, Isig_buffer1 and BG_buffer1 can be used for calibration. However, when the sensor is operating in scenarios 3 and 4, i.e., during conversion, Isig_buffer2 and BG_buffer2 can be used for calibration.

[0385] In another embodiment, an EIS-based dynamic slope method can be used for calibration purposes, wherein EIS is used to adjust the slope. Figure 38A This illustrates an example of how this method can be used to improve sensor accuracy. In this graph, data points 1 through 4 are discrete blood glucose values. From Figure 38A As can be seen, a sensor drop of 3810 occurs between data points 1 and 3. This drop can be detected using the aforementioned sensor state vector V. During the drop, the slope can be adjusted upwards to reduce insufficient readings, such as... Figure 38A As shown by reference numeral 3820 in the attached figure.

[0386] In a further embodiment, EIS diagnostics can be used to determine when sensor calibration is needed, which is particularly useful for events such as low-start-up events, sensitivity loss events, and other similar situations. Most current methods known in the art require periodic calibration based on a preset schedule, such as four calibrations per day. However, using EIS diagnostics, calibration becomes event-driven, allowing calibration to be performed only as needed and when it is most effective. Similarly, the state vector V can be used to determine when the sensor state changes and, if so, to request calibration.

[0387] More specifically, in an exemplary embodiment, Figure 38B This illustrates an EIS-assisted sensor calibration procedure including low-start-up detection. The 3850 is checked using the Nyquist slope, 1kHz real impedance, and limit checks (see, for example, the aforementioned). Figures 33A to 35 (Based on boundary checks related to the fusion algorithm and relevant thresholds based on EIS parameters), the reliability index 3853 can be developed for startup, such that when the 1kHz real impedance 3851 and the Nyquist slope 3852 are lower than their respective upper limits, RI_startup = 1, and the sensor is ready for calibration. In other words, if the reliability index 3853 is "high" (3854), the logic can proceed to calibration up to 3860.

[0388] On the other hand, when the 1kHz real impedance and Nyquist slope are above their respective upper limits (or thresholds), RI_startup = 0 (i.e., "low"), and the sensor is not ready for calibration (3856), meaning a low-startup problem may exist. Here, the trends of the 1kHz real impedance and Nyquist slope can be used to predict when these two parameters will be within range (3870). If the estimation only requires a very short time (e.g., less than 1 hour), the algorithm waits until the sensor is ready, i.e., until the aforementioned EIS-based parameters are within range (3874), at which point the algorithm performs calibration. However, if the waiting time is relatively long (3876), the sensor can be calibrated now, and the slope or offset can be gradually adjusted according to the trends of the 1kHz real impedance and Nyquist slope (3880). It should be noted that by making such adjustments, severe over-reading or under-reading caused by low startup can be avoided. As previously stated, the EIS-based parameters and related information used in this calibration algorithm are substantially independent of glucose.

[0389] It should be noted that, although the above regarding Figure 38BThe description illustrates a single working electrode and the calculation of its startup reliability index, but this is exemplary and not limiting. Therefore, in redundant sensors comprising two or more working electrodes, each of the multiple (redundant) working electrodes can be boundary-checked and a startup reliability index can be calculated. Then, based on their respective reliability indices, at least one working electrode can be identified as capable of continuing to operate to obtain glucose measurements. In other words, in a sensor with a single working electrode, if the working electrode exhibits a low startup, the actual use of the sensor (for measuring glucose) may be delayed until the low startup process has passed. This period is typically on the order of an hour or more, which is clearly disadvantageous. Conversely, in redundant sensors, the method described herein allows for adaptive startup or “smart” startup, where the electrode capable of data acquisition can be identified within a fairly short time, for example, on the order of a few minutes. This, in turn, reduces MARD, as a low startup typically increases MARD by approximately 1 / 2%.

[0390] In another implementation, EIS can help tune the calibration buffer. For existing calibration algorithms, the buffer size is typically 4, i.e., 4 Isig-BG pairs, and the weights are based on α and β, where α is an exponential function of the time constant and β is a function of the blood glucose variable, as mentioned earlier. Here, EIS can help determine when to flush the buffer, how to adjust the buffer weights, and what the appropriate buffer size is.

[0391] Embodiments of the present invention also relate to the application of EIS in the detection of interfering substances. Specifically, it is desirable to provide a drug infusion assembly comprising a combination of a sensor and a drug infusion catheter, wherein the sensor is placed within the infusion catheter. In this system, the physical position of the infusion catheter relative to the sensor needs to be considered, primarily due to the potential influence (i.e., interference with the sensor signal) on the sensor signal caused by the drug being infused and / or its inactive components.

[0392] For example, diluents used with insulin contain m-cresol as a preservative. In in vitro studies, if insulin (and thus m-cresol) is injected at a location immediately adjacent to the sensor, m-cresol has been found to adversely affect the glucose sensor. Therefore, systems that integrate the sensor and infusion catheter in a single needle must be able to detect or modulate the effect of m-cresol on the sensor signal. Because m-cresol affects the sensor signal, it is preferable to provide a method for detecting this interfering agent independently of the sensor signal itself.

[0393] Experiments have shown that the effect of m-cresol on the sensor signal is temporary and reversible. Nevertheless, when insulin infusion is too close to the sensor, m-cresol can easily "poison" the electrode, causing it to stop detecting glucose until the insulin (and m-cresol) are absorbed into the patient's tissues. In this respect, it has been found that the time interval between the start of insulin infusion and the sensor regaining its ability to detect glucose is typically about 40 minutes. However, advantageously, it has been found that within the same time interval, there is a significant increase in the 1 kHz impedance amplitude, completely independent of glucose concentration.

[0394] Specifically, Figure 39 The Isig and impedance data from the in vitro experiment are shown, where the sensor was placed in a 100 mg / dL glucose solution, and the impedance was measured at 1 kHz every 10 minutes, represented by the circled data point 3920. m-Cresol was then added to bring the solution to 0.35% m-Cresol (3930). It can be seen that Isig 3940 initially increases significantly upon the addition of m-Cresol, and then gradually decreases. The glucose concentration in the solution was doubled by adding another 100 mg / dL of glucose. However, this had no effect on Isig 3940 because the electrode could not detect glucose.

[0395] On the other hand, m-cresol has a significant effect on impedance amplitude and phase. Figure 40A The Bode plots of the phases before and after the addition of m-cresol are shown. Figure 40B The Bode plots of the impedance amplitudes before and after the addition of m-cresol are shown. As can be seen from the plot, after the addition of m-cresol, the impedance amplitude 4010 increases by at least one order of magnitude across the entire spectrum, starting from its initial value 4020. Simultaneously, the phase 4030 completely changes relative to its initial value 4040. Figure 40C In the Nyquist curves, curve 4050 before initialization and curve 4060 after initialization behave as expected, indicating a normally functioning sensor. However, curve 4070 is significantly different after the addition of m-cresol.

[0396] The above experiments identified a significant practical drawback of relying on Isig even after the addition of m-cresol. See again Figure 39Patients / users monitoring sensor signals may be misled into believing their glucose levels have suddenly spiked and they should receive a large dose of insulin. The user then administers the large dose, by which time the isig (insulin level) has begun to gradually decrease. In other words, everything may appear normal to the patient / user. However, what actually occurs is that the patient is receiving an unnecessary dose of insulin based on their pre-large dose glucose level, potentially putting them at risk of experiencing hypoglycemia. This hypothesis underscores the need for methods that are as independent as possible from glucose detection interferences.

[0397] Figure 41 Another experiment is shown, in which the sensor was initialized in a 100 mg / dL glucose solution, then the glucose level was increased to 400 mg / dL and maintained for one hour, before returning to 100 mg / dL. Subsequently, m-cresol was added to increase its concentration to 0.35%, and the sensor was maintained in this solution for 20 minutes. Finally, the sensor was placed in a 100 mg / dL glucose solution to allow Isig to recover after exposure to m-cresol. As can be seen from the figure, after initialization, the impedance amplitude 4110 at 1 kHz was approximately 2 kΩ. When m-cresol was added, Isig 4120 suddenly increased, and the impedance amplitude 4110 also suddenly increased. Moreover, when the sensor was returned to the 100 mg / dL glucose solution, the impedance amplitude 4110 also returned to near normal levels.

[0398] As can be seen from the above experiments, EIS can be used to detect the presence of interfering agents—in this example, m-cresol. Specifically, because the interfering agent affects the sensor in a way that increases the impedance amplitude across the entire frequency spectrum, the impedance amplitude can be used to detect the interference. Once interference is detected, the sensor operating voltage can be changed to a point where the interfering agent cannot be measured, or data reporting can be temporarily suspended. The sensor can then indicate to the patient / user that it cannot report data due to drug administration (until the measured impedance returns to pre-infusion levels). It should be noted that because the effect of the interfering agent is due to the preservative contained in insulin, the impedance amplitude will exhibit the same changes as described above, regardless of whether the insulin is injected rapidly or slowly.

[0399] Importantly, as mentioned above, the impedance amplitude is essentially independent of glucose, and the amplitude at 1 kHz is also essentially independent of glucose. See [link to relevant documentation]. Figure 41As can be seen, when the glucose concentration increases from 100 mg / dL to 400 mg / dL—a fourfold increase—the impedance amplitude at 1 kHz increases from approximately 2000 ohms to approximately 2200 ohms, or an increase of approximately 10%. In other words, the effect of glucose on the measured impedance amplitude appears to be about an order of magnitude smaller than the measured amplitude. This level of "signal-to-noise ratio" is typically small enough to filter out noise (i.e., the glucose effect), so that the remaining impedance amplitude is essentially independent of glucose. Furthermore, it should be emphasized that the impedance amplitude exhibits an even higher degree of glucose independence in actual human tissues compared to the buffer solutions used in the aforementioned in vitro experiments.

[0400] Embodiments of the present invention also relate to an Analog Front End Integrated Circuit (AFEIC), a custom application-specific integrated circuit (ASIC) that provides the necessary analog electronics to: (i) support multiple potentiometers and multi-terminal glucose sensor interfaces based on oxygen or peroxides; (ii) connect to a microcontroller to form a low-power sensor system; and (iii) perform EIS diagnostics, fusion algorithms, and other EIS-based methods based on measurements of EIS parameters. More specifically, the ASIC incorporates the sensor's diagnostic capabilities for measuring both real and virtual impedance parameters over a wide frequency range, along with a digital interface circuit for bidirectional communication with the microprocessor chip. Furthermore, the ASIC includes power control circuitry, a real-time clock, and a crystal oscillator capable of operating at very low standby and operating power, allowing the external microprocessor power supply to be turned off.

[0401] Figure 42A and 42B A block diagram of the ASIC is shown below, and Table 1 provides a description of the pad signals (in...). Figure 42A and 42B (As shown on the left), some of these signals are multiplexed on a single contact.

[0402]

[0403]

[0404]

[0405] Reference Figure 42A and 42B And Table 1 describes the ASIC.

[0406] Power Plane

[0407] The ASIC has a power supply board powered via a power supply contact VBAT (4210) with an operating input ranging from 2.0 volts to 4.5 volts. The power supply board has a regulator to reduce the voltage used by some of the circuitry on the board. The power supply is called VDDBU (4212) and has output contacts for testing and shunt operation. The circuitry on the VBAT power supply includes an RC oscillator, a real-time clock (RC osc) 4214, battery protection circuitry, regulator control, power-on reset (POR) circuitry, and various inputs / outputs. The contacts on the VBAT power supply board are configured to handle less than 75 nA at 40°C and VBAT = 3.50 V.

[0408] The ASIC also features a VDD power supply for providing logic. The VDD power supply voltage range is programmable from at least 1.6 volts to 2.4 volts. The circuitry on the VDD power board includes most of the digital logic, timers (32kHz), and a real-time clock (32kHz). The VDD power board includes level shifters that connect to other voltage boards as needed. The level shifters further have interfaces that are regulated to prevent any power board from generating a current increase greater than 10nA if the other power board is not powered.

[0409] The ASIC includes an on-board regulator (with shutdown control) and an external VDD source selection. The regulator input is a separate contact, REG_VDD_IN (4216), which, along with other I / O on VBAT, has electrostatic discharge (ESD) protection. The on-board regulator has an output contact, REG_VDD_OUT (4217). The ASIC also has an input contact for VDD, independent of the REG_VDD_OUT contact.

[0410] The ASIC includes an analog power supply board, called VDDA (4218), which is powered by a regulator on the VDD board or an external source, and is typically powered by a filtered VDD. The VDDA power supply circuitry is configured to operate within a 0.1-volt VDD range, thus eliminating the need for level conversion between the VDDA and VDD power supply boards. The VDDA power supply powers sensor analog circuitry, analog measurement circuitry, and any other noise-sensitive circuitry.

[0411] The ASIC includes a contact power supply and a VPAD for specifying digital interface signals. The contact power supply operates at a voltage ranging from at least 1.8V to 3.3V. These contacts have individual power supply contacts and are powered by an external source. The contacts also incorporate level shifters into other on-board circuitry to make the flexible contact power supply range independent of the VDD logic supply voltage. The ASIC can adjust the VPAD contact loop signal so that when the VPAD power supply is disabled, other supply currents do not increase by more than 10nA.

[0412] bias generator

[0413] The ASIC features a bias generator circuit, BIAS_GEN(4220), which is powered by a VBAT supply and generates a bias current that is stable when the system is supplied with voltage. The output current has the following specific characteristics: (i) supply sensitivity of <±2.5% under a supply voltage range of 1.6V to 4.5V; and (ii) current accuracy after regulation of <±3%.

[0414] The BIAS_GEN circuit generates both converted and unconverted output currents to power circuits that require bias current to operate. The operating current consumption of the BIAS_GEN circuit, at 25°C with VBAT ranging from 2.5V to 4.5V, is less than 0.3µA (excluding any bias output current). Ultimately, the temperature coefficient of the bias current is typically between 4,000ppm / °C and 6,000ppm / °C.

[0415] Reference voltage

[0416] As described herein, the ASIC is configured to have a low reference supply voltage, powered by the VBAT power supply. The reference voltage has an enable input that accepts a signal from logic powered by either VBAT or VDDBU. The ASIC is designed such that when VBAT is powered on, the enable signal does not cause any current increase exceeding 10nA from any power supply from the signal interface.

[0417] The reference voltage has the following specific characteristics: (i) Output voltage: 1.220 ± 3 mV after regulation; (ii) Supply sensitivity: < ± 6 mV under 1.6 V to 4.5 V input conditions; (iii) Temperature sensitivity: < ± 5 mV under 0 °C to 60 °C conditions; and (iv) Default output voltage accuracy (unregulated): 1.220 V ± 50 mV. Furthermore, the source current is less than 800 nA at 4.5 V and 40 °C. In this embodiment, when the reference is disabled, the reference output is driven to VSSA, thereby preventing the VDD voltage regulator from overshooting to a level exceeding the logic breakdown voltage.

[0418] 32kHz oscillator

[0419] The ASIC includes a low-power 32.768kHz crystal oscillator 4222, which is powered by the VDDA supply and whose capacitance at the crystal oscillator contacts (XTALI, XTALO) can be adjusted via software. Specifically, the frequency adjustment range is at least -50ppm to +100ppm, using a maximum step size of 2ppm across the entire adjustment range. Here, the crystal can be assumed to have a load capacitance of 7pF, Ls = 6.9512kH, Cs = 3.3952fF, Rs = 70kΩ, a parallel capacitance of 1pF, and a PCB parasitic capacitance of 2pF on each crystal terminal.

[0420] The ASIC provides a VPAD-level output on the CLK_32kHz contact, which can be disabled via software and logic control. The default value drives a 32kHz oscillator. The OSC32K_BYPASS (4224) input pin disables the 32kHz oscillator (without power consumption) and enables the XTALI contact digital input. The circuitry associated with this function is configured not to increase any ASIC current exceeding 10nA in any state of the OSC32K_BYPASS signal, which differs from the oscillator current, when OSC32K_BYPASS is low.

[0421] The 32kHz oscillator is required to always operate when the VDDA board is powered on, except in shunt mode. If OSC32K_BYPASS is true, the 32kHz oscillator analog circuitry enters a low-power state, and the XTALI contact is configured to receive digital inputs with levels from 0 to VDDA. It should be noted that the 32kHz oscillator output has a duty cycle of 40% to 60%.

[0422] timer

[0423] The ASIC includes a timer 4226, which times the clock by dividing by 2 using a 32kHz oscillator. This timer is presettable and has two programmable timeout settings. It has 4 programmable bits, providing a total count of 17 minutes and 4 seconds. The timer also has a programmable delay to disable the clock on the CLK_32kHz contact and to set the microprocessor (uP) interface signals on the VPAD board to a predetermined state (see the Microprocessor Wake-up Control Signals section below). This allows the microprocessor to enter a pause mode without an external clock. However, this function can be disabled via software using the programmable bits.

[0424] The timer also includes a programmable delay to wake the microprocessor by enabling the CLK_32kHz clock output and setting UP_WAKEUP high. The transition of POR2 (VDD POR) from a low power state to an OK power state enables the 32kHz oscillator, outputs the CLK_32kHz clock, and sets UP_WAKEUP high. Power-off and power-on are configured to be controlled by programmable control bits.

[0425] Real-time clock (RTC)

[0426] The ASIC also features a 48-bit readable / writable binary counter, which operates via a non-gated, free-running 32kHz oscillator. Write operations to the Real-Time Clock 4228 require writing the key address before the clock can be written. The clock write path is configured to stop 1 millisecond to 20 milliseconds after writing the key address.

[0427] The real-time clock 4228 is configured to be reset to half-count (MSB = 1, all other bits are 0) upon power-on reset via POR1_IN (VBATPOR) or POR2_IN (VDD_POR). In embodiments of the invention, the real-time clock has programmable interrupt capability and is designed to be stable against single-event upsets (SEUs), which can be achieved, if desired, through layout techniques or by adding capacitance to appropriate nodes.

[0428] RC oscillator

[0429] The ASIC also includes an RC clock powered by a VBAT supply or a VBAT-derived supply. The RC oscillator operates continuously, except that it can be bypassed by writing to a register bit and applying a signal to GPIO_VBAT at levels 0 to VBAT in analog test mode (see Digital Test section). The RC oscillator is non-adjustable and includes the following specific characteristics: (i) a frequency of 750Hz to 1500Hz; (ii) a duty cycle of 50% ± 10%; (iii) a current consumption of less than 200nA at 25°C; (iv) a frequency variation of less than ±2% under a VBAT supply of 1V to 4.5V, and a frequency variation greater than 1% under a VBAT supply of 1.8V to 4.5V; and (v) a frequency variation of less than +2% and -2% at temperatures of 15°C to 40°C and VBAT = 3.5V. The RC frequency can be measured using a 32kHz crystal oscillator or an external frequency source (see Oscillator Calibration Circuit).

[0430] Real-time RC clock (based on RC oscillator)

[0431] The ASIC includes a 48-bit readable / writable binary ripple counter based on an RC oscillator. Writing to the RC real-time clock requires writing the key address before writing the clock. The clock write path stops 1ms to 20ms after writing the key address, where the guard window time is configured to be generated by the RC clock.

[0432] If the crystal oscillator is off, the real-time RC clock allows for the generation of a relative timestamp and is configured to be reset to half-count (MSB = 1, all others 0) via POR1_IN (BAT POR). The real-time RC clock is designed to be stable relative to a single-event flip (SEU), which can be achieved through wiring techniques or by adding capacitance to appropriate nodes (if needed). On the falling edge of POR2_IN, or if the ASIC enters a low battery state, the RC real-time clock value can be captured into a register that can be read via the SPI port. This register and associated logic reside on the VBAT or VDDBU power board.

[0433] Battery protection circuit

[0434] The ASIC includes a battery protection circuit 4230 that uses a comparator to monitor the battery voltage and is powered by a power supply from the VBAT power board. The battery protection circuit is configured to operate continuously using the power applied to the VBAT power supply. The battery protection circuit may use an RC oscillator for timing signals and may have an average current consumption of less than 30 nA. The battery protection circuit includes an external voltage divider with a total resistance of 3 M ohms.

[0435] The battery protection circuit uses an external voltage divider with a ratio of 0.421 to the 2.90V battery threshold. The ASIC also has an internal voltage divider with a ratio of 0.421 ± 0.5%. This divider is connected between BATT_DIV_EN (4232) and VSSA (4234), and its output is a pin called BATT_DIV_INT (4236). To save package pins, in this implementation, BATT_DIV_INT is internally connected to BATT_DIV within the package. Furthermore, in this configuration, BATT_DIV_EN does not need to extend from the package, saving two package pins.

[0436] The battery protection circuit is configured to sample the voltage on the input pin BATT_DIV (4238) approximately twice per second, where the sampling time is generated by an RC oscillator. The ASIC can adjust the voltage divider of the RC oscillator, thereby adjusting the sampling interval to 0.500 seconds ± 5 milliseconds while the RC oscillator is operating within its tolerance range. In a preferred embodiment, the ASIC has a test mode that allows for more frequent sampling intervals during testing.

[0437] The comparator input is configured to accept inputs from 0 volts to VBAT volts. For inputs from 0 volts to VBAT volts, the input current BATT_DIV at the comparator input is less than 10 nA. The comparator sampling circuit outputs a positive pulse to the contact BATT_DIV_EN, which can be used by external circuitry to enable the on-chip external resistor divider only during sampling time to save power. The high logic level is VBAT voltage, and the low level is VSS level.

[0438] With VBAT = 3.0V, the output resistance of the BATT_DIV_EN contact can be less than 2k ohms. This allows the voltage divider to be directly driven by this output. After a programmable number of consecutive samples indicate a low battery condition, the comparator control circuit triggers an interrupt signal to the interrupt output contact UP_INT. The default number of samples is 4, although the number of consecutive samples is programmable from 4 to 120.

[0439] After a programmable number of consecutive samples indicate low battery power and the aforementioned UP_INT is generated, the comparator control circuitry is configured to generate a signal that allows the ASIC to enter a low-power mode: the VDD regulator can be disabled and a low signal can be sent to the VPAD_EN contact. This can be referred to as a low battery state. Furthermore, the number of consecutive samples is programmable from 4 to 120 samples, with a default of 4 samples.

[0440] The comparator has separate programmable thresholds for lowering and raising the voltage on BATT_DIV. This is implemented in the digital logic to multiplex the two values ​​in the circuit based on the low battery state. Therefore, if the low battery state is low, the lower threshold is used, and if the low battery state is high, the higher threshold is used. Specifically, the comparator has 16 programmable thresholds, ranging from 1.22 to 1.645 ± 3%, where the DNL of the programmable thresholds is set to less than 0.2 LSB.

[0441] The comparator threshold varies by less than + / -1% between 20°C and 40°C. The default threshold for lowering the voltage is 1.44V (the VBAT threshold of the rated voltage divider is 3.41V), and the default threshold for raising the voltage is 1.53V (the VBAT threshold of the rated voltage divider is 3.63V). After the ASIC enters a low battery state, if the comparator detects four consecutive OK battery flags, the ASIC can initiate the microprocessor boot sequence.

[0442] Battery power board power-on reset

[0443] If the input VBAT swings greater than 1.2 volts within a 50-microsecond time period, or if the VBAT voltage is lower than 1.6 ± 0.3 volts, a power-on reset (POR) output is generated on the contact nPOR1_OUT (4240). This POR is extended into a minimum pulse with a width of 5 milliseconds. The output of the POR circuit is set to active low and goes to the contact nPOR1_OUT on the VBAT power board.

[0444] The IC has an input contact, nPOR1_IN (4242), for the battery power board POR. This input contact has RC filtering so that pulses shorter than 50 nanoseconds will not cause a logic reset. In this implementation, nPOR1_OUT is externally connected to nPOR1_IN during normal operation, thus separating the analog circuitry from the digital circuitry for testing. nPOR1_IN causes all logic on any power board to reset and initializes all registers to their default values. Therefore, the POR bit of the reset status register is set and all other reset status register bits are cleared. The POR reset circuit is configured to consume no more than 0.1uA of VBAT power for a period of more than 5 seconds after power-on.

[0445] VDD Power-On Reset (POR)

[0446] The ASIC also includes a voltage comparator circuit that generates a VDD voltage plate reset signal upon power-up or when VDD drops below a programmable threshold. Several voltage thresholds are programmable, with a default value of 1.8V-15% (1.53V). The POR2 has a programmable threshold for raising the voltage, which introduces hysteresis. This raise threshold is also programmable, with a default value of 1.60V ± 3%.

[0447] The POR signal is active low and has an output contact nPOR2_OUT(4244) on the VDD power board. The ASIC also has an active low POR open-drain output on the VBAT power board, nPOR2_OUT_OD(4246). This can be used to apply POR to other system components.

[0448] The VDD power supply logic has a POR derived from the input contact nPOR2_IN (4248). The nPOR2_IN contact is located on the VDD power supply board and has RC filtering so that pulses shorter than 50 nanoseconds will not cause a logic reset. nPOR2_OUT is configured to be externally connected to the nPOR2_IN input contact under normal operating conditions, thereby isolating the analog circuit from the digital circuit.

[0449] The resulting reset is extended to a valid time of at least 700 milliseconds after VDD rises above the programmable threshold, thus ensuring crystal oscillator stability. The POR reset circuit consumes no more than 0.1uA of VDD power for more than 5 seconds after power-on, and no more than 0.1uA of VBAT power for more than 5 seconds after power-on. The register storing the POR threshold is powered by the VDD power supply board.

[0450] Sensor interface electronics

[0451] In embodiments of the invention, in any combination of peroxide or oxygen sensors, the sensor circuitry supports up to five sensor work electrodes (4310), although a greater number of such electrodes may be used in other embodiments. The peroxide sensor work electrode draws a source current, while the oxygen sensor work electrode draws a sink current. For this embodiment, the sensor may be configured as follows: Figure 43 The constant potential configuration is shown.

[0452] The sensing electronics have programmable power control for each electrode interface circuit, minimizing current consumption by shutting off current flowing to unused sensing electronics. The sensor electronics also include electronics driving a counter electrode 4320 that uses feedback information from a reference electrode 4330. Current flowing to this circuit when not in use is programmable to save power. The interface electronics include a multiplexer 4250 so that the counter electrode and the reference electrode can be connected to either of the (redundant) work electrodes.

[0453] The ASIC is configured to provide the following sensor interfaces: (i) RE: a reference electrode that establishes a reference potential for the solution for setting the WORK voltage; (ii) WORK1-WORK5: sensor working electrodes in which the desired reduction / oxidation (redox) reaction occurs; and (iii) COUNTER: an output from this contact that maintains a known voltage on the RE electrode relative to the system VSS. In this embodiment of the invention, the ASIC is configured to individually set the WORK voltage of up to five WORK electrodes with a resolution and accuracy greater than or equal to 5 mV.

[0454] In oxygen mode, the WORK voltage is programmable between at least 0 and 1.22V relative to the VSSA. In peroxide mode, the WORK voltage is programmable between at least 0.6V and 2.054V relative to the VSSA. If VDDA is less than 2.15V, the WORK voltage can operate up to VDDA-0.1V. The ASIC includes current measurement circuitry for measuring the WORK electrode current in peroxide sensor mode. This can be achieved, for example, by a current-to-voltage or current-to-frequency converter, which may have the following specific characteristics: (i) current range: 0-300nA; (ii) voltage output range: same as the WORK electrode in peroxide / oxygen mode; (iii) output compensation voltage: ±5mV maximum; and (iv) uncalibrated resolution: ±.25nA.

[0455] After applying a calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0456] 5pA-1nA: ±3% ±20pA

[0457] 1nA-10nA: ±3% ±20pA

[0458] 10nA-300nA: ±3% ±.2nA

[0459] For the current-to-frequency converter (IF) only, the frequency range is 0Hz to 50kHz. In peroxide mode, the current converter must operate within a specific voltage range relative to the VSS of the WORK electrode. Here, for each converter, including the digital-to-analog (DAC) current, the current consumption to the 2.5V supply is less than 2uA and the WORK electrode current is less than 10nA.

[0460] The current converter can be enabled or disabled via software control. When disabled, the WORK electrode will exhibit a very high impedance value, i.e., greater than 100M ohms. Furthermore, for the IF converter only, the IF converter can output to a 32-bit counter, which can be read, written, and cleared by the microprocessor and test logic. During counter readings, the counter clock pauses to ensure accurate readings.

[0461] In embodiments of the invention, the ASIC further includes a current measurement circuit for measuring the WORK electrode current in oxygen sensor mode. The circuit can be implemented as a current-to-voltage or current-to-frequency converter, and programmable bits can be used to configure the current converter to operate in oxygen mode. As previously mentioned, in oxygen mode, the current converter must operate within a specific voltage range of the WORK electrode relative to VSS. Here, again, the current range is 3.7 pA to 300 nA, the voltage output range is the same as that of the WORK electrode in oxygen mode, the output compensation voltage is ±5 mV at most, and the uncalibrated resolution is 3.7 pA ± 2 pA.

[0462] After applying the calibration factor to the gain and assuming an acquisition time of 10 seconds or less, the current measurement accuracy is:

[0463] 5pA–1nA: ±3% ±20pA

[0464] 1nA–10nA: ±3% ±20pA

[0465] 10nA–300nA: ±3% ±.2nA

[0466] For the current-to-frequency converter (IF) only, the frequency range is 0Hz to 50kHz, and for each converter, including the DAC current, the current consumption to the 2.5V supply is less than 2uA and the WORK electrode current is less than 10nA. The current converter can be enabled or disabled via software control. When disabled, the WORK electrode can exhibit a very high impedance value, i.e., greater than 100M ohms. Furthermore, for the IF only, the IF converter output is sent to a 32-bit counter, which can be read, written, and cleared by a microprocessor and test logic. During counter readings, the counter clock is paused to ensure accurate readings.

[0467] In an embodiment of the invention, the input bias current of the reference electrode (RE) 4330 is less than 0.05 nA at 40°C. The COUNTER electrode adjusts its output to maintain the desired voltage on the RE electrode. This is accomplished by an amplifier 4340, whose output to the COUNTER electrode 4320 attempts to minimize the difference between the actual RE electrode voltage and the target RE voltage, which is set by a DAC.

[0468] The RE setting voltage is programmable and can be set between at least 0 and 1.80V, and the normal mode input range of the COUNTER amplifier includes at least .20 to (VDD - .20)V. Register bits can be used to select the normal mode input range (if needed) and to program the COUNTER's operating mode. The WORK voltage is set to a resolution and accuracy greater than or equal to 5mV. It should be noted that in normal mode, the COUNTER voltage seeks to maintain the RE voltage at the level of the programmed RE target value. However, in drive-relative mode, the COUNTER electrode voltage is driven to the programmed RE target voltage.

[0469] All electrode drive circuits are configured to drive electrodes to electrode loads without oscillation for any application scenario. Figure 44 An embodiment of the invention is shown. Figure 43 The equivalent AC circuit between the electrodes in the constant potential configuration shown. Figure 44 The equivalent circuit shown can be used between any electrodes (i.e., WORK1 to WORK5, COUNTER, and RE electrodes), and the range of values ​​for each circuit element is as follows:

[0470] Ru = [200-5k] Ohms

[0471] Cc = [10-2000]pF

[0472] Rpo = [1-20] k ohms

[0473] Rf = [200-2000] k ohms

[0474] Cf = [2-30]uF.

[0475] During initialization, the drive current for the WORK and COUNTER electrodes needs to be higher than that required for normal constant-potential operation. This allows the programmable register bits to be used to program the electrode drive circuitry to a higher power state if additional drive is needed. It is important to achieve low-power operation in normal constant-potential mode, where the electrode current is typically less than 300 nA.

[0476] In a preferred embodiment, during initialization, the WORK1 to WORK5 electrodes can be programmed from 0 to VDD volts in steps equal to or less than 5mV, and their drive or sink current output capability is a minimum of 20µA in the range from .20V to (VDD-.20V). Furthermore, during initialization, the ASIC is typically configured to measure up to 20µA of current from a single WORK electrode with an accuracy of ±2% ±40nA. Also, during initialization, the RE setting voltage can be programmed as described above, the COUNTER drive circuit output must be able to use the COUNTER electrode to draw or sink a minimum of 50µA of current in the range from .20V to (VDD-.20V), and the source current (VDD and VDDA) flowing to the initialization circuit must be less than 50µA above any provided output current.

[0477] Current calibrator

[0478] In embodiments of the invention, the ASIC has a reference current that can be applied to any work electrode for calibration purposes. In this regard, the calibrator includes a programmable bit that causes a current output to either sink or pull current. Assuming a zero-tolerance external precision resistor, the programmable current includes at least 10nA, 100nA, and 300nA, with an accuracy greater than ±1% ±1nA. The calibrator uses a 1-megaohm precision resistor connected to the contact TP_RES (4260) as a reference resistor. Furthermore, the reference current can be used for initialization and / or sensor state purposes on either the COUNTER or RE electrode. A constant current can be applied to either the COUNTER or RE electrode, and the electrode voltage can be measured by an ADC.

[0479] High-speed RC oscillator

[0480] Referring to Figure 42, the ASIC also includes a high-speed RC oscillator 4262, which provides an analog-to-digital converter (ADC) 4264, an ADC sequence generator 4266, and other digital functions requiring a high-speed clock above 32 kHz. The high-speed RC oscillator is phase-locked to a 32 kHz clock (32.768 kHz) to provide a programmable output frequency from 524.3 kHz to 1048 kHz. Furthermore, the high-speed RC oscillator features a duty cycle of 50% ± 10%, phase jitter of less than 0.5% rms, current consumption of less than 10 μA, and a stable frequency over the VDD operating range (a voltage range of 1.6 V to 2.5 V). The high-speed RC oscillator is "off" (i.e., disabled) by default, in which case current consumption is less than 10 nA. However, the ASIC has a programmable bit to enable the high-speed RC oscillator.

[0481] Analog-to-digital converter

[0482] The ASIC includes a 12-bit ADC (4264) with the following features: (i) the ability to affect the conversion in less than 1.5 milliseconds when operated by a 32kHz clock; (ii) the ability to perform fast conversion when timed by a high-speed RC oscillator; (iii) an accuracy of at least 10 bits (12-bit ± 4 counts); (iv) a 1.220V reference voltage input with a temperature sensitivity of less than 0.2mV / ℃ at 20°C to 40°C; (v) full-scale input ranges of 0 to 1.22V, 0 to 1.774V, 0 to 2.44V, and 0 to VDDA, wherein the 1.774V and 2.44V ranges have programmable bits that reduce the conversion range to lower values ​​to accommodate lower VDDA voltages; (vi) current consumption to the power supply of less than 50uA; (vi) a converter capable of operating by a 32kHz clock or a high-speed RC clock; (vii) a DNL of less than 1 LSB; and (viii) an interrupt signal at the end of the conversion.

[0483] like Figure 42A and 42B As shown, the ASIC has an analog multiplexer 4268 at the input of the ADC4264, both of which are software-controlled. In a preferred embodiment, at least the following signals are connected to the multiplexer:

[0484] (i) VDD – Core voltage and regulator output

[0485] (ii) VBAT – Battery Power

[0486] (iii) VDDA – Analog Power Supply

[0487] (iv) RE – Reference electrode of the sensor

[0488] (v)COUNTER – Counter electrode of the sensor

[0489] (vi)WORK1-WORK5—Working electrodes of the sensor

[0490] (vii) Temperature sensor

[0491] (viii) At least two external pins for analog signal input

[0492] (ix) EIS Integrated Output

[0493] (x)IV Current converter output.

[0494] For inputs COUNTER, RE, WORK1-WORK5, temperature sensor, and any other inputs susceptible to adverse load effects, the ASIC is configured such that the ADC load does not exceed ±0.01nA. The multiplexer includes a voltage divider and a buffer amplifier. The voltage divider is used for any input with a voltage higher than the ADC's input voltage range, and for load-sensitive inputs, the buffer amplifier reduces the input resistance of the split input to less than 1nA. The buffer amplifier further has a normal-mode input range of at least 0.8V to VDDA and a compensation of less than 3mV compared to an input range of 0.8V to VDDA-.1V.

[0495] In a preferred embodiment, the ASIC has a mode for acquiring ADC measurements in a programmed sequence. Therefore, the ASIC includes a programmable sequence generator 4266 that monitors measurements from up to eight ADC measurement input sources, having the following programmable parameters:

[0496] (i) ADC MUX input

[0497] (ii) ADC range

[0498] (iii) Measure the prior delay, wherein the delay can be programmed in a range of 0 to 62 msec with a pitch of 0.488 msec.

[0499] (iv) The number of measurements for each input is between 0 and 255.

[0500] (v) Number of measurement loops: 0-255, where a measurement loop refers to repeating the sequence of up to 8 input measurements multiple times (e.g., as an outer loop in a program).

[0501] (vi) Measure the delay between cycles, wherein the delay can be programmed in a range of 0 to 62 msec with a pitch of 0.488 msec.

[0502] The sequence generator 4266 is configured to start upon receiving an automatic measurement start command, and the measured values ​​can be stored in the ASIC for retrieval on the SPI interface. It should be noted that the sequence generator's time base can be designed between a 32kHz clock and a high-speed RC oscillator 4262.

[0503] Sensor diagnostic methods

[0504] As described in detail above, embodiments of the present invention relate to the application of impedance and impedance-related parameters in, for example, sensor diagnostic procedures and Isig / SG fusion algorithms. To this end, in a preferred embodiment, the ASIC described herein has the capability to measure the impedance amplitude and phase angle between any WORK sensor electrode and the RE and COUNTER electrodes in a constant-potential configuration. This is achieved, for example, by measuring the amplitude and phase of a sinusoidal current waveform that responds to the WORK electrode voltage. See, for example, Figure 42B The diagnostic circuit 4255 is included.

[0505] The ASIC has the capability to measure the resistive and capacitive components between any electrodes using, for example, an electrode multiplexer 4250. It should be noted that such measurements can be affected by sensor balancing and may require a set time or sensor initialization to record stable electrode currents. As discussed earlier, while the ASIC can be used for impedance measurements over a wide frequency range, a relatively narrow frequency range is suitable for the purposes of embodiments of the invention. Specifically, the ASIC's sinusoidal measurement capability can include test frequencies from about 0.10 Hz to about 8192 Hz. When performing these measurements, the minimum frequency resolution according to embodiments of the invention can be defined as shown in Table 2 below:

[0506] Table 2

[0507]

[0508] The sinusoidal wave amplitude can be programmed in 5mV increments between at least 10mVp-p and 50mVp-p, and in 10mV increments between 60mVp-p and 100mVp-p. In a preferred embodiment, the amplitude accuracy is greater than ±5% or ±5mV, whichever is greater. Furthermore, the ASIC can measure electrode impedance with accuracy specified in Table 3 below.

[0509]

[0510]

[0511] In embodiments of the invention, the ASIC can measure the phase of an input waveform relative to a time baseline, which can be used for impedance calculations to provide accuracy. The ASIC may also have on-chip resistors for calibrating the aforementioned electrode impedance circuitry. These on-chip resistors can then be calibrated by comparison with a known 1-megohm off-chip precision resistor.

[0512] The waveform data sampling can also be used to determine the impedance. The data can be transmitted to an external microprocessor with a Serial External Interface (SPI) for calibration and processing. The converted current data is adequately buffered to enable the transmission of 2000 ADC conversion data to external devices via the SPI interface without data loss. This assumes a maximum latency of 8 ms to accommodate interruptions in data transmission requests.

[0513] In embodiments of the invention, alternatively, or in addition to measuring the electrode impedance of a sinusoidal wave, the ASIC can measure the electrode current of a step input. Here, the ASIC can provide the electrode with a programmable amplitude pitch of 10 to 200 mV with a resolution greater than 5 mV and sample (measure) the resulting current waveform. The sampling time can be programmed to be at least 2 seconds, with a pitch of 0.25 seconds, and the sampling interval for measuring the current can include at least five programmable binary-weighted pitches, approximately 0.5 msec to 8 msec.

[0514] Electrode voltage sampling values ​​have a resolution of less than 1 mV and a range up to ±.25 volts. This measurement can be correlated with a suitable stable voltage to reduce the required data conversion dynamic range. Similarly, electrode current sampling values ​​have a resolution of less than .04 μA and a range up to 20 μA. If the measurement polarity is programmable, the current measurement can be unipolar.

[0515] In embodiments of the invention, current measurement can be performed using an IV converter. Furthermore, the ASIC can have an on-chip resistor for calibrating the current measurement. This on-chip resistor can then be calibrated by comparison with a known 1 megohm precision off-chip resistor. The current measurement sampling accuracy is greater than ±3% or ±10nA, whichever is greater. As previously stated, the converted current data can be sufficiently buffered to enable the transmission of 2000 ADC conversion data to external devices via the SPI interface without data loss. This assumes a maximum latency of 8 milliseconds for data transmission request interrupts.

[0516] Calibration voltage

[0517] The ASIC includes a precise reference voltage for calibrating the ADC. The output voltage is 1.000V ±3%, exhibiting a variation of less than ±1.5% during production and stability exceeding ±3mV over a temperature range of 20°C to 40°C. This precise calibration voltage can be calibrated during production via the ADC by comparing it to an external precision voltage. During production, the calibration factor can be stored in the system's nonvolatile memory (not on the ASIC) to achieve high accuracy.

[0518] The current consumption of the calibration voltage circuit is preferably less than 25uA. Moreover, when not in use, the calibration voltage circuit can reduce the power consumption to less than 10nA to conserve battery power.

[0519] Temperature sensor

[0520] The ASIC features a temperature sensor with a sensitivity of 9mV / ℃ to 11mV / ℃ over a temperature range of -10℃ to 60℃. The output voltage of the temperature sensor allows the ADC to measure temperature-dependent voltages within an ADC input range of 0 to 1.22V. The current consumption of the temperature sensor is preferably less than 25uA, and when not in use, the temperature sensor can reduce power consumption to less than 10nA to conserve battery power.

[0521] VDD voltage regulator

[0522] The ASIC has a VDD voltage regulator, which has the following characteristics:

[0523] (i) Minimum input voltage range: 2.0V–4.5V.

[0524] (ii) Minimum output voltage: 1.6-2.5V±5%, default 2.0V.

[0525] (iii) Voltage difference: Under the conditions of Iload (load current) = 100uA and Vin = 2.0V, Vin – Vout < 0.15V.

[0526] (iv) The output voltage is programmable with an accuracy within 2% of the values ​​shown in Table 4 below:

[0527] Table 4

[0528]

[0529]

[0530] (v) The regulator provides a 2.5V, 1mA output with an input voltage of 2.8V.

[0531] (vi) The regulator also has input and output contacts that can disconnect the circuit when using an external regulator. The current consumption of the regulator circuit is preferably less than 100 nA in this non-operating mode.

[0532] (vii) The variation in output voltage from a load ranging from 10uA to 1mA is preferably less than 25mV.

[0533] (viii) Except for the output current under a 1mA load, the current consumption of the power supply is less than 100uA.

[0534] (ix) Except for the output current under a 0.1mA load, the current consumption of the power supply is less than 10uA.

[0535] (x) Except for the output current under a 10uA load, the current consumption of the power supply is less than 1uA.

[0536] General Comparator

[0537] The ASIC includes at least two comparators 4270 and 4271 powered by VDDA. The comparators use 1.22V as a reference to generate thresholds. The comparator outputs are readable by the processor and can generate maskable interrupt signals on either a rising or falling edge, as determined by a configuration register.

[0538] The comparator features power control to reduce power when not in use, and the current supply is less than 50 nA per comparator. The comparator's response time is preferably less than 50 microseconds for a 20 mV overspeed signal, and the compensation voltage is less than ±8 mV.

[0539] The comparators also feature programmable hysteresis, with hysteresis options including threshold = 1.22V + Vhyst on the rising input, threshold = 1.22 - Vhyst on the falling input, or no hysteresis (Vhyst = 25 ± 10mV). The output from either comparator can be supplied to any GPIO on any power board (see the GPIO section).

[0540] The sensor on RE is connected to the detection circuit.

[0541] An analog switching capacitor circuit monitors the impedance of the RE connection to determine whether the sensor is connected. Specifically, an approximately 20pF capacitor is switched at a frequency of 16Hz driven by an inverter, with the output oscillating between VSS and VDD. A comparator detects the voltage oscillation on the RE contact; if the voltage oscillation is less than a threshold, the comparator output indicates a connection. This comparison can be performed over two pulse transitions. A oscillation below the threshold on both transitions indicates a connection, while a comparison showing a high oscillation on either phase indicates a disconnection. The connection / disconnection signal is debounced for at least 1 / 2 second so that its state transition is stable and indicates a new state.

[0542] The circuit has six threshold values ​​defined by the following resistors connected in parallel with a 20pF capacitor: 500kΩ, 1MΩ, 2MΩ, 4MΩ, 8MΩ, and 16MΩ. This parallel equivalent circuit is located between the RE contact and a virtual ground, which can be any voltage between power rails. The threshold accuracy is greater than ±30%.

[0543] The output of the sensor connection detection circuit can be programmably used to generate an interrupt signal (if the sensor is not connected) or a processor start signal (if the sensor is connected). The circuit is active regardless of whether nPOR2_IN is high or VDD and VDDA are present. The circuit's current consumption is less than 100nA on average.

[0544] WAKEUP (wake-up) contact

[0545] The WAKEUP circuit is powered by a VDD supply and has an input range from 0V to VBAT. The WAKEUP contact 4272 has a weak pull-down of 80±40nA. This current can be drawn from the output of the BIAS_GEN4220. The circuit consumes less than 50nA of average current at 0V input.

[0546] The WAKEUP input has a rising input voltage threshold, Vih, of 1.22 ± 0.1 V, and a falling input threshold of -25 mV ± 12 mV relative to the rising threshold. In a preferred embodiment, the circuitry associated with the WAKEUP input consumes no more than 100 nA for any input value from -0.2 to VBAT volts (excluding input pull-down current). The WAKEUP contacts are debounced for at least 1 / 2 second.

[0547] The output of the WAKEUP circuit can be programmably configured to generate an interrupt signal or a processor start signal if the WAKEUP contact changes state (see the Event Handler section). Importantly, it should be noted that if the battery protection circuit indicates a low battery state, the WAKEUP contact circuit is configured to use a low current of <1nA.

[0548] UART WAKEUP

[0549] The ASIC is configured to monitor the nRX_EXT contact 4274. If the nRX_EXT level remains high (UART interrupt) for longer than 1 / 2 second, a UARTWAKEUP event can be generated. This is because a sustained high-sampled UARTWAKEUP event can be generated for as short as 1 / 4 second. The UARTWAKEUP event is programmable to generate an interrupt signal, a WAKEUP signal, and / or a microprocessor reset signal (nRESET_OD). (See the Event Handler section).

[0550] In a preferred embodiment, the circuitry associated with the UARTWAKEUP input consumes no more than 100nA, and the UARTWAKEUP contact circuitry is configured to operate at a low current of <1nA if the battery protection circuitry indicates a low battery state. The UART WAKEUP input has a rising input voltage threshold Vih of 1.22±0.1V. The falling input threshold is -25mV±12mV relative to the rising threshold.

[0551] Microprocessor WAKEUP control signal

[0552] ASICs can generate signals to help control the power management of microprocessors. Specifically, ASICs can generate the following signals:

[0553] (i) nSHUTDN controls the power supply that enables the external VDD regulator. The nSHUTDN contact is located on the VBAT power rail. If the battery protection circuit indicates a low battery level, then nSHUTDN is low; otherwise, nSHUTDN is high.

[0554] (ii) VPAD_EN – VPAD_EN controls the power supply to the external regulator that provides power to the VPAD. An internal signal responding to this external signal ensures that inputs from the VPAD contact do not draw additional current due to the floating input when VPAD power is disabled. The VPAD_EN contact is the output on the VBAT power rail. The VPAD_EN signal is low if the battery protection signal indicates low battery. The VPAD_EN signal can be set low by a software instruction to start a timer; the timer stopping the count causes VPAD_EN to decrease. If the battery protection signal indicates a good battery, the following events can cause the VPAD_EN signal to rise (see the Event Handlers section for a more detailed description): nPOR2_IN transitioning from low to high; SW / timer (programmable); WAKEUP transition; low-to-high and / or high-to-low (programmable); sensor connection transition; low-to-high and / or high-to-low (programmable); UART interrupt; and RTC time events (programmable).

[0555] (iii) UP_WAKEUP - UP_WAKEUP can be connected to the microprocessor wake-up contact. It is intended to wake the microprocessor from sleep mode or a similar low-power mode. The UP_WAKEUP contact is an output on the VPAD power rail. The UP_WAKEUP signal can be programmed to be low-active, high-active, or pulsed. The UP_WAKEUP signal can be set low by a software instruction to start a timer; the timer stops counting, driving UP_WAKEUP down. If the battery protection signal indicates a good battery, the following events can cause the UP_WAKEUP signal to rise (see the Event Handlers section for a more detailed description): nPOR2_IN transitions from low to high, SW / timer (programmable); WAKEUP transition; low to high, and / or high to low (programmable); sensor connection transition; low to high, and / or high to low (programmable); UART interrupt; and RTC time events (programmable). The WAKEUP signal can be delayed by a programmable amount. If WAKEUP is programmed as a pulse, the pulse width is programmed.

[0556] (iv) The CLK_32kHz contact can be connected to a microprocessor to provide a low-speed clock. This clock is on-off programmable and programmably enabled until a wake-up event occurs. The CLK_32kHz contact is an output on the VPAD power rail. The CLK_32kHz signal is low if the battery protection signal indicates low battery. The CLK_32kHz output can be programmed off via a programmable bit. The default is ON. The CLK_32kHz signal can be disabled by a software instruction to start a timer. Stopping the timer's count causes CLK_32kHz to decrease. If the battery protection signal indicates that the battery is good, the following events may enable the CLK_32KHZ signal (see the Event Handler section for a more detailed description): nPOR2_IN transition from low to high; SW / timer (programmable); WAKEUP transition; low to high, and / or high to low (programmable); sensor connection transition; low to high, and / or high to low (programmable); UART interrupt; RTC time event (programmable); and battery protection circuitry detecting battery power.

[0557] (v) nRESET_OD - nRESET_OD can be connected to a microprocessor to reset the microprocessor. nRESET_OD is programmable as a wake-up event. The nRESET_OD contact is the output on the VPAD power rail. This contact is open-drain (nFET output). If the battery protection signal indicates low battery, then the nRESET_OD signal is low. The nRESET_OD active time is programmable from 1 ms to 200 ms. The default is 200 ms. The following events can cause the nRESET_OD signal to be activated low (see the Event Controllers section for a more detailed description): nPOR2_IN; SW / timer (programmable); WAKEUP transition; low to high, and / or high to low (programmable); sensor connection transition; low to high, and / or high to low (programmable); UART interrupt; and RTC time event (programmable).

[0558] (vi) UP_INT - UP_INT can be connected to a microprocessor to transmit interrupt signals. UP_INT is programmable as a wake-up event. The UP_INT contact is an output on the VPAD power rail. If the battery protection signal indicates low battery, then the UP_INT signal is low. The UP_INT signal can be set high by a software instruction to start a timer; the timer's stop count drives UP_INT to rise. If the battery protection signal indicates good battery, then the following events can cause the UP_INT signal to be activated high (see the Event Controllers section for a more detailed description): SW / timer (programmable); WAKEUP transition; low to high, and / or high to low, (programmable); sensor connection transition; low to high and / or high to low, (programmable); UART interrupt; RTC time event (programmable); low battery detection by the battery protection circuitry; and any ASIC interrupt signal when not masked.

[0559] The ASIC has GPIO1 and GPIO0 contacts that act as bootmode controls for the microprocessor. A POR2 event resets a 2-bit counter whose bits are mapped to GPIO1 and GPIO0 (MSB and LSB, respectively). A rising edge of the UART interrupt signal increments the counter, which has a modulus of 4, and resets to zero if the count increments in state 11. The bootmode counter can be preset via SPI.

[0560] Event handlers / monitors

[0561] The ASIC incorporates an event handler to define responses to events, including changes in system state and input signals. Events include all interrupt sources (e.g., UART_BRK, WAKE_UP, sensor connections, etc.). The event handler responses to stimuli are programmed in software via an SPI interface. However, some responses can be fixed (not programmable).

[0562] The event handler functions include enabling / disabling VPAD_EN, enabling / disabling CLK_32kHz, activating nRESET_OD, activating UP_WAKEUP, and activating UP_INT. Event monitor timers 1 through 5 can be independently programmed in 250ms increments from 250ms to 16,384 seconds. The timeout settings for event monitor timers 6 through 8 are hard-coded. Timers 6 and 7 have a timeout of 1 minute, and timer 8 has a timeout of 5 minutes.

[0563] The ASIC also features a monitoring function that monitors the microprocessor's response when an event is triggered. The event monitor can be activated if the microprocessor fails to respond to an event-induced activity. Once activated, the event monitor executes a programmable series of actions for event monitor timers 1 through 5, followed by a series of fixed actions for event monitor timers 6 through 8, to obtain a response from the microprocessor again. The series of actions includes interrupting, resetting, waking up, sending a 32kHz clock, and shutting down and starting the microprocessor.

[0564] During the series of actions, if the microprocessor regains its ability to acknowledge recorded activity, the event monitor is reset. If the ASIC fails to receive an acknowledgment from the microprocessor, the event monitor shuts down the microprocessor and activates an alarm, allowing UART_BRK to restart the microprocessor. When the alarm is activated, the alarm state generates a square wave at approximately 1 kHz on the contact ALARM in a programmable repeating pattern. The programmable pattern has two programmable sequences with programmable burst on and off times. The alarm also has another programmable pattern, which can be programmed via the SPI port. This other programmable pattern has two programmable sequences with programmable burst on and off times.

[0565] Digital-to-Analog (D / A)

[0566] In a preferred embodiment, the ASIC has two 8-bit D / A converters 4276 and 4278, which have the following features:

[0567] (i) The D / A is regulated in less than 1 msec with a load of less than 50 pF.

[0568] (ii) The D / A converter has at least 8 bits of precision.

[0569] (iii) The programmable output range is 0 to 1.22V or 0 to VDDA.

[0570] (iv) The temperature sensitivity of the D / A reference voltage is less than 1 mV / ℃.

[0571] (v) DNL is less than 1 LSB.

[0572] (vi) The current consumption of the D / A on the VDDA power supply is less than 2uA.

[0573] (vii) Each D / A has an output 1 to the contact.

[0574] (viii) The output of the D / A converter has high impedance. The load current must be less than 1 nA.

[0575] (ix) The D / A contacts can be programmable to output digital signals from registers. The output swing is from VSSA to VDDA.

[0576] Charger / Data Download Interface

[0577] The TX_EXT_OD4280 is an open-drain output, whose input is the signal on the TX_UP input contact. This allows the TX_EXT_OD contact to be open in the UART idle state. The TX_EXT_OD contact has a comparator that monitors its voltage. If the voltage is higher than the comparator's threshold voltage for the duration of debouncing (1 / 4 second), the output nBAT_CHRG_EN (4281) will decrease. This comparator and other related circuitry with this function are located on the VBAT and / or VDDBU boards.

[0578] The circuitry associated with this function must generate a low level on the TX_EXT_OD contact without disabling the validity of nBAT_CHRG_EN, due to normal communication with external devices. If POR1 is valid, then nBAT_CHRG_EN will be high (inactive). The comparator's threshold voltage ranges from 0.50V to 1.2V. The comparator may have hysteresis. Lowering the threshold is approximately 25mV lower than raising the threshold.

[0579] The nRX_EXT contact inverts the signal on it and outputs it to RX_UP. This allows the nRX_EXT signal to decrease slowly. nRX_EXT must accept input voltages up to VBAT. The nRX_EXT threshold is 1.22V ± 3%. The comparator's output can be acquired via the SPI bus for microprocessor reading.

[0580] The nRX_EXT contact also incorporates a programmable current acquisition method, which can be 80±30nA with a maximum voltage VBAT. The ASIC arrangement has a shielded programmable option to regulate the current from 30nA to 200nA in pitches of less than 50nA, with minimal shielding variation. A programmable bit can be used to block UART interrupt detection and raise RX_UP. During normal operation, this bit can be set high before current acquisition to nRX_EXT is initiated and low after current acquisition is disabled to ensure no minor faults or UART interrupt events occur on RX_UP. It should be noted that with a wetted connection detector, although current acquisition into nRX_EXT is valid, the RX comparator output, indicating a low input voltage, can indicate leakage current. The ASIC includes an approximately 100k ohm pull-down resistor on the nRX_EXT contact. This pull-down resistor is not connected when current acquisition is valid.

[0581] Sensor connection switch

[0582] The ASIC may include a contact SEN_CONN_SW (4282) capable of detecting low resistance to VSS (4284). SEN_CONN_SW draws a current of 5 to 25 µA at SEN_CONN_SW = 0V and has a maximum open-circuit voltage of 0.4V. The ASIC arrangement features a programmable shielding option to adjust this current from 1 µA to 20 µA in pitches of less than 5 µA, with minimal shielding variation. SEN_CONN_SW includes associated circuitry to detect the presence of resistance between SEN_CONN_SW and VSSA (4234) with a threshold value of 2 k ohms to 15 k ohms. The circuitry draws a maximum average current of 50 nA. Sampling must be used to achieve this low current.

[0583] Oscillator calibration circuit

[0584] The ASIC has counters whose inputs can be booted to an internal or external clock source. One counter generates a programmable gating interval for use with other counters. Depending on the 32kHz oscillator, the gating interval can range from 1 second to 15 seconds. Clocks bootable to any counter can be 32kHz, an RC oscillator, a high-speed RC oscillator, or inputs from any GPIO contact.

[0585] Oscillator bypassing

[0586] The ASIC can use an external clock to replace each of the oscillator's outputs. The ASIC has a register that can be written to only when a specific TEST_MODE is active. This register contains bits that enable the external inputs to the RC oscillator and can be shared with other analog detection control signals. However, if TEST_MODE is not active, this register will not enable any oscillator bypass bits.

[0587] The ASIC also features an input contact for an external clock to bypass the RC oscillator. The GPIO_VBAT contact is located on the VBAT power board. The ASIC also includes a bypass enable contact OSC32K_BYPASS for the 32kHz oscillator. When OSC32K_BYPASS is high, the 32kHz oscillator output is provided by driving the OSC32kHz_IN contact. It should be noted that the OSC32kHz_IN contact is typically connected to a crystal.

[0588] The ASIC has an input for an external clock to bypass HS_RC_OSC. The bypass is enabled via a programmable register bit. HS_RC_OSC can be programmably powered via GPIO on the VDD board or via GPIO on the VPAD board.

[0589] SPI slave port

[0590] The SPI slave port consists of an interface comprised of a chip select input (SPI_nCS) 4289, a clock input (SPI_CK) 4286, a serial data input (SPI_MOSI) 4287, and a serial data output (SPI_MISO) 4288. The chip select input (SPI_nCS) is active low and is activated by an external SPI master to start and limit SPI transactions. When SPI_nCS is active low, the SPI slave port configures itself as an SPI slave and processes data based on the clock input (SPI_CK). When SPI_nCS is inactive, the SPI slave port resets itself and remains in reset mode. Because this SPI interface supports block transfers, the master should maintain the SPI_nCS level until the transfer is complete.

[0591] The SPI clock input (SPI_CK) is always active through the SPI master. The SPI slave port uses the rising edge of SPI_CK to latch input data on the SPI_MOSI input and uses the falling edge of SPI_CK to drive output data on the SPI_MISO output. The serial data input (SPI_MOSI) is used to transfer data from the SPI master to the SPI slave. All data bits are active after the falling edge of SPI_CK. The serial data output (SPI_MISO) is used to transfer data from the SPI slave to the SPI master. All data bits are active after the falling edge of SPI_CK.

[0592] SPI_nCS, SPI_CK, and SPI_MOSI are typically driven by the SPI master unless the SPI master is disabled. If VPAD_EN is low, these inputs are regulated so that the current consumption associated with these inputs is less than 10nA and the SPI circuitry remains reset or inactive. SPI_MISO is only driven by the SPI slave port when SPI_nCS is active; otherwise, SPI_MISO is tri-state.

[0593] The Chip Select (SPI_nCS) function defines and constructs the data transmission packet for SPI data processing. The data transmission packet consists of three parts: a 4-bit command portion, followed by a 12-bit address portion, and then any number of 8-bit data bytes. Command bit 3 is used as an indicator bit. "1" indicates a write operation, and "0" indicates a read operation. Combinations of command bits 2, 1, and 0 have the following definitions. Unused combinations are not defined.

[0594] (i)0000: Read data and increment address

[0595] (ii)0001: Read data without changing the address

[0596] (iii)0010: Read data, reduce address

[0597] (iv)1000: Write data and increment address

[0598] (v)1001: Write data without changing the address.

[0599] (vi)1010: Write data without incrementing the address.

[0600] (vii)x011: Test port addressing

[0601] The 12-bit address portion defines the starting byte address. If SPI_nCS remains valid after the first data byte to indicate a multi-byte transfer, the address is incremented by one after each byte is transferred. The address bits... <11> (Address <11:0>) represents the highest address bit. The address wraps around after reaching the boundary.

[0602] Data is in byte form and can be transferred in blocks by extending SPI_nCS so that all bytes are transmitted as a single data packet.

[0603] Microprocessor interrupt

[0604] The ASIC features a VPAD logic level output, UP_INT, used to send interrupt signals to the main microprocessor. The microprocessor interrupt module consists of an interrupt status register, an interrupt mask register, and the ability to convert all interrupt statuses into a single microprocessor interrupt using a logical OR operation. Interrupt execution supports both edge-sensitive and level-sensitive modes. The interrupt signal polarity is programmable. The default interrupt polarity is TBD.

[0605] In a preferred embodiment, all interrupt signal sources on the AFE ASIC can be recorded in the interrupt status register. Writing a "1" to the corresponding interrupt status bit clears the corresponding waiting interrupt signal. All interrupt signal sources on the AFE ASIC can be masked via the interrupt mask register. Writing a "1" to the corresponding interrupt signal mask bit enables the masking of the corresponding waiting interrupt signal. Writing a "0" to the corresponding interrupt signal mask bit disables the masking of the corresponding interrupt signal. The default state of the interrupt signal mask register is TBD.

[0606] General Purpose Input / Output (GPIO) / Parallel Test Port

[0607] In embodiments of the present invention, the ASIC may have eight GPIOs that operate at the VPAD level. The ASIC has one GPIO that operates at the VBAT level and one GPIO that operates at the VDD level. All GPIOs have at least the following characteristics:

[0608] (i) Register bits control the selection and indication of each GPIO.

[0609] (ii) The ASIC has a way to configure GPIO as inputs that can be read through the SPI interface.

[0610] (iii) The ASIC has a way to configure GPIO as an input to generate an interrupt signal.

[0611] (iv) The ASIC has the ability to configure each GPIO as an output controlled by a register bit that can be written via the SPI interface.

[0612] (v) Programmably, the ASIC can output input signals applied to GPIO_VBAT or GPIO_VDD to GPIO (VPAD power board). (Level shifting function).

[0613] (vi) The ASIC has a way of configuring each GIPO as an input to the oscillator calibration circuit.

[0614] (vii) The ASIC has a way of configuring the output of each general-purpose comparator to at least one GIPO on each power board. The polarity of the comparator output can be programmed via programmable bits.

[0615] (viii) GPIO has the ability to generate microprocessor interrupt signals.

[0616] (ix)GPIO is programmable to open-drain output.

[0617] (x) The GPIO on the VPAD power board can be configured to implement microprocessor boot control.

[0618] The parallel test port shares an 8-bit GPIO on the VPAD voltage board. This test port can be used to observe register contents and various internal signals. In normal mode, the port's output is controlled by the port configuration register. Writing 8'hFF to the GPIO_O1S_REG & GPIO_O2S_REG registers enables test port data on the GPIO output, while writing 8'h00 to the GPIO_ON_REG register disables test port data and enables GPIO data on the GPIO output.

[0619] The registers and pre-grouped internal signals can be observed through this test port by addressing the target register via SPI. The SPI packet has a command bit set to 4'b0011, followed by the 12-bit target register address. The parallel test port continuously displays the contents of the address register until the next test port addressing command is received.

[0620] Simulate test port

[0621] The IC features a multiplexer feed contact TP_ANAMUX (4290) that provides visibility into the internal analog circuit nodes for testing. The IC also features a multiplexer feed contact TP_RES (4260) that provides visibility into the internal analog circuit nodes for testing. These contacts can also accommodate precision 1MΩ resistors for various system calibrations in common applications.

[0622] Chip ID

[0623] The ASIC includes a 32-bit masked programmable ID. The microprocessor can read this ID using an SPI interface. This ID is placed within the analog electronic device block so that changes to the ID do not require chip rerouting. The design should require only a change in one metal element or contact mask to alter the ID.

[0624] Alternate test output

[0625] The ASIC has 16 spare digital output signals that can be multiplexed to 8-bit GPIOs under commands sent via the SPI interface. These signals can be organized into two 8-bit bytes and can be communicated with the VSS if not in use.

[0626] Digital Test

[0627] The ASIC has a test mode controller that uses two input pins, TEST_CTL0 (4291) and TEST_CTL1 (4292). The test controller generates signals by combining test control signals, which have the following functionalities (TEST_CTL<1:0>):

[0628] (i) 0 is the normal operating mode;

[0629] (ii)1 is the simulation test mode;

[0630] (iii)2 is the scanning mode;

[0631] (iv)3 is a simulation test mode with VDD_EN controlled by the input of GPIO_VBAT.

[0632] The test controller logic is separated between the VDD and VDDBU power boards. During scan mode, the test LT_VBAT should be active high to adjust the analog output to suit the digital logic. The ASIC has a scan chain that executes across as many digital logic components as possible for fast digital testing.

[0633] Leakage test pin

[0634] The ASIC has a pin called LT_VBAT, which, when high, puts all analog blocks into a non-enabled mode, so that only leakage current is drawn from the power supply. LT_VBAT causes all digital outputs from the analog blocks to be in a stable high or low state, thus not affecting the interface logic current consumption. The LT_VBAT contact is located on the VBAT board and has pull-down resistors between 10k ohms and 40k ohms.

[0635] Electricity demand

[0636] In embodiments of the invention, the ASIC includes a low-power mode in which, to a minimum, the microprocessor clock is off, a 32kHz real-time clock runs, and activation circuitry detects sensor connections, level changes on the WAKE_UP pin, or nRX_EXT input break (interrupt). This mode consumes a maximum total current of 4.0uA for VBAT (VDDBU), VDD, and VDDA. When the battery protection circuitry detects low battery power (see description of the battery protection circuitry), the ASIC enters a mode where only the VBAT and VDDBU power supplies are active. This is referred to as the low battery state. The VBAT current in this mode is less than 0.3uA.

[0637] With the ASIC programmed to the following constant-potential configuration, the average current consumption of all power supplies is less than 7uA, wherein the constant-potential configuration is as follows: any one WORK electrode is active in H2O2 (peroxide) mode with its voltage set to 1.535V; the COUNTER amplifier is powered on with VSET_RE set to 1.00V; a 20MEG load resistor is connected between WORK and COUNTER; COUNTER and RE are connected together; and it is assumed that the current of one working electrode is measured every minute. The current measured after calibration should be 26.75nA ± 3%. Activating an additional working electrode increases the combined current consumption by less than 2uA, where the current of the working electrode is 25nA.

[0638] With regard to ASIC programming for a constant potential configuration, wherein the diagnostic function is initiated to measure the impedance relative to one of the work electrodes of the COUNTER electrode, the ASIC is configured to meet the following characteristics:

[0639] (i) Test frequencies: 0.1, 0.2, 0.3, 0.5 Hz, 1.0, 2.0, 5.0, 10, 100, 1000 and 4000 Hz.

[0640] (ii) The measurement of the above frequencies shall not exceed 50 seconds.

[0641] (iii) The total charge supplied to the ASIC is less than 8 millicos.

[0642] environment

[0643] In a preferred embodiment of the present invention, ASIC:

[0644] (i) Operate and comply with all regulations within the commercial temperature range of 0°C to 70°C.

[0645] (ii) It functions within the range of -20°C to 80°C, but its accuracy decreases.

[0646] (iii) Expected to operate after storage in a temperature range of -30°C to 80°C.

[0647] (iv) Expected to operate in a relative humidity range of 1% to 95%.

[0648] (v) ESD protection is above ±2KV. Unless otherwise specified, when packaged in a TBD package, the Human Body Model is located on all pins.

[0649] (vi) Configure WORK1-WORK5, COUNTER, RE, TX_EXT_OD, and nRX_EXT contacts to withstand human body modes higher than ±4KV.

[0650] (vii) Configured so that the leakage current of WORK1-WORK5 and RE contacts is less than 0.50 nA at 40°C.

[0651] In an embodiment of the present invention, the ASIC can be manufactured using a .25-micron CMOS process and the spare data of the ASIC is located on a DVD 916-TBD.

[0652] As described in detail above, the ASIC provides the necessary analog electronics to: (i) support various potentiometers and connect to oxygen- or peroxide-based multi-terminal glucose sensors; (ii) connect to a microprocessor to form a low-power sensor system; and (iii) perform EIS diagnostics based on measurements of EIS-based parameters. The measurement and calibration of EIS-based parameters are described herein according to embodiments of the invention.

[0653] As mentioned above, impedance in the frequency range of 0.1 Hz to 8 kHz previously provided information as the state of the sensor electrodes. The AFE IC circuit incorporates circuitry for generating the measurement drive signal and for performing the measurement, which is used to calculate the impedance. Design considerations for this circuit include current consumption, accuracy, measurement speed, processing requirements, and the amount of time required to control the microprocessor.

[0654] In a preferred embodiment of the invention, the technique used by the AFE IC to measure electrode impedance involves superimposing a sinusoidal voltage onto the DC voltage of the driving electrode and measuring the phase and amplitude of the resulting AC current. To generate the sinusoidal wave, a digitally synthesized sinusoidal current is incorporated into the AFE IC. This digital technique is used because the frequency and phase can be precisely controlled via a time baseline generated by a crystal, and it can easily generate frequencies from DC to 8 kHz. The sinusoidal current is applied through a resistor connected in series with a voltage source, thereby adding the AC component to the electrode voltage. This voltage is the AC driving voltage. This voltage is then buffered by an amplifier that drives the selected sensor electrode.

[0655] The current at the driving electrode contains the AC current component derived from the driving sine wave and is converted into a voltage. This voltage is then processed by multiplying it with a square wave, which has a fixed phase relative to the synthesized sine wave. This multiplied voltage is then integrated. After a programmable number of integration intervals—each interval being the number of integrations over half a cycle of the driving sine wave—the voltage is measured by an ADC. The real and imaginary parts of the impedance can be obtained by ca...

Claims

1. A method for performing diagnostics on a subcutaneous or implanted sensor having at least one working electrode, comprising: Define a vector that contains values ​​associated with one or more parameters based on electrochemical impedance spectroscopy (EIS) and one or more non-EIS parameters, wherein the one or more EIS-based parameters include at least one of membrane resistance (Rmem), membrane capacitance (Cmem), and double-layer capacitance (Cdl). For each of the EIS-based parameters and each of the non-EIS-based parameters, a corresponding threshold is defined; Perform a first EIS detection process to generate a first set of data relating to one or more EIS-based parameters; After a predetermined time interval, a second EIS detection process is performed to generate a second set of data related to the values ​​of one or more EIS-based parameters; Update the vector using the first set of data and the second set of data; as well as Monitor the vector value and determine whether the sensor has lost sensitivity based on a comparison of the vector value with the corresponding threshold.

2. The method according to claim 1, wherein, The one or more non-EIS-based parameters include at least one of the variability of the sensor current Isig, the rate of change of Isig, and the time period during which Isig is low.

3. The method according to claim 1, further comprising: The vector values ​​of each of the non-EIS-based parameters are updated periodically.

4. The method according to claim 3, wherein, Sensitivity loss is determined by comparing one of the monitored vector values ​​after updating the vector using the first set of data and the second set of data with the corresponding threshold.

5. The method according to claim 3, wherein, When multiple vector values ​​exceed their corresponding thresholds, it is determined that the sensor has lost sensitivity.

6. The method according to claim 5, wherein, The plurality of vector values ​​includes at least one EIS-based parameter value and at least one non-EIS-based parameter value.

7. The method according to claim 1, wherein, Each of the thresholds is an absolute threshold.

8. The method according to claim 1, wherein, Each of the thresholds is a relative threshold.

9. The method of claim 1, further comprising a second threshold for the combination of the EIS-based parameters, wherein, If the combined vector value of the EIS-based parameters exceeds the second threshold, it is determined that the sensor has lost sensitivity.

10. The method of claim 1, further comprising a second threshold for the combination of the EIS-based parameters and the non-EIS-based parameters, wherein, If the combined vector value of the EIS-based parameters and the non-EIS-based parameters exceeds the second threshold, it is determined that the sensor has lost sensitivity.

11. The method of claim 1, further comprising: A weighting coefficient is applied to each of the vector values ​​to produce a weighted vector value, and the sensitivity loss is determined based on the weighted vector value.

12. The method according to claim 11, wherein, The weighting coefficients are different for each of the EIS-based parameters and for each of the non-EIS-based parameters.

13. The method according to claim 12, wherein, For each of the EIS-based parameters and each of the non-EIS-based parameters, a weighting coefficient is calculated based on the respective threshold of each parameter.

14. The method of claim 13, further comprising a second threshold for the combination of the EIS-based parameters, wherein, If the combined weighted vector value of the EIS-based parameters exceeds the second threshold, it is determined that the sensor has lost sensitivity.

15. The method according to claim 14, wherein, The second threshold is an absolute threshold.

16. The method of claim 14, wherein, The second threshold is a relative threshold.

17. The method of claim 13, further comprising a second threshold for the combination of the EIS-based parameters and the non-EIS-based parameters, wherein, If the combined weighted vector value of the EIS-based parameters and the non-EIS-based parameters exceeds the second threshold, it is determined that the sensor has lost sensitivity.

18. The method according to claim 17, wherein, The second threshold is an absolute threshold.

19. The method according to claim 17, wherein, The second threshold is a relative threshold.

Images