An isolated power converter

Abstract

The embodiment of the present application discloses an isolated power converter. In the embodiment of the present application, the isolated power converter comprises a primary side circuit, a transformer and a secondary side circuit, wherein the secondary side circuit comprises a current conversion circuit and a feedback control circuit, the current conversion circuit excites the secondary side leakage inductance of the secondary side winding via a secondary side switch component in a secondary side switch network to realize current conversion, and the feedback control circuit is configured to control the switching state of the secondary side switch component in the secondary side switch network according to an output feedback signal representing an output signal. Thus, the embodiment can directly adjust the secondary side switch component in the secondary side switch network, so that the secondary side leakage inductance can serve as a power storage element, realize magnetic integration of the transformer, and thus can eliminate the feedback signal path of the primary side and the secondary side, reduce the control complexity and manufacturing cost.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of electronic power technology, and more particularly, to an isolated power converter. BACKGROUND

[0002] High frequency transformer is indispensable in the application of converter which needs electrical isolation. The traditional high frequency transformer has a relatively high coupling coefficient (k>0.9) because of the high magnetic permeability magnetic core which can bind the magnetic force line distribution of the winding in physical structure, and the leakage inductance between the primary and secondary sides is also small. However, in some specific applications, such as wireless charging or high power density module power supply, etc., because the physical position of the primary and secondary winding of the transformer is relatively fixed, or the loss caused by the magnetic core material is too large to be acceptable at a super high switching frequency, the method of adding magnetic core elements to the primary and secondary sides of the transformer to achieve high coupling coefficient cannot be continued. After removing the magnetic core, the transformer is loosely coupled, and the coupling coefficient will be greatly reduced (k<0.5), at this time the leakage inductance of the primary and secondary sides will increase sharply, and in the order of magnitude, it is equal to the magnetizing inductance of the transformer. The increased leakage inductance will cause serious problems to the power electronic converter, on the one hand, it may cause the voltage stress of the components to rise, on the other hand, it will worsen the output characteristics of the converter, causing the conversion efficiency to decrease, so it needs to be eliminated or compensated at the circuit and control level. SUMMARY

[0003] Therefore, the embodiment of the present application provides an isolated power converter, which directly adjusts the secondary side switching component, so that the secondary side leakage inductance can be used as a power storage element, realizes the magnetic integration of the transformer, and thus can eliminate the feedback signal path of the primary and secondary sides, reduces the control complexity and manufacturing cost.

[0004] In a first aspect, the embodiment of the present application provides an isolated power converter, which comprises:

[0005] a primary side circuit configured to generate an alternating current input signal;

[0006] a transformer comprising a primary side winding and a secondary side winding, configured to process the alternating current input signal to generate a secondary side input signal;

[0007] a secondary side circuit comprising a conversion circuit and a feedback control circuit, the conversion circuit is configured to convert the secondary side input signal to generate an output signal; wherein the conversion circuit magnetizes the secondary side leakage inductance of the secondary side winding via a secondary side switching component in a secondary side switching network to realize the conversion, and the feedback control circuit is configured to control the switching state of the secondary side switching component in the secondary side switching network according to an output feedback signal representing the output signal.

[0008] In the embodiment of the present application, the isolated power converter comprises a primary circuit, a transformer and a secondary circuit, wherein the secondary circuit comprises a current conversion circuit and a feedback control circuit, the current conversion circuit magnetizes the secondary leakage inductance of the secondary winding via a secondary switch component in a secondary switch network to realize current conversion, and the feedback control circuit is configured to control the switching state of the secondary switch component in the secondary switch network according to an output feedback signal representing an output signal. Thus, the embodiment can directly adjust the secondary switch component in the secondary switch network, so that the secondary leakage inductance can serve as a power storage element, realize magnetic integration of the transformer, and thus can eliminate the feedback signal path between the primary and secondary sides, reduce the control complexity and manufacturing cost. BRIEF DESCRIPTION OF DRAWINGS

[0009] The above and other objects, features and advantages of the present application will become more apparent from the following description of the embodiments of the present application taken with reference to the accompanying drawings, in which:

[0010] Figure 1 is a circuit diagram of the isolated power converter of the embodiment of the present application;

[0011] Figure 2 is a schematic diagram of the secondary circuit of the isolated power converter of the first embodiment of the present application;

[0012] Figure 3 is a schematic diagram of the secondary circuit of the isolated power converter of the second embodiment of the present application;

[0013] Figure 4 is a schematic diagram of the driving signal of the secondary switch component of the second embodiment of the present application;

[0014] Figures 5-8 is an equivalent circuit diagram of the secondary circuit of the second embodiment of the present application;

[0015] Figure 9 is a schematic diagram of the secondary circuit of the third embodiment of the present application;

[0016] Figure 10 is a schematic diagram of the driving signal of the secondary switch component of the third embodiment of the present application;

[0017] Figures 11-12 is an equivalent circuit diagram of the secondary circuit of the third embodiment of the present application;

[0018] Figure 13 is a schematic diagram of the secondary circuit of the fourth embodiment of the present application;

[0019] Figure 14 is a schematic diagram of the driving signal of the secondary switch component of the fourth embodiment of the present application;

[0020] Figure 15 is a schematic diagram of the secondary circuit of the fifth embodiment of the present application;

[0021] Figure 16 is a schematic diagram of the driving signal of the secondary side switch assembly of the embodiment five of the present application;

[0022] Figure 17 is a schematic diagram of the secondary side circuit of the embodiment six of the present application;

[0023] Figure 18 is a schematic diagram of the driving signal of the secondary side switch assembly of the embodiment six of the present application;

[0024] Figure 19 is a schematic diagram of the secondary side circuit of the embodiment seven of the present application;

[0025] Figure 20 is a schematic diagram of the driving signal of the secondary side switch assembly of the embodiment seven of the present application;

[0026] Figures 21-22 is an equivalent circuit diagram of the secondary side circuit of the embodiment seven of the present application;

[0027] Figure 23 is a schematic diagram of the secondary side circuit of the embodiment eight of the present application;

[0028] Figure 24 is a schematic diagram of the driving signal of the secondary side switch assembly of the embodiment eight of the present application;

[0029] Figure 25 is a flow chart of the secondary side voltage regulation control method of the embodiment of the present application. DETAILED DESCRIPTION

[0030] The present application is described in detail below based on the embodiments, but the present application is not limited to these embodiments only. In the following detailed description of the present application, some specific details are described in detail. The present application can also be completely understood without the description of these details by those skilled in the art. In order to avoid confusion of the essence of the present application, the well-known methods, processes, flows, elements and circuits are not described in detail.

[0031] In addition, those skilled in the art should understand that the drawings provided herein are for illustrative purposes only, and the drawings are not necessarily drawn to scale.

[0032] At the same time, it should be understood that in the following description, "circuit" refers to a conductive loop composed of at least one element or sub-circuit through electrical or electromagnetic connection. When an element or circuit is said to be "connected to" another element or said to be "connected between" two nodes, it can be directly coupled or connected to another element or there can be intermediate elements between the elements, and the connection between the elements can be physical, logical, or a combination thereof. On the contrary, when an element is said to be "directly coupled to" or "directly connected to" another element, it means that there is no intermediate element between the two.

[0033] Unless the context clearly requires otherwise, throughout the description, the words "comprise," "comprising," and the like are to be construed in an inclusive sense as opposed to an exclusive or exhaustive sense; that is to say, in the sense of "including, but not limited to."

[0034] In the description of the present application, it should be understood that the terms "first", "second" and the like are used to describe purposes only, and should not be understood as indicating or implying relative importance. In addition, in the description of the present application, unless otherwise stated, the meaning of "a plurality of" is two or more.

[0035] The existing leakage inductance compensation method generally adds capacitors to the original and secondary sides respectively, constructs two independent LC resonant cavities in combination with the leakage inductance, and then adjusts the switching frequency (PFM) of the converter to match or mismatch the impedance of the two resonant cavities to modulate the output energy. However, for occasions with wide input and output ranges, the above-mentioned resonant cavity construction method needs to adjust the switching frequency in a large range in the pure PFM control mode to achieve stable voltage. After combining PFM and pulse width modulation (PWM), the switching frequency adjustment range can be reduced, but the soft switching characteristics of the switching tube may be lost in some cases, which has an adverse effect on the switching device. In addition, the two resonant capacitors added to the original and secondary sides respectively bear the main power energy storage function, so the volume is large, which has a negative impact on the improvement of power density to some extent.

[0036] Figure 1 is a circuit diagram of an isolation type power converter according to an embodiment of the present application. In the embodiment of the present application, the isolation type power converter 1 includes a primary side circuit 11, a transformer 12 and a secondary side circuit 13. The primary side circuit 11 is configured to generate an alternating current input signal Ui. Optionally, the primary side circuit 11 includes a primary side impedance network 111. The primary side impedance network 111 is an impedance network including the leakage inductance of the primary side of the transformer 12. Optionally, the primary side impedance network 111 can also include a resonant capacitor added for compensating or adjusting the overall impedance of the transformer 12. In other optional implementation manners, the primary side impedance network 111 can also not include an additional resonant capacitor, so as to further reduce the complexity and cost of the circuit. The transformer 12 includes a primary side winding and a secondary side winding, which can be an ideal transformer circuit, and only includes a part in which the primary and secondary side cross-linking magnetic fluxes are completely coupled.

[0037] In the embodiment, the isolation type power converter 1 is divided into the primary and secondary sides of the transformer 12, the primary side inverter part in the isolation type power converter 1 is taken as a half stage, and all power devices of the secondary side of the power converter 1 are taken as a complete converter of one stage. At this time, the leakage inductance of the secondary side of the transformer is functionalized as a power energy storage element of the latter stage converter, and the magnetic integration of the transformer is realized.

[0038] In one alternative implementation, the primary-side circuit 11 may include an AC source to provide AC input. Optionally, the AC source may be a voltage source or a current source, and this embodiment is not limited to this.

[0039] In another optional implementation, the primary-side circuit 11 may include a DC input source and a DC-AC inverter network formed by the primary-side switching network 112. This DC-AC inverter network is used to convert the DC input source into an AC source. Optionally, the DC-AC inverter network can be configured as a forward converter, flyback converter, full-bridge, half-bridge, or push-pull structure, etc. It should be understood that this embodiment does not limit the structure of the DC-AC inverter network, as long as it can convert the DC source into an AC source. It should also be understood that this embodiment does not limit the AC input of the AC source; that is, the primary-side input of the isolated power converter 1 can be any AC waveform with unlimited amplitude and shape, and the pulse widths of the positive and negative half-cycles do not need to be equal. Since this invention uses closed-loop control of the secondary-side circuit, the primary-side switching network 112 in the primary-side circuit 11 can be controlled in an open-loop manner with a fixed duty cycle or a fixed frequency, without closed-loop modulation.

[0040] The secondary circuit 13 includes a converter circuit 131 and a feedback control circuit 132. The converter circuit 131 is configured to receive the secondary input signal Ui to generate an output signal Uo.

[0041] In this embodiment, the converter circuit 131 is formed by the secondary-side switching network 131a and the secondary-side leakage inductance Lks of the secondary winding of the transformer 12. The converter circuit 121 achieves current conversion by energizing the secondary-side leakage inductance Lks of the secondary winding via the secondary-side switching components of the secondary-side switching network 131a. The feedback control circuit 132 is configured to control the switching state of the secondary-side switching components in the secondary-side switching network 131a according to the output feedback signal characterizing the output signal Uo, thereby adjusting the output signal Uo to tend towards a desired value. Therefore, the isolated power converter 1 in this embodiment can utilize the secondary leakage inductance in the secondary circuit to achieve secondary closed-loop control. That is, the switching state of the secondary switching components in the secondary switching network is adjusted through negative feedback in the secondary circuit to achieve secondary self-regulation of voltage or current, thereby reducing the size of the secondary circuit. At the same time, the primary switching network in the primary circuit can be open-loop controlled with a fixed duty cycle or fixed frequency. This enables the adjustment of the output signal without primary-secondary communication, thereby achieving primary-secondary decoupling, reducing the control complexity of the system, and reducing the circuit structure and cost.

[0042] In an alternative implementation, the embodiment adopts different frequencies with primary and secondary energy transmission time difference to control the primary and secondary decoupling strategy. In the embodiment, the primary side switching frequency of the isolation type power converter 1 is not greater than the secondary side switching frequency, and the duty cycle of the primary side switching component is independent of the duty cycle of the secondary side switching component. Thus, the embodiment sets the secondary side switching frequency to be not less than the primary side switching frequency, so that the primary and secondary energy transmission time has difference, to achieve the secondary side self-voltage stabilization / current stabilization, while for the switching component in the primary side circuit, it can be controlled in open loop with a fixed duty cycle or fixed frequency, without setting the feedback signal path of the primary and secondary sides, to realize the primary and secondary decoupling.

[0043] The embodiment adjusts the switching state of the secondary side switching component based on the negative feedback of the output voltage of the isolation type power converter, to achieve voltage stabilization or current stabilization. That is, the embodiment realizes the secondary side self-voltage stabilization / current stabilization function of the isolation type power converter, without adjusting the switching frequency and duty cycle of the primary side switching component, and further without setting the feedback signal path of the primary and secondary sides, to improve the power density of the power supply, and reduce the control complexity of the system and the system cost.

[0044] The embodiment mainly describes the adjustment of the duty cycle of the secondary side switching component in PWM mode, taking the stabilization of the output voltage as an example. It should be understood that the embodiment does not limit the adjustment mode of the secondary side switching component.

[0045] Figure 2 is the schematic diagram of the secondary side circuit of the isolation type power converter of the embodiment. As shown in Figure 2 the secondary side circuit 2 of the isolation type power converter of the embodiment includes a current conversion circuit 21 formed by a secondary side leakage inductance Lks and a secondary side switching network 211, and a feedback control circuit 22. The secondary side switching network 211 includes at least one secondary side switching component. In the embodiment, the at least one secondary side switching component is controlled to adjust the switching state according to the corresponding driving signal, to adjust the output voltage and achieve voltage stabilization / current stabilization. The driving signal of the secondary side switching component is determined according to the error signal corresponding to the output voltage. Alternatively, the switching frequency of the secondary side switching component is not less than the switching frequency of the primary side switching component, so that the primary and secondary energy transmission time has difference, to achieve the secondary side self-voltage stabilization / current stabilization, while without setting the feedback signal path of the primary and secondary sides, to realize the primary and secondary decoupling.

[0046] In the embodiment, the feedback control circuit 22 is configured to control the switching state of the secondary-side switching component in the secondary-side switching network 211 according to the output feedback signal representing the output signal. Further, the feedback control circuit 22 adjusts the duty cycle of each secondary-side switching component according to the error between the output feedback signal and the reference signal, so that the output signal tends to the desired value. In an alternative implementation, the feedback control circuit 22 further comprises an error acquisition circuit 221 and a driving circuit 222. The error acquisition circuit 221 is configured to acquire an error signal according to the error between the output feedback signal of the output voltage and the reference signal. Alternatively, the error acquisition circuit 221 is configured to acquire the error signal Ve according to the difference between the output feedback signal Vf acquired from the output terminal and the reference signal Vref. Alternatively, the output feedback signal Vf is acquired by sampling the voltage across the output equivalent resistance Ro. Further alternatively, the secondary-side circuit 2 further comprises an output capacitor Co connected in parallel with the output equivalent resistance Ro, so as to further stabilize the output voltage, current and other electrical signals. In the embodiment, the reference signal Vref can be the desired value of the output signal, or can be another signal representing the desired value of the output signal.

[0047] In an alternative implementation, the driving signal 222 is configured to generate a PWM signal according to the error signal Ve, so as to determine the driving signal of each secondary-side switching component, thereby controlling the switching state of the corresponding secondary-side switching component. Thus, the embodiment can adjust the output voltage based on the negative feedback of the output voltage, so as to achieve voltage stabilization / current stabilization. Further alternatively, the driving circuit 222 generates the PWM signal based on the error signal Ve and a ramp signal RAMP, so as to determine the driving signal of each secondary-side switching component. It should be understood that the embodiment is not limited to the ramp signal, and other signals, such as a square wave signal, can also be applied to the generation of the driving signal in the embodiment, and the embodiment is not limited in this regard. It should be understood that when the secondary-side switching component is controlled by the PWM signal, the switching frequency is determined by the frequency of the ramp signal RAMP.

[0048] In an alternative implementation, the current conversion circuit 21 can be formed as an AC-DC rectifier circuit or an AC-AC rectifier circuit. Further alternatively, when the current conversion circuit 21 is formed as an AC-DC rectifier circuit, it can be a full-wave rectifier circuit or a half-wave rectifier circuit, and can also be a step-up current conversion circuit or a step-down current conversion circuit. In the embodiment, the rectifier circuit is not limited, as long as it can achieve self-stabilization of the output electrical signal through current conversion.

[0049] In an alternative implementation, if the switching frequency of the secondary-side switching component in the secondary-side switching network is an integer multiple of the switching frequency of the primary-side switching component, the ramp signal RAMP can also be synchronized by detecting the transition edge (e.g., rising edge) of the secondary-side input signal, so that it is synchronized with the switching frequency of the primary side, to further improve the signal regulation efficiency and reduce the loss.

[0050] In an alternative implementation, in the operation period of the secondary-side switching network, the secondary-side leakage inductance is magnetized by performing the chopper control on at least one of the secondary-side switching components, to achieve the current conversion. Specifically, the feedback control circuit is configured to perform the chopper control on the corresponding control switching component in the positive half cycle or the negative half cycle of the secondary-side input signal, so that the secondary-side leakage inductance is magnetized to achieve the current conversion, and further adjust the output signal to make it tend to the expected value, thereby achieving the secondary-side self-regulation of voltage or current.

[0051] In the embodiment of the present application, the isolated power converter includes a primary-side circuit, a transformer and a secondary-side circuit, wherein the secondary-side circuit includes a current conversion circuit and a feedback control circuit, the current conversion circuit magnetizes the secondary-side leakage inductance of the secondary-side winding via the secondary-side switching component in the secondary-side switching network to achieve the current conversion, and the feedback control circuit is configured to control the switching state of the secondary-side switching component in the secondary-side switching network according to the output feedback signal representing the output signal. Thus, the embodiment can directly adjust the secondary-side switching component in the secondary-side switching network, so that the secondary-side leakage inductance can be used as a power storage element, to achieve the magnetic integration of the transformer, and further to reduce the control complexity and manufacturing cost without the feedback signal path between the primary side and the secondary side.

[0052] The embodiment of the present application mainly takes the current conversion circuit as an example of the boost AC-DC rectifier circuit. The boost circuit means that the output signal is always greater than the input signal.

[0053] Figure 3 is a schematic diagram of a secondary-side circuit of an isolated power converter according to the second embodiment of the present application. In the embodiment, the current conversion circuit is a boost full-wave rectifier circuit. Optionally, the switching frequency of the secondary-side switching component is not less than 2 times the switching frequency of the primary-side switching component, to make the primary-side and secondary-side energy transmission time have a difference. For example, Figure 3As shown, in this embodiment, the secondary circuit 3 includes a converter circuit composed of a secondary leakage inductance Lks and a secondary switching network. The secondary switching network includes secondary switching components 31-34. Secondary switching component 31 is coupled between the first input terminal i1 of the secondary switching network and the first output terminal o1 of the secondary circuit 3; secondary switching component 32 is coupled between the first input terminal i1 of the secondary switching network and the second output terminal o2 of the secondary circuit 3; secondary switching component 33 is coupled between the second input terminal i2 of the secondary switching network and the first output terminal o1 of the secondary circuit 3; and secondary switching component 34 is coupled between the second input terminal i2 of the secondary switching network and the second output terminal o2 of the secondary circuit 3.

[0054] In one alternative implementation, the secondary-side switching assembly includes a switching element and a diode connected in parallel. Further, the switching element can be an active switching device. For example... Figure 3 As shown, secondary-side switching assembly 31 includes a switching element S1 and a diode D1; secondary-side switching assembly 32 includes a switching element S2 and a diode D2; secondary-side switching assembly 33 includes a switching element S3 and a diode D3; and secondary-side switching assembly 34 includes a switching element S4 and a diode D4. In other optional implementations, the secondary-side switching assembly can also be a transistor element with a parasitic diode, such as a MOSFET. Further, in this embodiment, the direction of current conduction in the secondary-side switching assembly serving as the control switching assembly is from the cathode to the anode of the diode. That is, when the control switching assembly is turned on, the switching element in the secondary-side switching assembly is in the on state, and the body diode is not conducting. The direction of current conduction in the secondary-side switching assembly serving as the non-control switching assembly is from the anode to the cathode of the diode, that is, when the non-control switching assembly is turned on, it can be based on the switching element or on the corresponding diode. Here, the non-control switching assembly refers to other secondary-side switching assemblies besides the control switching assembly during operation.

[0055] In one alternative implementation, under different control methods, the secondary switch components 31-34 in this embodiment can all be used as control switch components.

[0056] In the embodiment, the secondary side switch assemblies 32 and 34 are taken as the control switch assemblies. In the positive half cycle of the secondary side input signal, the second secondary side switch assembly 32 is taken as the control switch assembly, and the first secondary side switch assembly 31, the third secondary side switch assembly 33 and the fourth secondary side switch assembly 34 are taken as the non-control switch assemblies. In the negative half cycle of the secondary side input signal, the fourth secondary side switch assembly 34 is taken as the control switch assembly, and the first secondary side switch assembly 31, the second secondary side switch assembly 32 and the third secondary side switch assembly 33 are taken as the non-control switch assemblies. The feedback control circuit 30 is configured to perform the chopping control on the second secondary side switch assembly 32 in the positive half cycle of the secondary side input signal so that the secondary side leakage inductance is excited in the positive half cycle, and perform the chopping control on the fourth secondary side switch assembly 34 in the negative half cycle of the secondary side input signal so that the secondary side leakage inductance is excited in the negative half cycle, to realize the current conversion.

[0057] Further, in the embodiment, the feedback control circuit 30 comprises an error acquisition circuit 35 and a driving circuit 36. The corresponding driving signals are acquired by the error acquisition circuit 35 and the driving circuit 36 to perform the chopping control on the secondary side switch assemblies 32 and 34. Further, the driving signal of the secondary side switch assembly 31 is complementary to the secondary side switch assembly 32, and the driving signal of the secondary side switch assembly 33 is complementary to the secondary side switch assembly 34. Thus, the secondary side leakage inductance Lks of the embodiment is short-circuited and excited by the secondary side switch assemblies 32 and 34, and discharged to the output end through the secondary side switch assemblies 31 and 34 and 33 and 32 respectively, so that the secondary side circuit 3 plays a role of the boost rectifier, thereby realizing the secondary side voltage stabilization. In an optional implementation, in the embodiment, the error acquisition circuit 35 can comprise a sampling circuit 351 and a comparison circuit A1. The sampling circuit 351 is used to sample and acquire an output feedback signal Vf representing the output voltage. The comparison circuit A1 is used to compare the output feedback signal Vf and a reference signal Vref to acquire an error signal Ve. The reference signal Vref is used to represent the expected value of the output voltage. The driving circuit 36 further comprises a comparator A2 used to generate complementary PWM signals SW and SW' according to the error signal Ve and a predetermined ramp signal RAMP to control the secondary side switch assemblies 31-34. The PWM signal SW is positively selected to obtain the driving signal SW2 of the secondary side switch assembly 32, wherein the driving signal SW2 keeps in the always-on state in the negative half cycle. The PWM signal SW is negatively selected to obtain the driving signal SW4 of the secondary side switch assembly 34, wherein the driving signal SW4 keeps in the always-on state in the positive half cycle. The driving signal SW1 of the secondary side switch assembly 31 is complementary to the driving signal SW2, and the driving signal SW3 of the secondary side switch assembly 33 is complementary to the driving signal SW4.

[0058] In the embodiment, the secondary side circuit 3 further comprises a positive and negative phase detection circuit used to detect the positive and negative phases of the secondary side input signal to select the working secondary side switch assembly to switch different working modes.

[0059] Specifically, when the secondary-side input signal is in the positive half cycle, the driving signal controls the secondary-side switching component to be in the first working mode; when the secondary-side input signal is in the negative half cycle, the driving signal is switched to control the secondary-side switching component to be in the second working mode.

[0060] Optionally, the positive and negative phase detection circuit can be realized by zero-crossing detection, and the embodiment is not limited in this regard. Optionally, the positive and negative phase detection circuit can detect the signal at the endpoint a1, or the endpoint a2, or the endpoint a3 to obtain the positive and negative phases of the input signal, and the embodiment is not limited in the phase detection point.

[0061] In an optional embodiment, if the switching frequency of the secondary-side switching component is an integer multiple of the switching frequency of the primary-side switching component, the rising edge (for example, the rising edge) of the secondary-side input signal can also be detected to synchronize the ramp signal RAMP, so that it is synchronized with the switching frequency of the primary side.

[0062] Figure 4 is a schematic diagram of a driving signal of a secondary-side switching component of embodiment two. The secondary-side switching component 32 is selected in the positive half cycle of the secondary-side input signal, that is, the secondary-side switching component 32 is controlled in the positive half cycle of the secondary-side input signal. The secondary-side switching component 34 is selected in the negative half cycle of the secondary-side input signal, that is, the secondary-side switching component 34 is controlled in the negative half cycle of the secondary-side input signal. The secondary-side switching component 32 is high in the non-selected period (that is, in the negative half cycle of the secondary-side input signal), that is, it is in the always-on state. The secondary-side switching component 34 is high in the non-selected period (that is, in the positive half cycle of the secondary-side input signal), that is, it is in the always-on state. Further, the driving signals SW1 and SW2 of the secondary-side switching component 31 and the secondary-side switching component 32 are complementary, and the driving signals SW3 and SW4 of the secondary-side switching component 33 and the secondary-side switching component 34 are complementary.

[0063] As shown in Figure 4 The gate signal SW on Pri is positively high in the positive half cycle (t0-t1) of the secondary-side input signal and negatively high in the negative half cycle (t1-t2) of the secondary-side input signal, so that the secondary-side switching component 32 is selected in the positive half cycle of the secondary-side input signal, and the secondary-side switching component 34 is selected in the negative half cycle of the secondary-side input signal.

[0064] In the time period of t0-t1 (i.e. the positive half cycle of the secondary input signal), the driving signal controls the secondary switching components 31-34 to be in the first operation mode, specifically, the secondary switching component 32 and the secondary switching component 31 are alternately turned on, the secondary switching component 34 is kept on, and the secondary switching component 33 is kept off. In the time period of t1-t2 (i.e. the negative half cycle of the secondary input signal), the driving signal controls the secondary switching components 31-34 to be in the second operation mode, specifically, the secondary switching component 32 is kept on, the secondary switching component 31 is kept off, and the secondary switching component 34 and the secondary switching component 33 are alternately turned on. Thus, the feedback control circuit 30 generates the corresponding driving signal to control the second secondary switching component 32 and the fourth secondary switching component 34 to be on in the positive half cycle of the secondary input signal to excite the secondary leakage inductance Lks, control the first secondary switching component 31 and the fourth secondary switching component 34 to be on to release energy to the output end of the secondary circuit, control the fourth secondary switching component 34 and the second secondary switching component 32 to be on in the negative half cycle of the secondary input signal to excite the secondary leakage inductance, and control the third secondary switching component 33 and the second secondary switching component 32 to be on to release energy to the output end of the secondary circuit, so as to realize the boosting rectification of the current conversion circuit, and further realize the secondary self-voltage stabilization and current stabilization.

[0065] Figures 5-8 is the equivalent circuit diagram of the secondary circuit of the second embodiment of the present application. In the positive half cycle of the secondary input signal, when the secondary switching components 32 and 34 are turned on and the secondary switching components 31 and 33 are turned off (for example, in the time period of t0-t01), the equivalent circuit diagram of the secondary circuit is as shown in Figure 5 When the secondary switching components 32 and 34 are turned on and the secondary switching components 31 and 33 are turned off, the secondary leakage inductance Lks, the secondary switching component 32, the secondary switching component 34 and the secondary coil form a current loop. Wherein, the turning on of the secondary switching components 32 and 34 short-circuits the secondary leakage inductance Lks to be excited, and the excitation time is determined by the common turning-on time of the secondary switching components 32 and 34. In the current loop, the current direction of the secondary switching component 32 as the control switching component is from the cathode to the anode of the diode D2, and the current direction of the secondary switching component 34 as the non-control switching component is from the anode to the cathode of the diode D4.

[0066] When the secondary switching components 32 and 33 are turned off and the secondary switching components 31 and 34 are turned on (for example, in the time period of t01-t02), the equivalent circuit diagram of the secondary circuit is as shown in Figure 6As shown. When secondary-side switching components 32 and 33 are off and secondary-side switching components 31 and 34 are on, the secondary-side leakage inductance Lks, secondary-side switching component 31, output equivalent resistance Ro, secondary-side switching component 34, and secondary-side coil form a current loop, and the secondary-side leakage inductance Lks freewheels through secondary-side switching components 31 and 34. In the current loop, the current direction of secondary-side switching component 31, which is a non-control switching component, is from the anode to the cathode of diode D1, and the current direction of secondary-side control switch 34 is from the anode to the cathode of diode D4.

[0067] During the positive half-cycle of the secondary input signal, when the secondary switch assembly 32 is on, the secondary switch assemblies 32 and 34 are on, causing the secondary leakage inductance Lks to be short-circuited and energized. When the secondary switch assembly 32 is off, the secondary leakage inductance Lks is freewheeled through the secondary switch assemblies 31 and 34, thereby realizing the boost converter.

[0068] During the negative half-cycle of the secondary input signal, when secondary-side switching components 32 and 34 are turned on and secondary-side switching components 31 and 33 are turned off (e.g., at times t1-t11), the equivalent circuit diagram of the secondary circuit is as follows: Figure 7 As shown. When secondary-side switching components 32 and 34 are turned on and secondary-side switching components 31 and 33 are turned off, the secondary-side leakage inductance Lks, the secondary-side coil, secondary-side switching component 34, and secondary-side switching component 32 form a current loop. Specifically, the turning on of secondary-side switching components 32 and 34 causes the secondary-side leakage inductance Lks to be short-circuited and energized; the energizing time is determined by the combined on-time of secondary-side switching components 32 and 34. In the current loop, the current direction of secondary-side switching component 34, which acts as a control switching component, is from the cathode to the anode of diode D4, while the current direction of secondary-side switching component 32, which acts as a non-control switching component, is from the anode to the cathode of diode D2.

[0069] When secondary-side switching components 31 and 34 are off and secondary-side switching components 32 and 33 are on (e.g., at times t11-t12), the equivalent circuit diagram of the secondary-side circuit is as follows: Figure 8 As shown. When secondary-side switching components 31 and 34 are off and secondary-side switching components 32 and 33 are on, the secondary-side leakage inductance Lks, the secondary-side coil, the secondary-side switching component 33, the output equivalent resistance Ro, and the secondary-side switching component 32 form a current loop. The secondary-side leakage inductance Lks freewheels through the secondary-side switching components 32 and 33. In the current loop, the current direction of the secondary-side switching component 32, which is a non-control switching component, is from the anode to the cathode of diode D2, and the current direction of the secondary-side control switch 33 is from the anode to the cathode of diode D3.

[0070] In the negative half cycle of the secondary side input signal, during the conduction of the secondary side switching component 34, the secondary side leakage inductance Lks is short-circuited through the secondary side switching components 32 and 34 to excite the magnetism, and during the turn-off of the secondary side switching component 34, the secondary side leakage inductance Lks is freewheeling through the secondary side switching components 32 and 33, thereby realizing the boost current conversion.

[0071] In summary, in the positive and negative half cycles of the secondary side input signal, the secondary side leakage inductance Lks is short-circuited through the secondary side switching components 32 and 34 to excite the magnetism, and is freewheeling through the secondary side switching components 31 and 34 and the secondary side switching components 32 and 33, respectively, thereby realizing the full-wave boost rectification, and thus realizing the self-stabilization of the secondary side.

[0072] Figure 9 is the schematic diagram of the secondary side circuit of the third embodiment of the present application. As shown in the figure, Figure 9 the secondary side circuit 9 in the present embodiment is similar to the secondary side circuit 3 in the second embodiment of the present application, and the specific components are not described here. In the present embodiment, the secondary side switching components 33 and 31 are taken as the control switching components. In the positive half cycle of the secondary side input signal, the third secondary side switching component 33 is the control switching component, and the first secondary side switching component 31, the second secondary side switching component 32 and the fourth secondary side switching component 34 are non-control switching components. In the negative half cycle of the secondary side input signal, the first secondary side switching component 31 is the control switching component, and the second secondary side switching component 32, the third secondary side switching component 33 and the fourth secondary side switching component 34 are non-control switching components. The feedback control circuit 30 is configured to perform chopper control on the third secondary side switching component 33 in the positive half cycle of the secondary side input signal to excite the secondary side leakage inductance in the positive half cycle, and perform chopper control on the first secondary side switching component 31 in the negative half cycle of the secondary side input signal to excite the secondary side leakage inductance in the negative half cycle, so as to realize the current conversion.

[0073] The feedback control circuit 30 obtains an error signal according to the output feedback signal Vf and the reference signal Vref, and generates complementary PWM signals SW and SW' based on the error signal and the ramp signal RAMP to control the secondary side switching components 31-34. The PWM signal SW is selected in positive phase to obtain the driving signal SW3 of the secondary side switching component 33, and the driving signal SW3 is always on in the negative half cycle. The PWM signal SW is selected in negative phase to obtain the driving signal SW1 of the secondary side switching component 31, and the driving signal SW1 is always on in the positive half cycle. Further, the driving signal SW2 of the secondary side switching component 32 is complementary to the driving signal SW1, and the driving signal SW4 of the secondary side switching component 34 is complementary to the driving signal SW3. Thus, the secondary side leakage inductance Lks is short-circuited by the secondary side switching components 33 and 31 to excite the magnetic field, and discharged to the output terminal through the secondary side switching components 31 and 34 and 33 and 32 respectively, so that the secondary side circuit 9 functions as a boost rectifier, thereby realizing secondary side voltage stabilization.

[0074] In an optional implementation, in the embodiment, if the switching frequency of the secondary side switching components is an integer multiple of the switching frequency of the primary side switching components, the ramp signal RAMP can also be synchronized by detecting the transition edge (e.g. rising edge or falling edge) of the secondary side input signal, so that it is synchronized with the switching frequency of the primary side, further improving the voltage stabilization efficiency.

[0075] In an optional implementation, the working secondary side switching components can also be selected by detecting the positive and negative phases of the secondary side input signal to switch different working modes, that is, the driving signals are synchronized to switch the driving logic according to the positive and negative phases of the secondary side input signal. Specifically, when the secondary side input signal is in the positive half cycle, the driving signals control the secondary side switching components to be in the first working mode; when the secondary side input signal is in the negative half cycle, the driving signals are switched to control the secondary side switching components to be in the second working mode. Further, the secondary side circuit 9 can also include a positive and negative phase detection circuit (not shown in the figure) for detecting the positive and negative phases of the secondary side input signal to realize the phase synchronization of the driving signals of the secondary side switching components and the secondary side input signal, so that each secondary side switching component switches the corresponding working mode. Figure 9

[0076] Optionally, the positive and negative phase detection circuit can be realized by zero-crossing detection, which is not limited in the embodiment. Optionally, the positive and negative phase detection circuit can detect the signal at the endpoint a1, or the endpoint a2, or the endpoint a3 to obtain the positive and negative phases of the input signal, and the phase detection point is not limited in the embodiment.

[0077] Figure 10 ​is a schematic diagram of the driving signal of the secondary side switch assembly of embodiment three of the present application. In this embodiment, the secondary side switch assembly 33 is gated in the positive half cycle of the secondary side input signal, i.e. the secondary side switch assembly 33 is chopper controlled in the positive half cycle of the secondary side input signal, the secondary side switch assembly 31 is gated in the negative half cycle of the secondary side input signal, i.e. the secondary side switch assembly 31 is chopper controlled in the negative half cycle of the secondary side input signal, the secondary side switch assembly 33 is high in the non-gated period (i.e. in the negative half cycle of the secondary side input signal), i.e. keeps the always-on state, the secondary side switch assembly 31 is high in the non-gated period (i.e. in the positive half cycle of the secondary side input signal), i.e. keeps the always-on state. Further, the driving signal SW2 and SW1 of the secondary side switch assembly 32 is complementary to the secondary side switch assembly 31, the driving signal SW4 and SW3 of the secondary side switch assembly 34 is complementary to the secondary side switch assembly 33.

[0078] As shown in Figure 10 , the gate signal SW on Pri is positively high in the positive half cycle of the secondary side input signal (t0-t1), and is negatively high in the negative half cycle of the secondary side input signal (t1-t2), so that the secondary side switch assembly 33 is gated in the positive half cycle of the secondary side input signal, and the secondary side switch assembly 31 is gated in the negative half cycle of the secondary side input signal.

[0079] In t0-t1 (i.e. the positive half cycle of the secondary side input signal), the driving signal controls the secondary side switch assembly 31-34 to be in the first working mode, specifically, the secondary side switch assembly 33 and the secondary side switch assembly 34 are alternately turned on, the secondary side switch assembly 31 keeps on, and the secondary side switch assembly 32 keeps off. In t1-t2 (i.e. the negative half cycle of the secondary side input signal), the driving signal controls the secondary side switch assembly 31-34 to be in the second working mode, specifically, the secondary side switch assembly 33 keeps on, the secondary side switch assembly 34 keeps off, and the secondary side switch assembly 31 and the secondary side switch assembly 32 are alternately turned on. Thus, the feedback control circuit 30 generates corresponding driving signals to control the third secondary side switch assembly 33 and the first secondary side switch assembly 31 to be turned on in the positive half cycle of the secondary side input signal to excite the secondary side leakage inductance Lks, and to control the first secondary side switch assembly 31 and the fourth secondary side switch assembly 34 to be turned on to release energy to the output end of the secondary side circuit, and to control the third secondary side switch assembly 33 and the first secondary side switch assembly 31 to be turned on in the negative half cycle of the secondary side input signal to excite the secondary side leakage inductance, and to control the third secondary side switch assembly 33 and the second secondary side switch assembly 32 to be turned on to release energy to the output end of the secondary side circuit, so as to realize the boost rectification of the current conversion circuit, and further realize the secondary side self-stabilized voltage and current.

[0080] Figures 11-12is the equivalent circuit diagram of the secondary side circuit of the second embodiment of the present application. In the positive half cycle of the secondary side input signal, when the secondary side switching components 31 and 33 are turned on and the secondary side switching components 32 and 34 are turned off (for example, at time t0-t01), the equivalent circuit diagram of the secondary side circuit is as shown in Figure 11 When the secondary side switching components 32 and 34 are turned off and the secondary side switching components 31 and 33 are turned on, the secondary side leakage inductance Lks, the secondary side switching component 31, the secondary side switching component 33, and the secondary side coil form a current loop. Among them, the turn-on of the secondary side switching components 31 and 33 shorts the secondary side leakage inductance Lks to excite the magnetism, and the excitation time is determined by the common turn-on time of the secondary side switching components 31 and 33. In the current loop, the current direction of the secondary side switching component 33 as the control switching component is from the cathode to the anode of the diode D3, and the current direction of the secondary side switching component 31 as the non-control switching component is from the anode to the cathode of the diode D1.

[0081] When the secondary side switching components 32 and 33 are turned off and the secondary side switching components 34 and 31 are turned on (for example, at time t01-t02), the equivalent circuit diagram of the secondary side circuit is as shown in Figure 6 When the secondary side switching components 32 and 33 are turned off and the secondary side switching components 34 and 31 are turned on, the secondary side leakage inductance Lks, the secondary side switching component 31, the output equivalent resistance Ro, the secondary side switching component 34, and the secondary side coil form a current loop, and the secondary side leakage inductance Lks flows through the secondary side switching components 34 and 31. In the current loop, the current direction of the secondary side switching component 31 as the non-control switching component is from the anode to the cathode of the diode D1, and the current direction of the secondary side control switching component 34 is from the anode to the cathode of the diode D4.

[0082] In the positive half cycle of the secondary side input signal, during the turn-on of the secondary side switching component 33, the turn-on of the secondary side switching components 33 and 31 shorts the secondary side leakage inductance Lks to excite the magnetism, and during the turn-off of the secondary side switching component 33, the secondary side leakage inductance Lks flows through the secondary side switching components 34 and 31 to achieve the boost current conversion.

[0083] In the negative half cycle of the secondary side input signal, when the secondary side switching components 31 and 33 are turned on and the secondary side switching components 32 and 34 are turned off (for example, at time t1-t11), the equivalent circuit diagram of the secondary side circuit is as shown in Figure 12As shown in the figure. When the secondary side switch assembly 31 and 33 are turned on, and the secondary side switch assembly 32 and 34 are turned off, the secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 33, and the secondary side switch assembly 31 form a current loop. Among them, the secondary side switch assembly 31 and 33 are turned on to short circuit the secondary side leakage inductance Lks to excite the magnetism, and the excitation time is determined by the common conduction time of the secondary side switch assembly 31 and 33. Among them, the current direction of the secondary side switch assembly 31 as the control switch assembly is from the cathode to the anode of the diode D4, and the current direction of the secondary side switch assembly 33 as the non-control switch assembly is from the anode to the cathode of the diode D3.

[0084] When the secondary side switch assembly 31 and 34 are turned off, and the secondary side switch assembly 32 and 33 are turned on (for example, at t11-t12), the equivalent circuit diagram of the secondary side circuit is as shown in the figure. Figure 8 When the secondary side switch assembly 31 and 34 are turned off, and the secondary side switch assembly 32 and 33 are turned on, the secondary side leakage inductance Lks, the secondary side coil, the secondary side switch assembly 33, the output equivalent resistance Ro, and the secondary side switch assembly 32 form a current loop, and the secondary side leakage inductance Lks flows through the secondary side switch assembly 32 and 33. In the current loop, the current direction of the secondary side switch assembly 32 as the non-control switch assembly is from the anode to the cathode of the diode D2, and the current direction of the secondary side control switch 33 is from the anode to the cathode of the diode D3.

[0085] In the negative half cycle of the secondary side input signal, during the conduction period of the secondary side switch assembly 31, the secondary side leakage inductance Lks flows through the secondary side switch assembly 31 and 33 to short circuit and excite the magnetism, and during the turn-off period of the secondary side switch assembly 31, the secondary side leakage inductance Lks flows through the secondary side switch assembly 32 and 33 to achieve boost rectification.

[0086] In summary, in the positive half cycle and the negative half cycle of the secondary side input signal, the secondary side leakage inductance Lks flows through the secondary side switch assembly 31 and 33 to short circuit and excite the magnetism, and flows through the secondary side switch assembly 31, 34, and the secondary side switch assembly 32 and 33 respectively to achieve full-wave boost rectification, thereby realizing the self-voltage stabilization of the secondary side.

[0087] Figure 13 The figure is a schematic diagram of the secondary side circuit of the fourth embodiment of the application. Figure 13Another control strategy of the boost full-wave rectification is shown, in this embodiment, the second auxiliary side switch assembly (auxiliary side switch assembly 32) and the first auxiliary side switch assembly (auxiliary side switch assembly 31) are control switch assemblies. Among them, in the positive half cycle of the auxiliary side input signal, the second auxiliary side switch assembly 32 is the control switch assembly, and the first auxiliary side switch assembly 31, the third auxiliary side switch assembly 33 and the fourth auxiliary side switch assembly 34 are non-control switch assemblies. In the negative half cycle of the auxiliary side input signal, the first auxiliary side switch assembly 31 is the control switch assembly, and the second auxiliary side switch assembly 32, the third auxiliary side switch assembly 33 and the fourth auxiliary side switch assembly 34 are non-control switch assemblies. The feedback control circuit 30 is configured to perform chopping control on the second auxiliary side switch assembly 32 in the positive half cycle of the auxiliary side input signal so that the auxiliary side leakage inductance is excited in the positive half cycle, and perform chopping control on the first auxiliary side switch assembly 31 in the negative half cycle of the auxiliary side input signal so that the auxiliary side leakage inductance is excited in the negative half cycle, to realize full-wave boost rectification, and further realize auxiliary side self-voltage stabilization.

[0088] As shown in Figure 13 In this embodiment, the comparator A2 in the drive circuit generates complementary PWM signals SW and SW' according to the error signal Ve and the predetermined ramp signal RAMP to control the auxiliary side switch assemblies 31-34. Among them, the PWM signal SW is positively selected to obtain the drive signal SW2 of the auxiliary side switch assembly 32, and the PWM signal SW is negatively selected to obtain the drive signal SW1 of the auxiliary side switch assembly 31, wherein the drive signals SW1 and SW2 are complementary in the non-selected period. The auxiliary side switch assembly 34 is always on in the positive half cycle of the auxiliary side input signal; the auxiliary side switch assembly 33 is always on in the negative half cycle of the auxiliary side input signal. Therefore, the auxiliary side switch assembly 32 is controlled to be turned on in the positive half cycle of the auxiliary side input signal to realize leakage inductance Lks excitation, and the auxiliary side switch assembly 31 is controlled to be turned on in the negative half cycle of the auxiliary side input signal to realize leakage inductance Lks excitation.

[0089] In an optional implementation, in order to maintain the modulation logic effectiveness of the auxiliary side switch assemblies 31 and 32, the drive signals of the auxiliary side switch assemblies are synchronized with the primary side switch assemblies. Specifically, the ramp signal RAMP can be synchronized with the switching frequency of the primary side by detecting the jump edge of the auxiliary side input signal.

[0090] Further optionally, the auxiliary side circuit 13 can further include a positive and negative phase detection circuit (not shown in Figure 15 for detecting the positive and negative phases of the auxiliary side input signal to control the switching of different working modes of each auxiliary side switch assembly.

[0091] Figure 14 is a schematic diagram of the drive signals of the auxiliary side switch assemblies of the fourth embodiment of the present application. As Figure 14As shown, the selection signal SW on Pri is high in the positive half cycle (t0'-t1') of the secondary input signal and is low in the negative half cycle (t1'-t2') of the secondary input signal, so that the secondary switching assembly 32 is turned on in the positive half cycle of the secondary input signal and the secondary switching assembly 31 is turned on in the negative half cycle of the secondary input signal.

[0092] In the positive half cycle of the secondary input signal, when the secondary switching assemblies 32 and 34 are turned on and the secondary switching assemblies 31 and 33 are turned off (for example, at time t0'-t01'), the equivalent circuit diagram of the secondary circuit can refer to Figure 5 . In the equivalent circuit diagram, the secondary leakage inductance Lks, the secondary switching assembly 32, the secondary switching assembly 34, and the secondary coil form a current loop. The secondary switching assemblies 32 and 34 are turned on to short-circuit the secondary leakage inductance Lks to excite the magnetic field, and the excitation time is determined by the common conduction time of the secondary switching assemblies 32 and 34. In the current loop, the current direction of the secondary switching assembly 32 as the control switching assembly is from the cathode to the anode of the diode D2, and the current direction of the secondary switching assembly 34 as the non-control switching assembly is from the anode to the cathode of the diode D2.

[0093] When the secondary switching assemblies 32 and 33 are turned off and the secondary switching assemblies 31 and 34 are turned on (for example, at time t01'-t02'), the equivalent circuit diagram of the secondary circuit can refer to Figure 6 . In the equivalent circuit diagram, the secondary leakage inductance Lks, the secondary switching assembly 31, the output equivalent resistance Ro, the secondary switching assembly 34, and the secondary coil form a current loop, and the secondary leakage inductance Lks is freewheeling through the secondary switching assemblies 31 and 34. In the current loop, the current direction of the secondary switching assembly 31 as the non-control switching assembly is from the anode to the cathode of the diode D1, and the current direction of the secondary control switching assembly 34 is from the anode to the cathode of the diode D4.

[0094] In the positive half cycle of the secondary input signal, during the conduction of the secondary switching assembly 32, the secondary leakage inductance Lks is freewheeling through the secondary switching assemblies 32 and 34 to short-circuit the secondary leakage inductance Lks to excite the magnetic field, and during the turn-off of the secondary switching assembly 32, the secondary leakage inductance Lks is freewheeling through the secondary switching assemblies 31 and 34, thereby realizing the boost current conversion.

[0095] In the negative half cycle of the secondary input signal, when the secondary switching assemblies 31 and 33 are turned on and the secondary switching assemblies 32 and 34 are turned off (for example, at time t1'-t11'), the equivalent circuit diagram of the secondary circuit can refer to Figure 12The equivalent circuit diagram of the secondary side circuit when the secondary side switch component 31 and 34 are off, and the secondary side switch component 32 and 33 are on (for example, at the moment of t11'-t12') can refer to

[0096] The equivalent circuit diagram of the secondary side circuit when the secondary side switch component 31 and 34 are off, and the secondary side switch component 32 and 33 are on (for example, at the moment of t11'-t12') can refer to Figure 8 The equivalent circuit diagram of the secondary side circuit when the secondary side switch component 31 and 34 are off, and the secondary side switch component 32 and 33 are on (for example, at the moment of t11'-t12') can refer to

[0097] The equivalent circuit diagram of the secondary side circuit when the secondary side switch component 31 and 34 are off, and the secondary side switch component 32 and 33 are on (for example, at the moment of t11'-t12') can refer to

[0098] The equivalent circuit diagram of the secondary side circuit when the secondary side switch component 31 and 34 are off, and the secondary side switch component 32 and 33 are on (for example, at the moment of t11'-t12') can refer to

[0099] In the above embodiment, the driving signals SW4 and SW3 of the secondary side switch components 34 and 33 are respectively placed high according to the phase of the primary side in time, and synchronous rectification is performed to reduce the loss. In other optional implementation manners, the driving signals of the secondary side switch components 34 and 33 can also be kept as 0 at all times, and the rectification is performed by the diodes connected in parallel thereto.

[0100] Figure 15 is the schematic diagram of the secondary side circuit of the fifth embodiment of the present application. As Figure 15As shown, the secondary side circuit 15 in the embodiment is similar to the secondary side circuit 13 in the fourth embodiment of the present application, and the specific components are not described here. In the embodiment, the secondary side switch components 33 and 34 are taken as examples of the control switch. In the positive half cycle of the secondary side input signal, the third secondary side switch component 33 is the control switch component, and the first secondary side switch component 31, the second secondary side switch component 32 and the fourth secondary side switch component 34 are non-control switch components. In the negative half cycle of the secondary side input signal, the fourth secondary side switch component 34 is the control switch component, and the first secondary side switch component 31, the second secondary side switch component 32 and the third secondary side switch component 33 are non-control switch components. The feedback control circuit 30 is configured to perform chopper control on the third secondary side switch component 33 in the positive half cycle of the secondary side input signal so that the secondary side leakage inductance is excited in the positive half cycle, and perform chopper control on the fourth secondary side switch component 34 in the negative half cycle of the secondary side input signal so that the secondary side leakage inductance is excited in the negative half cycle, to achieve current conversion.

[0101] In the embodiment, the feedback control circuit 30 obtains an error signal Ve according to the output feedback signal Vf obtained by sampling and the reference signal Vref, and generates complementary PWM signals SW and SW' based on the error signal Ve and the ramp signal RAMP, to control the secondary side switch components 31-34. The PWM signal SW is positively selected to obtain the driving signal SW3 of the third secondary side switch component 33, and the PWM signal SW is negatively selected to obtain the driving signal SW4 of the fourth secondary side switch component 34. In the non-selected period, the driving signals SW1 and SW2 are complementary. The secondary side switch component 31 is always on in the positive half cycle of the secondary side input signal; and the secondary side switch component 32 is always on in the negative half cycle of the secondary side input signal. Therefore, the secondary side switch component 33 is controlled to be turned on in the positive half cycle of the secondary side input signal to excite the leakage inductance Lks, and the secondary side switch component 34 is controlled to be turned on in the negative half cycle of the secondary side input signal to excite the leakage inductance Lks.

[0102] In an optional implementation, in order to maintain the modulation logic effectiveness of the secondary side switch components 31 and 32, the driving signals of the secondary side switch components are synchronized with the primary side switch components. Specifically, the embodiment can synchronize the ramp signal RAMP by detecting the jump edge of the secondary side input signal, so that the ramp signal RAMP is synchronized with the switching frequency of the primary side.

[0103] Further optionally, the secondary side circuit 15 can further include a positive and negative phase detection circuit (not shown in the figure) for detecting the positive and negative phases of the secondary side input signal, to control the secondary side switch components to switch different working modes. Figure 15

[0104] Figure 16 is a schematic diagram of the driving signals of the secondary side switch components in the fifth embodiment of the present application. As shown in the figure, the driving signals of the secondary side switch components 31-34 are synchronized with the driving signals of the primary side switch components 11-14. Figure 16 ​As shown, the selection signal SW on Pri is positively set high in the positive half cycle (t0'-t1') of the secondary-side input signal and negatively set high in the negative half cycle (t1'-t2') of the secondary-side input signal, so that the secondary-side switching assembly 33 is turned on to realize excitation in the positive half cycle of the secondary-side input signal, and the secondary-side switching assembly 34 is turned on to realize excitation in the negative half cycle of the secondary-side input signal.

[0105] In the positive half cycle of the secondary-side input signal, when the secondary-side switching assemblies 31 and 33 are turned on and the secondary-side switching assemblies 32 and 34 are turned off (for example, at time t0'-t01'), the equivalent circuit diagram of the secondary-side circuit can refer to Figure 11 . Among them, the secondary-side leakage inductance Lks, the secondary-side switching assembly 31, the secondary-side switching assembly 33, and the secondary-side coil form a current loop. The secondary-side leakage inductance Lks is short-circuited to excite by the common conduction of the secondary-side switching assemblies 31 and 33, and the excitation time is determined by the common conduction time of the secondary-side switching assemblies 31 and 33. In the current loop, the current direction of the secondary-side switching assembly 33 as the control switching assembly is from the cathode to the anode of the diode D3, and the current direction of the secondary-side switching assembly 31 as the non-control switching assembly is from the anode to the cathode of the diode D1.

[0106] When the secondary-side switching assemblies 33 and 32 are turned off and the secondary-side switching assemblies 31 and 34 are turned on (for example, at time t01'-t02'), the equivalent circuit diagram of the secondary-side circuit can refer to Figure 6 . Among them, the secondary-side leakage inductance Lks, the secondary-side switching assembly 31, the output equivalent resistance Ro, the secondary-side switching assembly 34, and the secondary-side coil form a current loop, and the secondary-side leakage inductance Lks is freewheeling through the secondary-side switching assemblies 31 and 34. In the current loop, the current direction of the secondary-side switching assembly 31 as the non-control switching assembly is the anode to the cathode of the diode D1, and the current direction of the secondary-side control switching 34 is the anode to the cathode of the diode D4.

[0107] In the positive half cycle of the secondary-side input signal, during the conduction of the secondary-side switching assembly 33, the secondary-side leakage inductance Lks is short-circuited to excite by the conduction of the secondary-side switching assemblies 33 and 31, and during the turn-off of the secondary-side switching assembly 33, the secondary-side leakage inductance Lks is freewheeling through the secondary-side switching assemblies 31 and 34, thereby realizing boost current conversion.

[0108] In the negative half cycle of the secondary-side input signal, when the secondary-side switching assemblies 34 and 32 are turned on and the secondary-side switching assemblies 31 and 33 are turned off (for example, at time t1'-t11'), the equivalent circuit diagram of the secondary-side circuit is as shown in Figure 7The equivalent circuit diagram of the secondary side circuit when the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on (for example, at the time of t11'-t12') can refer to

[0109] The equivalent circuit diagram of the secondary side circuit when the secondary side switch components 31 and 34 are turned off and the secondary side switch components 32 and 33 are turned on (for example, at the time of t11'-t12') can refer to Figure 8 . In the current loop, the current direction of the secondary side switch component 32 as the non-control switch component is from the anode to the cathode of the diode D2, and the current direction of the secondary side control switch 33 is from the anode to the cathode of the diode D3.

[0110] In the negative half cycle of the secondary side input signal, during the conduction of the secondary side switch component 34, the secondary side leakage inductance Lks is conducted through the secondary side switch components 34 and 32 to short circuit excitation, and during the turn-off of the secondary side switch component 34, the secondary side leakage inductance Lks is conducted through the secondary side switch components 32 and 33 to freewheel, thereby realizing boost rectification.

[0111] In summary, in the positive half cycle and the negative half cycle of the secondary side input signal, the secondary side leakage inductance Lks is conducted through the secondary side switch components 33 and 34 to short circuit excitation, and is conducted through the secondary side switch components 32 and 33, and 31 and 34 to freewheel, realizing full-wave boost rectification, thereby realizing self-stabilization of the secondary side.

[0112] In the above embodiment, the driving signals SW2 and SW1 of the secondary side switch components 32 and 31 are respectively placed high according to the phase of the primary side in time, and synchronous rectification is performed to reduce loss. In other optional implementation manners, the driving signals of the secondary side switch components 32 and 31 can also be kept as 0 at all times, and the freewheeling is performed by the diodes connected in parallel thereto.

[0113] The driving signals of each secondary side switch component in the above-mentioned embodiments two to five can be synchronized with the primary side switch frequency. In other optional implementation manners, synchronization can also not be performed. The following takes the second secondary side switch component 32 and the fourth secondary side switch component 34 as control switch components as an example for specific description. It should be understood that the above-mentioned other control modes can also not be synchronized, and the control modes are similar to this, and are not described herein again.

[0114] Figure 17 is the schematic diagram of the secondary side circuit of the sixth embodiment of the present application. Figure 17 Another control strategy of the boost full-wave rectification is shown, which is relative to Figure 3 The secondary side circuit 17 of the present embodiment does not need a phase detection circuit, nor does it need to phase-gate the secondary side switching components 32 and 34, which further simplifies the circuit and the control strategy.

[0115] Figure 18 is the schematic diagram of the driving signals of the secondary side switching components of the sixth embodiment of the present application. As Figure 18 shown, in the present embodiment, the secondary side switching components 32 and 34 have the same driving signal, which is PWM signal SW as the driving signal in the whole cycle of the secondary side input signal, and the secondary side switching components 31 and 33 have the same driving signal, which is PWM signal SW' as the driving signal in the whole cycle of the secondary side input signal. Wherein, the driving signal SW and the driving signal SW' are complementary.

[0116] In the positive half cycle of the secondary side input signal, when the secondary side switching components 32 and 34 are turned on and the secondary side switching components 31 and 33 are turned off, the equivalent circuit diagram of the secondary side circuit can refer to Figure 5 . As Figure 5 shown, in the positive half cycle of the secondary side input signal, the secondary side leakage inductance Lks, the secondary side switching component 32, the secondary side switching component 34 and the secondary side coil form a current loop. The turn-on of the secondary side switching components 32 and 34 makes the secondary side leakage inductance Lks short-circuit excitation, and the excitation time is determined by the common turn-on time of the secondary side switching components 32 and 34. When the secondary side switching components 32 and 34 are turned off, the secondary side leakage inductance Lks continues to flow through the turn-on of the secondary side switching components 31 and 33, and the equivalent circuit diagram can refer to Figure 11 , which realizes full-wave boost rectification, thereby realizing the self-voltage stabilization of the secondary side.

[0117] In the negative half cycle of the secondary side input signal, when the secondary side switching components 32 and 34 are turned on and the secondary side switching components 31 and 33 are turned off, the equivalent circuit diagram of the secondary side circuit can refer to Figure 7 . As Figure 7 shown, in the negative half cycle of the secondary side input signal, the secondary side leakage inductance Lks, the secondary side switching component 34, the secondary side switching component 32 and the secondary side coil form a current loop, and the turn-on of the secondary side switching components 32 and 34 makes the secondary side leakage inductance Lks short-circuit excitation, and the excitation time is determined by the common turn-on time of the secondary side switching components 32 and 34. When the secondary side switching components 32 and 34 are turned off, the secondary side leakage inductance Lks continues to flow through the turn-on of the secondary side switching components 31 and 33, and the equivalent circuit diagram can refer to Figure 13 , which realizes full-wave boost rectification, thereby realizing the self-voltage stabilization of the secondary side.

[0118] In other optional implementations, the drive signals of the secondary side switch assemblies 31 and 33 can also be kept low at all times to continue to flow through the diodes D1 and D3 therein, which still can achieve full-wave boost rectification and further achieve self-stabilization of the secondary side voltage.

[0119] Further optionally, in the embodiment, when the secondary side switch frequency is much greater than the primary side switch frequency, that is, the ratio of the secondary side switch frequency to the primary side switch frequency is greater than a predetermined value (or the difference between the secondary side switch frequency and the primary side switch frequency is greater than a predetermined value), there is no need to detect the positive and negative phases of the secondary side input signal and synchronize the drive signals of the secondary side switch assemblies, which avoids load energy backflow and further improves the regulation efficiency. At the same time, this further simplifies the control strategy while ensuring voltage stabilization, as there is no need for a phase detection circuit, which further simplifies the circuit and reduces the cost.

[0120] In summary, from the equivalent circuit diagrams of the various control strategies of the boost full-wave rectification circuit described above, it can be concluded that, whether the control strategy is to take the secondary side switch assemblies 32 and 34 as the control switch assemblies (or the secondary side switch assemblies 31 and 33 as the control switch assemblies) or to take the secondary side switch assemblies 31 and 32 as the control switch assemblies (or the secondary side switch assemblies 33 and 34 as the control switch assemblies), the secondary side leakage inductance Lks can be in the boost rectification mode. The difference between the two control strategies is that, in the entire primary side working period, the excitation paths of the secondary side leakage inductance Lks are different. In the first control strategy, corresponding to Embodiment Two and Embodiment Three, the excitation paths include one excitation path when the secondary side switch assemblies 32 and 34 (or the secondary side switch assemblies 31 and 33) are turned on (see Figure 5 ), while in the second control strategy, corresponding to Embodiment Four and Embodiment Five, the excitation paths include one excitation path when the secondary side switch assemblies 32 and 34 are turned on (see Figure 5 ) and another excitation path when the secondary side switch assemblies 31 and 33 are turned on (see Figure 11 ). In addition, when the secondary side frequency is much greater than the primary side frequency, the control mode of Embodiment Six can be selected, in which case there is no need to detect the positive and negative phases of the secondary side input signal and synchronize the drive signals of the secondary side switch assemblies.

[0121] Figure 19 is a schematic diagram of the secondary side circuit of Embodiment Seven of the present application. In this embodiment, the current conversion circuit is a boost half-wave rectification circuit. Optionally, the switching frequency of the secondary side switch assembly is not less than the switching frequency of the primary side switch assembly, so that the primary and secondary side energy transmission times have differences. For example, Figure 19As shown, in an embodiment, the secondary side circuit 19 of the present embodiment comprises a current conversion circuit composed of a secondary side leakage inductance Lks' and a secondary side switch network, and the secondary side switch network comprises secondary side switch components 41-43. Among them, the secondary side switch component 41 is connected between the first input terminal i1' of the secondary side switch network and the first output terminal o1' of the secondary side circuit 29, the secondary side switch component 42 is connected between the first input terminal i1' of the secondary side switch network and the second output terminal o2' of the secondary side circuit 29, and the secondary side switch component 43 is connected between the second input terminal i2' of the secondary side switch network and the first output terminal o2' of the secondary side circuit 29.

[0122] In an optional implementation, the secondary side switch component comprises a switch element and a diode connected in parallel. Further, the switch element can be an active switch device. As shown in the figure, Figure 19 As shown, the secondary side switch component 41 comprises a switch element S1' and a diode D1', the secondary side switch component 42 comprises a switch element S2' and a diode D2', and the secondary side switch component 43 comprises a switch element S3' and a diode D3'. In other optional implementations, the secondary side switch component can also be a transistor element with a parasitic diode, such as a MOS tube, etc.

[0123] In an optional implementation, the secondary side switch component 42 in the present embodiment serves as a control switch component. That is, in the present embodiment, the corresponding driving signal is obtained by the error acquisition circuit 44 and the driving circuit 45 in the feedback control circuit to drive the secondary side switch component 42 to turn on or turn off. Further, during the positive half cycle of the secondary side input signal, the driving signal of the secondary side switch component 41 is complementary to the secondary side switch component 42, and the driving signal of the secondary side switch component 43 remains in a constant on state. Since Figure 19 As shown in the figure, the secondary side circuit comprises a boost half-wave rectifier circuit, so the secondary side switch components 41-43 are all low during the negative half cycle of the secondary side input signal, that is, remain in a turn-off state. And the diode in the secondary side switch component 43 of the present embodiment has a reverse blocking effect, so that no energy is transmitted to the secondary side, thereby achieving half-wave rectification. The half-wave rectification of the present embodiment takes the positive phase selection as an example, and it should be understood that the control strategy of the present embodiment can also be applied to the half-wave rectification of the negative phase selection (the secondary side switch components 41-43 remain in a turn-off state during the positive half cycle of the secondary side input signal), and the present embodiment does not limit this.

[0124] In the present embodiment, the secondary side leakage inductance Lks is short-circuited by the secondary side switch component 42 and the secondary side switch component 43 for excitation, and discharged to the output terminal by the secondary side switch component 41 and the secondary side switch component 43, so that the secondary side circuit plays a role of boost rectification, thereby realizing secondary side voltage stabilization.

[0125] In an alternative implementation, in the embodiment, the error acquisition circuit 44 can include a sampling circuit 441 and a comparison circuit A3. The sampling circuit 441 is configured to sample an output feedback signal Vf representing the output voltage. The comparison circuit A3 is configured to compare the output feedback signal Vf with a reference signal Vref representing the desired value of the output voltage, and acquire an error signal Ve. The drive circuit 45 further includes a comparator A4 configured to generate complementary PWM signals SW and SW' according to the error signal Ve and a predetermined ramp signal RAMP. In the embodiment, the PWM signal is positive-phase gated to acquire the drive signal SW2 of the secondary-side switching component 42. The drive signal SW1 of the secondary-side switching component 41 is complementary to the drive signal SW2 of the secondary-side switching component 42.

[0126] In an alternative implementation, in the embodiment, the switching frequency of the secondary-side switching component is an integer multiple of the switching frequency of the primary-side switching component, and the drive signal of the secondary-side switching component is synchronized with the primary-side switching component. Specifically, the embodiment can synchronize the ramp signal RAMP by detecting the rising edge of the secondary-side input signal, so as to keep the ramp signal RAMP synchronized with the switching frequency of the primary-side switching component.

[0127] Further alternatively, the secondary-side circuit 19 can further include a positive-negative phase detection circuit (not shown in the figure) configured to detect the positive-negative phase of the secondary-side input signal, so as to control the secondary-side switching components to switch different operation modes. Figure 19

[0128] Alternatively, the positive-negative phase detection circuit can be implemented by zero-crossing detection, which is not limited in the embodiment. Alternatively, the positive-negative phase detection circuit can detect the signal at the endpoint b1, or the endpoint b2, or the endpoint b3, so as to acquire the positive-negative phase of the input signal, and the phase detection point is not limited in the embodiment.

[0129] Figure 20 is a schematic diagram of the drive signal of the secondary-side switching component of the seventh embodiment of the application. In the positive half cycle of the secondary-side input signal, the drive signal SW1 of the secondary-side switching component 41 is complementary to the drive signal SW2 of the secondary-side switching component 42, and the drive signal SW3 of the secondary-side switching component 43 is kept high, i.e., kept always on. In the negative half cycle of the secondary-side input signal, the drive signals SW1-SW3 of the secondary-side switching components 41-43 are all kept low, i.e., kept always off.

[0130] As shown in Figure 20 , the gate signal SW on Pri is positively set high in the positive half cycle of the secondary-side input signal, and is negatively set high in the negative half cycle of the secondary-side input signal, so as to gate the PWM signal SW as the drive signal of the secondary-side switching component 42 in the positive half cycle of the secondary-side input signal. ​

[0131] Figures 21-22 is the equivalent circuit diagram of the secondary side circuit of the seventh embodiment of the present application. In the positive half cycle of the secondary side input signal, when the secondary side switching components 42 and 43 are turned on and the secondary side switching component 41 is turned off (for example, at time t3-t31), the equivalent circuit diagram of the secondary side circuit 19 is as shown in Figure 21 . In which, the secondary side leakage inductance Lks', the secondary side switching component 42, the secondary side switching component 43 and the secondary side coil form a current loop. In which, the secondary side leakage inductance Lks' is short-circuited by the secondary side switching component 42 and the secondary side switching component 43 to excite the magnetism, and the excitation time is determined by the turn-on time of the secondary side switching component 42.

[0132] When the secondary side switching component 42 is turned off and the secondary side switching components 41 and 43 are turned on (for example, at time t31-t32), the equivalent circuit diagram of the secondary side circuit 19 is as shown in Figure 22 . In which, the secondary side leakage inductance Lks', the secondary side switching component 41, the output equivalent resistance Ro', the secondary side switching component 43 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks' is freewheeling through the secondary side switching component 41 and the secondary side switching component 43.

[0133] In the positive half cycle of the secondary side input signal, during the turn-on period of the secondary side switching component 42, the secondary side leakage inductance Lks' is short-circuited by the secondary side switching component 42 and the secondary side switching component 43 to excite the magnetism, during the turn-off period of the secondary side switching component 42, the secondary side leakage inductance Lks' is freewheeling through the secondary side switching component 41 and the secondary side switching component 43, and the secondary side switching components 41-43 are kept turned off in the negative half cycle of the secondary side input signal, thereby realizing half-wave rectification, and further realizing voltage stabilization and reducing loss.

[0134] In other optional implementations, the primary side switching component can also not be synchronized, which further simplifies the circuit and control strategy relative to Figure 19 the secondary side circuit shown in

[0135] Figure 23 is the schematic diagram of the secondary side circuit of the eighth embodiment of the present application. Figure 23 shows another control strategy of half-wave rectification, and relative to the secondary side circuit shown in Figure 19 , the secondary side circuit of the present embodiment does not need a phase detection circuit, and also does not need to phase-gate the secondary side switching component 42, which further simplifies the circuit and control strategy.

[0136] Figure 24 is the schematic diagram of the driving signal of the secondary side switching component of the eighth embodiment of the present application. As shown in Figure 24 , in the present embodiment, the driving signal of the secondary side switching component 42 is SW2, and the secondary side switching components 41 and 43 are kept in the turned-off state, and the secondary side switching components 41 and 43 can use respective anti-parallel diodes D1', D3' for unidirectional conduction.

[0137] When the secondary side switch assembly 42 is turned on (for example, at time t3-t31), the equivalent circuit diagram of the secondary side circuit can refer to Figure 21 . Among them, the secondary side leakage inductance Lks, the secondary side switch assembly 32, the diode D3' in the secondary side switch assembly 34 and the secondary side coil form a current loop. Among them, the secondary side leakage inductance Lks' is excited by the short circuit of the secondary side switch assembly 42, and the excitation time is determined by the conduction time of the secondary side switch assembly 42. When the secondary side switch assembly 42 is turned off (for example, at time t31-t32), the equivalent circuit diagram of the secondary side circuit can refer to Figure 22 . Among them, the secondary side leakage inductance Lks', the diode D1' in the secondary side switch assembly 41, the output equivalent resistance Ro', the diode D3' in the secondary side switch assembly 43 and the secondary side coil form a current loop, and the secondary side leakage inductance Lks' is freewheeling through the diodes D1' and D3' in the secondary side switch assemblies 41 and 43. Thus, the present embodiment can realize half-wave voltage boosting rectification, and further realize self-stabilization of the secondary side and reduce power consumption.

[0138] Further optionally, in the present embodiment, when the secondary side switch frequency is much larger than the primary side switch frequency, that is, the ratio of the secondary side switch frequency to the primary side switch frequency (or the difference between the secondary side switch frequency and the primary side switch frequency) is greater than a predetermined value, it is not necessary to detect the positive and negative phases of the secondary side input signal and synchronize the driving signals of the secondary side switch assembly, which avoids load energy backflow and further improves regulation efficiency. At the same time, this further simplifies the control strategy while ensuring voltage stabilization, as there is no need for a phase detection circuit, the circuit is further simplified, and the cost is reduced.

[0139] In summary, the present embodiment uses leakage inductance as an energy storage element, and performs closed-loop modulation control on the secondary side switch assembly, while the primary side switch assembly can perform open-loop energy transmission at a fixed duty ratio or fixed frequency, thereby avoiding the problems of rising of component voltage stress and reduction of current conversion efficiency caused by the increase of leakage inductance in high-frequency transformer applications. In addition, compared with the traditional two-stage circuit, that is, the primary side uses closed-loop control and the secondary side uses switching control, the present application changes the control object from the primary side to the secondary side, which reduces the primary and secondary side signal isolation devices in the system, and reduces the system cost and complexity.

[0140] Figure 25 is a flow chart of the secondary side voltage stabilization control method of the present embodiment. As shown in Figure 25 , the secondary side voltage stabilization control method of the present embodiment comprises the following steps:

[0141] Step S110, sampling and acquiring an output feedback signal. The output feedback signal is used to represent the output signal.

[0142] Step S120, comparing the output feedback signal and the reference signal to obtain an error signal of the output signal. The reference signal is used to represent the expected value of the output signal.

[0143] Step S130, obtaining a driving signal of the secondary side switching component in the secondary side variable current circuit according to the error signal and a pre-set ramp signal. The ramp signal is used to determine the switching frequency of the secondary side switching component.

[0144] Step S140, adjusting the switching state of the corresponding secondary side switching component according to the driving signal to regulate the output signal.

[0145] The leakage inductance is used as an energy storage element in the embodiment of the present application, and the secondary side switching component is controlled in a closed loop. The primary side switching component can be used for open loop energy transmission with a fixed duty ratio or a fixed frequency, so that the problems of the increase of the voltage stress of the components and the decrease of the variable current efficiency caused by the increase of the leakage inductance in the high frequency transformer are avoided. In addition, compared with the traditional two-stage circuit, i.e., the primary side is controlled in a closed loop and the secondary side is controlled in a switching mode, the control object is changed from the primary side to the secondary side in the present application, so that the signal isolation device of the primary side and the secondary side is reduced, and the system cost and complexity are reduced.

[0146] The above description is only the preferred embodiment of the present application, and is not used to limit the present application. Any modification, equivalent replacement, improvement, etc. within the spirit and principle of the present application should be included in the protection scope of the present application.

Claims

1. An isolated power converter, characterized in that, The isolated power converter includes: The primary circuit is configured to generate an AC input signal; A transformer, including a primary winding and a secondary winding, is configured to process the AC input signal to generate a secondary input signal; The secondary circuit includes a converter circuit and a feedback control circuit. The converter circuit is configured to convert the secondary input signal to generate an output signal. The converter circuit achieves this conversion by energizing the secondary leakage inductance of the secondary winding via a secondary switching component in the secondary switching network. The feedback control circuit is configured to control the switching state of the secondary switching component in the secondary switching network based on an output feedback signal characterizing the output signal. The feedback control circuit is also configured to generate an error signal based on the output feedback signal and a reference signal, and to generate a drive signal based on the error signal and a ramp signal to control the switching state of each of the secondary switching components. The switching frequency of the secondary-side switch assembly is greater than that of the primary-side switch assembly. The duty cycle of the primary-side switch assembly is independent of the duty cycle of the secondary-side switch assembly. The switching frequency of the secondary-side switch assembly is determined based on the frequency of the ramp signal.

2. The isolated power converter according to claim 1, characterized in that, The primary-side circuit includes a primary-side impedance network and a primary-side switching network, wherein the primary-side switching components in the primary-side switching network are subjected to open-loop control with a fixed duty cycle or a fixed frequency.

3. The isolated power converter according to claim 1, characterized in that, The switching frequency of the secondary-side switch assembly is an integer multiple of the switching frequency of the primary-side switch assembly.

4. The isolated power converter according to claim 1, characterized in that, The feedback control circuit is further configured to adjust the duty cycle of each of the secondary-side switching components based on the error between the output feedback signal and the reference signal, so that the output signal tends toward the desired value.

5. The isolated power converter according to any one of claims 1-4, characterized in that, The converter circuit is an AC-DC rectifier circuit.

6. The isolated power converter according to claim 5, characterized in that, The secondary-side switch assembly includes at least one control switch assembly; The feedback control circuit is configured to perform chopping control on the corresponding control switch component during the positive or negative half-cycle of the secondary input signal, so that the secondary leakage current is magnetized to achieve current conversion.

7. The isolated power converter according to claim 6, characterized in that, The converter circuit is a full-wave rectifier circuit, and the switching frequency of the secondary-side switching component is not less than twice the switching frequency of the primary-side switching component. The secondary-side switch network includes a first secondary-side switch assembly, a second secondary-side switch assembly, a third secondary-side switch assembly, and a fourth secondary-side switch assembly; Wherein, the first secondary-side switch assembly is coupled between the first input terminal of the secondary-side switch network and the first output terminal of the secondary-side circuit, the second secondary-side switch assembly is coupled between the first input terminal of the secondary-side switch network and the second output terminal of the secondary-side circuit, the third secondary-side switch assembly is coupled between the second input terminal of the secondary-side switch network and the first output terminal of the secondary-side circuit, and the fourth secondary-side switch assembly is coupled between the second input terminal of the secondary-side switch network and the second output terminal of the secondary-side circuit.

8. The isolated power converter according to claim 7, characterized in that, The converter circuit is a boost full-wave rectifier circuit. The feedback control circuit is configured to perform chopping control on the second or third secondary-side switching component during the positive half-cycle of the secondary-side input signal and on the first or fourth secondary-side switching component during the negative half-cycle of the secondary-side input signal, so that the secondary-side leakage current is magnetized to achieve converter operation.

9. The isolated power converter according to claim 8, characterized in that, During the positive half-cycle of the secondary input signal, the second secondary-side switching component acts as a control switching component, and during the negative half-cycle of the secondary input signal, the fourth secondary-side switching component acts as a control switching component. The feedback control circuit is configured to control the second and fourth secondary-side switching components to conduct during the positive half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, control the first and fourth secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit, control the fourth and second secondary-side switching components to conduct during the negative half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, and control the third and second secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit.

10. The isolated power converter according to claim 8, characterized in that, During the positive half-cycle of the secondary input signal, the third secondary switch component acts as a control switch component, and during the negative half-cycle of the secondary input signal, the first secondary switch component acts as a control switch component. The feedback control circuit is configured to control the first and third secondary-side switching components to conduct during the positive half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, control the first and fourth secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit, control the third and first secondary-side switching components to conduct during the negative half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, and control the third and second secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit.

11. The isolated power converter according to claim 8, characterized in that, During the positive half-cycle of the secondary input signal, the second secondary switch component acts as a control switch component, and during the negative half-cycle of the secondary input signal, the first secondary switch component acts as a control switch component. The feedback control circuit is configured to control the second and fourth secondary-side switching components to conduct during the positive half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, control the first and fourth secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit, control the third and first secondary-side switching components to conduct during the negative half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, and control the third and second secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit.

12. The isolated power converter according to claim 8, characterized in that, During the positive half-cycle of the secondary input signal, the third secondary switch component acts as a control switch component, and during the negative half-cycle of the secondary input signal, the fourth secondary switch component acts as a control switch component. The feedback control circuit is configured to control the first and third secondary-side switching components to conduct during the positive half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, control the first and fourth secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit, control the fourth and second secondary-side switching components to conduct during the negative half-cycle of the secondary-side input signal to magnetize the secondary-side leakage current, and control the third and second secondary-side switching components to conduct to release energy to the output terminal of the secondary-side circuit.

13. The isolated power converter according to any one of claims 9-12, characterized in that, The drive signal of the first secondary-side switch assembly is complementary to the drive signal of the second secondary-side switch assembly, and the drive signal of the third secondary-side switch assembly is complementary to the drive signal of the fourth secondary-side switch assembly.

14. The isolated power converter according to any one of claims 9-12, characterized in that, The secondary-side switching assembly includes a switching element and a diode connected in parallel, or the secondary-side switching assembly is a transistor element with a parasitic diode; Wherein, when the control switch assembly is turned on, the current direction is from the corresponding diode cathode to the anode, and when the non-control switch assembly is turned on, the current direction is from the corresponding diode anode to the cathode. The non-control switch assembly is a secondary switch assembly other than the control switch assembly during operation.

15. The isolated power converter according to claim 7, characterized in that, The drive signals for the control switch components corresponding to the positive half-cycle of the secondary input signal and the control switch components corresponding to the negative half-cycle of the secondary input signal are the same.

16. The isolated power converter according to claim 6, characterized in that, The converter circuit is a boost half-wave rectifier circuit, and the converter circuit includes a fifth secondary-side switching assembly, a sixth secondary-side switching assembly, and a seventh secondary-side switching assembly. The fifth secondary-side switch assembly is coupled between the first input terminal of the secondary-side switch network and the first output terminal of the secondary-side circuit; the sixth secondary-side switch assembly is coupled between the first input terminal of the secondary-side switch network and the second output terminal of the secondary-side circuit; and the seventh secondary-side switch assembly is coupled between the second input terminal of the secondary-side switch network and the second output terminal of the secondary-side circuit.

17. The isolated power converter according to claim 16, characterized in that, During the positive half-cycle of the secondary input signal, the sixth secondary switch assembly acts as a control switch assembly. The secondary leakage inductance is energized via the sixth and seventh secondary switch assemblies, and energy is released to the output terminal via the fifth and seventh secondary switch assemblies.

18. The isolated power converter according to claim 17, characterized in that, The sixth secondary-side switch component performs chopping control during the positive half-cycle of the secondary-side input signal. The drive signal of the fifth secondary-side switch component is complementary to that of the sixth secondary-side switch component. The seventh secondary-side switch component remains normally on during the positive half-cycle of the secondary-side input signal. The fifth, sixth, and seventh secondary-side switch components all remain off during the negative half-cycle of the secondary-side input signal.

19. The isolated power converter according to claim 16, characterized in that, The sixth secondary-side switch assembly performs chopping control throughout the entire cycle of the secondary-side input signal, while the drive signals of the fifth and seventh secondary-side switch assemblies remain low.

20. The isolated power converter according to claim 1, characterized in that, The feedback control circuit further includes: An error acquisition circuit is configured to acquire an error signal based on the difference between the feedback signal and a reference signal; and The driving circuit is configured to generate a PWM signal based on the error signal and the ramp signal for chopping control, thereby determining the driving signal for each of the secondary-side switching components and controlling the switching state of the corresponding secondary-side switching components according to the driving signal.

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