IPMSM magnetic weakening control system and control method based on new sliding mode control

Abstract

This invention proposes a novel IPMSM field weakening control system and method based on sliding mode control, relating to the technical field of motor field weakening control. Building upon traditional field weakening control for embedded permanent magnet synchronous motors, a sliding mode speed control loop is designed based on a novel sliding mode approach law. This loop is applied to the field weakening control of embedded permanent magnet synchronous motors, improving the robustness, dynamic response speed, and resistance to external disturbances of the system. Simultaneously, it enhances the sliding mode approach rate and suppresses system chattering.

Description

Technical Field

[0001] The present invention relates to the technical field of motor weak magnetic field control, and more specifically, to an IPMSM weak magnetic field control system and a control method based on a novel sliding mode control. Background Art

[0002] Interior Permanent Magnet Synchronous Motor (IPMSM) has a series of advantages such as high torque density, high power density and high efficiency, and is widely used in electric transmission fields such as electric vehicles.

[0003] The operation control of built-in permanent magnet synchronous motors is crucial. Commonly used methods such as maximum torque per current ratio (MTPA) control are suitable for operation below the base speed. In an environment where the motor is required to run at a high speed, a flux-weakening (FW) control strategy must be adopted to achieve this. During the flux-weakening speed expansion process of built-in permanent magnet synchronous motors, the speed loop controller must meet the requirements of fast response, small error, high accuracy, and strong anti-interference ability.

[0004] In order to meet the requirements of the speed control loop of the built-in permanent magnet synchronous motor for weak magnetic field control, scholars have proposed methods such as sliding mode control, adaptive control, and neural network control. Among them, sliding mode control has the advantages of being insensitive to parameter changes, strong anti-interference ability, and good robustness, and is widely used in the design of the speed loop of the built-in permanent magnet synchronous motor. However, there are problems such as slow speed of state variables approaching the sliding surface and severe jitter in sliding mode control. To address this problem, many researchers have optimized it by improving the sliding mode reaching law. For example, in the published article on weak magnetic field vector control of built-in permanent magnet synchronous motor based on sliding mode variable structure, it is proposed to use sliding mode variable structure to design the speed loop to improve the weak magnetic performance of the built-in permanent magnet synchronous motor. The speed loop uses a sliding mode controller to replace the traditional PI controller, which improves the system response speed and stability to a certain extent. However, due to the introduction of sliding mode control, the system has severe jitter; in the public patent "Variable Exponential Power Reaching Law Sliding Mode and Its PMSM Control Application", a variable exponential reaching law is used to design a sliding mode speed controller. Although this scheme improves the sliding mode approach speed to a certain extent, the approach speed is limited when the system state is far away from the sliding surface, and it cannot meet the requirements of the permanent magnet synchronous motor. The requirement for fast response of the magnetic synchronous motor's weak magnetic algorithm; in the published patent "Sliding mode speed control method for built-in permanent magnet synchronous motor with improved reaching law", the reaching speed of the sliding mode speed controller is improved by improving the sliding mode reaching law, but the constant speed arrival term introduced in the improved reaching law leads to increased system jitter when the system state approaches the sliding surface; in the patent "A PMSM speed control method based on a new reaching law", a new sliding mode speed controller is designed, and the introduction of system state variables and sliding mode function power terms in the reaching law improves the sliding mode reaching speed to a certain extent, but there is still a large switching gain when the system state approaches the sliding surface, and the system jitter problem is not solved. Summary of the invention

[0005] In order to solve the problems of slow sliding mode approach speed and severe jittering during the weak magnetic field control of the built-in permanent magnet synchronous motor, the present invention proposes an IPMSM weak magnetic field control system and a control method based on a novel sliding mode control, and designs a sliding mode speed control loop with a novel sliding mode approach law, thereby improving the robustness, dynamic response speed and ability to resist external interference of the weak magnetic field control system of the built-in permanent magnet synchronous motor, while increasing the sliding mode approach rate and suppressing system jittering.

[0006] In order to achieve the above technical effects, the technical solution of the present invention is as follows:

[0007] An IPMSM weak magnetic control system based on a novel sliding mode control, comprising:

[0008] The speed detection module is used to detect the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor. m , electrical angle θ and electrical angular velocity ω e ;

[0009] Clark transformation module is used to perform Clark transformation on the three-phase current of the stator to obtain the current i α andi β ;

[0010] Park transformation module is used to convert the current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q ;

[0011] The sliding mode speed controller SMC is designed based on the new sliding mode reaching law to design the sliding mode speed control loop and to give a mechanical angular velocity ω * m And the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m The difference is taken as input, and the electromagnetic torque T of the output motor is e ;

[0012] MTPA control module, with electromagnetic torque T e is the input, output current given value i′ d and i′ q ;

[0013] The weak magnetic speed regulating unit includes a weak magnetic speed regulating module, a voltage feedforward compensation module and a PI regulator. The weak magnetic speed regulating module, the voltage feedforward compensation module and the PI regulator form a closed-loop feedback. The weak magnetic speed regulating unit uses a current given value i′ d and i′ q , Motor DC voltage U dc , electrical angular velocity ω e , the actual value of the current in the rotating coordinate system i d andi q As input, the internal closed loop updates the q-axis voltage signal and the d-axis voltage signal, and uses the q-axis voltage signal and the d-axis voltage signal as inputs of the park inverse transformation module;

[0014] Park inverse transformation module, which performs Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal to obtain the voltage in the stationary coordinate system;

[0015] The SVPWM module takes the voltage in the stationary coordinate system as input and outputs a switching signal;

[0016] The inverter controls the speed of the built-in permanent magnet synchronous motor with the switching signal output by the SVPWM module.

[0017] Preferably, the obtained stator three-phase current of the interior permanent magnet synchronous motor is represented as i a ,i b ,i c, Clark transformation is performed on the stator three-phase current to obtain:

[0018]

[0019] The current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q :

[0020]

[0021] Preferably, a sliding mode speed control loop is designed based on a novel sliding mode reaching law to output the electromagnetic torque T of the motor. e The process is:

[0022] Establish two state variables x1 and x2 of the system:

[0023]

[0024] in, is the derivative of x1; ω m The derivative of

[0025] Motor electromagnetic torque equation, motor motion equation:

[0026]

[0027]

[0028] Among them, L d , L q is the d and q axis inductance and L d ≠L q ;P n is the pole pair number; is the permanent magnet flux; i d is the d-axis component of the current; i q is the q-axis component of the current; J is the moment of inertia of the motor shaft end; B is the viscous friction coefficient; T L is the load torque;

[0029] Combining the motor electromagnetic torque equation and the motor motion equation, we get:

[0030]

[0031] The sliding surface s of the sliding mode speed controller SMC is designed as:

[0032] s=x1+c∫0 t x1 dτ

[0033] in, is a state variable of the system, c is the design parameter of the sliding surface, and c>0;

[0034] Taking the derivative of s, we get:

[0035]

[0036] Combining the exponential reaching law and the power reaching law, the sliding mode reaching law of the new sliding mode speed controller SMC is obtained:

[0037]

[0038] Among them, ε, k, α, β>0, γ>1, x1 in the above formula is the system state variable, and k and ε are reaching law parameters, s is the sliding surface function; sgn(s) is the sign function; γ is the power coefficient of the absolute value of the sliding surface; λ is the power term exponent, and α and β are the constant term coefficients for adjusting the rate of change of λ.

[0039] Replace the sign function with a saturation function to satisfy:

[0040]

[0041] Where σ represents a small positive constant; the control law of the sliding mode speed controller SMC, that is, the electromagnetic torque T e :

[0042]

[0043] Here, a sliding mode speed control loop is designed based on a new sliding mode reaching law, wherein the state variable x1 is introduced into the designed reaching law, and the speed approaching the sliding surface can be adaptively adjusted according to the size of the state variable x1 and the distance of the system from the sliding surface s. In addition, the saturation function is used instead of the sign function, which is conducive to quickly tracking the given speed signal, weakening the system's jitter, and improving the system's dynamic response speed and robustness.

[0044] Preferably, the electromagnetic torque T e is the input, output current given value i′ d and i′ q The calculation process satisfies:

[0045]

[0046]

[0047] Preferably, the weak magnetic speed control module, the voltage feedforward compensation module and the PI regulator of the weak magnetic speed control unit form a closed-loop feedback to output the q-axis voltage signal and the d-axis voltage signal; the initial q-axis voltage signal and the d-axis voltage signal, the motor DC voltage U dc And the current set value i′ output by the MTPA control module d and i′ q Input the weak magnetic speed control module, and output the d and q axis current values ​​after weak magnetic speed control Will As the input of the q-axis current loop PI controller, As the input of the d-axis current loop PI controller, the output voltage signal u d and u q ; and the d and q axis current values ​​after weak magnetic speed regulation and electrical angular velocity ω e Input the voltage feedforward compensation module together and output the compensation voltage signal u′ q , u′ d , the voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is obtained by adding and Compensated voltage signal and The initial q-axis voltage signal and d-axis voltage signal are the q-axis voltage signal and d-axis voltage signal after closed-loop update by the weak magnetic speed control unit, and serve as the input of the weak magnetic speed control module for the next feedback.

[0048] Preferably, the initial q-axis voltage signal and d-axis voltage signal, the motor DC voltage U dc When the current given values ​​i′d and i′q output by the MTPA control module are input into the weak magnetic speed control module, the following are used:

[0049]

[0050]

[0051] Determine whether the built-in permanent magnet synchronous motor needs to be subjected to magnetic field weakening speed regulation. If so, the d-axis and q-axis current values ​​after magnetic field weakening speed regulation are expressed as:

[0052]

[0053] Otherwise, the d and q axis currents remain unchanged;

[0054] Among them, U smax is the voltage limit value, I smax is the current limit value, Udc is the inverter DC bus voltage, θ′ is the weak magnetic regulation output angle.

[0055] Preferably, the d and q axis current values ​​after weak magnetic speed regulation are and electrical angular velocity ω e Input the voltage feedforward compensation module together and output the compensation voltage signal u′ q , u′ d The expression is:

[0056]

[0057] Preferably, the voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is obtained by adding and The expression is:

[0058]

[0059] The Park inverse transformation module performs Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal to obtain the voltage in the stationary coordinate system. The expression is:

[0060]

[0061] in, The voltage in the stationary coordinate system is obtained by inverse transformation of Park.

[0062] The present application also proposes an IPMSM weak magnetic control method based on a novel sliding mode control, the method comprising the following steps:

[0063] S1. Obtain the stator three-phase current of the built-in permanent magnet synchronous motor and detect the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m , electrical angle θ and electrical angular velocity ω e ;

[0064] S2. Perform Clark transformation on the three-phase current of the stator to obtain the current i α andi β , the current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q ;

[0065] S3. With a given mechanical angular velocity ω * m And the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motorm The difference is taken as input, and the electromagnetic torque T of the output motor is e ;

[0066] S4. With electromagnetic torque T e is the input, output current given value i′ d and i′ q ;

[0067] S5. The weak magnetic speed control unit uses the current given value i' d and i′ q , Motor DC voltage U dc , electrical angular velocity ω e , the actual value of the current in the rotating coordinate system i d andi q As input, the internal closed loop updates the q-axis voltage signal and the d-axis voltage signal;

[0068] S6. Perform a Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal using a Park inverse transformation module to obtain the voltage in a stationary coordinate system;

[0069] S7. Use the voltage in the stationary coordinate system as the input of the SVPWM module, output a switching signal, and control the speed of the built-in permanent magnet synchronous motor with the switching signal output by the SVPWM module.

[0070] Preferably, in step S3, the electromagnetic torque T of the output motor is e When , two state variables x1 and x2 of the system are established:

[0071]

[0072] in, is the derivative of x1; ω m The derivative of

[0073] Establish the motor electromagnetic torque equation and motor motion equation:

[0074]

[0075]

[0076] Among them, L d , L q is the d and q axis inductance and L d ≠L q ;P n is the pole pair number; is the permanent magnet flux; i d is the d-axis component of the current; i qis the q-axis component of the current; J is the moment of inertia of the motor shaft end; B is the viscous friction coefficient; T L is the load torque;

[0077] Combining the motor electromagnetic torque equation and the motor motion equation, we get:

[0078]

[0079] The sliding surface s of the sliding mode speed controller SMC is designed as:

[0080] s=x1+c∫0 t x1 dτ

[0081] in, x1 is a state variable of the system, c is the design parameter of the sliding surface, and c>0;

[0082] Taking the derivative of s, we get:

[0083]

[0084] Combining the exponential reaching law and the power reaching law, the sliding mode reaching law of the new sliding mode speed controller SMC is obtained:

[0085]

[0086] Among them, ε, k, α, β>0, γ>1, x1 in the above formula is the system state variable, and ε and k are reaching law parameters, s is the sliding surface function; sgn(s) is the sign function; γ is the power coefficient of the absolute value of the sliding surface; λ is the power term exponent, and α and β are the constant term coefficients that adjust the rate of change of λ.

[0087] Replace the sign function with a saturation function to satisfy:

[0088]

[0089] Where σ represents a small positive constant; the control law of the sliding mode speed controller SMC, that is, the electromagnetic torque T e :

[0090]

[0091] Compared with the prior art, the technical solution of the present invention has the following beneficial effects:

[0092] The present invention proposes an IPMSM weak magnetic field control system and a control method based on a novel sliding mode control. On the basis of the traditional built-in permanent magnet synchronous motor weak magnetic field control, a sliding mode speed control loop is designed based on a novel sliding mode reaching law. The sliding mode speed control loop is applied to the built-in permanent magnet synchronous motor weak magnetic field control, which improves the robustness, dynamic response speed and ability to resist external interference of the built-in permanent magnet synchronous motor weak magnetic field control system, while improving the sliding mode reaching rate and suppressing system chattering. BRIEF DESCRIPTION OF THE DRAWINGS

[0093] Figure 1 A schematic diagram showing a magnetic field weakening control system of the IPMSM proposed in Example 1 of the present invention;

[0094] Figure 2 A schematic flow chart showing the magnetic field weakening control method of the IPMSM proposed in Embodiment 3 of the present invention;

[0095] Figure 3 A schematic diagram showing the rotation speed of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the magnetic field weakening control of a traditional sliding mode controller;

[0096] Figure 4 A schematic diagram showing the rotation speed of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the magnetic field weakening control of the IPMSM proposed in the present invention;

[0097] Figure 5 A schematic diagram showing the d-axis current of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the magnetic field weakening control of the conventional sliding mode controller;

[0098] Figure 6 A schematic diagram showing the d-axis current of the interior permanent magnet synchronous motor proposed in Example 4 of the present invention based on the weak magnetic field control of the IPMSM proposed in the present invention;

[0099] Figure 7 A schematic diagram showing the q-axis current of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the magnetic field weakening control of the conventional sliding mode controller;

[0100] Figure 8 A schematic diagram showing the q-axis current of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the magnetic field weakening control of the IPMSM proposed in the present invention;

[0101] Fig. 9 A schematic diagram showing the output torque of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the magnetic weakening control of a traditional sliding mode controller;

[0102] Fig.10 A schematic diagram showing the output torque of the interior permanent magnet synchronous motor proposed in Embodiment 4 of the present invention based on the field weakening control of the IPMSM proposed in the present invention; DETAILED DESCRIPTION

[0103] The drawings are for illustrative purposes only and should not be construed as limiting the present patent;

[0104] In order to better illustrate the present embodiment, some parts of the drawings may be omitted, enlarged or reduced, and do not represent the actual size;

[0105] It is understandable to those skilled in the art that descriptions of certain well-known contents in the drawings may be omitted.

[0106] The technical solution of the present invention is further described below in conjunction with the accompanying drawings and embodiments.

[0107] The positional relationships described in the drawings are only for illustrative purposes and should not be construed as limiting the present patent;

[0108] Example 1

[0109] like Figure 1 As shown, this embodiment proposes an IPMSM weak magnetic control system based on a new sliding mode control, see Figure 1 , the system comprising:

[0110] The speed detection module is used to detect the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor. m , electrical angular velocity ω e In this embodiment, the mechanical angular velocity ω is detected by an encoder m , electrical angular velocity ω e and electrical angle θ;

[0111] Clark transformation module 1 is used to perform Clark transformation on the stator three-phase current to obtain the current i α andi β ;

[0112] Park conversion module 2 is used to convert the current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q ;

[0113] In this embodiment, the obtained stator three-phase current of the interior permanent magnet synchronous motor is represented as i a ,i b ,i c , Clark transformation is performed on the stator three-phase current to obtain:

[0114]

[0115] The current i α andiβ Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q :

[0116]

[0117] The sliding mode speed controller SMC is designed based on the new sliding mode reaching law to design the sliding mode speed control loop and to give a mechanical angular velocity ω * m And the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m The difference is taken as input, and the electromagnetic torque T of the output motor is e ;

[0118] MTPA control module, with electromagnetic torque T e is the input, output current given value i′ d and i′ q ;Here, the MTPA control module is based on the MTPA strategy;

[0119] Among them, the electromagnetic torque T e is the input, output current given value i′ d and i′ q The calculation process satisfies:

[0120]

[0121]

[0122] Weak field speed control unit 3, see Figure 1 The unit includes a weak magnetic speed control module, a voltage feedforward compensation module and a PI regulator, and a closed-loop feedback is formed between the weak magnetic speed control module, the voltage feedforward compensation module and the PI regulator; the weak magnetic speed control unit takes the current given value i′ d and i′ q、 Motor DC voltage U dc , electrical angular velocity ω e , the actual value of the current in the rotating coordinate system i d andi q As input, the internal closed loop updates the q-axis voltage signal and the d-axis voltage signal, and uses the q-axis voltage signal and the d-axis voltage signal as inputs of the park inverse transformation module;

[0123] The weak magnetic speed control module, the voltage feedforward compensation module and the PI regulator of the weak magnetic speed control unit form a closed-loop feedback to output the q-axis voltage signal and the d-axis voltage signal; set the initial q-axis voltage signal and the d-axis voltage signal, and convert the initial q-axis voltage signal and the d-axis voltage signal, the motor DC voltage U dc And the current set value i′ output by the MTPA control moduled and i′ q Input the weak magnetic speed control module, and output the d and q axis current values ​​after weak magnetic speed control Will As the input of the q-axis current loop PI controller, As the input of the d-axis current loop PI controller, the output voltage signal u d and u q ; and the d and q axis current values ​​after weak magnetic speed regulation and electrical angular velocity ω e Input the voltage feedforward compensation module together and output the compensation voltage signal u′ q , u′ d , the voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is obtained by adding and Compensated voltage signal and The initial q-axis voltage signal and d-axis voltage signal are the q-axis voltage signal and d-axis voltage signal after being updated by the weak magnetic speed control unit in a closed loop, and serve as the input of the weak magnetic speed control module for the next feedback;

[0124] After the initial q-axis voltage signal and d-axis voltage signal, the motor DC voltage U dc And the current set value i′ output by the MTPA control module d and i′ q When inputting the weak magnetic speed control module, through:

[0125]

[0126]

[0127] Determine whether the built-in permanent magnet synchronous motor needs to be subjected to magnetic field weakening speed regulation. If so, the d-axis and q-axis current values ​​after magnetic field weakening speed regulation are expressed as:

[0128]

[0129] Otherwise, the d and q axis currents remain unchanged;

[0130] Among them, U smax is the voltage limit value, I smax is the current limit value, U dc is the inverter DC bus voltage, θ′ is the weak magnetic regulation output angle.

[0131] D and Q axis current values ​​after magnetic field weakening speed regulation and electrical angular velocity ω eInput the voltage feedforward compensation module together and output the compensation voltage signal u′ q , u′ d The expression is:

[0132]

[0133] The voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is obtained by adding and The expression is:

[0134]

[0135] The park inverse transformation module 4 performs park inverse transformation on the q-axis voltage signal and the d-axis voltage signal to obtain the voltage in the stationary coordinate system; the specific process is:

[0136] The Park inverse transformation module performs Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal to obtain the voltage in the stationary coordinate system. The expression is:

[0137]

[0138] in, The voltage in the stationary coordinate system is obtained by inverse transformation of Park.

[0139] The SVPWM module takes the voltage in the stationary coordinate system as input and outputs a switching signal;

[0140] The inverter 5 controls the speed of the built-in permanent magnet synchronous motor with the switching signal output by the SVPWM module.

[0141] Example 2

[0142] In this embodiment, the sliding mode speed control loop designed by the proposed novel sliding mode reaching law is described. First, two state variables x1 and x2 of the system are established:

[0143]

[0144] in, is the derivative of x1; ω m The derivative of m is the actual mechanical angular velocity;

[0145] Establish the motor electromagnetic torque equation and motor motion equation:

[0146]

[0147]

[0148] Among them, L d and L q are the d-axis and q-axis inductances, and L d ≠L q ; P n is the number of pole pairs; is the permanent magnet flux linkage; i d is the component of the current on the d-axis; i q is the component of the current on the q-axis; J is the moment of inertia at the motor shaft end; B is the viscous friction coefficient; T L is the load torque;

[0149] Combining the motor electromagnetic torque equation and the motor motion equation, we get:

[0150]

[0151] The sliding mode surface s of the sliding mode speed controller SMC is designed as:

[0152] s = x1 + c∫0 t x1 dτ

[0153] Among them, x1 is a state variable of the system, c is the design parameter of the sliding mode surface, and c > 0;

[0154] Taking the derivative of s, we get:

[0155]

[0156] In a typical reaching law, the power reaching law is where k > 0, 0 < a < 1. By introducing the power term |s| a it ensures a smaller switching gain when the system state approaches the sliding mode surface, weakens the chattering, but the approaching speed is slower when it is far from the sliding mode surface. The exponential reaching law is where ε, k > 0. Its approaching speed is faster, but the chattering is larger when it is close to the sliding mode surface. Combining the exponential reaching law and the power reaching law, we obtain the sliding mode reaching law of the new sliding mode speed controller SMC:

[0157]

[0158] where ε, k, α, β > 0, γ > 1. In the above formula, x1 is the system state variable, and ε and k are the reaching law parameters, s is the sliding mode surface function; sgn(s) is the sign function; γ is the power coefficient of the absolute value of the sliding mode surface; λ is the power term exponent, and α, β are the constant term coefficients for adjusting the change rate of λ;

[0159] Replace the sign function with a saturation function to satisfy:

[0160]

[0161] Where σ represents a small positive constant; the control law of the sliding mode speed controller SMC, that is, the electromagnetic torque T e :

[0162]

[0163] Here, in the specific implementation process, given the mechanical angular velocity and the actual mechanical angular velocity ω m The difference is used as the input signal, and the speed loop outputs the electromagnetic torque T of the motor. e Introducing the state variable x1 in the design of the sliding surface can associate the approaching law with the system state and reduce the steady-state error of the system; in the design of the approaching law, the advantages of the power approaching law are retained, and the state variable x1 and the integral sliding surface s are further introduced. The approaching speed can be adaptively adjusted according to the size of the state variable x1 and the distance between the system and the sliding surface s, and finally the origin is stabilized, the system chattering is reduced, and the robustness of the system is improved; at the same time, the continuous saturation function is used It replaces the sign function sgn(s) in the conventional reaching law to further reduce the system chattering.

[0164] Example 3

[0165] like Figure 2 As shown, this embodiment proposes an IPMSM weak magnetic control method based on a new sliding mode control, see Figure 2 , the method comprises the following steps:

[0166] S1. Obtain the stator three-phase current of the built-in permanent magnet synchronous motor and detect the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m , electrical angle θ and electrical angular velocity ω e ;

[0167] S2. Perform Clark transformation on the three-phase current of the stator to obtain the current i α andi β , the current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q ;

[0168] S3. With a given mechanical angular velocity ω * m And the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor mThe difference is taken as input, and the electromagnetic torque T of the output motor is e ;

[0169] S4. With electromagnetic torque T e is the input, output current given value i′ d and i′ q ;

[0170] S5. The weak magnetic speed control unit uses the current given value i' d and i′ q , Motor DC voltage U dc , electrical angular velocity ω e , the actual value of the current in the rotating coordinate system i d andi q As input, the internal closed loop updates the q-axis voltage signal and the d-axis voltage signal;

[0171] S6. Perform a Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal using a Park inverse transformation module to obtain the voltage in a stationary coordinate system;

[0172] S7. Use the voltage in the stationary coordinate system as the input of the SVPWM module, output a switching signal, and control the speed of the built-in permanent magnet synchronous motor with the switching signal output by the SVPWM module.

[0173] In step S3, the electromagnetic torque T of the output motor is e When , two state variables x1 and x2 of the system are established:

[0174]

[0175] in, is the derivative of x1; ω m The derivative of

[0176] Motor electromagnetic torque equation, motor motion equation:

[0177]

[0178]

[0179] Among them, L d , L q is the d and q axis inductance and L d ≠L q ;P n is the pole pair number; is the permanent magnet flux; i d is the d-axis component of the current; i q is the q-axis component of the current; J is the moment of inertia of the motor shaft end; B is the viscous friction coefficient; T Lis the load torque;

[0180] Combining the motor electromagnetic torque equation and the motor motion equation, we get:

[0181]

[0182] The sliding surface s of the sliding mode speed controller SMC is designed as:

[0183] s=x1+c∫0 t x1 dτ

[0184] in, x1 is a state variable of the system, c is the design parameter of the sliding surface, and c>0;

[0185] Taking the derivative of s, we get:

[0186]

[0187] Combining the exponential reaching law and the power reaching law, the sliding mode reaching law of the new sliding mode speed controller SMC is obtained:

[0188]

[0189] Among them, ε, k, α, β>0, γ>1, x1 in the above formula is the system state variable, and x1=ω * -ω, ε and k are reaching law parameters, s is the sliding surface function; sgn(s) is the sign function; γ is the power coefficient of the absolute value of the sliding surface; λ is the power term exponent, α and β are the constant term coefficients for adjusting the rate of change of λ;

[0190] Replace the sign function with a saturation function to satisfy:

[0191]

[0192] Where σ represents a small positive constant; the control law of the sliding mode speed controller SMC, that is, the electromagnetic torque T e :

[0193]

[0194] Compared with the traditional sliding mode speed controller, it can ensure that the system has good approaching performance and chattering suppression effect when the system state is far away from and close to the sliding surface. According to the distance of the system state from the sliding surface, the approaching stage can be divided into two stages: far away from the sliding surface and close to the sliding surface.

[0195] (1) Stay away from the sliding surface

[0196] It can be seen from the new reaching law that when the system state is far away from the sliding mode, x1 is relatively large. When λ approaches 0, λ approaches α. At this time, the sliding mode reaching law is similar to the double power reaching law. The reaching speed of the double power reaching law is higher than that of the exponential reaching law, which greatly improves the sliding mode reaching speed and solves the problem that the conventional power reaching law has a low reaching speed when it is far away from the sliding surface.

[0197] (2) Close to the sliding surface

[0198] When the system state approaches the sliding mode, the power term |s| γ This ensures a smaller switching gain and weakens the chattering. In order to further suppress the chattering, the new reaching law The term has a variable speed approaching characteristic, which reduces the switching gain and further weakens the chattering.

[0199] Example 4

[0200] In this embodiment, in order to verify the control effect of the new sliding mode controller, Matlab / Simulink is used to simulate and analyze the new sliding mode controller. The simulation motor parameters are set as follows: bus voltage U dc =192V, carrier time T = 0.0001s, direct axis inductance L d =0.0002H, quadrature axis inductance L q =0.00047H, stator resistance R s =0.029Ω, rotor flux ψ f =0.062Wb, moment of inertia J = 0.003kg·m 2 , viscous damping coefficient B = 0.0003N·m·s, pole pair number p n =4, maximum voltage U smax =110.85V, maximum current I smax =110A, load torque T L The step is from 0N·m to 6N·m at 0.5s, and the simulation time is set to 2s. The traditional sliding mode controller weakening magnetic method is simulated and compared with the new sliding mode controller weakening magnetic method. Figure 3 A schematic diagram showing the speed of an internal permanent magnet synchronous motor based on magnetic field weakening control using a traditional sliding mode controller. Figure 4 The figure shows the speed of the built-in permanent magnet synchronous motor proposed in this embodiment based on the weak magnetic field control of the IPMSM proposed in the present invention; the speed of the permanent magnet synchronous motor gradually increases from 0 r / min, and the speed reaches 4000 r / min at 1 second, and the motor has entered the weak magnetic field stage. Figure 3 and Figure 4 It can be seen from the comparison that according to the method proposed by the present invention, the corresponding Figure 4 The controller has a faster response speed and smaller jitter in the approach process than the traditional sliding mode controller.

[0201] Figure 5The schematic diagram of the d-axis current of the interior permanent magnet synchronous motor based on the weak magnetic control of the traditional sliding mode controller is shown. Figure 6 A schematic diagram showing the d-axis current of the interior permanent magnet synchronous motor proposed in this embodiment based on the weak magnetic field control of the IPMSM proposed in the present invention, Figure 7 The schematic diagram of the q-axis current of the interior permanent magnet synchronous motor based on the weak magnetic control of the traditional sliding mode controller is shown. Figure 8 A schematic diagram showing the q-axis current of the interior permanent magnet synchronous motor proposed in this embodiment based on the weak magnetic field control of the IPMSM proposed in the present invention; Figure 6 and Figure 5 The comparison, Figure 8 and Figure 7 From the comparison, it can be concluded that the d-axis and q-axis current waveforms of the weak magnetic control based on the IPMSM proposed in the present invention tend to be stable before 1.4s, which is faster than the response of the traditional sliding mode weak magnetic algorithm; after 1.4s, the d-axis and q-axis current waveforms output by the new sliding mode weak magnetic algorithm are smoother, with fewer burrs, smaller jitter and stronger stability.

[0202] Fig. 9 The schematic diagram shows the output torque of the built-in permanent magnet synchronous motor based on the weak magnetic control of the traditional sliding mode controller. Fig.10 A schematic diagram showing the output torque of the interior permanent magnet synchronous motor proposed in this embodiment 4 based on the weak magnetic control of the IPMSM proposed in the present invention is shown as follows: Fig.10 and Fig. 9 It can be seen from the comparison that, by using the method proposed in this embodiment, the output torque has a small overshoot, fewer burrs, and is more stable and smooth.

[0203] Obviously, the above embodiments of the present invention are only examples for clearly illustrating the present invention, and are not intended to limit the embodiments of the present invention. For those skilled in the art, other different forms of changes or modifications can be made based on the above description. It is not necessary and impossible to list all the embodiments here. Any modifications, equivalent substitutions and improvements made within the spirit and principles of the present invention should be included in the protection scope of the claims of the present invention.

Claims

1. An IPMSM weak magnetic control system based on a new sliding mode control, characterized in that: include: The speed detection module is used to detect the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor. m , electrical angle θ and electrical angular velocity ω e ; Clark transformation module is used to perform Clark transformation on the three-phase current of the stator to obtain the current i α andi β ; Park transformation module is used to convert the current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q ; The sliding mode speed controller SMC is designed based on the new sliding mode reaching law to design the sliding mode speed control loop and to give a mechanical angular velocity ω * m And the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m The difference is taken as input, and the electromagnetic torque T of the output motor is e ; MTPA control module, based on electromagnetic torque T e is the input, output current given value i′ d and i′ q ; The weak magnetic speed regulating unit includes a weak magnetic speed regulating module, a voltage feedforward compensation module and a PI regulator. The weak magnetic speed regulating module, the voltage feedforward compensation module and the PI regulator form a closed-loop feedback. The weak magnetic speed regulating unit uses a current given value i′ d and i′ q , Motor DC voltage U dc , electrical angular velocity ω e , the actual value of the current in the rotating coordinate system i d andi q As input, the internal closed loop updates the q-axis voltage signal and the d-axis voltage signal, and uses the q-axis voltage signal and the d-axis voltage signal as inputs of the park inverse transformation module; The weak magnetic speed control module, voltage feedforward compensation module and PI regulator of the weak magnetic speed control unit form a closed-loop feedback to output the q-axis voltage signal and the d-axis voltage signal; the initial q-axis voltage signal and the d-axis voltage signal, the motor DC voltage U dc And the current set value i′ output by the MTPA control module d and i′ q Input the weak magnetic speed control module, and output the d and q axis current values ​​after weak magnetic speed control Will As the input of the q-axis current loop PI controller, As the input of the d-axis current loop PI controller, the output voltage signal u q and u d ; and the d and q axis current values ​​after weak magnetic speed regulation and electrical angular velocity ω e Input the voltage feedforward compensation module together and output the compensation voltage signal u′ q , u′ d , the voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is added and Compensated voltage signal and The initial q-axis voltage signal and d-axis voltage signal are the q-axis voltage signal and d-axis voltage signal after being updated by the weak magnetic speed control unit in a closed loop, and serve as the input of the next weak magnetic speed control module; Park inverse transformation module, which performs Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal to obtain the voltage in the stationary coordinate system; The SVPWM module takes the voltage in the stationary coordinate system as input and outputs a switching signal; The inverter controls the speed of the built-in permanent magnet synchronous motor with the switching signal output by the SVPWM module.

2. The IPMSM weak magnetic control system based on the novel sliding mode control according to claim 1 is characterized in that: The obtained stator three-phase current of the internal permanent magnet synchronous motor is expressed as i a ,i b ,i c , Clark transformation is performed on the stator three-phase current to obtain: The current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q :

3. The IPMSM weak magnetic control system based on the novel sliding mode control according to claim 1 is characterized in that: The sliding mode speed control loop is designed based on the new sliding mode reaching law, and the electromagnetic torque T of the output motor is e The process is: Establish two state variables x1 and x2 of the system: in, is the derivative of x1; ω m The derivative of Motor electromagnetic torque equation, motor motion equation: Among them, L d , L q is the d and q axis inductance and L d ≠L q ;P n is the pole pair number; is the permanent magnet flux; i d is the d-axis component of the current; i q is the q-axis component of the current; J is the moment of inertia of the motor shaft end; B is the viscous friction coefficient; T L is the load torque; Combining the motor electromagnetic torque equation and the motor motion equation, we get: The sliding surface s of the sliding mode speed controller SMC is designed as: in, x1 is a state variable of the system, c is the design parameter of the sliding surface, and c>0; Taking the derivative of s, we get: Combining the exponential reaching law and the power reaching law, the sliding mode reaching law of the new sliding mode speed controller SMC is obtained: Among them, ε, k, α, β>0, γ>1, x1 in the above formula is the system state variable, and ε and k are reaching law parameters, s is the sliding surface function; sgn(s) is the sign function; γ is the power coefficient of the absolute value of the sliding surface; λ is the power term exponent, and α and β are the constant term coefficients that adjust the rate of change of λ. Replace the sign function with a saturation function to satisfy: Where σ represents a small positive constant; the control law of the sliding mode speed controller SMC, that is, the electromagnetic torque T e :

4. The IPMSM weak magnetic control system based on the novel sliding mode control according to claim 3 is characterized in that: The electromagnetic torque T e is the input, output current given value i′ d and i′ q The calculation process satisfies:

5. The IPMSM weak magnetic control system based on the novel sliding mode control according to claim 4 is characterized in that: The q-axis voltage signal, d-axis voltage signal, and motor DC voltage U dc And the current set value i′ output by the MTPA control module d and i′ q When inputting the weak magnetic speed control module, through: Determine whether the built-in permanent magnet synchronous motor needs to be subjected to magnetic field weakening speed regulation. If so, the d-axis and q-axis current values ​​after magnetic field weakening speed regulation are expressed as: Otherwise, the d and q axis currents remain unchanged; Among them, U smax is the voltage limit value, I smax is the current limit value, U dc is the inverter DC bus voltage, θ′ is the weak magnetic regulation output angle.

6. The IPMSM weak magnetic control system based on the novel sliding mode control according to claim 5 is characterized in that: D and Q axis current values ​​after magnetic field weakening speed regulation and electrical angular velocity ω e Input the voltage feedforward compensation module together and output the compensation voltage signal u′ q , u′ d The expression is:

7. The IPMSM weak magnetic control system based on the novel sliding mode control according to claim 5 is characterized in that: The voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is added and The expression is: The Park inverse transformation module performs Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal to obtain the voltage in the stationary coordinate system. The expression is: in, The voltage in the stationary coordinate system is obtained by inverse transformation of Park.

8. An IPMSM weak magnetic control method based on a novel sliding mode control, characterized in that: The method comprises the following steps: S1. Obtain the stator three-phase current of the built-in permanent magnet synchronous motor and detect the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m , electrical angle θ and electrical angular velocity ω e ; S2. Perform Clark transformation on the three-phase current of the stator to obtain the current i α andi β , the current i α andi β Perform park transformation to obtain the actual value i of the current in the rotating coordinate system d andi q ; S3. Design a sliding mode speed control loop based on the new sliding mode reaching law, and use a given mechanical angular velocity ω * m And the actual mechanical angular velocity ω of the built-in permanent magnet synchronous motor m The difference is taken as input, and the electromagnetic torque T of the output motor is e ; S4. With electromagnetic torque T e is the input, output current given value i′ d and i′ q , S5. The weak magnetic speed control unit uses the current given value i' d and i′ q、 Motor DC voltage U dc , electrical angular velocity ω e , the actual value of the current in the rotating coordinate system i d andi q As input, the internal closed loop updates the q-axis voltage signal and the d-axis voltage signal; The weak magnetic speed control module, voltage feedforward compensation module and PI regulator of the weak magnetic speed control unit form a closed-loop feedback to output the q-axis voltage signal and the d-axis voltage signal; the initial q-axis voltage signal and the d-axis voltage signal, the motor DC voltage and the current given values ​​i′d and i′q output by the MTPA control module are input into the weak magnetic speed control module to output the d and q axis current values ​​after weak magnetic speed control. Will As the input of the q-axis current loop PI controller, As the input of the d-axis current loop PI controller, the output voltage signal u q and u d ; and the d and q axis current values ​​after weak magnetic speed regulation , and the electrical angular velocity ωe are input into the voltage feedforward compensation module, and the compensation voltage signal u′ is output q , u′ d , the voltage signal u d and u q and compensation voltage signal u′ q , u′ d The voltage signal after compensation is added and Compensated voltage signal and The initial q-axis voltage signal and d-axis voltage signal are the q-axis voltage signal and d-axis voltage signal after being updated by the weak magnetic speed control unit in a closed loop, and serve as the input of the next weak magnetic speed control module; S6. Perform a Park inverse transformation on the q-axis voltage signal and the d-axis voltage signal using a Park inverse transformation module to obtain the voltage in a stationary coordinate system; S7. Use the voltage in the stationary coordinate system as the input of the SVPWM module, output a switching signal, and control the speed of the built-in permanent magnet synchronous motor with the switching signal output by the SVPWM module.

9. The IPMSM weak magnetic control method based on the novel sliding mode control according to claim 8 is characterized in that: In step S3, a sliding mode speed control loop is designed based on the new sliding mode reaching law to output the electromagnetic torque T of the motor. e The process is: Establish two state variables x1 and x2 of the system: in, is the derivative of x1; ω m The derivative of Motor electromagnetic torque equation, motor motion equation: Among them, L d , L q is the d and q axis inductance and L d ≠L q ;P n is the pole pair number; is the permanent magnet flux; i q is the d-axis component of the current; i w is the q-axis component of the current; J is the moment of inertia of the motor shaft end; B is the viscous friction coefficient; T L is the load torque; Combining the motor electromagnetic torque equation and the motor motion equation, we get: The sliding surface s of the sliding mode speed controller SMC is designed as: in, x1 is a state variable of the system, c is the design parameter of the sliding surface, and c>0; Taking the derivative of s, we get: Combining the exponential reaching law and the power reaching law, the sliding mode reaching law of the new sliding mode speed controller SMC is obtained: Among them, ε, k, α, β>0, γ>1, x1 in the above formula is the system state variable, and ε and k are reaching law parameters, s is the sliding surface function; sgn(s) is the sign function; γ is the power coefficient of the absolute value of the sliding surface; λ is the power term exponent, and α and β are the constant term coefficients that adjust the rate of change of λ. Replace the sign function with a saturation function to satisfy: Where σ represents a small positive constant; the control law of the sliding mode speed controller SMC, that is, the electromagnetic torque T e :

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