Model predictive device and control method for multi-phase llc-based dc converter

Abstract

The application provides a kind of multi-phase LLC-based direct current converter model prediction device and control method, specifically relates to direct current converter energy conversion technical field.The device includes two-stage converter cascade, input port has S, output port has N.The front stage is composed of S Buck circuits in parallel, each Buck circuit contains an inductor and two MOS tubes;The back stage is composed of N-phase LLC resonant converter, connected with the output port of the front stage Buck circuit.The front bridge of N-phase LLC resonant converter adopts half-bridge circuit, including two MOS tubes, resonant inductor, resonant capacitor and excitation inductor.The device adopts model prediction control method, the output voltage of next cycle is predicted to adjust relevant variables to maintain the purpose of output voltage.The application can access multiple voltage levels, model prediction control can be performed on direct current converter, voltage of specified voltage level can be output, voltage output can be effectively adjusted, and the demand of multi-scene application can be met.

Description

Technical Field

[0001] This invention relates to the field of DC-DC converter energy conversion technology, and in particular to a DC-DC converter model prediction device and control method based on multiphase LLC. Background Technology

[0002] With increasing greenhouse gas emissions and the proposed goal of "carbon neutrality," research on DC / DC converters has become increasingly urgent. Currently, high efficiency, high frequency, and high power density are the development trends of power electronic converters, and the demand for DC / DC converters in various fields is gradually increasing, thus placing higher requirements on them.

[0003] However, existing DC / DC converters and their control methods can only regulate single-port input and multi-port output, or multi-port input and single-port output, and cannot simultaneously meet the energy regulation requirements of multi-port output and multi-port input scenarios. Furthermore, they have limited current carrying capacity, high transmission losses, and high costs. In practice, traditional PI control is often used, which has a slow response speed and phase lag, making it difficult to apply to current DC / DC converters. Summary of the Invention

[0004] The technical problem to be solved by the present invention is to address the shortcomings of the prior art by providing a DC-DC converter model prediction device and control method based on multiphase LLC, which can realize multi-voltage level output, solve the problem of slow dynamic response, improve the power output quality, and can be applied to a variety of occasions.

[0005] To solve the above-mentioned technical problems, the technical solution adopted by the present invention is as follows:

[0006] This invention provides a model prediction device and control method for a DC-DC converter based on a multiphase LLC circuit. The device is configured as follows: the multiphase LLC-based DC-DC converter consists of two cascaded converter stages. It has S input ports, designated as input port 1, input port 2, input port 3…input port S; and N output ports, designated as output port 1, output port 2, output port 3…output port N. The front stage consists of S parallel Buck circuits, named Buck1, Buck2,…, BuckS. The rear stage consists of a multiphase LLC resonant converter.

[0007] The preceding S Buck circuits have a total of S input ports, each with independent input voltages, and the input voltages are respectively V. in1 V in2 V in3 , ...V inS The output side is connected in parallel. Each Buck circuit includes an inductor L. SAnd two MOSFETs, the output voltage of the Buck circuit's common output port 0 is V. OUT0 Output current i OUT0 Output capacitor C S1 .

[0008] The subsequent multiphase LLC resonant converter has one input port, connected to the output port of the preceding Buck circuit. The positive terminal of the multiphase LLC resonant converter is connected to the positive terminal of the Buck circuit's output port, and the negative terminal is connected to the negative terminal of the Buck circuit's output port. The front bridge of the multiphase LLC resonant converter uses a half-bridge circuit, containing two MOSFETs and a resonant inductor L. r Resonant capacitor C r Magnetizing inductance L M The transformer is a matrix transformer T, with one primary side port and N secondary side ports, namely N1, N2, N3, ..., N. N The multiphase LLC's rear bridge circuit uses full-wave rectification, with N output ports and output voltages of V. OUT1 V OUT2 V OUT3 , ..., V OUTN The output currents are i OUT1 i OUT2 i OUT3 , ..., i OUTN .

[0009] A model prediction device and control method for DC-DC converters based on multiphase LLC are presented. The main steps of the control method are as follows:

[0010] Step 1 uses integration over one cycle to discretize the continuous-time state equation of the multiphase LLC DC converter to obtain the average state-space model.

[0011]

[0012] Among them, L S1 L represents the inductance in Buck1. S2 This represents the inductance in Buck2, and so on, L. SS This represents the inductance in BuckS; k is the switching period, i L1 (k) refers to the inductance L flowing through Buck1 during the kth switching cycle. S1 The current value, i LS (k) refers to the current flowing through inductor L in BuckS during the k-th switching cycle. SS The current value; C S1 It is the capacitor at the final output port 0 of the Buck circuit, L r C rThese are the resonant inductor and resonant capacitor of the subsequent multiphase LLC resonant converter; V OUTN (k+1) refers to the output voltage of output port N during k+1 cycles; K1, K2, ..., K N Represents the turns ratio of a matrix transformer; i OUTN (k) refers to the output current value of the Nth output port in the kth switching cycle; T S It is the switching cycle.

[0013] Step 2: Input voltage V in1 V in2 V in3 , ...V inS Output voltage V OUT0 V OUT1 V OUT2 , ..., V OUTN Output current i OUT0 i OUT1 i OUT2 , ..., i OUTN Filtering and sampling are performed to obtain the predicted value for the next cycle based on the current switching cycle; sampling will be performed using Hall elements, and the size of the sampling resistor is related to the voltage level being sampled; the filtering method is Kalman filtering.

[0014] Step 3 adopts a closed-loop control structure to obtain the optimal duty cycle of the front-stage MOSFET based on the predicted value of the next cycle and the input and output values ​​based on the current switching cycle.

[0015] Step 3.1 The common output voltage reference value V of the front-end Buck circuit in the (k+1)th switching cycle OUT0ref (k+1) is:

[0016]

[0017] K S Indicates the compensation coefficient; i OUT0 (k) refers to the output current of the common output port of the preceding Buck circuit during the kth cycle.

[0018] Step 3.2 Based on the V obtained in Step 3.1 OUT0ref The duty cycle d can be calculated from the sampled values ​​obtained by Kalman filtering in step 2 (k+1):

[0019]

[0020]

[0021] Among them, V OUT0(k+1) represents the output voltage at the common output port of the preceding Buck circuit during the (k+1)th switching cycle, K I It is the integral coefficient of the corrected reference voltage; V OUT0ref (k+1) is a given reference value at k+1 cycles; V inS (k) represents the input voltage of input port S in the kth switching cycle; σ represents the product coefficient under different deviation results.

[0022] Step 3.3 Obtain the optimal duty cycle based on Step 3.2:

[0023]

[0024] The optimal duty cycle obtained by solving refers to the duty cycle of the MOSFET directly connected to the input port in the preceding Buck circuit, with another MOSFET conducting in a complementary manner to this MOSFET.

[0025] Step 4 employs a closed-loop control structure to obtain the switching frequency of the primary side of the subsequent LLC multiphase resonant converter based on the current model prediction value and the input and output values ​​of the current cycle.

[0026] Step 4.1 The reference values ​​for output port 1, output port 2, ..., output port N in the (k+1)th cycle can be represented as:

[0027]

[0028] Step 4.2 Using the output voltage V of output port 1 OUT1 Using the reference, the switching frequency f on the primary side of the subsequent multiphase LLC resonant converter can be obtained. s :

[0029]

[0030] K D It is the state coefficient.

[0031] In the latter stage, the two switches on the primary side of the multiphase LLC resonant converter each have a duty cycle of 50% and are complementary in conduction.

[0032] Step 5 employs a closed-loop control structure to obtain the synchronous rectification conduction time of the secondary side of the subsequent multi-phase LLC resonant converter based on the switching frequency of the primary side of the subsequent multi-phase LLC resonant converter and the input and output values ​​of the current cycle.

[0033] Step 5.1 Based on the expression for the reference value in the (k+1)th cycle and the sampled voltage value obtained in Step 4.1, obtain the synchronous rectification conduction time T1...T for output port 1, output port 2, ..., output port N. N :

[0034]

[0035] Step 5.2 Based on the simplification in Step 5.1, obtain the synchronous rectification conduction time:

[0036]

[0037] Wherein, the synchronous rectification conduction time T1 obtained according to step 5.2 is the conduction time of the MOSFET connected to the positive terminal of the first port of the secondary side of the matrix transformer T. The switching frequency of the MOSFET is the same as the switching frequency obtained in step 4, and the MOSFETs connected to the negative terminal of the first port of the secondary side of the matrix transformer T conduct complementaryly. T2, ..., T N And so on.

[0038] The beneficial effects of adopting the above technical solution are as follows: This invention integrates multiple voltages to output different voltage levels, standardizes the output voltage levels, employs a matrix transformer design to reduce costs, and can be applied in various scenarios. This invention optimizes the average state-space model of multi-port input and multi-port output, reducing errors, improving control accuracy, overcoming the bandwidth limitations of traditional PI control structures, and improving the system's anti-interference capability and dynamic response speed. Attached Figure Description

[0039] Figure 1 This is a topology diagram of an embodiment of the present invention;

[0040] Figure 2 This is the control flowchart of the present invention. Detailed Implementation

[0041] The specific embodiments of the present invention will be described in further detail below with reference to the accompanying drawings and examples. The following examples are for illustrative purposes only and are not intended to limit the scope of the invention.

[0042] This embodiment uses a four-phase two-stage LLC DC-DC converter. The front stage consists of four Buck circuits connected in parallel, and the rear stage is a half-bridge LLC four-phase output converter using a four-phase output matrix transformer. The specific topology is as follows: Figure 1 As shown. The relevant electrical parameters are set as follows: Buck1 inductor 20uH, Buck2 inductor 35uH, Buck3 inductor 24uH, Buck4 inductor 37uH, and the switching frequency is 100kHz for all. The LLC resonant inductor is set to 100uH, the resonant capacitor to 25nF, and the matrix transformer turns ratio is 8:1:1:2:2:4:4:6:6.

[0043] Figure 2This is a control block diagram of the model prediction device for a multiphase LLC-based DC-DC converter proposed in this invention, which includes triple closed-loop control to control the DC-DC converter variables. The main steps of this embodiment are as follows:

[0044] Step 1 uses integration over one cycle to discretize the continuous-time state equation of the multiphase LLC DC converter to obtain the average state-space model.

[0045]

[0046] Among them, L S1 L represents the inductance in Buck1. S2 This represents the inductance in Buck2, and so on, L. SS This represents the inductance in BuckS; k is the switching period, i L1 (k) refers to the inductance L flowing through Buck1 during the kth switching cycle. S1 The current value, i LS (k) refers to the current flowing through inductor L in BuckS during the k-th switching cycle. SS The current value; C S1 It is the capacitor at the final output port 0 of the Buck circuit, L r C r These are the resonant inductor and resonant capacitor of the subsequent multiphase LLC resonant converter; V OUTN (k+1) refers to the output voltage of output port N during k+1 cycles; K1, K2, ..., K N Represents the turns ratio of a matrix transformer; i OUTN (k) refers to the output current value of the Nth output port in the kth switching cycle; T S It is the switching cycle.

[0047] Step 2: Input voltage V in1 V in2 V in3 , ...V inS Output voltage V OUT0 V OUT1 V OUT2 , ..., V OUTN Output current i OUT0 i OUT1 i OUT2 , ..., i OUTN Filtering and sampling are performed to obtain the predicted value for the next cycle based on the current switching cycle; sampling will be performed using Hall elements, and the size of the sampling resistor is related to the voltage level being sampled; the filtering method is Kalman filtering.

[0048] Step 3 adopts a closed-loop control structure to obtain the optimal duty cycle of the front-stage MOSFET based on the predicted value of the next cycle and the input and output values ​​based on the current switching cycle.

[0049] Step 3.1 The common output voltage reference value V of the front-end Buck circuit in the (k+1)th switching cycle OUT0ref (k+1) is:

[0050]

[0051] K S Indicates the compensation coefficient; i OUT0 (k) refers to the output current of the common output port of the preceding Buck circuit during the kth cycle.

[0052] Step 3.2 Based on the V obtained in Step 3.1 OUT0ref The duty cycle d can be calculated from the sampled values ​​obtained by Kalman filtering in step 2 (k+1):

[0053]

[0054]

[0055] Among them, V OUT0 (k+1) represents the output voltage at the common output port of the preceding Buck circuit during the (k+1)th switching cycle, K I It is the integral coefficient of the corrected reference voltage; V OUT0ref (k+1) is a given reference value at k+1 cycles; V inS (k) represents the input voltage of input port S in the kth switching cycle; σ represents the product coefficient under different deviation results.

[0056] Step 3.3 Obtain the optimal duty cycle based on Step 3.2:

[0057]

[0058] V inS (k) represents the input voltage of input port S during the k-th switching cycle. The optimal duty cycle obtained by solving refers to the duty cycle of the MOSFET directly connected to the input port in the preceding Buck circuit, with another MOSFET conducting complementary to this MOSFET.

[0059] The optimal duty cycle obtained by solving refers to the duty cycle of the MOSFET directly connected to the input port in the preceding Buck circuit, with the other MOSFET conducting complementaryly to this MOSFET. Based on the above steps, the final calculations are as follows: the duty cycle of the MOSFET directly connected to the input port in Buck1 is 0.454, the duty cycle of the MOSFET directly connected to the input port in Buck2 is 0.376, the duty cycle of the MOSFET directly connected to the input port in Buck3 is 0.492, and the duty cycle of the MOSFET directly connected to the input port in Buck4 is 0.673. out0 It is 80V.

[0060] Step 4 employs a closed-loop control structure to obtain the switching frequency of the primary side of the subsequent LLC multiphase resonant converter based on the current model prediction value and the input and output values ​​of the current cycle.

[0061] Step 4.1 The reference values ​​for output port 1, output port 2, ..., output port N in the (k+1)th cycle can be expressed as:

[0062]

[0063] Step 4.2 Using the output voltage V of output port 1 OUT1 Using the reference, the switching frequency f on the primary side of the subsequent multiphase LLC resonant converter can be obtained. s :

[0064]

[0065] K D It is the state coefficient.

[0066] In this stage, the two switches on the primary side of the multiphase LLC resonant converter each have a 50% duty cycle and are complementary in conduction. Calculations show that when the output voltage is stable, the switching frequency of the two switches on the primary side of the multiphase LLC resonant converter is 102kHz.

[0067] Step 5 employs a closed-loop control structure to obtain the synchronous rectification conduction time of the secondary side of the subsequent multi-phase LLC resonant converter based on the switching frequency of the primary side of the subsequent multi-phase LLC resonant converter and the input and output values ​​of the current cycle.

[0068] Step 5.1 Based on the expression for the reference value in the (k+1)th cycle and the sampled voltage value obtained in Step 4.1, obtain the synchronous rectification conduction time T1...T for output port 1, output port 2, ..., output port N. N :

[0069]

[0070] Step 5.2 Based on the simplification in Step 5.1, obtain the synchronous rectification conduction time:

[0071]

[0072] Wherein, the synchronous rectification conduction time T1 obtained according to step 5.2 is the conduction time of the MOSFET connected to the positive terminal of the first port of the secondary side of the matrix transformer T. The switching frequency of the MOSFET is the same as the switching frequency obtained in step 4, and the MOSFETs connected to the negative terminal of the first port of the secondary side of the matrix transformer T conduct complementaryly. T2, ..., T N And so on. The output voltage V at output port 1. OUT1 It is 10V, the output voltage V of output port 2. OUT2 It is 20V, the output voltage V of output port 3. OUT3 It is 40V, the output voltage V of output port 4. OUT4 It is 60V.

[0073] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention, and not to limit them; although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some or all of the technical features therein; and these modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the scope defined by the claims of the present invention.

Claims

1. A DC-DC converter model prediction device based on multiphase LLC, characterized in that, The device is configured as follows: it consists of two cascaded converters with a total of S input ports, namely input port 1, input port 2, input port 3... input port S; and a total of N output ports, namely output port 1, output port 2, output port 3... output port N; the front stage consists of S parallel Buck circuits, named Buck1, Buck2,..., BuckS, and the rear stage consists of a multiphase LLC resonant converter; The preceding S Buck circuits have a total of S input ports, each with independent input voltages, and the input voltages are respectively V. in1 V in2 V in3 , ...V inS The output side is connected in parallel; each Buck circuit includes an inductor L. S And two MOSFETs, the output voltage of the Buck circuit's common output port 0 is V. OUT0 Output current i OUT0 Output capacitor C S1 ; The subsequent multiphase LLC resonant converter has one input port, connected to the output port of the preceding Buck circuit. The positive terminal of the multiphase LLC resonant converter is connected to the positive terminal of the Buck circuit's output port, and the negative terminal is connected to the negative terminal of the Buck circuit's output port. The front bridge of the multiphase LLC resonant converter uses a half-bridge circuit, containing two MOSFETs and a resonant inductor L. r Resonant capacitor C r Magnetizing inductance L M The transformer is a matrix transformer T, with one primary side port and N secondary side ports, namely N1, N2, N3, ..., N. N The multiphase LLC's rear bridge circuit uses full-wave rectification, with N output ports and output voltages of V respectively. OUT1 V OUT2 V OUT3 , ..., V OUTN The output currents are i OUT1 i OUT2 i OUT3 , ..., i OUTN .

2. A model predictive control method for DC-DC converters based on multiphase LLC, characterized in that, The method includes the following steps: Step 1 uses integration over one cycle to discretize the continuous-time state equation of the multiphase LLC DC converter to obtain the average state-space model. Step 2: Input voltage V in1 V in2 V in3 , ...V inS Output voltage V OUT0 V OUT1 V OUT2 , ..., V OUTN Output current i OUT0 i OUT1 i OUT2 , ..., i OUTN Filtering and sampling are performed to obtain the predicted value for the next cycle based on the current switching cycle; sampling will be performed using Hall elements, and the size of the sampling resistor is related to the voltage level being sampled; the filtering method is Kalman filtering. Step 3 adopts a closed-loop control structure to obtain the optimal duty cycle of the front-stage MOSFET based on the predicted value of the next cycle and the input and output values ​​based on the current switching cycle. Step 4 employs a closed-loop control structure to obtain the switching frequency of the primary side of the subsequent LLC multiphase resonant converter based on the current model prediction value and the input and output values ​​of the current cycle. Step 5 employs a closed-loop control structure to obtain the synchronous rectification conduction time of the secondary side of the subsequent multi-phase LLC resonant converter based on the switching frequency of the primary side of the subsequent multi-phase LLC resonant converter and the input and output values ​​of the current cycle.

3. The model predictive control method for a DC-DC converter based on multiphase LLC according to claim 2, characterized in that, The discrete mathematical model obtained in step 1 of the method is: Among them, L S1 L represents the inductance in Buck1. S2 This represents the inductance in Buck2, and so on, L. SS This represents the inductance in BuckS; k is the switching period, i L1 (k) refers to the inductance L flowing through Buck1 during the kth switching cycle. S1 The current value, i LS (k) refers to the current flowing through inductor L in BuckS during the k-th switching cycle. SS The current value; C S1 It is the capacitor at the final output port 0 of the Buck circuit, L r C r These are the resonant inductor and resonant capacitor of the subsequent multiphase LLC resonant converter; V OUTN (k+1) refers to the output voltage of output port N during k+1 cycles; K1, K2, ..., K N Represents the turns ratio of a matrix transformer; i OUTN (k) refers to the output current value of the Nth output port in the kth switching cycle; T S It is the switching cycle.

4. The model predictive control method for a DC-DC converter based on multiphase LLC according to claim 2, characterized in that, Step 3 specifically includes the following steps: Step 3.1 The common output voltage reference value V of the front-end Buck circuit in the (k+1)th switching cycle OUT0ref (k+1) is: K S Indicates the compensation coefficient; i OUT0 (k) refers to the output current of the common output port of the preceding Buck circuit in the kth cycle; Step 3.2 Based on the V obtained in Step 3.1 OUT0ref The duty cycle d can be calculated from the sampled values ​​obtained by Kalman filtering in step 2 (k+1): Among them, V OUT0 (k+1) represents the output voltage at the common output port of the preceding Buck circuit during the (k+1)th switching cycle, K I It is the integral coefficient of the corrected reference voltage; V OUT0ref (k+1) is a given reference value at k+1 cycles; V inS (k) represents the input voltage of input port S in the k-th switching cycle; σ represents the product coefficient under different deviation results; Step 3.3 Obtain the optimal duty cycle based on Step 3.2: The optimal duty cycle obtained by solving refers to the duty cycle of the MOSFET directly connected to the input port in the preceding Buck circuit, with another MOSFET conducting in a complementary manner to this MOSFET.

5. The model predictive control method for a DC-DC converter based on multiphase LLC according to claim 2, characterized in that, Step 4 specifically includes the following steps: Step 4.1 The reference values ​​for output port 1, output port 2, ..., output port N in the (k+1)th cycle can be expressed as: Step 4.2 Using the output voltage V of output port 1 OUT1 Using the reference, the switching frequency f on the primary side of the subsequent multiphase LLC resonant converter can be obtained. s : K D These are state coefficients; In the latter stage, the two switches on the primary side of the multiphase LLC resonant converter each have a duty cycle of 50% and are complementary in conduction.

6. The model predictive control method for a DC-DC converter based on multiphase LLC according to claim 2, characterized in that, Step 5 specifically includes the following steps: Step 5.1 Based on the expression for the reference value in the (k+1)th cycle and the sampled voltage value obtained in Step 4.1, obtain the synchronous rectification conduction time T1...T for output port 1, output port 2, ..., output port N. N : Step 5.2 Based on the simplification in Step 5.1, obtain the synchronous rectification conduction time: Wherein, the synchronous rectification conduction time T1 obtained according to step 5.2 is the conduction time of the MOSFET connected to the positive terminal of the first port of the secondary side of the matrix transformer T. The switching frequency of the MOSFET is the same as the switching frequency obtained in step 4. The MOSFETs connected to the negative terminal of the first port of the secondary side of the matrix transformer T conduct complementaryly. T2, ..., T N And so on.

7. The model predictive control method for a DC-DC converter based on multiphase LLC according to claim 4, characterized in that, The product coefficient σ under different deviations can be expressed as:

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