A high degree of freedom far-field customized compact circularly polarized planar lens antenna

Abstract

The application is a high degree of freedom far-field customized compact circularly polarized planar lens antenna. The application relates to the technical field of radiative metasurface antennas. Due to the limitation of space feed, it is difficult for metasurfaces in the microwave band to be applied to integrated systems. Here, the application realizes precise far-field radiation mode manipulation. Two E-shaped unit superimpositions constitute a geometric phase-based metasurface unit, realizing 2pi range transmission phase coverage and high transmittance. The application method realizes highly customized far-field mode control. Several proof-of-concept RTM prototypes are designed and manufactured, realizing energy-controllable multi-beam, wide-angle deflection and customized beamforming. Simulation and experimental results verify that the proposed method and RTM structure can provide a compact wavefront modulation platform and realize highly accurate and customized far-field beamforming.

Description

Technical Field

[0001] This invention relates to the field of metasurface lens antenna technology, and is a compact circularly polarized planar lens antenna with high degree of freedom and customizable far-field characteristics. Background Technology

[0002] With the rapid development of next-generation wireless communication and the widespread application of technologies such as the Internet of Things (IoT), the demand for customized beamforming and complex electromagnetic signal processing is constantly increasing to meet the needs of various application scenarios, such as base station communication, autonomous vehicles, vital sign recognition devices, and smart homes. Furthermore, complex application scenarios typically place demands on electromagnetic devices with low complexity, high integration, and miniaturization.

[0003] Metasurfaces provide an advanced platform for space-customized beam manipulation, enabling efficient beam deflection and energy control by adjusting the complex amplitude distribution on their metasurface units. Existing metasurface-based far-field manipulation devices can be divided into three main categories: phase gradient metasurfaces (PGMs), complex-amplitude metasurfaces (CAMs), and metagratings (MGs). Existing PGMs typically generate beam energy modulation in a single diffraction direction or 2-D direction, and are usually designed using optimization or analytical methods. CAMs can freely manipulate the far-field beam because they introduce additional degrees of freedom in the amplitude, allowing the far-field distribution to be expressed using explicit analytical expressions. By designing the complex amplitude of the output electric field of each unit, precise control over the far-field beam direction and energy can be achieved. However, CAMs inevitably result in some input energy loss and require complex design processes, which limits the widespread application of complex-amplitude metasurfaces. MGs achieve efficient, large-angle beam deflection in specific directions by introducing sparse polarization-sensitive units. However, MGs typically operate under a specific polarization, thus limiting beam manipulation to only one dimension. Furthermore, the fixed diffraction order reduces the control flexibility of MGs with large electrical dimensions. Therefore, achieving a high degree of customization of the energy, direction, and width of far-field beams remains a significant challenge.

[0004] Meanwhile, the aforementioned far-field modulation devices typically require an external feed, sacrificing overall compactness. In the microwave band, feed antennas hinder system integration and miniaturization. Recently, radiation-type metasurfaces (RTMs) have been reported, offering new prospects for designing compact and miniaturized beam modulation devices. By combining the feed network with metasurface elements without affecting the latter's phase modulation function, the metasurface and the feed antenna are integrated.

[0005] RTM metasurface elements not only achieve full-phase modulation but also realize a compact profile as radiators. The former is achieved through geometric phase modulation, while the latter relies on a feeding network. Compared to existing sequentially rotating arrays (SRAs) that effectively utilize the theory that the excitation phase is related to the rotation angle of linearly polarized (LP) array elements, RTM extends the traditional SRA by using a metasurface design mindset, focusing on the complex amplitude distribution and spatial polarization state of the element's radiated electric field. This further unlocks the platform's potential, enabling more advanced and diverse electromagnetic wave manipulation capabilities. Summary of the Invention

[0006] This invention aims to achieve precise far-field radiation mode manipulation on a compact feeding platform. Based on this, the invention provides a compact circularly polarized planar lens antenna with high degree of freedom for far-field customization.

[0007] It should be noted that, in this document, relational terms such as "first" and "second" are used only to distinguish one entity or operation from another, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Furthermore, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such process, method, article, or apparatus.

[0008] This invention provides a compact circularly polarized planar lens antenna with a highly flexible far-field customization method and a simplified feed network. The invention offers the following technical solutions:

[0009] A high-degree-of-freedom far-field customization method and a compact circularly polarized planar lens antenna with a simplified feed network are disclosed. The metasurface antenna includes a 16×16 element array along the x and y directions, respectively. The feed network of the 16×16 element array is obtained by cascading several 1-to-2 power dividers. A 50Ω feed microstrip extends inward and is connected to a 1-to-2 power divider. The two output ports of the power divider are connected to a new power divider, and then a 1-to-256 power divider network is formed in a loop.

[0010] Power dividers in a power supply network can be classified into three types, referred to as the first power divider, the second power divider, and the third power divider. The first power divider is located at the power supply end, with an input port impedance of 50Ω and an output port impedance of 100Ω. The second power divider has an input port impedance of 100Ω and an output port impedance of 100Ω. The third power divider has an input impedance of 100Ω and its output port is connected to the cell. The power divider at the power supply end has an input impedance of 50Ω and both output ports are designed to have an impedance of 100Ω.

[0011] Preferably, using 100Ω microstrip lines as the transmission lines TL, which make up the majority of the power supply network, helps to save wiring space.

[0012] Preferably, the 16×16 unit array is formed by stacking two e-shaped TE elements together to form a TE metasurface unit, with metal vias extending to the top layer, and the two microstrip lines are divided into two e-shaped structures, with the two microstrip lines connected in parallel.

[0013] Preferably, the e-shaped TE element is a three-layer structure, occupying an area of ​​p × p × h mm. 3 The top layer is an E-shaped patch, and the bottom layer contains a feed microstrip line for impedance matching, with an input impedance of 100Ω. The middle copper layer serves not only as the ground for the top and bottom layers but also as a connection via vias. The copper layer is 0.035 mm thick, and the dielectric substrates are h1 = 3 mm and h2 = 1.27 mm thick, respectively.

[0014] An antenna based on a compact circularly polarized planar lens antenna with high degree of freedom far-field customization.

[0015] A fast far-field reconstruction method based on far-field phase mask optimization, the method comprising:

[0016] Step 1: Input the amplitude and phase distribution of the far field. The target amplitude distribution is a 360×90 matrix, which corresponds to the projection of a three-dimensional far-field radiation mode onto a two-dimensional plane. When the target beam is a pencil beam, only the elements corresponding to that direction are assigned; when the target beam is a wide beam, the elements corresponding to the directions covered by the wide beam are assigned.

[0017] Step 2: Begin the iterative process to obtain the metasurface aperture field distribution;

[0018] Step 3: Optimize the metasurface design using the Particle Swarm Optimization (PSO) algorithm.

[0019] A computer-readable storage medium having a computer program stored thereon, which is executed by a processor to implement a fast far-field reconstruction method based on far-field phase mask optimization.

[0020] A computer device includes a memory and a processor, the memory storing a computer program, and the processor executing the computer program to implement a fast far-field reconstruction method based on far-field phase mask optimization.

[0021] The present invention has the following beneficial effects:

[0022] Compared with the prior art, the present invention:

[0023] This invention proposes a compact circularly polarized RTM that achieves precise far-field radiation mode manipulation. The two superimposed E-shaped elements exhibit 2π transmission phase coverage and high transmittance. Furthermore, a far-field reconstruction method is introduced to achieve customized manipulation of far-field modes based on phase mask optimization. Several RTM prototypes were designed and fabricated, enabling energy-controlled multi-beam, wide-angle deflection, and custom beamforming to verify the superior performance of the proposed RTM and far-field reconstruction method, which may be a potential candidate for compact wave modulation elements.

[0024] Due to space feed limitations, metasurfaces in the microwave band are often difficult to apply to integrated systems. This invention presents a compact circularly polarized radiating metasurface (RTM) that achieves precise far-field radiation mode manipulation. Two E-shaped units are stacked to form a geometrically phase-based metasurface unit, achieving transmission phase coverage and high transmittance over a 2π range. The method of this invention enables highly customized far-field mode control. Several proof-of-concept RTM prototypes were designed and fabricated, achieving energy-controlled multi-beaming, wide-angle deflection, and customized beamforming. Simulation and experimental results validate the proposed method and RTM structure, which provides a compact wavefront modulation platform capable of highly precise and customized far-field beamforming. Attached Figure Description

[0025] To more clearly illustrate the specific embodiments of the present invention or the technical solutions in the prior art, the drawings used in the description of the specific embodiments or the prior art will be briefly introduced below. Obviously, the drawings described below are some embodiments of the present invention. For those skilled in the art, other drawings can be obtained from these drawings without creative effort.

[0026] Figure 1 is a schematic diagram of an e-shaped metasurface unit;

[0027] Figure 2 is a schematic diagram of the TE unit;

[0028] Figure 3 is a schematic diagram of the far-field reconstruction algorithm;

[0029] Figure 4 is a schematic diagram of a 16×16 feeder network;

[0030] Figure 5 is a schematic diagram of the power divider;

[0031] Figure 6 Far-field simulation pattern for multi-beam directional manipulation;

[0032] Figure 7 Simulate 2D radiation pattern for eight beams;

[0033] Figure 8 The far-field simulation pattern for multi-beam energy modulation;

[0034] Figure 9 The far-field simulation pattern for complex beam modulation;

[0035] Figure 10 The far-field simulation pattern of the broadband radiation characteristics in the range of 9.6 GHz to 10.4 GHz is shown.

[0036] Figure 11 A schematic diagram of the far-field radiation pattern measurement device and front and rear views of the manufactured sample;

[0037] Figure 12 These are the experimental results. Detailed Implementation

[0038] The technical solution of the present invention will now be clearly and completely described with reference to the accompanying drawings. Obviously, the described embodiments are only some, not all, of the embodiments of the present invention. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0039] In the description of this invention, it should be noted that the terms "center," "upper," "lower," "left," "right," "vertical," "horizontal," "inner," and "outer," etc., indicate the orientation or positional relationship based on the orientation or positional relationship shown in the accompanying drawings. They are used only for the convenience of describing the invention and for simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, they should not be construed as limitations on the invention. Furthermore, the terms "first," "second," and "third" are used for descriptive purposes only and should not be construed as indicating or implying relative importance.

[0040] In the description of this invention, it should be noted that, unless otherwise explicitly specified and limited, the terms "installation," "connection," and "linking" should be interpreted broadly. For example, they can refer to a fixed connection, a detachable connection, or an integral connection; they can refer to a mechanical connection or an electrical connection; they can refer to a direct connection or an indirect connection through an intermediate medium; and they can refer to the internal connection of two components. Those skilled in the art can understand the specific meaning of the above terms in this invention based on the specific circumstances.

[0041] Furthermore, the technical features involved in the different embodiments of the present invention described below can be combined with each other as long as they do not conflict with each other.

[0042] The present invention will be described in detail below with reference to specific embodiments. Specific Implementation

[0043] according to Figures 1 to 12 As shown, the specific optimized technical solution adopted by the present invention to solve the above-mentioned technical problems is: The present invention relates to a high degree of freedom far-field customized compact circularly polarized planar lens antenna.

[0044] A highly customizable, compact circularly polarized planar lens antenna for far-field use, wherein the metasurface comprises a 16×16 element array along the x and y directions, respectively. The feed network of the 16×16 element array is obtained by cascading several 1 to 2 power dividers. A 50Ω feed microstrip extends inward and is connected to a 1 to 2 power divider. The two output ports of the power divider are connected to a new power divider, and then the process is repeated to form a 1 to 256 power divider network.

[0045] Power dividers in a power supply network can be classified into three types, referred to as the first power divider, the second power divider, and the third power divider. The first power divider is located at the power supply end, with an input port impedance of 50Ω and an output port impedance of 100Ω. The second power divider has an input port impedance of 100Ω and an output port impedance of 100Ω. The third power divider has an input impedance of 100Ω and its output port is connected to the cell. The power divider at the power supply end has an input impedance of 50Ω and both output ports are designed to have an impedance of 100Ω.

[0046] Using 100Ω microstrip lines as the transmission lines that make up the majority of the power supply network helps save wiring space.

[0047] A 16×16 unit array is formed by stacking two e-shaped TE elements together to form a TE metasurface unit. Metal vias extend to the top layer, and two microstrip lines are divided into two e-shaped structures, which are connected in parallel.

[0048] The e-shaped TE element is a three-layer structure, occupying an area of ​​p × p × h mm. 3 The top layer is an E-shaped patch, and the bottom layer contains a feed microstrip line for impedance matching, with an input impedance of 100Ω. The middle copper layer serves not only as the ground for the top and bottom layers but also as a connection via vias. The copper layer is 0.035 mm thick, and the dielectric substrates are h1 = 3 mm and h2 = 1.27 mm thick, respectively.

[0049] The present invention provides an antenna based on a custom far-field beamforming phase-modulated radiating metasurface.

[0050] This invention provides a fast far-field reconstruction method based on far-field phase mask optimization, the method comprising:

[0051] Step 1: Input the amplitude and phase distribution of the far field. The target amplitude distribution is a 360×90 matrix, which corresponds to the projection of a three-dimensional far-field radiation mode onto a two-dimensional plane. When the target beam is a pencil beam, only the elements corresponding to that direction are assigned; when the target beam is a wide beam, the elements corresponding to the directions covered by the wide beam are assigned.

[0052] Step 2: Begin the iterative process to obtain the metasurface aperture field distribution;

[0053] The present invention provides a computer-readable storage medium having a computer program stored thereon, characterized in that the program is executed by a processor to implement a fast far-field reconstruction method based on far-field phase mask optimization.

[0054] The present invention provides a computer device, including a memory and a processor, wherein the memory stores a computer program, characterized in that: when the processor executes the computer program, it implements a fast far-field reconstruction method based on far-field phase mask optimization. Specific Implementation

[0055] The only difference between Embodiment 2 and Embodiment 1 of this application is that:

[0056] In summary, the goal of RTM is to eliminate the bulky external feed of traditional metasurfaces to achieve high compactness, while maintaining the diverse electromagnetic wave modulation capabilities of traditional reflective and transmissive metasurfaces. RTM metasurface cells not only achieve full-phase modulation but also function as efficient radiators with compact profiles; the former is achieved through geometric phase modulation, while the latter relies on the feed network. Compared to existing sequentially rotating arrays (SRAs) that effectively utilize the theory that the excitation phase is related to the rotation angle of linearly polarized (LP) array elements, RTM extends the traditional SRA design by focusing on the complex amplitude distribution of radiation and the polarization of each cell, further unlocking the platform's potential for more advanced and diverse electromagnetic wave manipulation capabilities.

[0057] Figure 1The e-shaped metasurface cell of the phase-modulated RTM is shown. This cell is a three-layer structure occupying an area of ​​p × p × h mm³. The top layer is an E-shaped patch. The bottom layer contains the feed microstrip line for impedance matching, with an input impedance of 100Ω. The middle copper layer serves not only as the ground for the top and bottom layers but also connects them via vias. The copper thickness is 0.035 mm, while the thicknesses of the F4BM-2 dielectric substrates are h1 = 3 mm and h2 = 1.27 mm, respectively. The bottom dielectric substrate is thinner to facilitate a smaller feed network size.

[0058] First, the radiation and impedance characteristics of a single E-shaped (SE) cell were considered. To generate right-hand circularly polarized (RCP) radiation, the SE cell extends in a right-hand direction from one end of the top-layer feed microstrip line. Figure 1 (b) describes the specific parameters of the SE element. The radius r and opening angle α of the SE element determine the resonant frequency of the radiated electromagnetic wave. This invention requires adjusting r and α to achieve resonance at 10 GHz. The radiation response of the SE element is analyzed by the frequency domain solver in CST Microwave Studio.

[0059] Specific parameters of the SE cell (p = 12 mm, w1 = 0.4 mm, w2 = 1 mm, w3 = 1 mm, w4 = 0.8 mm, l1 = 1.5 mm, l2 = 4.5 mm) and surface current distribution. Dimensions α and r correspond to 10 GHz. S 11 Dimensions α and r correspond to the co-polarization radiation amplitude at 10 GHz. α and r This corresponds to the cross-polarization radiation amplitude at 10 GHz.

[0060] Periodic boundary conditions are set along the x and y directions, and a waveguide port excitation is added at the end of the bottom microstrip line. Figure 1 The surface current distribution on the SE element is also shown. It can be seen that although the SE element generates a right-handed (RH) current path due to the structural orientation, an opposite current path appears at the terminal reflection, resulting in undesirable left-handed (LH) radiation. Figure 1 The effects of α and r on radiation and impedance characteristics were analyzed. Common polarization (RCP) radiation can remain above 0.8 over a wide range, such as... Figure 1 Region 3 is shown. However, S11 and cross-polarized (LCP) radiation vary with α and r. Figure 1In the diagram, regions 1 and 4 correspond to each other, and regions 2 and 5 also correspond. It can be predicted that when dealing with SE metasurface cells, output impedance and low-level cross-polarization radiation cannot be simultaneously satisfied. Region 1 exhibits an S11 level below 20 dB, while region 4 has a cross-polarization radiation amplitude of approximately 0.5. Region 2 exhibits a cross-polarization radiation amplitude below 0.2, but region 5 exhibits an S11 level of approximately -10 dB. The former is unacceptable because high cross-polarization radiation causes it to exhibit linear polarization radiation, while for the latter, an S11 of around -10 dB may result in a large imaginary part or a small real part of the output impedance compared to the ideal output port of 100Ω, which would be detrimental to the feed network design of the RTM array. From another perspective, although most of the energy is fed to the cells, the resonant frequency of the matching point is not the resonant frequency of the SE structure.

[0061] To suppress cross-polarization and achieve excellent output impedance, two e-type (TE) elements are stacked together to form a TE metasurface unit. For example... Figure 2 As shown, a metal via extends to the top layer, and the two microstrip lines are each divided into two E-shaped structures. These two microstrip lines are connected in parallel, which makes the impedance of the overall structure easier to adjust. Figure 2 The surface current distribution of the TE cell is also given. In the case of the SE cell, both RH and LH currents exist. It should be noted that although there is still a backward current generating the LH path in the case of TE, the RH current and LH current between the two E-shaped structures will cancel each other out, which is... Figure 2 This can be clearly observed in the text. Figure 2 The common-polarization and cross-polarization radiation amplitudes of the TE cell are shown. It can be seen that when S11 is below -20 dB, the former remains at 0.9, while the latter is around 0.2. Impedance matching and cross-polarization suppression are both guaranteed in this case. The bandwidth spans from 7.6 to 11 GHz for S11 < -10 dB. Clearly, the TE cell exhibits enhanced performance compared to the SE cell.

[0062] Next, the TE cell is rotated to obtain the geometric phase response. As the cell rotates, the path of the surface current also rotates, introducing an additional phase that is only related to the geometric path. Figure 2 The diagram shows S11 and common-polarization transmission amplitude when the element rotates from 0° to 360°. During the element's rotation, the common-polarization transmittance remains stable at approximately 0.91, and S11 remains around -20 dB, clearly demonstrating that element rotation does not affect radiation performance. Furthermore, from... Figure 2As can be seen, the radiated phase covers the range of 0°–360°, thus achieving full-phase modulation. Meanwhile, the bandwidth of S11 < -10 dB remains almost unchanged between 7.7 and 10.7 GHz. Therefore, the TE cell exhibits better stability, making it possible to implement complex beamforming in RTM arrays that only utilize phase.

[0063] Existing far-field modulation methods focus on a specific modulation dimension, such as wide-angle deflection, multi-beaming, energy control, or fan-beam generation. While some works have considered two modulation dimensions, methods for simultaneously modulating beam direction, beam energy, and beamwidth have not been fully explored, making it challenging to meet the complex and varied application scenarios. Furthermore, in complex far-field constructions with only phase, the limitation of only one modulation degree of freedom often requires multiple iterations and optimizations to achieve the desired functionality. Therefore, it is urgent to obtain the phase distribution of the modulation device effectively and accurately according to specific requirements. Thus, a fast far-field reconstruction strategy based on far-field phase mask optimization is introduced, which can obtain adjustable parameters of the electromagnetic modulation device from any customized application scenario.

[0064] Figure 3 The principle of the proposed phase-modulated far-field reconstruction strategy is explained in detail. Metasurface aperture field ( Figure 3 The blue rectangular disk in the image is discretized in a Cartesian coordinate system, and the far field ( Figure 3 The orange disk in the image is also represented in polar coordinates. Therefore, both the metasurface aperture field and far field consist of an M×N grid with a discretized complex amplitude (CA) distribution. The amplitude and phase distributions of the far field correspond to the dashed and solid ellipses above the orange disk, respectively, while the amplitude and phase distributions of the elemental pore field correspond to the dashed and solid rectangles above the rectangular blue disk, respectively. The entire far-field reconstruction algorithm consists of three parts: input, iterative process, and output. Figure 3 The descriptions are in the three dashed boxes.

[0065] First, the input includes the amplitude and phase distribution of the far field. Target amplitude distribution. A target This represents the required far-field beamforming, phase distribution. P random These values ​​are randomly generated and used as initial values ​​for phase mask optimization. The target amplitude distribution is a 360×90 matrix, corresponding to the projection of a three-dimensional far-field radiation mode onto a two-dimensional plane. Each element of the matrix corresponds to an angle from 0 to 90° in the pitch plane (vertical plane). theta The changes, and the angles from 0 to 360° in the pitch plane. phiThe azimuth (horizontal) ranges from 0 to 360°. If an element of the matrix is ​​assigned a value, it means that a target beam is generated in the direction corresponding to that element (theta, phi). The element values ​​range from 0 to 1, corresponding to the normalized intensity values. When the target beam is a pencil beam, only the element corresponding to that direction is assigned; when the target beam is a wide beam, the element corresponding to the direction covered by the wide beam is assigned. Based on this principle, flexible control of the target beam can be achieved.

[0066] Next, the iterative process begins. To obtain the metasurface aperture field distribution, a discrete two-dimensional inverse Fourier transform is performed on the far-field CA distribution, thereby obtaining the initial CA distribution of the metasurface aperture field. A meta And Pmeta. Phase-modulated metasurfaces tend to ensure high transmittance to improve transmission efficiency; therefore, for the proposed phase-only platform, Ameta is reset to 1, and the phase distribution is preserved.

[0067] Due to its adaptability and accuracy, the Particle Swarm Optimization (PSO) algorithm is widely used in metasurface design, such as reducing radar cross-section (RCS), improving antenna gain, and achromatic focusing. Existing work tends to use PSO to directly optimize the aperture field distribution of the metasurface, such as the arrangement of elements. Here, we apply the PSO algorithm iteratively to the above process, using the far-field phase distribution as the phase mask optimization variable to minimize the loss function, which provides freedom for the modulation of the far-field amplitude distribution. In each iteration, the far-field amplitude distribution remains the target amplitude distribution, while the far-field phase distribution is replaced by the previously calculated phase. The entire iteration is terminated when the MSE tends to stabilize with the number of iterations.

[0068] Finally, the desired phase distribution of the metasurface aperture field can be obtained from the phase distribution obtained in the previous iteration. Through the above process, far-field beamforming can be easily manipulated arbitrarily simply by adjusting the target far-field amplitude distribution.

[0069] In summary, the algorithm of this invention exhibits great flexibility and simplicity. The desired phase distribution can be obtained by simply presetting the target far-field amplitude and performing multiple iterative calculations until convergence. This direct manipulation of the far-field amplitude distribution significantly reduces design complexity and provides considerable design freedom for handling various complex application scenarios.

[0070] The feed network and cells are combined to form the basic structure of the RTM. Each RTM consists of 16×16 cells, located along the x and y directions. Therefore, a 1-to-256 power divider network was designed to ensure power distribution among the cells.

[0071] The 16×16 array feed network is obtained by cascading several 1 to 2 power dividers. A 50Ω feed microstrip extends inward and is connected to a 1 to 2 power divider. The two output ports of this power divider are connected to a new power divider, and this process is repeated to form a 1 to 256 power divider network, as shown below. Figure 4 As shown, the power dividers in the feed network can be divided into three types, referred to as the first power divider #1 (PD#1), the second power divider #2 (PD#2), and the third power divider #3 (PD#3). PD#1 is located at the feed end, with an input port impedance of 50Ω and an output port impedance of 100Ω. PD#2 has an input port impedance of 100Ω and an output port impedance of 100Ω. PD#3 has an input impedance of 100Ω, and its output port is connected to the cell. For ease of subsequent measurement requirements, the input impedance of the power divider at the feed end is 50Ω, and the impedance of both output ports is designed to be 100Ω. Since the characteristic impedance of a microstrip line is inversely proportional to its width, using a 100Ω microstrip line as the transmission line (TL) that occupies most of the feed network helps save wiring space and avoids overlap caused by excessively wide microstrip lines.

[0072] First power divider #1 (W4 = 0.8 mm, L4 = 21 mm, W5 = 4 mm, W6 = 2.1 mm), second power divider #2 (L5 = 5.7 mm, W7 = 1.9 mm), third power divider #3 (W3 = 1 mm, W8 = 1.6 mm, L6 = 3.9 mm, L7 = 3 mm, L8 = 4.5 mm)

[0073] Here, the characteristics of three power dividers are analyzed in detail. Figure 5 Three power divider structures are illustrated, demonstrating how adding a transition line (TL) between the input and output achieves good impedance matching. Notably, a curved transition line is added to PD#3 because the space available for the feeding structure is limited when feeding the cells; the curved transmission line avoids TL overlap, improving impedance performance. Figure 5 As shown, all three power dividers exhibit excellent impedance characteristics, achieving a -10dB impedance bandwidth of 8GHz-12GHz. Meanwhile, S21 and S31 show a power distribution performance of around -3dB, indicating a 1 to 2 power distribution capability. Therefore, this structure can serve as the basis for designing a 16×16 feeder network, based on... Figure 4 The design strategy shown can effectively distribute equal amounts of energy among the cells, achieving the desired far-field radiation.

[0074] B. Numerical Results

[0075] In summary, the proposed far-field reconstruction method is universal for most metasurface platforms with only phase characteristics. Using the aforementioned elements as phase modulators, an RTM was constructed in CST Microwave Studio for simulation verification. Each RTM consists of 16×16 elements along both the x and y directions. A carefully designed feed network ensures comprehensive power feeding of the elements, significantly reducing the longitudinal dimension of the metasurface. Furthermore, the far-field distribution of the RCP was observed at 10 GHz.

[0076] First, the effectiveness of the proposed method in manipulating beam direction is demonstrated. Eight RTMs were designed to prove the ability to control beam direction and beam number. Figure 6 As shown, eight samples were simulated, with the number of equal-energy beams gradually increasing from one to eight, θ=30°, and intervals of 45°. From Figure 6 As can be seen, the direction and number of main beams are consistent with the expected settings. However, the results deteriorate slightly with an increase in the number of beams. With fewer beams, such as a single or dual beam, the direction of the main beam is very clear, and the energy of the sidelobes is almost invisible. With an increase in the number of beams, especially seven or eight beams, the sidelobes increase, which often becomes non-negligible. Figure 6 The eight bundles in the middle show the greatest deterioration of the sidelobe. Figure 7 The diagram shows the two-dimensional radiation modes in four planes for two array sizes in the eight-beam configuration. It is clear that higher sidelobes appear in the phi=0° plane, and the difference between the main lobe and sidelobes is approximately 8 dB, which is an acceptable but not superior result. Figure 7 As shown. Furthermore, the beams in other phi planes (45° to 135°) exhibit sidelobe levels below 15 dB. Additionally, fluctuations in the main beam energy were observed in the phi=45° plane. These sidelobe and main beam energy fluctuations may originate from changes in the boundary conditions of the elements on the metasurface. In element simulations, the elements are set to periodic boundary conditions. When dealing with metasurface arrays, the elements are surrounded by different elements, which inevitably affects the phase and amplitude of the working radiation. The coupling between elements is not negligible, which also leads to the generation of sidelobes and energy fluctuations. As... Figure 7 As shown, the performance of multi-beams can be further improved by increasing the size of the array. Furthermore, this invention notes that in the phi=0° plane, the back lobe exhibits significant back-radiation, differing from the main lobe by approximately 8 dB.

[0077] Table 1. Simulation gain and aperture efficiency of 8 multibeam samples.

[0078] type 1 2 3 4 5 6 7 8 <![CDATA[ G peak (dBi)]]> 24.4 20.5 18.5 17.3 16.8 15.3 14.8 14.3 <![CDATA[ η sim (%)]]> 53.5 42.3 41.3 41.7 44.3 39.1 39.0 38.4

[0079] This primarily originates from the radiation from the bottom microstrip feed network. The feed network consists of multiple cascaded 1-to-2 power dividers. To some extent, the feed microstrip also functions as a microstrip patch antenna array, operating at the same frequency as the metaplane, thus exhibiting symmetrical dual-beam radiation characteristics. The high back lobe issue can be mitigated by appropriately selecting the number of array elements and further improving the feed network design.

[0080] Next, six samples were designed to demonstrate the ability to control beam direction, beam energy, and beamwidth, namely Sample I to Sample VI. Figure 8 The modulation of the wide-angle beam is shown. Sample I has beams preset at θ=20° and θ=60° on the same plane with an energy ratio of 1:1. Sample I has a peak gain of 20.3 dBi and an aperture efficiency of 42.3%. Simulation results show that the main direction meets the requirements with only a deviation of 0.1 dB. The cross-polarization level in the direction of maximum radiation is below -20 dB. In addition, there are sidelobes of nearly 6 dB in other directions, which are caused by the large impedance shift of the element at large angles, resulting in phase instability.

[0081] Furthermore, the ability to manipulate multi-beam energy modulation in different directions was verified. Sample II consists of four beams in a plane with preset energy ratios of -40°, -15°, 25°, 55° to -4, 0, 0, -6 (normalized dB values). Figure 8 The simulation results of the three-dimensional far-field radiation pattern and two-dimensional radiation pattern of Sample II in the phi=0° plane at 10 GHz are shown. The simulation results of Sample II are four main beams with beam directions of -40°, -15°, 26°, and 54°, and energy ratios of -4, -0.2, 0, and -5.9. It should be noted that the simulation results are basically consistent with the preset values, and the sidelobes are well suppressed below 10 dB. Sample II achieves a peak gain of 20.0 dBi and an aperture efficiency of 51.4%. The cross-polarization level in the direction of maximum radiation is below -20 dB. Figure 8 As shown, the simulation results for Sample III are four main beams with beam directions of -50°, -20°, 20°, and 45°, and energy ratios of -0, -6, 0, and -6. As shown in the figure, the simulation results for Sample III are also four main beams with beam directions of -49°, -22°, 20°, and 45°, and energy ratios of -0.4, -5.8, 0, and -5.2. Sample III achieves a peak gain of 20.2 dBi and an aperture efficiency of 51.1%. The cross-polarization level in the direction of maximum radiation is below -20 dB. The overall trend is consistent with the expected target. Although there are relatively moderate deviations in the preset energy and angle, all these deviations are within a reasonable and acceptable range (angle deviation less than 2°, energy deviation less than 1 dB).

[0082] The preceding analysis has sufficiently demonstrated the modulation capability of the method of this invention for arbitrary beam directions. Here, we consider beam energy modulation on different planes to achieve full-space beam energy control, and present two samples. Sample four involves four beams generated on two planes, where the beam directions on the phi=0° plane are preset to -60°, 20°, with an energy ratio of 0, 0; and the beam directions on the phi=90° plane are preset to -45°, 25°, with an energy ratio of 0, -3. Figure 9 The simulated far-field patterns of the four main beams are shown. In the phi=0° plane, the beam directions are -62° and 22°, with energy ratios of -1.1 and 0. In the phi=90° plane, the beam directions are -45° and 24°, with energy ratios of -2.9 and 0. The peak gain of sample IV is 18.8 dBi, and the aperture efficiency is 48.1%. The cross-polarization level in the direction of maximum radiation is below -20 dB. Figure 9 Simulated far-field patterns are shown for two beams with preset beam directions of -50° and 30° and preset energy ratios of -4 and 0 in the phi=0° plane, and for four main beams with preset beam directions of -40° and 20° and preset energy ratios of -6 and 0 in the phi=90° plane. Sample IV has a peak gain of 19.3 dBi and an aperture efficiency of 42.3%. The cross-polarization level in the direction of maximum radiation is below -20 dB. All simulation results are in excellent agreement with the preset results. It is noted that the performance of spatial multi-beam energy modulation is good even at large angles and with significant energy ratio differences. The simulated sidelobes are all below 8 dB. This demonstrates the excellent performance of the proposed far-field reconstruction method for beam energy control in space based on RTM.

[0083] Finally, combining the proposed far-field optimization algorithm for beam manipulation, beamwidth modulation was achieved. Precise modulation of the fan-shaped beam is crucial in far-field holography and wireless communication. Sample VI has three pencil beams with preset beam directions of -40°, -25°, and 15°, preset energy ratios of --3, 0, and --3, and a 30° wide fan-shaped beam. Figure 9As shown, the simulation results for sample VI are in good agreement with the preset values, with beam directions of -42°, -27°, and -14°, energy ratios of -3.2, 0, and -2.8, and a fan beam width of 31°. Sample VI achieves a peak gain of 17.7 dBi and an aperture efficiency of 33.2%. The cross-polarization level in the maximum radiation direction is below -20 dB. For complex beams, it is difficult to obtain accurate results using traditional resolution methods, so the method of this invention has a significant advantage. Through algorithm optimization, the main lobe of samples I-VI is consistent with the far-field distribution of the target, but there are still obvious sidelobes and some power diffusion. Among them, the large backlobes in samples IV-VI still originate from the radiation of the feed network. The 8 dB sidelobes in samples IV-V are caused by element coupling and phase inconsistency. By introducing sidelobe suppression to improve the algorithm, the overall sidelobe level can be effectively controlled below -10 dB. In addition, the algorithm of this invention has great flexibility and improvement potential; for example, further introducing a sidelobe optimization algorithm can improve the overall radiation performance.

[0084] Figure 9 These are the simulated S11 results for samples I through VI. S11 is less than -10 dB across the desired frequency bands. Furthermore, broadband radiation characteristics have been considered. Figure 10 Simulated radiation profiles for samples III, IV, and VI from 9.6 GHz to 10.4 GHz are provided. Within the 9.6 to 10.4 GHz range, deflection stability is generally guaranteed. However, this invention notes the problem of energy distribution instability. The larger the deflection angle, the greater the energy instability, primarily due to frequency-dependent phase fluctuations and uneven energy distribution in the feed network. Furthermore, the proposed algorithm's applicability to a single frequency point with broadband instability also contributes to broadband radiation fluctuations. Optimizing the feed network design is also a feasible method to improve the radiation bandwidth, for example, by using substrate integrated coaxial lines (SICL). Overall, the proposed device maintains essentially stable radiation characteristics within the 9.6 GHz to 10.4 GHz range, which is largely consistent with the simulated S11 < -10 dB bandwidth.

[0085] Therefore, the proposed method enables flexible control over the beam direction, beam energy, and beamwidth generated by a phase-only RTM. Furthermore, the RTM, as a highly integrated and compact platform, holds great promise for miniaturizing electromagnetic devices.

[0086] Here, samples III, IV, and VI were selected for experimental verification in a microwave anechoic chamber. (Samples 216 × 216 mm) 2The coaxial terminal is fabricated using printed circuit board (PCB) etching technology. For ease of testing, the inner conductor of the coaxial terminal is connected to the output 50Ω microstrip line of the feed network, exposing a small portion of the metal ground line on the dielectric substrate, connecting the outer conductor to the metal ground line of the intermediate layer. For example... Figure 11 As shown, a broadband horn antenna, acting as a receiver, is placed far from the sample. Both the receiving horn antenna and the RTMs are connected to a vector network analyzer via coaxial cables to measure the transmission coefficient, which reflects the antenna radiation pattern as the RTM mounting platform rotates. The far-field patterns in the phi=0° and phi=90° planes are measured by rotating the worktable and adjusting the orientation of the metasurface antenna.

[0087] Figure 12 The two-dimensional radiation patterns of samples III, IV, and VI measured in the phi=0° and phi=90° planes are shown. For sample III, the measured radiation pattern shows beam directions of -48°, -20°, 19°, and 46°, with energy ratios of --1, --5.7, 0, and --5.1. For sample VI, the measured radiation pattern shows a fan-shaped beam with a width of 31° and multiple pencil beams with preset directions of -42°, -26°, and -16°, and energies of --2.3, 0, and --2.6. For sample IV, the measured radiation pattern shows that in the phi=0° plane, the dual-beam beam pointing at -62° and 20° with energy ratios of --1 and 0, and in the phi=90° plane, the dual-beam beam pointing at -45° and 24° with energy ratios of --3.5 and 0. The discrepancy between the measured and simulated sidelobes may originate from the metal rotating platform. The measured peak gains of samples III, IV, and VI were 19.9 dBi, 18.4 dBi, and 17.4 dBi, respectively. The aperture efficiencies of samples III, IV, and VI were 50.3%, 47.2%, and 30.8%, respectively. The cross-polarization level measured in the direction of maximum radiation was below -20 dB. The proposed far-field optimization algorithm achieves customizable adjustment of the far-field distribution through phase-only design, and the RTMs exhibit miniaturized integration performance, thus realizing efficient automatic arbitrary beam modulation synthesis for multi-beam antenna design applications.

[0088] This invention proposes a general far-field beam manipulation method validated by compact RTMs, enabling energy-customizable multi-beam, wide-angle deflection, and customized beamforming. Two E-shaped elements construct fully phase-modulated RTMs, achieving superior radiation performance. Several proof-of-concept RTM prototypes were designed to verify the beam manipulation capability. Furthermore, three typical RTMs were fabricated and measured. Simulations and measurements showed excellent agreement with the pre-set results. The proposed method is not limited to metasurface demonstration platforms and can be extended to higher frequency bands, providing new insights for the design of beam manipulation devices.

[0089] In the description of this specification, references to terms such as "one embodiment," "some embodiments," "example," "specific example," or "some examples," etc., indicate that a specific feature, structure, material, or characteristic described in connection with that embodiment or example is included in at least one embodiment or example of the present invention. In this specification, the illustrative expressions of the above terms do not necessarily refer to the same embodiment or example. Moreover, the specific features, structures, materials, or characteristics described may be combined in any suitable manner in one or N embodiments or examples. Furthermore, those skilled in the art can combine and integrate the different embodiments or examples described in this specification and the features of different embodiments or examples without contradiction. Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of indicated technical features. Thus, a feature defined as "first" or "second" may explicitly or implicitly include at least one of that feature. In the description of the present invention, "N" means at least two, such as two, three, etc., unless otherwise explicitly specified. Any process or method described in the flowcharts or otherwise herein can be understood as representing a module, segment, or portion of code comprising one or more N executable instructions for implementing custom logical functions or processes, and the scope of preferred embodiments of the invention includes additional implementations in which functions may be performed not in the order shown or discussed, including substantially simultaneously or in reverse order according to the functions involved, as will be understood by those skilled in the art to which embodiments of the invention pertain. The logic and / or steps represented in the flowcharts or otherwise described herein, for example, can be considered as a ordered list of executable instructions for implementing logical functions, and can be embodied in any computer-readable medium for use by, or in conjunction with, an instruction execution system, apparatus, or device (such as a computer-based system, a processor-included system, or other system that can fetch and execute instructions from, an instruction execution system, apparatus, or device). For the purposes of this specification, "computer-readable medium" can be any means that can contain, store, communicate, propagate, or transmit programs for use by, or in conjunction with, an instruction execution system, apparatus, or device. More specific examples of computer-readable media (a non-exhaustive list) include the following: electrical connections having one or N wires (electronic devices), portable computer disks (magnetic devices), random access memory (RAM), read-only memory (ROM), erasable and editable read-only memory (EPROM or flash memory), fiber optic devices, and portable optical disc read-only memory (CDROM).Furthermore, the computer-readable medium can even be paper or other suitable media on which the program can be printed, since the program can be obtained electronically, for example, by optically scanning the paper or other medium, followed by editing, interpreting, or otherwise processing as necessary, and then stored in a computer memory. It should be understood that various parts of the invention can be implemented in hardware, software, firmware, or a combination thereof. In the above embodiments, the N steps or methods can be implemented in software or firmware stored in memory and executed by a suitable instruction execution system. For example, if implemented in hardware, as in another embodiment, it can be implemented using any one or a combination of the following techniques known in the art: discrete logic circuits having logic gates for implementing logical functions on data signals, application-specific integrated circuits (ASICs) having suitable combinational logic gates, programmable gate arrays (PGAs), field-programmable gate arrays (FPGAs), etc.

[0090] The above description is merely a preferred embodiment of a high-degree-of-freedom, far-field customized compact circularly polarized planar lens antenna. The scope of protection for such an antenna is not limited to the above embodiments; all technical solutions falling within this conceptual framework are within the protection scope of this invention. It should be noted that for those skilled in the art, any improvements and variations made without departing from the principles of this invention should also be considered within the protection scope of this invention.

Claims

1. A high degree of freedom far field customized compact circularly polarized planar lens antenna, characterized by: The 16*16 unit array is superimposed by two e-shaped TE elements along the x direction and the y direction respectively to form a TE metasurface unit, and the feeding network of the 16*16 unit array is obtained by cascading several 1 / 2 power dividers; a 50Ω feeding microstrip extends inward and is connected to a 1 / 2 power divider, the two output ports of the power divider are connected to a new power divider, and then a 1 / 256 power divider network is formed by circulation; The power dividers in the feeding network can be divided into three types, which are respectively referred to as a first power divider, a second power divider and a third power divider, wherein the first power divider is located at the feeding end, the input port impedance is 50Ω, and the output port impedance is 100Ω; the input port impedance of the second power divider is 100Ω, and the output port impedance is 100Ω; the input impedance of the third power divider is 100Ω, and the output port is connected to the unit; the input impedance of the power divider at the feeding end is 50Ω, and the impedance of the two output ports is designed as 100Ω; The 100Ω microstrip line is used as the transmission line which occupies most of the feeding network, which is beneficial to save wiring space; The e-shaped TE element is a three-layer structure with a size of p x p x h mm 3 The top layer is an e-shaped patch, and the bottom layer includes a feed microstrip line for impedance matching, wherein the input impedance is 100 Ω. The middle copper layer not only serves as the ground of the top and bottom layers, but also is connected through a through hole. The thickness of the copper is 0.035 mm, the metal through hole extends to the top layer, and the two microstrip lines are respectively divided into two e-shaped structures, and the two microstrip lines are connected in parallel.

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