Permanent magnet motor model predictive current control method and system

Abstract

The application discloses a kind of permanent magnet motor model prediction current control method and system, it is equipped with speed outer ring control and current inner ring control, wherein current outer ring control: the difference of speed reference value and speed feedback value is obtained by PI controller to obtain stator current q-axis component reference value, stator current d-axis component reference value is given as 0;Current inner ring control: using the stator voltage sample value and current sample value collected at k time to obtain stator current feedback value, and then calculate current error, current error is obtained after cost function minimization module to obtain the initial switching signal duty ratio of three-phase two-level inverter, and d max , d mid And d min It is obtained by sorting from big to small, and finally reconstruction module obtains final switching signal duty ratio d A ', d B ' and d C ', finally using pulse width modulation technology generates the driving signal of inverter. The steady-state performance of model predictive control (MPC) can be effectively improved by the technical scheme of the application.

Description

Technical Field

[0001] The present invention belongs to the technical field of motor control, and in particular relates to a permanent magnet motor model prediction current control method and system. Background Art

[0002] Permanent magnet synchronous motors (PMSMs) have been widely used in industry due to their high efficiency, high reliability, and high torque and power density. In recent years, with the rapid and stable development of microprocessors such as DSPs and FPGAs, finite set model predictive control (FCS-MPC) has become a mature control strategy for high-performance motor drives. FCS-MPC offers advantages such as a simple control concept, fast dynamic response, the ability to handle nonlinear constraints, and ease of multivariable control.

[0003] Because traditional FCS-MPC applies only one voltage vector per control cycle, this can lead to high steady-state ripple caused by a low average switching frequency. Therefore, to achieve better control performance, traditional FCS-MPC requires increasing the sampling frequency, which undoubtedly increases the hardware requirements. Therefore, it is necessary to improve the steady-state performance of traditional FCS-MPC without increasing the sampling frequency.

[0004] To address these issues, commonly used methods include multi-step predictive control (MPC) and dual-vector predictive control (DVC). Multi-step predictive control improves the control performance of the FCS-MPC algorithm by increasing its prediction range. However, as the prediction range increases, the computational burden of the control algorithm also increases. Dual-vector predictive control applies two voltage vectors within a control cycle, improving steady-state performance. However, DVC cannot theoretically eliminate steady-state errors and requires an enumeration process to confirm the two voltage vectors, increasing the algorithmic complexity. Summary of the Invention

[0005] In response to the shortcomings of the existing technology, especially the steady-state errors and overly complex algorithms of traditional dual-vector predictive control, the present invention provides a permanent magnet motor model predictive current control method and system, which reduces the computational complexity of predictive control without increasing the sampling frequency and obtaining good steady-state performance.

[0006] In one aspect, the present invention provides a permanent magnet motor model prediction current control method, which includes the following steps:

[0007] In each sampling period, the permanent magnet motor is sampled to obtain at least the stator voltage sampling value and stator current sampling value k corresponds to the sampling moment or sampling point;

[0008] Using the stator voltage sampling value The stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment

[0009] Get the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system

[0010] Based on the stator current feedback values ​​of the d-axis and q-axis The stator current reference values ​​of the d-axis and q-axis Calculate the current error in the dq rotating coordinate system

[0011] Based on minimizing the control cost function, the current error Get the initial switching signal duty cycle d of the three-phase two-level inverter A d B and d C , and sort them from largest to smallest to get d max d mid and d min ;

[0012] Based on the sorted three-phase duty cycle d max d mid and d min The duty cycle is reconstructed to obtain the final switching signal duty cycle d of the three-phase two-level inverter. A '、d B ' and d C ';

[0013] According to the final switch duty cycle d A '、d B ' and d C 'Generate driving signals for three-phase two-level inverter.

[0014] The technical solution of the present invention starts from the perspective of the three-phase duty cycle, sets the three-phase duty cycle as the key control variable, and directly obtains the duty cycle of each phase of the inverter through vector synthesis and cost function minimization. Compared with the traditional predictive current control method, it omits the process of determining the voltage vector, calculating the vector action time, and then converting it into a three-phase duty cycle, eliminates the enumeration process of minimizing the cost function, and reduces the complexity of the algorithm. In addition, by reconstructing the duty cycle, smaller current ripple and better steady-state performance are obtained.

[0015] Further optionally, the final switching signal duty cycle d of the three-phase two-level inverter A '、d B ' and d C 'Calculated by the following formula:

[0016]

[0017] Where, d0=1-(d max -d mid )-(d mid -d min )=1-(d max -d min ).

[0018] Further optionally, the initial switching signal duty cycle is calculated by the following formula:

[0019]

[0020] Where, d xi ,i∈{1,2,3} is the voltage vector u selected in the horizontal direction x x The three-phase duty cycle; d yi , i∈{1,2,3} is the voltage vector u selected in the vertical direction y y The three-phase duty cycle, d α d β are the duty cycles in the horizontal direction x and the vertical direction y, respectively, which are obtained by taking the partial derivative of the cost function.

[0021] Further optionally, the voltage vector u selected in the horizontal direction x x And the voltage vector u selected in the vertical direction y y The selection process is as follows: first, the current error After coordinate transformation, we can get and Then, based on and The positive and negative signs of are selected and determined from the voltage vector space according to the rules in the table below;

[0022]

[0023] The voltage vector space is composed of two effective voltage vectors (u a ,u c ), two virtual voltage vectors (u b ,u d ) and two zero vectors (u0, u7), the two effective voltage vectors (u a ,u c ) and the two virtual voltage vectors (u b ,u d The three-phase duty ratio of ) is shown in the following table:

[0024]

[0025] Among them, d i1 di2 d i3 is the three-phase duty cycle, and The current error The corresponding d-axis and q-axis current errors are: and is the error in the α and β directions of the two-phase stationary coordinate system obtained through coordinate transformation; u dc is the voltage amplitude on the DC side of the inverter.

[0026] Further optionally, the current error The formula is as follows:

[0027]

[0028] Among them, R s is the stator resistance; L d , L q are the direct-axis component and quadrature-axis component of the stator inductance respectively; ω e is the angular velocity; T s is the sampling time; ψ f is the permanent magnet flux.

[0029] Further optionally, the obtained q-axis stator current reference value The speed reference value and the speed sampling value The difference is input into the PI controller to obtain the stator current reference value Set to 0;

[0030] Among them, the speed sampling value It is obtained by sampling the motor.

[0031] Further optionally, performing the step of using the stator voltage sampling value The stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq coordinate system at the next moment , calculate according to the following formula:

[0032]

[0033]

[0034]

[0035]

[0036] in, The stator current feedback value and voltage value before the next compensation in the dq rotating coordinate system; They are stator current feedback values The corresponding d-axis and q-axis components; The voltage values ​​are Corresponding d-axis and q-axis components; C 3s / 2r is the transformation matrix from the three-phase stationary coordinate system to the two-phase rotating coordinate system; R s is the stator resistance; L d , L q are the direct-axis component and quadrature-axis component of the stator inductance respectively; ω e is the electrical angular velocity; T s is the sampling time; ψ f is the permanent magnet flux; θ is the rotor position angle.

[0037] In a second aspect, the present invention provides a system for predicting current control method based on the permanent magnet motor model, comprising:

[0038] The sampling module is used to sample the permanent magnet motor in each sampling period to obtain at least the stator voltage sampling value. and stator current sampling value k corresponds to the sampling moment or sampling point;

[0039] The stator current feedback value acquisition module is used to use the stator voltage sampling value Stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment

[0040] Stator current reference value acquisition module, used to obtain the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system

[0041] Current error calculation module, used for stator current feedback value based on the d-axis and q-axis Stator current reference values ​​of d-axis and q-axis Calculate the current error in the dq rotating coordinate system

[0042] The initial duty cycle calculation module is used to minimize the control cost function based on the current error Get the initial switching signal duty cycle d of the three-phase two-level inverter A d B and d C , and sort by size to get d max d mid and d min ;

[0043] Reconstruction module for the three-phase duty cycle d maxd mid and d min , the three-phase duty cycle is reconstructed to obtain the final switching signal duty cycle d of the three-phase two-level inverter A '、d B ' and d C ';

[0044] A driving module for switching the output voltage according to the final switching duty cycle d A '、d B ' and d C 'Generate driving signals for three-phase two-level inverter.

[0045] In a third aspect, the present invention provides a system based on the method, which includes a permanent magnet motor and a control subsystem. The control subsystem uses the method to generate a drive signal for a three-phase two-level inverter, thereby controlling the permanent magnet motor.

[0046] In a fourth aspect, the present invention provides a computer-readable storage medium storing a computer program, wherein the computer program is called by a processor to implement:

[0047] Steps of a permanent magnet motor model predictive current control method

[0048] Beneficial effects

[0049] 1. The predictive current control method provided by the technical solution of the present invention calculates the current error using the stator current feedback value and the stator current reference value, and then obtains the initial switching duty cycle by minimizing the control cost function. Compared with traditional predictive current control methods, this method eliminates the process of determining the voltage vector, calculating the vector action time, and converting it to a three-phase duty cycle. It also eliminates the enumeration process of minimizing the cost function, reducing the complexity of the algorithm. In addition, to achieve smaller current ripple and better steady-state performance, the present invention reconstructs the three-phase duty cycle, effectively resolving / mitigating the steady-state error existing in traditional dual-vector predictive control.

[0050] 2. The present invention modifies the traditional voltage vector space and establishes a voltage vector selection table that only considers the current deviation signal. It utilizes the characteristics of zero-beat control to directly calculate the duty cycle of the three-phase switch. That is, from the perspective of the three-phase duty cycle, through vector synthesis, the control cost function is minimized to directly obtain the duty cycle of each phase of the inverter. The whole process is simpler and more effective. Experiments have also proved that compared with the traditional dual-vector predictive control, the technical solution of the present invention achieves a good steady-state effect at a lower sampling frequency. BRIEF DESCRIPTION OF THE DRAWINGS

[0051] Figure 1 The voltage vector space modified for the present invention;

[0052] Figure 2 A structural block diagram of a permanent magnet motor model predictive current control provided by an embodiment of the present invention;

[0053] Figure 3 The experimental results are shown for traditional dual vector predictive control with a sampling frequency of 10kHz and a motor running at 200rpm with rated load.

[0054] Figure 4 The present invention provides an example of a permanent magnet motor model prediction current control method at a sampling frequency of 4kHz, with the motor running at 200rpm with rated load experimental results;

[0055] Figure 5 The experimental results are for traditional dual vector predictive control with a sampling frequency of 10kHz and a motor running at 1000rpm with rated load.

[0056] Figure 6 The present invention provides an example of a permanent magnet motor model prediction current control method at a sampling frequency of 4kHz, with the motor running at 1000rpm with rated load experimental results. DETAILED DESCRIPTION

[0057] The technical solution of the present invention provides a method for predictive current control of a permanent magnet motor model, which reduces the computational complexity of predictive control without increasing the sampling frequency and obtaining good steady-state performance. and speed feedback value Subtract the difference, and pass it through the PI controller to obtain the reference value of the stator current q-axis component Stator current d-axis component reference value Given as 0; Current inner loop control: the stator voltage sampling value of the permanent magnet motor collected at time k Current sampling value After coordinate transformation and one-step delay compensation module, the stator current feedback value in the dq rotating coordinate system at the next moment is obtained Depend on and Calculate the current error of the d-axis and q-axis The initial switching signal duty cycle of the three-phase two-level inverter is obtained by minimizing the control cost function module and sorted from large to small to obtain d max d mid and d min The sorted three-phase duty cycle is passed through the three-phase duty cycle reconstruction module to obtain the final switching signal duty cycle d of each phase of the inverter. A '、d B ' and d C', and finally the pulse width modulation technology is used to generate the driving signal of the inverter.

[0058] The present invention modifies the voltage vector space from the perspective of duty cycle. The modified voltage vector space is as follows: Figure 1 As shown, it includes two effective voltage vectors (u a ,u c ), two virtual voltage vectors (u b ,u d ) and two zero vectors (u0, u7). Two effective voltage vectors (u a ,u c ) and two virtual voltage vectors (u b ,u d The three-phase duty ratios of the zero vectors u0 and u7 are shown in Table 1. The duty ratios corresponding to the two zero vectors u0 and u7 are expressed as: (0, 0, 0) and (1, 1, 1). The present invention will be further described below with reference to embodiments, using a permanent magnet synchronous motor as an example.

[0059] Table 1

[0060]

[0061] Example 1:

[0062] A method for controlling a permanent magnet motor current by model prediction is provided in accordance with a first embodiment of the present invention, and includes the following steps:

[0063] 1) In each sampling period, the permanent magnet synchronous motor is sampled to obtain the stator voltage sampling value Stator current sampling value and speed sampling value

[0064] 2) Sample the stator voltage Stator current sampling value Perform coordinate transformation and obtain the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment through a one-step delay compensation module

[0065] Among them, this embodiment calculates the stator current feedback values ​​of the d-axis and q-axis according to the following formula:

[0066]

[0067]

[0068]

[0069]

[0070] in, The stator current feedback value and voltage value before the next compensation in the dq rotating coordinate system; They are stator current feedback values The corresponding d-axis and q-axis components; The voltage values ​​are Corresponding d-axis and q-axis components; C 3s / 2r is the transformation matrix from the three-phase stationary coordinate system to the two-phase rotating coordinate system; R s is the stator resistance; L d , L q are the direct-axis component and quadrature-axis component of the stator inductance respectively; ω e is the electrical angular velocity; T s is the sampling time; ψ f is the permanent magnet flux; θ is the rotor position angle.

[0071] 3) Obtain the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system About speed reference value and speed sampling value Subtract the difference, and pass it through the PI controller to obtain the stator current reference value of the q-axis component. And the stator current reference value of the d-axis component Given as 0.

[0072] Among them, the stator current reference value of the q-axis component is obtained through the speed outer loop PI regulator The stator current reference value of the q axis is calculated using the following formula:

[0073]

[0074] Where k p 、k i are the proportional gain coefficient and integral gain coefficient of PI regulation respectively; s is the integral sign.

[0075] 4) By and Calculating current error Specifically, the current error is calculated using the following formula: Corresponding d-axis and q-axis current errors and

[0076]

[0077] 5) Based on the minimization of the cost function, the current error is used Get the initial switching signal duty cycle d of the three-phase two-level inverter A dB and d C , and sort them from largest to smallest to get d max d mid and d min .

[0078] In this embodiment, the following control cost function of the traditional model for predicting current is taken as an example. The selected control cost function J is expressed as:

[0079]

[0080] Step 2) Substitute into the above formula and combine with step 4) Calculation formula, get the new control cost function J:

[0081] Then, according to the above current error Coordinate transformation error and Then, based on its positive and negative signs, the sector can be determined and the voltage vector can be selected. The relationship between its value sign and the sector and voltage vector is shown in Table 2.

[0082] Table 2

[0083]

[0084] The voltage vector to be synthesized is decomposed into the following in the x and y directions:

[0085]

[0086] Where u x The voltage vector selected for the horizontal direction; u y The voltage vector selected in the vertical direction, u α syn 、u β syn is the synthesized voltage vector, and the voltage vector is selected according to Table 2.

[0087] In this embodiment, the partial derivative of the control cost function is taken to minimize the control cost function, and u is directly obtained. α syn 、u β syn Corresponding duty cycle in the x and y directions:

[0088]

[0089] Where, d α d βare the duty cycles in the horizontal direction x and the vertical direction y, respectively. It should be noted that, according to the aforementioned technical approach, the use of other types of cost functions is also applicable to the technical approach of the present invention and is not limited to the traditional model prediction current control cost function selected in this embodiment (different types of model prediction algorithms use different functions to evaluate control behavior to obtain corresponding model prediction control laws; these functions are collectively referred to as "value functions / cost functions").

[0090] Then, the initial switching signal duty cycle of each phase of the inverter is:

[0091]

[0092] Among them, d A d B and d C Arrange in descending order to get d max d mid and d min .d xi ,i∈{1,2,3} is the three-phase duty cycle of the horizontal voltage vector; d yi ,i∈{1,2,3} is the three-phase duty cycle of the vertical voltage vector, and the specific value is determined according to Table 1.

[0093] 6) Sorted three-phase duty cycle d max d mid and d min ,The final switching signal of each phase of the inverter is obtained through the three-phase duty cycle reconstruction module.

[0094] Among them, the duty cycle of the switching signal of each phase should be greater than 0 and less than 1. However, the duty cycle of the switching signal constructed by step 5) may exceed this range. Therefore, in step 6), in order to limit the range of the duty cycle of the switching signal of each phase, the final duty cycle of the switching signal of each phase of the inverter d A '、d B ' and d C ' can be calculated by the following formula:

[0095]

[0096] Where, d0=1-(d max -d mid )-(d mid -d min )=1-(d max -d min ).

[0097] For example: For d A >d B >d CIn the case of , the equivalent effective voltage vectors are u1(1,0,0) and u2(1,1,0). The action time of the two equivalent effective voltage vectors is respectively (d A -d B ), (d A -d B ), so when the duty cycle is modified, the two values ​​are kept unchanged, and the action time of the actual effective voltage vector will be the same after modification.

[0098] The specific modification process is as follows:

[0099] Three-phase duty cycle d A d B and d C First, follow the order from maximum to minimum d max d mid and d min The two effective voltage vector action times are respectively and the difference (d max -d mid ) and (d mid -d min ) is proportional to . Therefore d0=1-(d max -d mid )-(d mid -d min )=1-(d max -d min ) is proportional to the action time of the zero voltage vector. 0.5d0 is the average action time of the zero vector. The new duty cycle d A '、d B ' and d C ' can be expressed as:

[0100]

[0101] For example, d max =d A ,d mid =d B ,d min =d C ,d0=1-(d max -d min )=1-(d A -d C ),but:

[0102]

[0103] 7) According to the three-phase duty cycle d A '、d B ' and d C ', using pulse width modulation technology to generate the drive signal of the inverter.

[0104] In order to verify the effectiveness of the method described in the present invention, the present invention conducted a comparative experiment. The effectiveness of the above method can be verified by comparing Figures 3 to 6 The experimental results shown are as follows. The conventional dual-vector model predictive control experiment was conducted at a sampling frequency of 10 kHz, while the permanent magnet motor model predictive current control experiment provided by the present invention was conducted at a sampling frequency of 4 kHz (this is only for illustration, and other sampling frequencies are also applicable). Figures 3 to 6 In each figure, the waveforms from top to bottom are the speed, the stator current q-axis component, the stator current d-axis component, and the stator A-phase current. Figures 3 to 6 is the steady-state experimental result of the motor, Figure 3 and Figure 4 The experimental results of the motor running at 200rpm with rated load are respectively corresponding to the traditional dual vector predictive control and the three-vector model predictive current control method based on duty cycle reconstruction provided by the example of the present invention. Figure 5 and Figure 6 The experimental results of the conventional dual vector predictive control and the permanent magnet motor model predictive current control method provided by the present invention are respectively corresponding to the motor running at 1000rpm with rated load. Figures 3 to 6 From the comparison, it can be found that compared with the traditional dual-vector predictive control, at a lower sampling frequency, the method used in the example of the present invention achieves a good steady-state effect. At the same time, it can also be seen that the permanent magnet motor model predictive current control provided by the example of the present invention can obtain lower stator current d, q-axis component pulsation and a more sinusoidal stator current.

[0105] Example 2:

[0106] Based on the method described in Example 1, this embodiment provides a system for predicting current control method based on the permanent magnet motor model, which includes: a sampling module, a stator current feedback value acquisition module, a stator current reference value acquisition module, a current error calculation module, an initial duty cycle calculation module, a reconstruction module and a drive module.

[0107] The sampling module is used to sample the motor in each sampling period to obtain at least the stator voltage sampling value. and stator current sampling value k corresponds to the sampling moment or sampling point; the stator current feedback value acquisition module is used to use the stator voltage sampling value Stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment The stator current reference value acquisition module is used to obtain the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system. The current error calculation module is used to calculate the stator current feedback value based on the d-axis and q-axis Stator current reference values ​​of d-axis and q-axis Calculate the current error in the dq rotating coordinate system The initial duty cycle calculation module is used to minimize the control cost function based on the current error Get the initial switching signal duty cycle d of the three-phase two-level inverter A d B and d C , and sort by size to get d max d mid and d min ; The reconstruction module is used to calculate the duty cycle of the three phases based on the sorted max d mid and d min , the three-phase duty cycle is reconstructed to obtain the final switching signal duty cycle d of the three-phase two-level inverter A '、d B ' and d C '; The driving module is used to determine the final switch duty cycle d A '、d B ' and d C 'Generate driving signals for three-phase two-level inverter.

[0108] Please refer to the above-mentioned methods for the specific implementation process of each module, and will not be elaborated here. It should be understood that the above-mentioned division of functional modules is merely a division of logical functions. In actual implementation, other division methods can be used. For example, multiple units or components can be combined or integrated into another system, or some features can be ignored or not implemented. At the same time, the above-mentioned integrated units can be implemented in the form of hardware or software functional units.

[0109] Example 3:

[0110] This embodiment provides a system based on the permanent magnet motor model predictive current control method, which includes a permanent magnet synchronous motor and a control subsystem. The control subsystem uses the permanent magnet synchronous motor model-free predictive control method to generate a drive signal for a two-level inverter, thereby controlling the permanent magnet synchronous motor.

[0111] The formation of the control subsystem may be achieved by referring to the method described in Example 2, or a control terminal device storing a program corresponding to the permanent magnet motor model prediction current control method may be used to generate a switch control signal.

[0112] Example 4:

[0113] The present invention provides a computer-readable storage medium storing a computer program, wherein the computer program is called by a processor to implement:

[0114] In each sampling period, the permanent magnet synchronous motor is sampled to obtain the stator voltage sampling value Stator current sampling value and speed sampling value

[0115] The stator voltage sampling value Stator current sampling value Perform coordinate transformation and obtain the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment through a one-step delay compensation module

[0116] Get the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system About speed reference value and speed sampling value Subtract the difference, and pass it through the PI controller to obtain the stator current reference value of the q-axis component. And the stator current reference value of the d-axis component Given as 0.

[0117] Depend on and Calculating current error

[0118] Based on minimizing the control cost function, the current error Get the initial switching signal duty cycle d of the three-phase two-level inverter A d B and d C , and sort them from largest to smallest to get d max d mid and d min .

[0119] Sorted three-phase duty cycle d max d mid and d min ,The final switching signal of each phase of the inverter is obtained through the three-phase duty cycle reconstruction module.

[0120] According to the sorted three-phase duty cycle d A '、d B ' and d C ', using pulse width modulation technology to generate the drive signal of the inverter.

[0121] It should be understood that the implementation process of some steps and whether to execute some steps and the execution order can refer to the implementation process of the above-mentioned embodiment.

[0122] The readable storage medium is a computer-readable storage medium, which can be the internal storage unit of the controller described in any of the aforementioned embodiments, such as the hard disk or memory of the controller. For example, the terrain element model constructed in the present invention exists in the hard disk, and then the computer program that performs the fusion step is stored in the memory, so that the fusion process is implemented based on the memory. The readable storage medium can also be an external storage device of the controller, such as a plug-in hard disk, a smart memory card (Smart Media Card, SMC), a secure digital (Secure Digital, SD) card, a flash card (Flash Card), etc. equipped on the controller. Furthermore, the readable storage medium can also include both the internal storage unit of the controller and an external storage device. The readable storage medium is used to store the computer program and other programs and data required by the controller. The readable storage medium can also be used to temporarily store data that has been output or is to be output.

[0123] Based on this understanding, the technical solution of the present invention, or the portion that contributes to the prior art, or all or part of the technical solution, can be embodied in the form of a software product. This computer software product is stored in a storage medium and includes instructions for causing a computer device (which may be a personal computer, server, or network device, etc.) to execute all or part of the steps of the method described in each embodiment of the present invention. The aforementioned readable storage medium includes various media capable of storing program code, such as a USB flash drive, a mobile hard drive, a read-only memory (ROM), a random access memory (RAM), a magnetic disk, or an optical disk.

[0124] It should be emphasized that the examples described in the present invention are illustrative rather than restrictive. Therefore, the present invention is not limited to the examples described in the specific embodiments. Any other embodiments derived by those skilled in the art based on the technical solution of the present invention, which do not depart from the purpose and scope of the present invention, whether modified or replaced, also fall within the scope of protection of the present invention.

[0125] It should be emphasized that the examples described in the present invention are illustrative rather than restrictive. Therefore, the present invention is not limited to the examples described in the specific embodiments. Any other embodiments derived by those skilled in the art based on the technical solution of the present invention that do not depart from the purpose and scope of the present invention, whether modified or replaced, also fall within the scope of protection of the present invention.

Claims

1. A permanent magnet motor model prediction current control method, characterized by: The following steps are involved: In each sampling period, the permanent magnet motor is sampled to obtain at least the stator voltage sampling value and stator current sampling value k corresponds to the sampling moment or sampling point; Using the stator voltage sampling value The stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment Get the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system Based on the stator current feedback values ​​of the d-axis and q-axis The stator current reference values ​​of the d-axis and q-axis Calculate the current error in the dq rotating coordinate system Based on minimizing the control cost function, the current error Get the initial switching signal duty cycle d of the three-phase two-level inverter A d B and d C , and sort them from largest to smallest to get d max d mid and d min ; Based on the sorted three-phase duty cycle d max d mid and d min The duty cycle is reconstructed to obtain the final switching signal duty cycle d of the three-phase two-level inverter. A '、d B ' and d C '; According to the final switch duty cycle d A '、d B ' and d C 'Generate driving signals for three-phase two-level inverter; The final switching signal duty cycle d of the three-phase two-level inverter A '、d B ' and d C 'Calculated by the following formula: Where, d0=1-(d max -d mid )-(d mid -d min )=1-(d max -d min ); The initial switching signal duty cycle is calculated by the following formula: Where, d xi ,i∈{1,2,3} is the voltage vector u selected in the horizontal direction x x The three-phase duty cycle; d yi , i∈{1,2,3} is the voltage vector u selected in the vertical direction y y The three-phase duty cycle, d α d β are the duty cycles in the horizontal direction x and the vertical direction y, respectively, which are obtained by taking the partial derivative of the control cost function.

2. The method according to claim 1, wherein: The voltage vector u selected in the horizontal direction x x And the voltage vector u selected in the vertical direction y y The selection process is as follows: first, the current error After coordinate transformation, we get and Then, based on and The positive and negative signs of are selected and determined from the voltage vector space according to the rules in the table below; The voltage vector space is composed of two effective voltage vectors (u a ,u c ), two virtual voltage vectors (u b ,u d ) and two zero vectors (u0, u7), the two effective voltage vectors (u a ,u c ) and the two virtual voltage vectors (u b ,u d The three-phase duty ratio of ) is shown in the following table: Among them, d i1 d i2 d i3 is the three-phase duty cycle, and The current error The corresponding d-axis and q-axis current errors are: and is the error in the α and β directions of the two-phase stationary coordinate system obtained through coordinate transformation; u dc is the voltage amplitude on the DC side of the inverter.

3. The method according to claim 1, wherein: The current error Corresponding d-axis and q-axis current errors and The formula is as follows: Among them, R s is the stator resistance; L d , L q are the direct-axis component and quadrature-axis component of the stator inductance respectively; ω e is the angular velocity; T s is the sampling time; ψ f is the permanent magnet flux.

4. The method according to claim 1, wherein: Obtained q-axis stator current reference value The speed reference value and speed sampling value The stator current reference value of the d-axis is obtained by inputting the difference into the PI controller. Set to 0; Among them, the speed sampling value It is obtained by sampling the motor.

5. The method according to claim 1, wherein: Execute the stator voltage sampling value The stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq coordinate system at the next moment , calculate according to the following formula: in, The stator current feedback value and voltage value before the next compensation in the dq rotating coordinate system; They are stator current feedback values The corresponding d-axis and q-axis components; The voltage values ​​are Corresponding d-axis and q-axis components; C 3s / 2r is the transformation matrix from the three-phase stationary coordinate system to the two-phase rotating coordinate system; R s is the stator resistance; L d , L q are the direct-axis component and quadrature-axis component of the stator inductance respectively; ω e is the electrical angular velocity; T s is the sampling time; ψ f is the permanent magnet flux; θ is the rotor position angle.

6. A system based on the method according to any one of claims 1 to 5, characterized in that: include: The sampling module is used to sample the permanent magnet motor in each sampling period to obtain at least the stator voltage sampling value. and stator current sampling value k corresponds to the sampling moment or sampling point; The stator current feedback value acquisition module is used to use the stator voltage sampling value Stator current sampling value Get the stator current feedback values ​​of the d-axis and q-axis in the dq rotating coordinate system at the next moment Stator current reference value acquisition module, used to obtain the stator current reference values ​​of the d-axis and q-axis in the dq rotating coordinate system Current error calculation module, used for stator current feedback value based on the d-axis and q-axis Stator current reference values ​​of d-axis and q-axis Calculate the current error in the dq rotating coordinate system The initial duty cycle calculation module is used to minimize the control cost function based on the current error Get the initial switching signal duty cycle d of the three-phase two-level inverter A d B and d C , and sort by size to get d max d mid and d min ; Reconstruction module for the three-phase duty cycle d max d mid and d min , the three-phase duty cycle is reconstructed to obtain the final switching signal duty cycle d of the three-phase two-level inverter A '、d B ' and d C '; A driving module for switching the output voltage according to the final switching duty cycle d A '、d B ' and d C 'Generate driving signals for three-phase two-level inverter.

7. A system based on the method according to any one of claims 1 to 5, characterized in that: The system comprises a permanent magnet motor and a control subsystem. The control subsystem adopts the method to generate a driving signal of a three-phase two-level inverter, thereby controlling the permanent magnet motor.

8. A computer-readable storage medium, characterized in that: A computer program is stored, which is called by a processor to implement: The steps of the method according to any one of claims 1 to 5.

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