A method for iq imbalance image rejection of a zero intermediate frequency receiving array

Abstract

The application discloses an IQ imbalance mirror suppression method of a zero intermediate frequency receiving array, and comprises the following steps: calculating a radio frequency signal received by each array element; obtaining M radio frequency signals after phase shifting; obtaining M intermediate frequency signals output by mixing and filtering; obtaining M baseband analog in-phase signals and M baseband analog quadrature signals; the M baseband analog in-phase signals and the M baseband analog quadrature signals are respectively subjected to A / D sampling to obtain M baseband digital complex signals; before beam synthesis of a digital array, amplitude and phase consistency of M receiving channels is calibrated to obtain M receiving channel amplitude and phase error compensation values; the M baseband digital complex signals of the digital array are subjected to channel error compensation and beam weighting compensation, and summation is performed to obtain a final beam forming output complex signal; the application has the advantages that the IQ imbalance mirror suppression capability of a multi-channel zero intermediate frequency receiving digital array is improved.

Description

Technical Field

[0001] This invention relates to the field of IQ unbalanced image suppression, and more specifically to a method for IQ unbalanced image suppression of a zero-IF receiver array. Background Technology

[0002] A zero-IF receiver is a receiver that directly converts radio frequency signals into the original transmitted signal without requiring an intermediate frequency (IF). Zero-IF receivers have gained widespread attention and application due to their simple circuit structure, low power consumption, ease of integration, small size, and low cost. However, in practical applications, due to limitations in current device manufacturing processes, the quadrature local oscillator frequency sources used in the in-phase (I-branch) and quadrature (Q-branch) branches of a zero-IF receiver chip cannot guarantee absolute quadrature. Furthermore, the amplitude-frequency responses of analog devices such as mixers and low-pass filters in each branch cannot be guaranteed to be completely consistent. This results in a certain amplitude-phase error between the in-phase and quadrature branches, manifesting as image components in the spectrum of the baseband signal synthesized from the in-phase and quadrature branches. When the energy of the image components is too high, it will cause severe distortion of the original signal, affecting the overall performance of the receiving channel. Therefore, research on IQ imbalance image suppression methods for zero-IF receivers has significant practical implications.

[0003] Currently, there is a wealth of research literature on IQ imbalance image suppression methods for single-channel zero-IF receivers, such as the image interference suppression system and method in zero-IF architecture software radio disclosed in Chinese Patent Publication No. CN115865115A. However, there is little discussion on IQ imbalance image suppression methods using multi-channel zero-IF receivers in digital arrays. If the single-channel IQ imbalance image suppression method is directly applied to digital arrays, estimating and compensating the IQ imbalance error for each channel separately, followed by beamforming and other processing, the system design complexity and computational load are extremely high, especially in large digital arrays. Furthermore, compared to single-channel zero-IF reception, the impact of IQ imbalance error on the beamforming output of a zero-IF digital receiver array is more complex. Its output not only exhibits image components in the frequency domain but also in the spatial domain, and the energy of the spatial and frequency domain image components varies with the beam pointing. Therefore, it is necessary to find an IQ imbalance image suppression method suitable for large digital arrays that can suppress both the frequency domain image component and the spatial domain image component caused by IQ imbalance error without increasing the complexity of digital array receiving channel design and channel correction, thus solving the IQ imbalance image suppression problem of zero-IF receiving digital arrays. Summary of the Invention

[0004] The technical problem to be solved by this invention is how to provide a multi-channel IQ imbalance image suppression method to solve the IQ imbalance image suppression problem of multi-channel zero-IF receiving digital arrays.

[0005] This invention solves the above-mentioned technical problems through the following technical means: a method for suppressing IQ imbalance image in a zero-IF receiver array, comprising the following steps:

[0006] Step 1: The digital array has a total of M array elements. Calculate the radio frequency signal received by each array element.

[0007] Step 2: After each RF signal is amplified by a low-noise amplifier, it is sent to a phase shifter to add a pre-configured random phase, resulting in M ​​phase-shifted RF signals.

[0008] Step 3: Mix the phase-shifted M RF signals with the M first-stage local oscillator signals and perform bandpass filtering to obtain M intermediate frequency analog signals from the mixing and filtering outputs;

[0009] Step 4: Mix and low-pass filter the M intermediate frequency analog signals with the M second-stage in-phase local oscillator signals and the M second-stage quadrature local oscillator signals respectively to obtain M baseband analog in-phase signals and M baseband analog quadrature signals;

[0010] Step 5: Perform A / D sampling on M baseband analog in-phase signals and M baseband analog quadrature signals respectively to obtain M baseband digital complex signals;

[0011] Step 6: Before beamforming, the amplitude and phase consistency of the M receiving channels are calibrated to obtain the amplitude and phase error compensation values ​​of the M receiving channels.

[0012] Step 7: Perform channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and sum them to obtain the final beamforming output complex signal.

[0013] Further, step 1 includes:

[0014] The radio frequency signal received by the m-th array element is

[0015] s m (t)=A(t)cos[Ω c (t-τ m )+φ(t)]m=1,…,M

[0016] Where A(t) represents amplitude modulation information, φ(t) represents phase modulation information, and Ω c τ is the analog angular frequency of the carrier wave for the received signal. m The time delay difference is determined by the coordinates of the array elements and the direction of signal incidence, where t is the current time and M is the total number of array elements.

[0017] Furthermore, step 2 includes:

[0018] Step 2-1: Generate M random phases that follow a uniform distribution in the range [0, π] beforehand. These M random phases are then assigned to the phase shifters of the M receiving channels, respectively.

[0019] Step 2-2, M radio frequency signals After being amplified by a low-noise amplifier, the signal is fed into a phase shifter for phase shifting, resulting in M ​​phase-shifted radio frequency signals. Right now

[0020]

[0021] in, and These are amplitude and phase errors caused by the low-noise amplifiers in each channel, respectively.

[0022] Furthermore, step 3 includes:

[0023] Step 3-1: The local oscillator frequency source of the digital array generates the first-stage local oscillator signal LoF1 = cos[Ω1t]. After passing through the power divider, the power is divided into M local oscillator signals. The first-stage local oscillator signal is updated to... Where Ω1 is the first-stage local oscillator angular frequency, and These are amplitude error and phase error, respectively, caused by the power divider;

[0024] Step 3-2: The center frequency of the M bandpass filters in the receiving channel is equal to Ω. IF =Ω c -Ω1, the amplitude and phase errors near the center frequency are respectively and m = 1, ..., M;

[0025] Step 3-3, the M phase-shifted radio frequency signals With M first-stage local oscillator signals Perform mixing and bandpass filtering separately to obtain M intermediate frequency analog signals from the mixing and filtering outputs.

[0026]

[0027] Among them, A cm For channel amplitude error and For channel phase error and

[0028] Furthermore, step 4 includes:

[0029] Step 4-1: The local oscillator frequency source of the digital array generates the second-stage in-phase local oscillator signal LoF2_I = cos[Ω]. IF t] and the second-order orthogonal local oscillator signal LoF2_Q=-sin[Ω IFThe second-stage in-phase local oscillator signal LoF2_I is divided into M in-phase local oscillator signals by a power divider. and in, and These are the amplitude and phase errors introduced by the power divider during the process. The second-stage quadrature local oscillator signal LoF2_Q is divided into M quadrature local oscillator signals by the power divider. and in, and These are the amplitude error and phase error introduced by the power divider during the process, respectively.

[0030] Step 4-2: Convert the M intermediate frequency analog signals With M second-stage in-phase local oscillator signals After mixing and low-pass filtering, M baseband analog in-phase signals are obtained. have

[0031]

[0032] in, Let be the real impulse response of the low-pass filter in the in-phase branch of the m-th channel. and These are the amplitude error and phase error caused by the low-pass filter of the in-phase branch, respectively.

[0033] Step 4-3: Convert the M intermediate frequency analog signals With M second-order orthogonal local oscillator signals After mixing and low-pass filtering, M baseband analog in-phase signals are obtained. have

[0034]

[0035] in, Let be the real impulse response of the low-pass filter in the orthogonal branch of the m-th channel. and These are the amplitude error and phase error caused by the low-pass filter of the orthogonal branch, respectively.

[0036] Furthermore, step 5 includes:

[0037] Step 5-1, A / D sampling period is T s The index of the sampled value is an integer n, and there are M baseband analog in-phase signals. and M baseband analog orthogonal signals A / D sampling is performed separately to obtain M baseband digital in-phase signals. and M baseband digital quadrature signals The formula is as follows

[0038]

[0039]

[0040] Step 5-2: Taking the in-phase branch as a reference, the amplitude error g of the quadrature branch relative to the in-phase branch. m and phase error θ m They are respectively

[0041]

[0042] Then M baseband digital complex signals The calculation formula is

[0043]

[0044] in,

[0045] Furthermore, step 6 includes:

[0046] Step 6-1: The single-frequency correction source is placed in the normal direction of the digital array, satisfying the array far-field condition. The single-frequency signal radiated by the correction source is s. c (t)=A0cos[(Ω c +Ω d )t+φ0], where φ0 is the initial phase, A0 is the amplitude of the single-frequency signal, and Ω d For the frequency offset of a single-frequency signal and Ω d >0;

[0047] Step 6-2: The digital array receives a single-frequency signal, and M receiving channels simultaneously acquire N samples of the baseband digital complex signal. Then there is

[0048]

[0049] Step 6-3: Process N samples from M channels Perform Fourier transforms on each channel and calculate the frequency Ω for each of the M channels. d Spectral value at have

[0050] Step 6-4: Using the spectrum value Sp1 of the first channel as a reference, calculate the amplitude and phase error compensation value of each receiving channel relative to the first channel. Right now

[0051] Furthermore, step 7 includes:

[0052] Step 7-1: Utilize the amplitude and phase error compensation values ​​of M channels For M baseband complex signals respectively Compensation will be provided.

[0053]

[0054] Step 7-2: Based on the delay difference τ generated when the signal is incident on the array surface m Constructing beam weighting coefficients Therefore, the final beamforming output complex signal z(n) is

[0055]

[0056] Where, α m This is the amplitude weighting value for the m-th element in the array.

[0057] Furthermore, when step 3 is omitted, the M phase-shifted radio frequency signals Replace the M intermediate frequency analog signals in step 4 Then proceed with steps 4 through 7.

[0058] Furthermore, when step 3 is omitted, the frequency Ω of the second-stage local oscillator signal generated in step 4... IF =Ω c .

[0059] The advantages of this invention are:

[0060] (1) Without increasing the complexity of array channel design and channel correction, this invention only adds channel phase shifters and pre-configured random phases, thereby realizing random phase configuration of the zero-IF receiver output baseband digital complex signal (including the target signal and its mirror interference signal). When estimating the channel compensation value, only the inter-channel amplitude and phase error compensation value corresponding to the target signal in the baseband digital complex signal is calculated, and this compensation value is used to compensate the M baseband digital complex signals of the digital array. During beamforming, the mirror interference signal is canceled out due to the doubling of random phase error, while the target signal in the baseband digital complex signal is retained after amplitude and phase error compensation, thereby solving the IQ unbalanced image suppression problem of multi-channel zero-IF receiver digital array and improving the IQ unbalanced image suppression capability.

[0061] (2) The channel phase shifters in the digital array of the present invention are flexibly configured and can be pre-set to a uniformly distributed random phase. No modification is required when the array is working, making it convenient to use.

[0062] (3) The principle of this invention is simple, the amount of computation is small, and it does not require complex IQ imbalance error estimation and compensation algorithms, which is easy to implement in engineering.

[0063] (4) The IQ imbalance image suppression capability of the digital array of the present invention is related to the number of array elements. As the array size increases, the image suppression capability increases. Therefore, the method provided by the present invention is very suitable for large-scale digital arrays.

[0064] (5) The method provided by the present invention can be used for both narrowband digital arrays and wideband digital arrays for IQ imbalance image suppression, and is not limited by the array structure. Attached Figure Description

[0065] Figure 1 This is a block diagram illustrating the principle of an IQ imbalance image suppression method for a zero-IF receiver array disclosed in an embodiment of the present invention.

[0066] Figure 2 The array beam diagram with 256 array elements is shown in the simulation experiment of the IQ unbalanced image suppression method of a zero intermediate frequency receiving array disclosed in the embodiment of the present invention, without the use of a phase shifter.

[0067] Figure 3 The array beam diagram with 1024 array elements is shown in the simulation experiment of the IQ unbalanced image suppression method of a zero intermediate frequency receiving array disclosed in the embodiment of the present invention without using a phase shifter.

[0068] Figure 4 The simulation experiment of the IQ imbalance image suppression method of the zero intermediate frequency receiving array disclosed in the embodiment of the present invention does not use a phase shifter, and the target and image beam diagrams are shown when the number of array elements is 256.

[0069] Figure 5 The target and image beam diagrams for a zero-IF receiver array IQ imbalance image suppression method disclosed in the embodiments of the present invention are shown without the use of a phase shifter and with 1024 array elements.

[0070] Figure 6 The simulation experiment of the IQ imbalance image suppression method of a zero intermediate frequency receiving array disclosed in the embodiment of the present invention uses a phase shifter, and the array beam diagram is shown when the number of array elements is 256.

[0071] Figure 7 The array beam diagram with 1024 array elements is shown in the simulation experiment of the IQ imbalance image suppression method of a zero intermediate frequency receiving array disclosed in the embodiment of the present invention, using a phase shifter.

[0072] Figure 8 The simulation experiment of the IQ imbalance image suppression method of a zero intermediate frequency receiving array disclosed in the embodiment of the present invention uses a phase shifter, and the target and image beam diagrams are shown when the number of array elements is 256.

[0073] Figure 9The simulation experiment of the IQ unbalanced image suppression method of a zero intermediate frequency receiving array disclosed in the embodiment of the present invention uses a phase shifter, and the target and image beam diagrams are shown when the number of array elements is 1024. Detailed Implementation

[0074] To make the objectives, technical solutions, and advantages of the embodiments of the present invention clearer, the technical solutions of the embodiments of the present invention will be clearly and completely described below in conjunction with the embodiments of the present invention. Obviously, the described embodiments are only some embodiments of the present invention, not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0075] like Figure 1 As shown, this invention provides a method for suppressing IQ imbalance image in a zero-IF receiver array, comprising the following steps:

[0076] Step 1: The digital array has M array elements. Calculate the radio frequency signal received by each array element. The specific process is as follows:

[0077] The radio frequency signal received by the m-th array element is

[0078] s m (t)=A(t)cos[Ω c (t-τ m )+φ(t)]m=1,…,M

[0079] Where A(t) represents amplitude modulation information, φ(t) represents phase modulation information, and Ω c τ is the analog angular frequency of the carrier wave for the received signal. m The time delay difference is determined by the coordinates of the array elements and the direction of signal incidence, where t is the current time and M is the total number of array elements.

[0080] Step 2: After each RF signal is amplified by a low-noise amplifier, it is sent to a phase shifter to add a pre-configured random phase, resulting in M ​​phase-shifted RF signals. The specific process is as follows:

[0081] Step 2-1: Generate M random phases that follow a uniform distribution in the range [0, π] beforehand. These M random phases are then assigned to the phase shifters of the M receiving channels, respectively.

[0082] Step 2-2, M radio frequency signals After being amplified by a low-noise amplifier, the signal is fed into a phase shifter for phase shifting, resulting in M ​​phase-shifted radio frequency signals. Right now

[0083]

[0084] in, and These are amplitude and phase errors caused by the low-noise amplifiers in each channel, respectively.

[0085] Step 3: Mix and bandpass filter the phase-shifted M RF signals with the M first-stage local oscillator signals to obtain M intermediate frequency analog signals from the mixing and filtering outputs. The specific process is as follows:

[0086] Step 3-1: The local oscillator frequency source of the digital array generates the first-stage local oscillator signal LoF1 = cos[Ω1t]. After passing through the power divider, the power is divided into M local oscillator signals. The first-stage local oscillator signal is updated to... Where Ω1 is the first local oscillating frequency. and These are amplitude error and phase error, respectively, caused by the power divider;

[0087] Step 3-2: The center frequency of the M bandpass filters in the receiving channel is equal to Ω. IF =Ω c -Ω1, the amplitude and phase errors near the center frequency are respectively and m = 1, ..., M;

[0088] Step 3-3, the M phase-shifted radio frequency signals With M first-stage local oscillator signals Perform mixing and bandpass filtering separately to obtain M intermediate frequency analog signals from the mixing and filtering outputs.

[0089]

[0090] Among them, A cm For channel amplitude error and For channel phase error and

[0091] Step 4: Mix and low-pass filter the M intermediate frequency analog signals with the M second-stage in-phase local oscillator signals and the M second-stage quadrature local oscillator signals respectively to obtain M baseband analog in-phase signals and M baseband analog quadrature signals. The specific process is as follows:

[0092] Step 4-1: The local oscillator frequency source of the digital array generates the second-stage in-phase local oscillator signal LoF2_I = cos[Ω]. IF t] and the second-order orthogonal local oscillator signal LoF2_Q=-sin[Ω IF The second-stage in-phase local oscillator signal LoF2_I is divided into M in-phase local oscillator signals by a power divider. and in, and These are the amplitude and phase errors introduced by the power divider during the process. The second-stage quadrature local oscillator signal LoF2_Q is divided into M quadrature local oscillator signals by the power divider. and in, and These are the amplitude error and phase error introduced by the power divider during the process, respectively.

[0093] Step 4-2: Convert the M intermediate frequency analog signals With M second-stage in-phase local oscillator signals After mixing and low-pass filtering, M baseband analog in-phase signals are obtained. have

[0094]

[0095] in, Let be the real impulse response of the low-pass filter in the in-phase branch of the m-th channel. and These are the amplitude error and phase error caused by the low-pass filter of the in-phase branch, respectively.

[0096] Step 4-3: Convert the M intermediate frequency analog signals With M second-order orthogonal local oscillator signals After mixing and low-pass filtering, M baseband analog in-phase signals are obtained. have

[0097]

[0098] in, Let be the real impulse response of the low-pass filter in the orthogonal branch of the m-th channel. and These are the amplitude error and phase error caused by the low-pass filter of the orthogonal branch, respectively.

[0099] Step 5: Perform A / D sampling on M baseband analog in-phase signals and M baseband analog quadrature signals respectively to obtain M baseband digital complex signals. The specific process is as follows:

[0100] Step 5-1, A / D sampling period is T s The index of the sampled value is an integer n, and there are M baseband analog in-phase signals. and M baseband analog orthogonal signals A / D sampling is performed separately to obtain M baseband digital in-phase signals. and M baseband digital quadrature signals The formula is as follows

[0101]

[0102]

[0103] Step 5-2: Taking the in-phase branch as a reference, the amplitude error g of the quadrature branch relative to the in-phase branch. m and phase error θ m They are respectively

[0104]

[0105]

[0106] Then M baseband digital complex signals The calculation formula is

[0107]

[0108] in,

[0109] Step 6: Before beamforming, the amplitude and phase consistency of the M receiving channels are calibrated to obtain the amplitude and phase error compensation values ​​for the M receiving channels. The specific process is as follows:

[0110] Step 6-1: The single-frequency correction source is placed in the normal direction of the digital array, satisfying the array far-field condition. The single-frequency signal radiated by the correction source is s. c (t)=A0cos[(Ω c +Ω d )t+φ0], where φ0 is the initial phase, A0 is the amplitude of the single-frequency signal, and Ω d For the frequency offset of a single-frequency signal and Ω d >0;

[0111] Step 6-2: The digital array receives a single-frequency signal, and M receiving channels simultaneously acquire N samples of the baseband digital complex signal. Then there is

[0112]

[0113] Step 6-3: Process N samples from M channels Perform Fourier transforms on each channel and calculate the frequency Ω for each of the M channels. d Spectral value at have

[0114] Step 6-4: Using the spectrum value Sp1 of the first channel as a reference, calculate the amplitude and phase error compensation value of each receiving channel relative to the first channel. Right now

[0115] Step 7: Perform channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and sum them to obtain the final beamforming output complex signal. The specific process is as follows:

[0116] Step 7-1: Utilize the amplitude and phase error compensation values ​​of M channels For M baseband complex signals respectively Compensation will be provided.

[0117]

[0118] Step 7-2: Based on the delay difference τ generated when the signal is incident on the array surface m Constructing beam weighting coefficients Therefore, the final beamforming output complex signal z(n) is

[0119]

[0120] Where, α m This is the amplitude weighting value for the m-th element in the array.

[0121] As a further improvement of the present invention, the first-stage mixing and bandpass filtering process described in step 3 can be omitted. Even without omitting step 3, the entire scheme can still achieve image suppression. Without omitting step 3, bandpass filtering can be performed, thereby improving the image suppression effect. When step 3 is omitted, the phase-shifted M RF signals... Replace the M intermediate frequency analog signals in step 4 Then, steps 4 through 7 are executed. If step 3 is omitted, the frequency Ω of the second-stage local oscillator signal generated in step 4... IF =Ω c .

[0122] The performance of the IQ imbalance image suppression method of the zero-IF receiving digital array of the present invention is verified through two simulation scenarios. In the simulation experiments, the system operates at a radio frequency of 8 GHz, the system bandwidth is 30 MHz, the digital array is a uniform linear array with an element spacing of 11 mm, and the number of elements is 256 and 1024 respectively. The uniform linear array uses a Taylor weight of -30 dB. The beamforming pointing angle is 60 degrees.

[0123] Considering that zero-IF receiver array channels (from the first-stage mixer and bandpass filter to the A / D output) are usually implemented using integrated chips in practical engineering, the amplitude A between array channels... m and phase The error is relatively small, so we take the channel amplitude error A here. m The channel phase error follows a uniform distribution within the range of 0dB to 0.5dB. It follows a uniform distribution within ±10 degrees. In integrated chip implementation, the IQ imbalance image suppression between the in-phase and quadrature branches is typically between 20dB and 40dB. Here, we take the amplitude error g of the IQ imbalance. m It follows a uniform distribution between 0.5dB and 1.5dB, with a phase error θ. m The IQ imbalance amplitude and phase error are uniformly distributed between 0 and 10 degrees, so the average image rejection ratio is 22.8 dB.

[0124] Scenario 1: Phase shifter without using low-noise amplifier

[0125] The phase shifter values ​​after the low-noise amplifier in each digital array channel are set to 0. First, the digital array is calibrated using a single-frequency signal (RF frequency of 8.001 GHz) to obtain the inter-channel amplitude and phase error compensation values. After error compensation, the array beam pattern is calculated. The array beam patterns for 256 and 1024 elements are shown below. Figure 2 and Figure 3 As shown.

[0126] The ideal beam pattern in the figure represents the beam pattern without any errors. The error pattern is the beam pattern after correction and compensation when channel and IQ imbalance amplitude and phase errors exist. Compared with the ideal pattern, the residual error after correction and compensation affects the sidelobes and has a peak in the image direction pointed to by the target beam. The suppression ratio between the target and the image is approximately 23.2 dB. This is basically consistent with the image suppression ratio of 22.8 dB caused by the average amplitude and phase error of IQ imbalance. Sidelobe weighting cannot suppress the peak in the image direction pointed to by the target.

[0127] After correction and compensation, if the target beam pattern and the mirror beam pattern are calculated separately, the beam patterns corresponding to 256 and 1024 array elements are as follows: Figure 4 and Figure 5 As shown in the figure, due to channel errors and IQ imbalance errors, the mirror beam is not completely synthesized. When the number of array elements is 256, the frequency domain image suppression reaches -51dB at the target beam pointing direction, but the spatial domain image suppression is only -23.3dB at the beam pointing mirror direction. When the number of array elements is 1024, the frequency domain image suppression reaches -57dB at the target beam pointing direction, but the spatial domain image suppression is only -23dB at the beam pointing mirror direction. Therefore, simply using sidelobe weighting cannot suppress the spatial domain image component. The system cannot simultaneously suppress both frequency and spatial domain image components.

[0128] Scenario 2: Phase shifter after using low-noise amplifier

[0129] The phase shifter values ​​after low-noise amplification in the digital array channels follow a uniform distribution within [0, π]. First, a single-frequency signal (RF frequency of 8.001 GHz) is used to correct the digital array, obtaining the inter-channel amplitude and phase error compensation values. After error compensation, the array beam pattern is calculated. The array beam patterns for 256 and 1024 array elements are shown below. Figure 6 and Figure 7 As shown.

[0130] The ideal beam pattern in the figure represents the beam pattern without any errors. The error pattern is the beam pattern after correction and compensation when channel and IQ unbalanced amplitude and phase errors exist. Compared with the ideal pattern, the residual error after correction and compensation affects the sidelobes. Due to the use of phase shifters, spatial image components are suppressed, and the sidelobe level of the error pattern is better than -30dB.

[0131] After correction and compensation, if the target beam pattern and the mirror beam pattern are calculated separately, the beam patterns corresponding to 256 and 1024 array elements are as follows: Figure 8 and Figure 9 As shown in the figure, when the number of array elements is 256, the frequency domain image suppression reaches -48.5dB at the target beam direction, and the spatial domain image suppression is better than -48dB. When the number of array elements is 1024, the frequency domain image suppression reaches -62dB at the target beam direction, and the spatial domain image suppression is better than -49.7dB. Because the array channels use phase shifters, both spatial and frequency domain image components are suppressed, and the increased array size further improves the spatial and frequency domain image suppression capabilities.

[0132] In summary, the IQ imbalance image suppression method for zero-IF receiver arrays is simple in principle, requires little computation, and is easy to implement in engineering. Its performance has been verified through simulation experiments. The scheme proposed in this invention does not increase the complexity of array channel design and correction, and can suppress not only frequency domain image components caused by IQ imbalance errors but also spatial domain image components, making it particularly suitable for large digital arrays.

[0133] The above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A method for suppressing IQ imbalance image in a zero-IF receiver array, characterized in that, Includes the following steps: Step 1: The digital array has M array elements. Calculate the radio frequency signal received by each array element; Step 1 includes: The radio frequency signal received by the m-th array element is s m (t)=A(t)cos[Ω c (t-τ m )+φ(t)]m=1,…,M Where A(t) represents amplitude modulation information, φ(t) represents phase modulation information, and Ω c τ is the analog angular frequency of the carrier wave for the received signal. m The time delay difference is determined by the coordinates of the array elements and the direction of signal incidence, where t is the current time and M is the total number of array elements; Step 2: Each RF signal, after being amplified by a low-noise amplifier, is fed into a phase shifter to add a pre-configured random phase, resulting in M ​​phase-shifted RF signals; Step 2 includes: Step 2-1: Generate M random phases that follow a uniform distribution in the range [0, π] beforehand. These M random phases are then assigned to the phase shifters of the M receiving channels, respectively. Step 2-2, M radio frequency signals After being amplified by a low-noise amplifier, the signal is fed into a phase shifter for phase shifting, resulting in M ​​phase-shifted radio frequency signals. Right now in, and These are respectively the amplitude error and phase error caused by the low-noise amplifiers in each channel; Step 3: Mix and bandpass filter the phase-shifted M RF signals with the M first-stage local oscillator signals to obtain M intermediate frequency analog signals output by mixing and filtering; Step 3 includes: Step 3-1: The local oscillator frequency source of the digital array generates the first-stage local oscillator signal LoF1 = cos[Ω1t]. After passing through the power divider, the power is divided into M local oscillator signals. The first-stage local oscillator signal is updated to... Where Ω1 is the angular frequency of the first-stage local oscillator. and These are amplitude error and phase error, respectively, caused by the power divider; Step 3-2: The center frequency of the M bandpass filters in the receiving channel is equal to Ω. IF =Ω c -Ω1, the amplitude and phase errors near the center frequency are respectively and m = 1, ..., M; Step 3-3, the M RF signals after phase shifting With M first-stage local oscillator signals Perform mixing and bandpass filtering separately to obtain M intermediate frequency analog signals from the mixing and filtering outputs. Among them, A cm For channel amplitude error and For channel phase error and Step 4: Mix and low-pass filter the M intermediate frequency analog signals with the M second-stage in-phase local oscillator signals and the M second-stage quadrature local oscillator signals respectively to obtain M baseband analog in-phase signals and M baseband analog quadrature signals; Step 5: Perform A / D sampling on M baseband analog in-phase signals and M baseband analog quadrature signals respectively to obtain M baseband digital complex signals; Step 6: Before beamforming, the amplitude and phase consistency of the M receiving channels are calibrated to obtain the amplitude and phase error compensation values ​​of the M receiving channels. Step 7: Perform channel amplitude and phase error compensation and beam weighting compensation on the M baseband digital complex signals of the digital array, and sum them to obtain the final beamforming output complex signal.

2. The method for suppressing IQ imbalance image in a zero-IF receiver array according to claim 1, characterized in that, Step 4 includes: Step 4-1: The local oscillator frequency source of the digital array generates the second-stage in-phase local oscillator signal LoF2_I = cos[Ω]. IF t] and the second-order orthogonal local oscillator signal LoF2_Q=-sin[Ω IF The second-stage in-phase local oscillator signal LoF2_I is divided into M in-phase local oscillator signals by a power divider. and in, and The first stage involves amplitude and phase errors introduced by the power divider during the power division of M in-phase local oscillator signals. The second-stage quadrature local oscillator signal LoF2_Q is then divided into M quadrature local oscillator signals by the power divider. and in, and These are the amplitude and phase errors introduced by the power divider during the process of dividing the power into M quadrature local oscillator signals; Step 4-2: Convert the M intermediate frequency analog signals With M second-stage in-phase local oscillator signals After mixing and low-pass filtering, M baseband analog in-phase signals are obtained. have in, Let be the real impulse response of the low-pass filter in the in-phase branch of the m-th channel. and These are the amplitude error and phase error caused by the low-pass filter of the in-phase branch, respectively. Step 4-3: Convert the M intermediate frequency analog signals With M second-order orthogonal local oscillator signals After mixing and low-pass filtering, M baseband analog in-phase signals are obtained. have in, Let be the real impulse response of the low-pass filter in the orthogonal branch of the m-th channel. and These are the amplitude error and phase error caused by the low-pass filter of the orthogonal branch, respectively.

3. The method for suppressing IQ imbalance image in a zero-IF receiver array according to claim 2, characterized in that, Step 5 includes: Step 5-1, A / D sampling period is T s The index of the sampled value is an integer n, and there are M baseband analog in-phase signals. and M baseband analog orthogonal signals A / D sampling is performed separately to obtain M baseband digital in-phase signals. and M baseband digital quadrature signals The formula is as follows Step 5-2: Taking the in-phase branch as a reference, the amplitude error g of the quadrature branch relative to the in-phase branch. m and phase error θ m They are respectively Then M baseband digital complex signals The calculation formula is in, 4. The method for suppressing IQ imbalance image in a zero-IF receiver array according to claim 3, characterized in that, Step 6 includes: Step 6-1: Place the single-frequency correction source in the normal direction of the digital array to obtain the single-frequency signal radiated by the correction source; Step 6-2: The digital array receives a single-frequency signal, and M receiving channels simultaneously acquire N samples of the baseband digital complex signal. Step 6-3: Process N samples from M channels Perform Fourier transforms on each channel and calculate the frequency Ω for each of the M channels. d Spectral value at Step 6-4: Using the spectrum value Sp1 of the first channel as a reference, calculate the amplitude and phase error compensation value of each receiving channel relative to the first channel. Right now 5. The method for suppressing IQ imbalance image in a zero-IF receiver array according to claim 4, characterized in that, Step 7 includes: Step 7-1: Utilize the amplitude and phase error compensation values ​​of M channels For M baseband complex signals respectively To provide compensation, there is z m (n)·ε m ; Step 7-2: Based on the delay difference τ generated when the signal is incident on the array surface m Constructing beam weighting coefficients Therefore, the final beamforming output complex signal is Where, α m This is the amplitude weighting value for the m-th element in the array.

6. The method for suppressing IQ imbalance image in a zero-IF receiver array according to claim 5, characterized in that, When step 3 is omitted, the M phase-shifted radio frequency signals Replace the M intermediate frequency analog signals in step 4 Then proceed with steps 4 through 7.

7. The method for suppressing IQ imbalance image in a zero-IF receiver array according to claim 5, characterized in that, When step 3 is omitted, the frequency Ω of the second-stage local oscillator signal generated in step 4 is... IF =Ω c .

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