Adaptive bandwidth virtual impedance LLC current sharing control method

Abstract

The application provides an adaptive-bandwidth virtual impedance LLC current-sharing control method, and a system composed of an input parallel output parallel system of N-phase LLC resonant converters, and a capacitor C connected in parallel at the output end o The control method feeds back the output voltage and the output current of each phase, introduces adaptive-bandwidth virtual impedance in the circulating current feedback loop, adjusts the LLC output closed-loop impedance of each phase to realize the current-sharing control of the overall system, and the bandwidth gain of the current-sharing current loop where the adaptive-bandwidth virtual impedance is located is controlled by the system working condition: when the system is current-sharing, the bandwidth gain of the current-sharing current loop is infinitesimal attenuation; when the system is not current-sharing, the gain of the current-sharing current loop is increased to inhibit the circulating current of the system. Therefore, the method has the advantages of simple structure, easy system module integration design, good current-sharing effect, excellent transient response, and solves the voltage static error problem existing in the traditional virtual impedance current-sharing method.

Description

Technical Field

[0001] This invention relates to the field of multi-module converter control, and more specifically to a current sharing control method based on adaptive bandwidth virtual impedance LLC. Background Technology

[0002] In many high-power power systems and industrial applications, efficient and reliable power solutions are needed to meet ever-increasing power demands. To address these challenges, LLC input-parallel output-parallel (IPOP) systems have become an important technological solution. For example, data centers require substantial power to support servers, storage devices, and network equipment. These devices require efficient power supplies and redundancy to ensure system availability. Multiple LLCs connected in parallel can provide high efficiency, scalability, and redundancy to meet the growing power demands of data centers. Simultaneously, with the proliferation of electric vehicles, charging infrastructure needs to provide high-power charging capabilities. LLC IPOP systems can help achieve fast charging while distributing current loads, reducing heat generation in charging equipment, and improving charging efficiency. Furthermore, communication base stations require stable power supplies to ensure the reliability of communication networks. LLC IPOP systems can provide efficient power solutions while distributing loads and reducing the risk of equipment failure.

[0003] However, when multi-phase LLCs are connected via IPOP, the offset of resonant component parameters between different modules leads to different gain characteristics. Unlike traditional PWM converters, LLC resonant topologies are highly sensitive to parameters. Even if multiple LLCs within the system operate at the same frequency, small parameter offsets can cause severe current imbalance, significantly impacting the overall system efficiency and stability. Therefore, implementing multi-phase current sharing in LLC IPOP systems is essential. Summary of the Invention

[0004] The purpose of this invention is to achieve current sharing in a multiphase LLC IPOP system, and compared with traditional current sharing methods, it greatly reduces the transient response time of the system and reduces the steady-state voltage error of the system.

[0005] To address the aforementioned technical problems, this invention provides an adaptive bandwidth virtual impedance LLC current sharing control method. The system consists of an N-phase LLC resonant converter with parallel input and parallel output, and a capacitor C is connected in parallel at the output. o ;

[0006] The control method achieves overall system current sharing control by feeding back the output voltage and the output current of each phase, and introducing an adaptive bandwidth virtual impedance in the circulating current feedback loop to adjust the closed-loop impedance of the LLC output of each phase. Furthermore, the bandwidth gain of the current sharing current loop containing the adaptive bandwidth virtual impedance is controlled by the system operating conditions.

[0007] When the system is sharing current, the bandwidth gain of the current sharing current loop is attenuated by an infinitesimal amount.

[0008] When the system experiences uneven current distribution, the gain of the current sharing loop increases, suppressing the circulating current in the system.

[0009] In a preferred embodiment: the control method includes the following steps:

[0010] 1) In each sampling period, the output voltage ν0 of the input-output parallel system and the output current i of each phase LLC resonant converter are measured. oj Sampling is performed; where j takes values ​​from 1 to N, and N is the number of LLC modules in the parallel input-parallel output system;

[0011] 2) Through i oj Calculate the ideal average value of the system output current.

[0012]

[0013] 3) Calculate the circulating current Δi of the LLC output for each phase sequentially. oj :

[0014]

[0015] 4) Set △i oj Feed into the flow equalization ring compensator G io Obtain the current sharing loop output, and then connect this output to the adaptive bandwidth.

[0016] The output control voltage V of the LLC current sharing loop in this phase is obtained by multiplying the virtual impedances. ic The adaptive bandwidth virtual impedance is the gain provided by the current sharing loop, as shown in the following equation:

[0017]

[0018] Where Zcs is the added virtual impedance, F1 is the output voltage, and v ref Reference voltage;

[0019] 5) Compare the output voltage ν0 with the reference voltage v ref Subtraction yields the system output voltage error signal Δv o :

[0020] Δv o =v o -vref

[0021] The obtained Δν0 is fed into the voltage loop compensator G. vo The system voltage loop output control voltage V of the parallel system with LLC input and parallel output is obtained. vc ;

[0022] 6) Finally, the output control voltage V of the current sharing loop obtained in steps 4 and 5 is... ic and voltage loop output control voltage V vc Subtraction is fed into the frequency programming unit V co Frequency programming unit V co The control voltage is converted into a frequency switching control signal and sent to the corresponding LLC module to control the switching of the switching transistors of each phase LLC half-bridge inverter in the system.

[0023] In a preferred embodiment: the flow equalization ring compensator G io It is a type I or type II compensator.

[0024] In a preferred embodiment, the expression for the Type II compensator is as follows:

[0025]

[0026] in, boost is the phase that needs to be boosted, ω c The system cutoff frequency ω c =2πf c , G c Let be the gain at the cutoff frequency of the open-loop transfer function.

[0027] Compared with the prior art, the technical solution of the present invention has the following beneficial effects:

[0028] 1) No additional hardware circuitry is required, reducing the hardware cost of the current sharing system.

[0029] 2) Easy to modularize and integrate. There are no power level connections between multi-phase LLC modules except for sampling and control signal lines, which greatly reduces the coupling and complexity of the multi-phase LLC IPOP system.

[0030] 3) It solves the voltage steady-state error problem of the traditional virtual impedance current sharing method, making it better suited for low-voltage, high-current applications, and the output voltage will not be significantly smaller due to the increase in output current.

[0031] 4) Excellent current sharing effect. When the system is operating at half load or above, its current imbalance can be reduced to below 5%, and the transient response time is also greatly shortened, with the voltage change rate increasing from 14.63V / s to 200V / s. Attached Figure Description

[0032] Figure 1 The topology diagram of the two-phase half-bridge LLC resonant converter IPOP is shown.

[0033] Figure 2 This is a diagram of the overall system control.

[0034] Figure 3 Bode plot of voltage loop;

[0035] Figure 4 This is a Bode plot of the equal flow loop;

[0036] Figure 5 The graph shows the relationship between the current sharing loop bandwidth, virtual impedance, and current error.

[0037] Figure 6 The waveform diagram for the output of a 200W single-voltage loop is shown below.

[0038] Figure 7 The output waveform diagram for 960W virtual impedance control;

[0039] Figure 8 This is a waveform diagram of the voltage jump output of the traditional virtual impedance current sharing method.

[0040] Figure 9 The output waveform diagram for voltage jump control using virtual impedance;

[0041] Figure 10 A graph showing the relationship between current imbalance and output power;

[0042] Figure 11 This is a schematic diagram of the control method. Detailed Implementation

[0043] The technical solutions of the present invention will be clearly and completely described below with reference to the accompanying drawings of the embodiments of the present invention. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. All other embodiments obtained by those skilled in the art based on the embodiments of the present invention without creative effort are within the scope of protection of the present invention.

[0044] In the description of this invention, it should be noted that the terms "upper," "lower," "inner," "outer," "top / bottom," etc., indicate the orientation or positional relationship based on the orientation or positional relationship shown in the accompanying drawings. They are used only for the convenience of describing the invention and for simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation. Therefore, they should not be construed as limitations on the invention. Furthermore, the terms "first" and "second" are used for descriptive purposes only and should not be construed as indicating or implying relative importance.

[0045] In the description of this invention, it should be noted that, unless otherwise explicitly specified and limited, the terms "installed", "equipped", "sleeved / connected", "connected", etc., should be interpreted broadly. For example, "connection" can be a wall-mounted connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be a direct connection or an indirect connection through an intermediate medium; it can be a connection within two components. For those skilled in the art, the specific meaning of the above terms in this invention can be understood according to the specific circumstances.

[0046] This embodiment provides a current sharing control method based on adaptive bandwidth virtual impedance LLC. The system consists of an N-phase LLC resonant converter with parallel input and parallel output, and a capacitor C is connected in parallel at the output. o ;

[0047] The control method achieves overall system current sharing control by feeding back the output voltage and the output current of each phase, and introducing an adaptive bandwidth virtual impedance in the circulating current feedback loop to adjust the closed-loop impedance of the LLC output of each phase. Furthermore, the bandwidth gain of the current sharing current loop containing the adaptive bandwidth virtual impedance is controlled by the system operating conditions.

[0048] When the system is sharing current, the bandwidth gain of the current sharing current loop is attenuated by an infinitesimal amount.

[0049] When the system experiences uneven current distribution, the gain of the current sharing loop increases, suppressing the circulating current in the system.

[0050] The control method includes the following steps:

[0051] 1) In each sampling period, the output voltage ν0 of the input-output parallel system and the output current i of each phase LLC resonant converter are measured. oj Sampling is performed; where j takes values ​​from 1 to N, and N is the number of LLC modules in the parallel input-parallel output system;

[0052] 2) Through i oj Calculate the ideal average value of the system output current.

[0053]

[0054] 3) Calculate the circulating current Δi of the LLC output for each phase sequentially. oj :

[0055]

[0056] 4) Set △i oj Feed into the flow equalization ring compensator G io The current sharing loop output is obtained, and this output is multiplied by the adaptive bandwidth virtual impedance to obtain the control voltage V of the LLC current sharing loop output for that phase.ic The adaptive bandwidth virtual impedance is the gain provided by the current sharing loop, as shown in the following equation:

[0057]

[0058] Where Zcs is the added virtual impedance, F1 is the output voltage, and v ref Reference voltage;

[0059] 5) Compare the output voltage ν0 with the reference voltage v ref Subtraction yields the system output voltage error signal Δv o :

[0060] Δv o =v o -v ref

[0061] The obtained Δν0 is fed into the voltage loop compensator G. vo The system voltage loop output control voltage V of the parallel system with LLC input and parallel output is obtained. vc ;

[0062] 6) Finally, the output control voltage V of the current sharing loop obtained in steps 4 and 5 is... ic and voltage loop output control voltage V vc Subtraction is fed into the frequency programming unit V co Frequency programming unit V co The control voltage is converted into a frequency switching control signal and sent to the corresponding LLC module to control the switching of the switching transistors of each phase LLC half-bridge inverter in the system.

[0063] The design of the voltage loop compensator is a crucial component of the voltage loop and will be discussed in detail here. Due to the bandwidth limitations of the voltage sampling feedback loop, the cutoff frequency of the voltage loop compensator should not exceed the bandwidth of the system's voltage sampling feedback loop. Otherwise, the voltage sampling feedback loop will introduce a significant phase delay, reducing the system's phase margin and potentially causing system instability. The voltage loop compensator can be either Type I or Type II. Type I compensators are characterized by their ease of design and simple structure, and can significantly reduce the computational burden on the digital controller if digital control is used. Type II compensators, compared to Type I, can provide up to 90° of phase compensation at the cutoff frequency, thereby increasing the system's cutoff frequency and resulting in a faster transient response. Therefore, if high system response time is required, a Type II compensator can be selected; conversely, if system stability is a greater concern, a Type I compensator can be used.

[0064] Here, we take the Type II compensator as an example. The expression for the Type II compensator is as follows:

[0065]

[0066] in, boost is the phase that needs to be boosted, ω c The system cutoff frequency ω c =2πf c , G c Let be the gain at the cutoff frequency of the open-loop transfer function.

[0067] Meanwhile, the closed-loop transfer function T of the system voltage loop vo-cl as follows:

[0068]

[0069] in H fvj Let G be the power stage transfer function of the j-th phase LLC. vo For voltage loop compensators, F1 is the voltage acquisition and feedback loop, V co For frequency programming units, Z olj Let C be the open-loop output impedance of the j-th phase LLC. o Z is the output capacitor of the IPOP system. olj It can be obtained from the LLC topology using the fundamental wave analysis method (FHA), and its expression is as follows:

[0070]

[0071] Where n is the transformer turns ratio T, C r L r These are the LLC resonant capacitor and resonant inductor, respectively. m This is the magnetizing inductance of the transformer.

[0072] This can be derived from the voltage loop closed-loop transfer function. The voltage loop closed-loop transfer function, besides being related to the designed voltage loop compensator G... vo In addition to the flow equalization loop G, io Therefore, the quality of the current sharing loop design will affect the voltage control performance of the system. Current sharing loop H ij Theoretically, the smaller the effect of H in the voltage loop, the better, as this would lead to more precise voltage loop control. However, H ij The opposite is true in the flow equalization ring, H ij A larger value results in better current sharing for the system. Here, the closed-loop transfer function T of the current sharing loop is given. io-cl :

[0073]

[0074] The F2 current sampling feedback stage and the other stages have already been described above and will not be repeated here. Here, the current sharing ring compensator G... io Design similar to G vo With the same design, either Type I or Type II compensators can be used to balance stability and transient response. The cutoff frequency of the compensator can be selected based on the frequency of the corresponding current sampling feedback loop to ensure system stability. Clearly, there is a certain trade-off between the design of the current sharing loop and the voltage loop, requiring a compromise between current sharing and constant voltage performance. When a better constant voltage effect is required, the current sharing loop G... io The cutoff frequency is relatively low; conversely, when the system requires higher current sharing performance, G can be used. vo The cutoff frequency is designed to be lower.

[0075] Example

[0076] See Figure 1 As can be seen, the two-phase half-bridge LLC resonant converter IPOP topology is used, where the single-phase LLC converter consists of L r C r Forming a resonant cavity, L m The magnetizing inductance of transformer T is used because the application scenario is low voltage and high current. Therefore, a full-wave rectification structure is adopted at the back end of the transformer, and the transformer is selected with a center tap structure and a turns ratio of n:1:1. S1 and S2 are the main switching transistors of the LLC front-end inverter. The control signals mentioned above are the control signals of these two switching transistors, which are composed of complementary switching signals with a duty cycle of 50%. The control signal frequencies of the LLC switching transistors of phases A and B are obtained by the control strategy mentioned above. D1 and D2 are full-wave rectifier diodes, and their selection requirements are high current stress and low voltage stress. C... o1 C o2 For phases A and B of the LLC, the output capacitors are C. o This is the output capacitor of the overall LLC IPOP system.

[0077] From the above topology, the DC gain of the LLC resonant converter can be derived as follows:

[0078]

[0079] in

[0080] To analyze the equivalent load change of a single-phase LLC under frequency modulation control, and based on this, to analyze the current sharing characteristics of the LLC IPOP system, the equivalent load distribution characteristics can be analyzed. Therefore, R can be obtained from the DC gain formula. L Simplifying, we get the following formula:

[0081]

[0082] make The following equation can be solved:

[0083]

[0084] In the above formula, f1 is the operating frequency achievable with the minimum equivalent load resistance at the system resonance operating point under a fixed DC gain, while f2 is the maximum equivalent load resistance at that fixed DC gain operating point. A negative single-phase equivalent load resistance indicates that at that operating frequency, it cannot provide the voltage gain required by the reference phase LLC; therefore, R... L-equ >0 can be used as a boundary condition for a single-phase system.

[0085] If we assume:

[0086] 1. All LLC modules in the system operate within the reference phase ZVS range;

[0087] 2. The system operates under a certain load;

[0088] Then we can obtain the following formula, where L r * C r * This represents the parameter offset of a certain phase LLC module relative to the reference phase LLC module in the system. This is the parameter boundary condition that enables the LLC IPOP system to operate in a current-sharing manner.

[0089]

[0090] Based on the above topology analysis, combined with the traditional virtual impedance current sharing method, the overall system control block diagram can be obtained. (See [link]). Figure 2 .

[0091] In the diagram, F1 and F2 are the output voltage and output current sampling feedback loops, respectively, and H... fv This is the small-signal model of the power stage of the LLC resonant converter derived by EDF. Since the LLC small-signal model is related to actual operating conditions, a large margin is left for the design of the compensator. Therefore, H is used here. fv Calculated under full system load conditions, where N is the number of parallel phases of the LLC IPOP, and Z... ol R is the open-loop output impedance. L For the load resistance, C o For the system output capacitor, G vo As a voltage loop compensator, its bandwidth determines the system voltage tracking speed. For easier understanding, please refer to further information. Figure 11 The green box represents the current sharing loop, used to achieve current sharing in the system and reduce the circulating current. G... ioIt is a current sharing ring compensator, and its bandwidth is determined by Figure 11 The steps in the blue box determine the process, where Z... cs This is the virtual impedance introduced in series; the larger its value, the larger the bandwidth of the current-sharing loop. When the system reaches current sharing, this bandwidth provides an infinitesimal attenuation to the current-sharing loop. At this point, the system approximates single-voltage loop control, that is... Figure 11 The red portion in the diagram represents increased system stability; however, when the system current imbalance is significant, this bandwidth provides a large gain to the current sharing loop, allowing the circulating current to decrease rapidly under the action of the current sharing loop. The output control voltage is obtained after compensation by the current sharing loop and voltage loop, and this control voltage is then passed through V... co The frequency programming stage obtains the control frequency of each phase LLC.

[0092] The transfer function of the overall system can be obtained from the above analysis:

[0093]

[0094] in

[0095] From the above formula, i can be... o Simplifying and making all its modules equal, we get the following formula:

[0096]

[0097] Because G vo G io With a first-order integrator, the system's H value can be approximately derived in the low-frequency range. i1 =H i2 =……=H iN H v1 =H v2 =……=H vN Substituting it into the above formula, we can obtain i o1 =i o2 =……=i oN Therefore, this control strategy can achieve current sharing in N-phase LLC.

[0098] The above has already been deduced If we take this as a prerequisite and substitute it into the system transfer function, we can obtain the following equation:

[0099]

[0100] Combining these equations, the output voltage error can be derived as follows:

[0101]

[0102] From the above equation, it can be deduced that the voltage error is approximately zero in the low-frequency range. Therefore, this method can also solve the problem of steady-state voltage error caused by traditional virtual impedance. Eliminating irrelevant input variables from the system transfer function yields the following closed-loop voltage transfer function:

[0103]

[0104] Similarly, the closed-loop transfer function of the flow-equalizing loop can be obtained as follows:

[0105]

[0106] The hardware circuit design has cutoff frequencies of 200Hz and 500Hz for F1 and F2, respectively. Therefore, the cutoff frequencies of the voltage loop and current sharing loop are both 100Hz. A type II compensator is used for compensation, with a boost phase design of 40°. The target compensator can be calculated according to the following formula, and the Bode plot can be drawn.

[0107] The expression for the Type II compensator is as follows:

[0108]

[0109] in, boost is the phase that needs to be boosted, ω c The system cutoff frequency ω c =2πf c , G c Let be the gain at the cutoff frequency of the open-loop transfer function.

[0110] From the above equation, we can deduce that G... io The following formula

[0111]

[0112] Since digital control is used, the above formula needs to be discretized. The Tustin method is used to discretize the voltage loop compensator obtained above. The Tustin method formula is as follows:

[0113]

[0114] Where T s Let z be the discrete period, i.e., the digital control period, and z be the discrete operator. Substituting the above equation into the current loop compensator calculated earlier, we can obtain the discrete form of the current loop compensator, which can be used for digital control implementation.

[0115] Here T s If we take 100 μs, then G io (s) The discrete equations can be calculated to obtain the following formula:

[0116]

[0117] Output voltage v o With reference voltage v ref Subtraction yields the system output voltage error signal Δv o V ref The output voltage is designed to meet the target, while v o To obtain the actual output voltage, the obtained Δv o Input voltage loop compensator G vo The LLCIPOP system voltage loop output control voltage V is obtained. vc The G here vo The design and G mentioned above io Similarly, based on the derivation of the power stage transfer function of an actual single-phase LLC module and the design of the sampling circuit, a Type II compensator is still selected here. The relevant parameters of this Type II compensator are designed with a boost of 40° and G. c It is 26.2dB, and the cutoff frequency is f. c If the frequency is 100Hz, then G can be derived from the formula above. vo The following formula

[0118]

[0119] Similarly, the discrete form of this voltage loop can be obtained using the bilinear sampling method, with the discrete period T. s Similarly, taking 100 μs as the value, the discrete formula for this voltage loop can be obtained as follows:

[0120]

[0121] The voltage loop Bode plot is as follows: Figure 3 As shown, the Bode plot of the equalization loop is as follows: Figure 4 As shown in the figure, it is clear that after compensation, the phase margin of the voltage loop and current sharing loop is greatly improved, from 19.4° to 123°.

[0122] The loop bandwidth of the current sharing loop is in G io Once the design is complete, it is solely determined by K, which in turn depends on the operating conditions. When the system load remains constant, increasing K increases the bandwidth, and vice versa. The degree of influence is similar to that of the serially inserted Z. cs It depends on the size. Figure 5The corresponding relationship diagram is provided. Under severe current imbalance, the current sharing loop can provide a gain of up to 40dB, while under complete current sharing, it provides a -40dB attenuation. The larger the virtual impedance introduced into the system, the larger the loop bandwidth of the current sharing loop; therefore, the more severe the current imbalance, the larger the bandwidth of the current sharing loop. Conversely, when the system achieves current sharing, the bandwidth of the current sharing loop is 0, meaning the current sharing loop no longer functions. At this point, the system operates approximately under a single voltage loop, which not only improves system reliability but also significantly improves the system's response speed under current imbalance.

[0123] Figure 6 The waveform diagram shows the system using single voltage loop control with an output power of 200W. It can be clearly seen from the diagram that phase B bears almost all the load, while phase A is approximately operating under no-load conditions. The current imbalance of the system is close to 100%.

[0124] Figure 7 The waveform diagram of the system under full load when using the adaptive bandwidth virtual impedance control strategy proposed in this paper is shown. It can be clearly seen that the current imbalance has been reduced to 2.85%. Compared with the traditional single voltage loop control above, the current sharing effect is very obvious, which verifies the effectiveness of the current sharing method proposed in this paper. Figure 8 , Figure 9 The figures show the voltage jump waveforms under light load (20%) for the system using adaptive bandwidth virtual impedance control and traditional virtual impedance current sharing control, respectively, with the jump voltage changing from 48V to 54V. The comparison clearly shows that the traditional virtual impedance current sharing method is slow and has a large voltage steady-state error, while the adaptive bandwidth virtual impedance method proposed in this paper not only significantly reduces the voltage steady-state error but also greatly improves the transient response speed.

[0125] Figure 10 The graph shows the relationship between current imbalance and output power. As the system output power gradually increases, the current imbalance also gradually decreases. When the system is approximately operating under no-load conditions, the current imbalance is 0.15, while when operating under full load, the current imbalance is only 0.02. This is consistent with the application scenario of low voltage and high current for IPOP systems.

[0126] In summary, this current sharing control strategy has several advantages: First, it eliminates the need for additional hardware circuitry, effectively reducing hardware costs; second, its modular and integrated implementation is simple, requiring only control signal lines to connect multi-phase LLC modules, significantly reducing system coupling and complexity; third, it solves the voltage steady-state error problem inherent in traditional virtual impedance current sharing methods, making it particularly suitable for low-voltage, high-current scenarios, where the output voltage does not decrease significantly with increasing output current; finally, it exhibits excellent current sharing performance, reducing current imbalance to below 5% under half-load and above load conditions, while significantly shortening transient response time and increasing voltage change rate from 14.63V / s to 200V / s.

[0127] The above description is merely a preferred embodiment of the present invention, but the design concept of the present invention is not limited thereto. Any non-substantial modifications made to the present invention by those skilled in the art within the scope of the technology disclosed in the present invention using this concept shall be deemed as an infringement of the protection scope of the present invention.

Claims

1. A method for adaptive bandwidth virtual impedance LLC current sharing control, characterized in that: The system is composed of N-phase LLC resonant converter input parallel output parallel system, and a capacitor C is connected in parallel at the output end o ; The control method introduces adaptive bandwidth virtual impedance in a circulating current feedback loop to regulate the LLC output closed-loop impedance of each phase by feeding back the output voltage and the output current of each phase to realize the current sharing control of the overall system, and the bandwidth gain of the current sharing current loop where the adaptive bandwidth virtual impedance is located is controlled by the system working condition: When the system is current sharing, the bandwidth gain of the current sharing current loop is infinitesimal attenuation; When the system is not current sharing, the gain of the current sharing current loop is increased to suppress the system circulating current; The control method comprises the following steps: 1) In each sampling period, the output voltage ν0 of the input-output parallel system and the output current i of each phase LLC resonant converter are measured. oj Sampling is performed; where j takes values ​​from 1 to N, and N is the number of LLC modules in the parallel input-parallel output system; 2) by i oj Computing the ideal mean value of the output current : ； 3) Calculate each phase LLC output loop current Δi in turn oj : ； 4) send in Δi oj into the current sharing loop compensator G io to get the current sharing loop output, multiply the output with the adaptive bandwidth virtual impedance to get the phase LLC current sharing loop output control voltage V ic ; wherein the adaptive bandwidth virtual impedance provides a gain to the current sharing loop as follows: ； where Zcs is the added virtual impedance, F1 is the output voltage, v ref is the reference voltage; 5) Compare the output voltage ν0 with the reference voltage v ref Subtraction yields the system output voltage error signal Δv o : ； The resulting AV0 is fed into a voltage loop compensator G vo The resulting LLC input parallel output parallel system voltage loop output control voltage V vc ; 6) Finally, the output control voltage V of the current sharing loop obtained in steps 4 and 5 is... ic and voltage loop output control voltage V vc Subtraction is fed into the frequency programming unit V co Frequency programming unit V co The control voltage is converted into a frequency switching control signal and sent to the corresponding LLC module to control the switching of the switching transistors of each phase LLC half-bridge inverter in the system.

2. The adaptive-bandwidth virtual impedance LLC current-sharing control method of claim 1, wherein: The current-sharing ring compensator G io is a type I or type II compensator.

3. The adaptive-bandwidth virtual impedance-based LLC current-sharing control method of claim 2, wherein: The expression of the type II compensator is as follows: ； wherein, , is the phase to be boosted, is the system cutoff frequency , , , is the gain at the cutoff frequency of the open loop transfer function.

Images