PWM modulation method and device for double three-phase permanent magnet synchronous motor

Abstract

The application discloses a PWM modulation method and device for a double three-phase permanent magnet synchronous motor, and solves the problem of how to suppress equivalent odd-order switching harmonics of a double three-phase permanent magnet synchronous motor driving system, and belongs to the field of motor driving noise suppression. The application adopts a six-phase CHB driving circuit to drive one-phase winding of the double three-phase permanent magnet synchronous motor; each phase CHB driving circuit is formed by cascading N single bodies, and each single body is a full-bridge circuit powered by an independent bus power supply; the six-phase CHB driving circuit is used for unipolar carrier phase-shifting modulation of the double three-phase permanent magnet synchronous motor, the unipolar carrier phase-shifting modulation decomposes the full-bridge circuit in each single body into two half-bridges, the modulation waves of the two half-bridges are staggered by 180 degrees, each carrier single body is unfolded in turn according to pi / N, and the phase of the carrier group of the second group of three-phase windings is delayed by theta c2 ; and zeta is a phase belt angle of the double three-phase permanent magnet synchronous motor. The application can realize 4N times equivalent switching frequency frequency multiplication in the sense of electromagnetic torque.

Description

TECHNICAL FIELD

[0001] The application relates to a dual three-phase permanent magnet synchronous motor PWM modulation method and device, and belongs to the field of motor drive noise suppression. BACKGROUND

[0002] For electric vehicles, all-electric aircraft, all-electric ships and other electrical systems that use energy storage as the main energy source, there is a demand for high-efficiency operation in a wide power output range. To ensure full utilization of the energy storage system, such driving systems are generally used in low-voltage high-power output situations. Compared with traditional three-phase motors, dual three-phase motors have the advantages of low back electromotive force, small single-phase current, large output power, and high operating efficiency, and have good application value in the field of low-voltage high-power efficient operation.

[0003] The characteristics of the electromagnetic torque noise spectrum are directly related to factors such as system switching frequency, modulation method, topology, etc. How to suppress noise so that the motor drive system meets the low noise requirement in a wide frequency band is also a major problem faced by the electric drive system. The core of noise suppression is to improve the switching frequency, but a higher switching frequency will increase the switching loss of the switching tube, and the system efficiency will decrease, so the upper limit of the switching frequency of the driver will be limited by the system efficiency requirement. Compared with traditional two-level circuits, multi-level circuits have the characteristics of low voltage stress, low switching loss, low harmonic content, small electromagnetic interference, and increased control dimension of devices, and are suitable for applications that are sensitive to torque ripple and noise suppression. With the same equivalent switching frequency, the more the number of levels, the lower the actual switching frequency, and the efficiency is generally higher, so multi-level circuits are an effective solution for high-efficiency, low-vibration motor drives. Cascaded H-bridge (CHB) converters are generally used as high-voltage and high-power power electronic devices, and are widely used in static var generators, active power filters and other power applications that require harmonic suppression and reactive power compensation in power systems. At the same time, CHB has high-voltage and high-power output characteristics and is also used in high-voltage and high-power transmission systems. CHB is rarely used in low-voltage motor drives, especially in low-voltage dual three-phase permanent magnet synchronous motor drives.

[0004] Currently, the increase in the equivalent switching frequency of CHB drivers is mainly achieved by increasing the number of cascades or increasing the switching frequency of the switching tube. The former requires an increase in the circuit, which is not conducive to high power density integration of the system, and the latter will bring additional device losses, which cannot guarantee the system efficiency. SUMMARY

[0005] In view of how to suppress the equivalent odd-order switching harmonics of a dual three-phase permanent magnet synchronous motor drive system, the application provides a dual three-phase permanent magnet synchronous motor PWM modulation method and device.

[0006] The application discloses a PWM modulation method of a double three-phase permanent magnet synchronous motor.

[0007] S1, six-phase voltage modulation waves of the CHB-driven double three-phase permanent magnet synchronous motor are generated, modulation waves of two half-bridges in each single body of the CHB driving circuit are staggered by 180 degrees, and given modulation wave voltages of each group of half-bridges of six phases are obtained.

[0008] S2, a phase shift angle is obtained according to the number of cascades of the CHB and a phase belt angle of the double three-phase permanent magnet synchronous motor.

[0009]

[0010] In the formula, θ is the phase shift angle; ζ is the phase belt angle of the double three-phase permanent magnet synchronous motor. c2

[0011] S3, each carrier single body of the two groups of three-phase windings is unfolded in turn according to π / N, and the phase of the carrier group of the second group of three-phase windings is delayed by θ c2 , and then the carrier groups of the two groups of three-phase windings are generated.

[0012] S4, three-phase decoupling modulation is performed on the given modulation wave voltages of each group of half-bridges of six phases and the generated carrier groups of the two groups of three-phase windings.

[0013] Preferably, S1 comprises:

[0014] The control loop of the double three-phase permanent magnet synchronous motor is controlled according to double dq vector control, the inverse transformation of the coordinate transformation matrix is used on the basis that the command voltages generated by the control loop are independent in pairs in the three-phase neutral point and the zero sequence component is 0, and six-phase voltage modulation waves are generated.

[0015] According to the principle of single-polarity modulation, the modulation waves of two half-bridges in each single body of the CHB driving circuit are staggered by 180 degrees, and the given modulation wave voltages of each group of half-bridges of six phases are generated.

[0016] The application further provides a PWM modulation device of a double three-phase permanent magnet synchronous motor, which comprises a six-phase CHB driving circuit and a modulation module, the six-phase CHB driving circuit drives one phase winding of the double three-phase permanent magnet synchronous motor respectively, each CHB driving circuit is cascaded by N single bodies, and each single body is a full-bridge circuit powered by an independent bus power supply.

[0017] ​The modulation module is used for single-polarity carrier phase modulation of the dual three-phase permanent magnet synchronous motor by using a six-phase CHB drive circuit, the single-polarity carrier phase modulation decomposes the full-bridge circuit in each single body into two half-bridges, the two half-bridge modulation waves are staggered by 180 degrees, and each carrier single body is modulated in turn according to π / N.

[0018] As preferred, the modulation module is further used for, on the basis of the carrier group phase of the second set of three-phase windings being staggered by θ c2 ;

[0019]

[0020] In the formula, θ c2 is a phase shift angle; ζ is a phase belt angle of the dual three-phase permanent magnet synchronous motor.

[0021] The present application has the advantages that the present application does not need to make hardware changes to the system, and retains the advantages of simple structure and reliable operation of the driver; the present application uses CHB for driving the dual three-phase permanent magnet synchronous motor, and realizes 2N times of equivalent switching frequency multiplication at the actual switching frequency; the present application combines the CHB output voltage and the radial electromagnetic force suppression characteristic of the dual three-phase permanent magnet synchronous motor to give a carrier harmonic suppression modulation method specific to the CHB driven dual three-phase permanent magnet synchronous motor, which suppresses the odd switching harmonic of the electromagnetic torque on the basis of the original 2N times of equivalent switching frequency, and further multiplies the equivalent switching frequency related to the system vibration. BRIEF DESCRIPTION OF DRAWINGS

[0022] Figure 1 It is a topology structure diagram of the CHB driven dual three-phase permanent magnet synchronous motor system, in which the neutral points of the two sets of three-phase windings are independent, the full-bridge single body is composed of two half-bridge single bodies, and the per-phase drive topology is composed of the full-bridge single body in cascade, the cascade label i is from 1 to N, ABC is the first set of three-phase windings, DEF is the second set of three-phase windings, and the voltage of each full-bridge bus is V dc ;

[0023] Figure 2 It is a full-bridge single body modulation waveform;

[0024] Figure 3 It is a multi-single body cascade modulation waveform;

[0025] Figure 4 In (a), it is an equivalent circuit of the motor synchronous rotating coordinate system, Figure 4 In (b), it is a high-order harmonic equivalent circuit of the synchronous rotating coordinate system, Figure 4 In (c), it is a high-order harmonic equivalent circuit of the natural coordinate system;

[0026] Figure 5 It is a control block diagram of the modulation method of the present application, in which Ud1 , U q1 , U d2 , U q2 modulation wave command voltage generated for the control loop of the dual three-phase permanent magnet synchronous motor, U a1 , U b1 , U c1 , U a2 , U b2 , U c2 is each phase winding phase voltage;

[0027] Figure 6 is the phase combination relationship of carrier group 1 and carrier group 2 when the cascade number of CHB is 3, wherein carrier 1, 2 and 3 belong to carrier group 1, carrier 4, 5 and 6 belong to carrier group 2, and the overall phase difference value of carrier group 1 and carrier group 2 is θ c2 ;

[0028] Figure 7 is a current waveform diagram and current Fourier decomposition diagram of the dual three-phase permanent magnet synchronous motor driven by the CHB drive circuit with a phase belt angle of 60°, wherein the horizontal coordinate is frequency and the vertical coordinate is current size, wherein Figure 7(a) is the current waveform of the first set of three-phase windings and its Fourier decomposition when the six-phase CHB drive circuit works independently without mutual cooperation, i.e., the two carrier groups coincide, and Figure 7(b) is the current waveform of the first set of three-phase windings and its Fourier decomposition after the carrier group 2 is phase-shifted by θ c2 according to the method of the present application;

[0029] Figure 8 is a Fourier decomposition diagram of electromagnetic torque of the dual three-phase permanent magnet synchronous motor driven by the CHB drive circuit, wherein Figure 8(a) is the Fourier decomposition of electromagnetic torque of the dual three-phase permanent magnet synchronous motor when the six-phase CHB drive circuit works independently without mutual cooperation, i.e., the two carrier groups coincide, and Figure 8(b) is the Fourier decomposition of electromagnetic torque of the dual three-phase permanent magnet synchronous motor after the carrier group 2 is phase-shifted by θ c2 according to the method of the present application;

[0030] Figure 9 is a current waveform diagram and current Fourier decomposition diagram of the dual three-phase permanent magnet synchronous motor driven by the CHB drive circuit with a phase belt angle of 30°, wherein Figure 9(a) is the current waveform of the first set of three-phase windings and its Fourier decomposition when the six-phase CHB drive circuit works independently without mutual cooperation, i.e., the two carrier groups coincide, and Figure 9(b) is the current waveform of the first set of three-phase windings and its Fourier decomposition after the carrier group 2 is phase-shifted by θ c2 according to the method of the present application;

[0031] Figure 10 shows the Fourier decomposition of the electromagnetic torque of the dual three-phase permanent magnet synchronous motor driven by the CHB drive circuit with a phase band angle of 30°. Figure 10(a) shows the Fourier decomposition of the electromagnetic torque of the dual three-phase permanent magnet synchronous motor when the six-phase CHB drive circuit operates independently without mutual coordination, i.e., when the two sets of carrier groups overlap. Figure 10(b) shows the phase shift θ of the carrier group 2 when the six-phase CHB drive circuit is mutually coordinated according to the method of this invention. c2 Subsequently, the electromagnetic torque of the dual three-phase permanent magnet synchronous motor was decomposed using Fourier transform. Detailed Implementation

[0032] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0033] It should be noted that, unless otherwise specified, the embodiments and features described in the present invention can be combined with each other.

[0034] The present invention will be further described below with reference to the accompanying drawings and specific embodiments, but this is not intended to limit the scope of the invention.

[0035] The PWM modulation method for the dual three-phase permanent magnet synchronous motor in this embodiment uses a six-phase CHB drive circuit. Each of the six-phase CHB drive circuits drives one phase winding of the dual three-phase permanent magnet synchronous motor. Each phase CHB drive circuit is composed of N cascaded units, and each unit is a full-bridge circuit powered by an independent bus power supply.

[0036] The topology of the CHB drive circuit driving a dual three-phase permanent magnet synchronous motor is as follows: Figure 1 As shown, each unit is composed of a full-bridge circuit powered by an independent bus power supply. Each phase consists of N units cascaded together. Taking phase A as an example, its output phase voltage is the sum of the cascaded full-bridge circuits:

[0037]

[0038] The expression for the output voltage of a single full-bridge unit depends on the modulation algorithm used in the CHB driver circuit. In this embodiment, the most common unipolar carrier phase-shift modulation for CHB driver circuits is selected. This modulation algorithm decomposes a single full-bridge unit into two half-bridges, with the modulation waves of the two half-bridges offset by 180 degrees. Then, each full-bridge carrier unit is modulated sequentially according to π / n.

[0039] The switching sub-harmonic band of the inverter output voltage needs to be determined for vibration analysis, and therefore the double Fourier decomposition expression of the CHB drive circuit output voltage needs to be written based on the determined modulation method. As can be seen from equation (1), the cascaded H-bridge phase voltage is the sum of the output voltages of each full-bridge unit, and therefore the double Fourier decomposition expression of the output voltage of each full-bridge unit needs to be solved. The output voltage of each full-bridge unit is composed of two half-bridges with a phase difference of 180°. The double Fourier decomposition expression of the output voltage of each half-bridge is as follows:

[0040]

[0041] wherein V dc is the bus voltage; M is the modulation ratio, i.e. the ratio of the peak value of the modulation wave to the peak value of the carrier wave; ω0is the angular frequency of the modulation wave; ω c is the angular frequency of the carrier wave; θ0is the initial phase angle of the modulation wave; θ c is the initial phase angle of the carrier wave; n is the harmonic number of the modulation wave; m is the harmonic number of the carrier wave; J n (x) is the n-th order Bessel function of the variable x.

[0042] Equation (2) is the double Fourier decomposition of the voltages at points a and z in the half-bridge unit as shown in the figure, wherein the first term represents the fundamental wave component generated by the modulation, and the second term represents the sideband harmonic (corresponding to n≠0) formed by the interaction of the carrier harmonic (corresponding to n=0) and the modulation harmonic. Figure 1 As can be seen from equations (3) and (4), the carrier phase-shift modulation CHB drive circuit full-bridge unit can be equivalent to the difference between the half-bridge units with a phase difference of 180°, and therefore the double Fourier decomposition expression of the output voltage of the full-bridge unit can be obtained by bringing θ0in equation (2) into the difference between 0 and π and then simplifying a series of equations:

[0043] Figure 2 Figure 3

[0044]

[0045] As can be seen from equation (3), this modulation method only retains the odd-numbered (2n-1) modulation wave sideband terms of the even-numbered (2m) carrier harmonic, and the rest are eliminated. The double Fourier decomposition expression of the output voltage of the CHB drive circuit can be obtained by bringing equation (3) into equation (1) and then simplifying:

[0046]

[0047] In equation (4), the harmonics of the output voltage of the CHB drive circuit are the 2n-1 sidebands around the 2m carrier harmonic of the full-bridge, and become the 2n-1 sidebands around the 2Nm after cascading, which is equivalent to achieving an equivalent switching frequency of 2N times at the actual switching frequency, which meets the equivalent switching frequency formula of the CHB drive circuit.

[0048] ​​​The motor vibration is mainly caused by the radial electromagnetic force from the carrier sideband harmonics. According to the Maxwell tensor method, the radial electromagnetic force can be expressed as:

[0049]

[0050] where B air-gap (θ, t) is the air-gap magnetic induction, and μ0 is the vacuum permeability. The air-gap magnetic field is generated by the interaction of the stator magnetic field and the rotor magnetic field. Here, only the vibration caused by the time harmonic is concerned, so the electrical angle is regarded as a constant value, and thus formula (9) can be expressed as:

[0051]

[0052] where f p (t) is the spatial magnetic motive force generated by the rotor permanent magnet, f s (t) is the stator armature reaction fundamental magnetic motive force, and f h (t) is the stator armature reaction harmonic magnetic motive force.

[0053] The vibration mainly concerns the harmonic magnetic motive force. The rotor magnetic motive force and the stator armature reaction fundamental magnetic motive force are combined as the main magnetic motive force f M (t) = f p (t) + f s (t), and formula (6) can be expressed as:

[0054]

[0055] Meanwhile, the harmonic magnetic motive force is decomposed by combining the Hopkinson law. The magnetic motive force can be expressed as the product of the turn function and the current, and thus the harmonic magnetic motive force can be expressed as:

[0056] f h (θ, t) = ΣN h (θ)(i 1h (t) + i 2h (t)) (8)

[0057] where N h (θ) is the equivalent turn function with the spatial electrical angle as the independent variable, i 1h (t), and i 2h (t) represent the sideband harmonics near the carrier of the first and second sets of three-phase windings, respectively.

[0058] Formula (8) is substituted into formula (7), and the radial electromagnetic force can be expressed as the following three parts:

[0059]

[0060] where i h1 , ih2 These correspond to the common carrier sideband harmonic currents in the two sets of three-phase windings. From the above equation, it can be seen that when the carrier sideband harmonic currents of the first and second sets of three-phase windings are out of phase, p in equation (9) can be... 2r p 3r Radial electromagnetic force is eliminated, thereby reducing motor noise. Therefore, the core of motor noise reduction is to construct an anti-phase carrier sideband harmonic current.

[0061] Based on the synchronous rotating coordinate coefficient mathematical model of the motor, and assuming the internal coupling of the windings is eliminated, the equivalent circuit can be simplified to... Figure 4 In (a), where Ls is the synchronous inductor corresponding to L d With L q U s1 U s2 These correspond to the direct and quadrature axis voltages of the two sets of three-phase windings, respectively, i s1 i s2 These correspond to the AC and DC axis currents of the two sets of three-phase windings, respectively, and e1 and e2 correspond to the AC and DC axis back EMFs of the two sets of three-phase windings, respectively.

[0062] Will Figure 4 (a) High-frequency processing can be obtained Figure 4 (b) Equivalent circuit of higher harmonics in synchronous rotating coordinate system, where U sh1 U sh2 These correspond to the direct and quadrature axis high-frequency harmonic voltages of the two sets of three-phase windings, i sh1 i sh2 These correspond to the AC and DC axis harmonic currents of two sets of three-phase windings, respectively.

[0063] By restoring the synchronously rotating coordinate system to the natural coordinate system using the inverse transformation matrix, we can obtain... Figure 4 In (c), where L1 is the stator self-inductance, M is the stator mutual inductance, and U... h1 U h2 These correspond to the common phase carrier sideband harmonic voltages in the two sets of three-phase windings.

[0064] right Figure 4 By writing Kirchhoff's voltage equations for loop (c), we can obtain:

[0065]

[0066] From equation (10), we can obtain that when u h1 =-u h2 At that time, i h1 +i h2 =0, so when the inverter outputs the sideband harmonic voltage of the two sets of three-phase motor stator windings in opposite phase, the radial electromagnetic force can be suppressed, thereby reducing motor noise.

[0067] The fundamental phase angle of the different three-phase windings of the dual three-phase permanent magnet synchronous motor with a phase belt angle of 60° is different, and is related to the spatial distribution of the phase belt angle of the motor. Assuming that the phase angle of the first set of three-phase windings is the fundamental phase angle of 0°, the fundamental phase angle of the second set of three-phase windings is the phase shift angle ζ. By default, when the carrier phase angle of the first set of three-phase windings is 0°, the output voltage expression of the first set of three-phase windings can be written as formula (11):

[0068]

[0069] The voltage expression of the second set of three-phase windings can be expressed as the following formula:

[0070]

[0071] The harmonic voltage of the second set of three-phase windings has an additional 2mNθ c2 +(2n-1)ζ in the harmonic phase angle compared with the harmonic voltage of the first set of three-phase windings. In order to make the harmonic voltage of the first set of three-phase windings and the harmonic voltage of the second set of three-phase windings opposite, it is necessary to ensure that:

[0072] 2mNθ c2 +(2n-1)ζ=kπ k=1,3,5,7··· (13)

[0073] It can be further known that the carrier phase angle of the second set of three-phase windings needs to satisfy:

[0074]

[0075] Because k and 2n-1 are both odd numbers, the carrier phase angle offset of the second set of three-phase windings can be further expressed as:

[0076]

[0077] According to formula (15), when the carrier phase angle of the second set of three-phase windings satisfies , the equivalent switching frequency related to vibration of the system can be changed to a high-frequency noise reduction of 4Nm.

[0078] The PWM modulation method of the dual three-phase permanent magnet synchronous motor of the embodiment comprises:

[0079] Step one, as shown in Figure 5 , the modulation voltage command generated by the control loop is combined with the electrical angle of the dual three-phase permanent magnet synchronous motor to perform coordinate transformation, and a six-phase voltage modulation wave is generated.

[0080] The control loop of the dual three-phase permanent magnet synchronous motor performs double-dq vector control, and the command voltage U d1 , U q1 , U d2 , U q2On the basis of three-phase neutral points being independent of each other, and zero sequence component being 0, the given voltage value of double-dq plane is converted into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 phase winding phase voltage by using inverse transformation of coordinate transformation matrix of double three-phase permanent magnet synchronous motor.

[0081] The modulation wave of two half-bridges in each single body of the CHB drive circuit is staggered by 180° to obtain the given modulation wave voltage of six-phase half-bridge in each group;

[0082] Step two, the phase shift angle is obtained according to the cascade number of the CHB drive circuit and the phase belt angle of the double three-phase permanent magnet synchronous motor:

[0083]

[0084] In the formula, θ c2 is the phase shift angle; ζ is the phase belt angle of the double three-phase permanent magnet synchronous motor;

[0085] Step three, each carrier wave group of the two groups of three-phase windings is unfolded in turn according to π / N, and the phase of the carrier wave group of the second group of three-phase windings is delayed by θ c2 , as shown in the formula, and then the carrier wave groups of the two groups of three-phase windings are generated. Figure 6

[0086] Step four, the given modulation wave voltage of six-phase half-bridge in each group is modulated by three-phase decoupling with the generated carrier wave groups of the two groups of three-phase windings.

[0087] The phase belt angle of the double three-phase permanent magnet synchronous motor is generally 30° or 60°, and the embodiment will be specifically described in the form of examples for the two cases.

[0088] Example 1: In this example, a double three-phase permanent magnet synchronous motor with 16 poles, a speed of 120 rpm and a phase belt angle of 60° is used for simulation experiment, the carrier wave center frequency is selected as 4 kHz, and the cascade number of the CHB drive circuit is 3, so the actual switching frequency satisfies the equivalent 1 / 2N times relationship, which is 667 Hz. In order to enhance the persuasiveness, the two sets of three-phase windings are compared by using independent single-polarity carrier wave phase shift modulation and the method proposed in this embodiment.

[0089] In order to suppress the electromagnetic torque odd carrier wave frequency harmonic component, the following technical scheme is adopted in this embodiment:

[0090] First step: the double-dq vector control is used in the control loop of the double three-phase permanent magnet synchronous motor, and the instruction voltage U d1 , U q1 , U d2 ​, U q2 On the basis of three-phase neutral points being independent of each other and zero sequence components being 0, the given voltage value in the double-dq plane is converted into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 Phase voltage of each winding.

[0091]

[0092] Second step: the number 3 of CHB drive circuit cascades and the phase belt angle 60° of the double three-phase permanent magnet synchronous motor (equivalent to two sets of three-phase windings in the same phase) are brought into the phase shift angle operation equation:

[0093]

[0094] Third step: according to the second step, the phase difference value of the two groups of three-phase winding carrier groups is 30°, each single CHB drive circuit is expanded according to the principle of unipolar carrier modulation, and then the three carriers of the second set of three-phase winding carrier group are lagged by 30°, and then the carrier groups of the two groups of three-phase windings are generated.

[0095] Fourth step, according to U a1 , U b1 , U c1 , U a2 , U b2 , U c2 Phase voltage of each winding, in order to further improve the equivalent switching frequency, the modulation waves of the two half-bridges in each single CHB drive circuit are staggered by 180° according to the principle of unipolar modulation, and the given modulation wave voltage of six-phase each half-bridge is generated. Then, the two groups of carrier groups generated in the third step are subjected to three-phase decoupling sinusoidal pulse width modulation, and finally the corresponding switching drive signals are generated to drive the switching tubes.

[0096] As shown in Figs. 8(a) and 8(b). It can be seen that the 60° phase belt angle double three-phase permanent magnet synchronous motor and the CHB drive circuit work independently and do not cooperate with each other, that is, the two groups of carrier groups coincide and operate, after the modulation scheme of the two sets of three-phase windings of the application is used, the reverse harmonic voltage can completely meet the electromagnetic torque odd switching sub-harmonic and its sideband harmonic elimination law to complete the equivalent frequency multiplication. As shown in Figs. 7(a) and 7(b), the first set of three-phase winding current does not have a coupling inductance in the inverter output, and the complete reverse harmonic voltage will form a larger differential mode voltage, which will eventually make the current harmonic quality slightly worse.

[0097] Example 2: This example uses a 16-pole, 120 rpm, 30° phase belt dual three-phase permanent magnet synchronous motor for simulation experiment, selects the carrier center frequency as 4 kHz, the CHB cascade number as 3, and thus the actual switching frequency meets the equivalent 1 / 2N times relationship, which is 667 Hz. In order to enhance the persuasiveness, this experiment compares two sets of three-phase windings using independent unipolar carrier phase shift modulation with the method proposed in this example.

[0098] In order to suppress the electromagnetic torque odd carrier frequency harmonic components, this example uses the following technical solutions:

[0099] First step: the control loop of the dual three-phase permanent magnet synchronous motor is controlled according to the dual dq vector control, and the instruction voltage U d1 , U q1 , U d2 , U q2 generated by the control loop is converted into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 of each phase winding voltage on the basis of three-phase neutral point two independent, zero sequence component is 0, using the inverse transformation of the 30° phase belt dual three-phase permanent magnet synchronous motor coordinate transformation matrix to convert the given voltage value of the dual dq plane into U a1 , U b1 , U c1 , U a2 , U b2 , U c2 of each phase winding voltage.

[0100]

[0101] Second step: the CHB drive circuit cascade number 3 and the 30° phase belt of the dual three-phase permanent magnet synchronous motor (equivalent to two sets of three-phase windings in the same phase) are brought into the phase shift angle operation equation:

[0102]

[0103] Third step: according to the second step, the phase difference value of the two groups of three-phase winding carrier groups is 25°, each CHB drive circuit is according to the principle of unipolar carrier modulation, and each group of three carriers is expanded according to π / N=π / 3 in turn, and then the three carriers of the second set of three-phase winding carrier group are lagged by 25°, and thus the carrier groups of the two groups of three-phase windings are generated.

[0104] Fourth step, according to U a1 , U b1 , U c1 , U a2 , U b2 , U c2The phase winding phase voltage, for further improving the equivalent switching frequency, the embodiment according to the principle of unipolar modulation will generate six-phase given modulation wave voltage of each half-bridge of two groups of three-phase windings by interlacing 180 degrees of each half-bridge modulation wave of two groups of three-phase windings. Then, three-phase decoupling sinusoidal pulse width modulation is performed with the two groups of carrier waves generated in the third step, and finally the corresponding switching driving signals are generated to drive the switching tubes.

[0105] As shown in FIG. 10(a) and FIG. 10(b), it can be seen that the 30° phase belt angle dual three-phase permanent magnet synchronous motor and the CHB driving circuit work independently and do not cooperate with each other, that is, the two groups of carrier waves coincide and operate, and after the modulation scheme of the two sets of three-phase windings of the embodiment is used, the reverse harmonic voltage can complete the electromagnetic torque odd switching sub-harmonic elimination and odd switching sub-band harmonic suppression. As shown in FIG. 9(a) and FIG. 9(b), because the 30° phase belt angle dual three-phase permanent magnet synchronous motor does not completely eliminate the odd switching sub-band harmonic, the differential mode voltage formed thereby is not as severe as that of the 60° phase belt angle dual three-phase permanent magnet synchronous motor, and finally the current switching sub-harmonic quality of the first set of three-phase windings is improved.

[0106] The embodiment is also applicable to the application occasions of the CHB driving circuit with other natural number cascade number of different phase belt angles (for example, 15°, etc.) of the dual three-phase motor.

[0107] The modulation method of the embodiment can also be realized by a modulation device, which comprises a six-phase CHB driving circuit and a modulation module, and the six-phase CHB driving circuit drives one phase winding of the dual three-phase permanent magnet synchronous motor; each CHB driving circuit is cascaded by N single bodies, and each single body is a full-bridge circuit powered by an independent bus power supply;

[0108] The modulation module is used for performing unipolar carrier wave phase shift modulation on the dual three-phase permanent magnet synchronous motor by using the six-phase CHB driving circuit, the unipolar carrier wave phase shift modulation decomposes the full-bridge circuit in each single body into two half-bridges, the two half-bridge modulation waves are interlaced by 180 degrees, and each carrier wave single body is modulated in turn according to π / N. On the basis of the π / N expansion in each carrier wave single body, the phase of the carrier wave group of the second group of three-phase windings is delayed by θ c2 ;

[0109]

[0110] In the formula, θ c2 is a phase shift angle; ζ is a phase belt angle of the dual three-phase permanent magnet synchronous motor. The principle is the same as the modulation method.

[0111] While the application has been described with reference to particular embodiments thereof, it is to be understood that these embodiments are merely illustrative of the principles and applications of the present application. It will be apparent to those skilled in the art that numerous modifications can be made within the scope of the present application as defined by the appended claims. It is intended that all such modification fall within the spirit and scope of the present application. It will be understood that the features described in connection with one embodiment can be used in connection with another embodiment.

Claims

1. A PWM modulation method for a dual three-phase permanent magnet synchronous motor, characterized in that, The six-phase CHB driving circuit drives one phase winding of the double three-phase permanent magnet synchronous motor; each CHB driving circuit is cascaded by N single bodies, and each single body is a full-bridge circuit powered by an independent bus power supply; The modulation method comprises: S1, generating a six-phase voltage modulation wave of the CHB-driven double three-phase permanent magnet synchronous motor, and mutually offsetting the modulation waves of two half-bridges in each single body of the CHB driving circuit by 180° to obtain a given modulation wave voltage of each group of half-bridges of the six phases; S2, obtaining a phase shift angle according to the number of cascaded CHBs and the phase belt angle of the double three-phase permanent magnet synchronous motor: In the formula, θ c2 is a phase shift angle; ζ is a phase belt angle of the dual three-phase permanent magnet synchronous motor; S3, each carrier group of the two groups of three-phase windings is unfolded in turn according to π / N, and the phase of the carrier group of the second group of three-phase windings is delayed by θ c2 , and further generating carrier groups of the two groups of three-phase windings; S4, according to the given modulation wave voltage of each group of half-bridges of the six phases, and the generated carrier group of the two groups of three-phase windings, three-phase decoupling modulation is carried out.

2. The dual three-phase permanent magnet synchronous machine PWM modulation method according to claim 1, characterized in that, The S1 comprises: The control loop of the double three-phase permanent magnet synchronous motor is controlled according to the double dq vector control, the command voltage generated by the control loop is independent in the three-phase neutral point, and the zero sequence component is 0, and the inverse transformation of the coordinate transformation matrix is used to generate a six-phase voltage modulation wave; According to the principle of single-polarity modulation, the modulation waves of two half-bridges in each single body of the CHB driving circuit are mutually offset by 180° to generate a given modulation wave voltage of each group of half-bridges of the six phases.

3. The dual three-phase permanent magnet synchronous machine PWM modulation method according to claim 2, characterized in that, The phase belt angle of the double three-phase permanent magnet synchronous motor is 30°, and the coordinate transformation matrix in S1 is: Wherein, θ represents the electrical angle of the double three-phase permanent magnet synchronous motor.

4. The dual three-phase permanent magnet synchronous machine PWM modulation method according to claim 2, characterized in that, The phase belt angle of the double three-phase permanent magnet synchronous motor is 60°, and the coordinate transformation matrix in S1 is: Wherein, θ represents the electrical angle of the double three-phase permanent magnet synchronous motor.

5. A storage device readable by a computer, the storage device storing a computer program, characterized in that, The computer program is executed by the processor to realize the steps of the double three-phase permanent magnet synchronous motor PWM modulation method according to any one of claims 1 to 4.

6. A magnetic disc satellite moment attitude control apparatus comprising a storage device, a processor, and a computer program stored in the storage device and operable on the processor, characterized in that, The processor executes the computer program to realize the steps of the double three-phase permanent magnet synchronous motor PWM modulation method according to any one of claims 1 to 4.

7. A computer program product comprising a computer program, characterized in that, The computer program is executed by the processor to realize the steps of the double three-phase permanent magnet synchronous motor PWM modulation method according to any one of claims 1 to 4.

8. A PWM modulator for a dual three-phase permanent magnet synchronous motor, characterized by The six-phase CHB driving circuit drives one phase winding of the double three-phase permanent magnet synchronous motor; each CHB driving circuit is cascaded by N single bodies, and each single body is a full-bridge circuit powered by an independent bus power supply; The modulation module is used for single-polarity carrier phase shift modulation of the double three-phase permanent magnet synchronous motor by the six-phase CHB driving circuit, the single-polarity carrier phase shift modulation decomposes the full-bridge circuit in each single body into two half-bridges, the modulation waves of the two half-bridges are mutually offset by 180 degrees, and each carrier single body is modulated in turn according to π / N; The modulation module is further configured to, on the basis of the fact that each carrier element is unfolded in turn according to π / N, make the carrier group phase of the second group of three-phase windings lag by θ c2 ; In the formula, θ c2 is a phase shift angle; ζ is a phase belt angle of the dual three-phase permanent magnet synchronous motor.

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