Impedance remodeling method and device of inverter and inverter

Abstract

The invention discloses an inverter impedance remodeling method and device and an inverter, and relates to the technical field of new energy. The impedance remodeling method of the inverter comprises the following steps: acquiring a first filter capacitor voltage and an initial three-phase modulation quantity of the inverter; performing differential operation on the first filter capacitor voltage to obtain differential voltage; determining a virtual resistance voltage of the inverter according to the differential voltage; according to the virtual resistance voltage and the initial three-phase modulation amount, a target pulse width modulation signal is determined, and the target pulse width modulation signal is used for controlling on-off of a three-phase converter bridge so as to achieve impedance remodeling of the inverter. According to the embodiment of the invention, the method can directly remodel the output impedance of the inverter, can improve the stability of the multi-inverter twin-trawling system, facilitates the improvement of the parallel group number of twin-trawling systems of an aging experiment plant, and accelerates the delivery speed of a product.

Description

Technical Field

[0001] The present application belongs to the field of new energy technology, and specifically relates to an impedance reshaping method and device of an inverter, and an inverter. Background Art

[0002] Most new energy power generation systems use grid-following inverters as grid-connected interface devices. Each inverter is connected to the common coupling point PCC in parallel, and then connected to the grid via a resistive-inductive transmission line. Ideally, the grid can provide sufficient voltage support for the PCC, and there is no interactive coupling between each inverter and the grid; however, when the transmission line is long, the grid becomes a weak grid characterized by "low short-circuit ratio SCR", and its ability to support PCC voltage is greatly reduced, resulting in strong interactive coupling between inverters and between inverters and the grid, which may cause oscillations in different frequency bands in the system.

[0003] Existing research mainly addresses the stability problem of multiple inverters directly connected to the grid. By establishing the ratio of grid impedance to inverter impedance, the stability of the system is analyzed based on the Nyquist stability criterion. Common methods for improving stability include reducing the bandwidth of the phase-locked loop, modifying the phase-locked loop structure, reducing the grid voltage feedforward coefficient, grid voltage weighted feedforward, online measurement of grid impedance, and various active damping methods.

[0004] However, in the aging test plant of the manufacturing enterprise, the inverter and rectifier are usually connected to the PCC after forming a pair-drag system. At this time, there is coupling on the DC side between the inverter and the rectifier, the system is more complex and the interactive coupling effect is stronger. Although most of the current in the pair-drag system flows through the inverter and rectifier in a circulating manner, and only a small part of the current flows into the power grid, in an extremely weak power grid environment, various synchronization stability problems may still occur due to the reduction of the PCC voltage support capacity. Therefore, there is still a lack of stability improvement methods for multi-inverter pair-drag systems, which is not conducive to increasing the number of parallel groups of pair-drag systems in aging test plants and accelerating the speed of product delivery. Summary of the invention

[0005] The technical problem to be solved by the present application is to provide an impedance reshaping method, device and inverter for the above-mentioned deficiencies in the prior art. The method can be used to directly reshape the output impedance of the inverter, thereby improving the stability of the multi-inverter drag system, which is beneficial to increase the number of parallel groups of the drag system in the aging test plant and speed up the delivery of products.

[0006] In a first aspect, an embodiment of the present application provides a method, comprising:

[0007] Acquiring a first filter capacitor voltage and an initial three-phase modulation amount of the inverter;

[0008] Performing a differential operation on the voltage of the first filter capacitor to obtain a differential voltage;

[0009] According to the differential voltage, the virtual resistance voltage of the inverter is determined;

[0010] According to the virtual resistance voltage and the initial three-phase modulation amount, the target pulse width modulation signal is determined, and the target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve the impedance reshaping of the inverter.

[0011] In some implementations of the first aspect, before performing a differential operation on the first filter capacitor voltage to obtain a differential voltage, the method further includes:

[0012] Performing low-pass filtering on the first filter capacitor voltage to obtain a second filter capacitor voltage;

[0013] Performing a differential operation on the voltage of the first filter capacitor to obtain a differential voltage includes:

[0014] A differential operation is performed on the second filter capacitor voltage to obtain a differential voltage.

[0015] In some implementations of the first aspect, low-pass filtering the first filter capacitor voltage to obtain the second filter capacitor voltage includes:

[0016] Performing low-pass filtering on the first filter capacitor voltage using a first-order Butterworth low-pass filter to obtain a second filter capacitor voltage;

[0017] Among them, the transfer function of the first-order Butterworth low-pass filter satisfies formula (1);

[0018] Formula (1) includes:

[0019]

[0020] Among them, G LPF (s) represents the transfer function of the first-order Butterworth low-pass filter; T LPF represents the time constant of the first-order Butterworth low-pass filter; s represents the frequency operation factor.

[0021] In some implementations of the first aspect, determining a virtual resistance voltage of the inverter according to the differential voltage includes:

[0022] Obtain the filter capacitor capacitance and virtual resistor resistance of the inverter;

[0023] Substitute the filter capacitor value, differential voltage and virtual resistor value into formula (2) to calculate and obtain the virtual resistor voltage of the inverter;

[0024] Formula (2) includes:

[0025]

[0026] Among them, G df (s) represents the virtual resistance voltage; C f Indicates the capacitance of the filter capacitor; Represents differential voltage; R v Indicates the virtual resistance value.

[0027] In some implementations of the first aspect, determining a target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount includes:

[0028] The virtual resistance voltage and the initial three-phase modulation amount are superimposed to obtain the target three-phase modulation amount;

[0029] The target three-phase modulation amount is pulse-width modulated to obtain a target pulse-width modulation signal.

[0030] In some implementations of the first aspect, obtaining an initial three-phase modulation amount includes:

[0031] Obtaining a DC voltage reference value, a DC voltage measured value, a q-axis current reference value, and a grid voltage phase angle of the inverter;

[0032] Integrate the difference between the DC voltage measured value and the DC voltage reference value to obtain the d-axis current reference value;

[0033] Determine the d-axis voltage modulation amount according to the d-axis current reference value;

[0034] Determine the q-axis voltage modulation amount according to the q-axis current reference value;

[0035] Based on the grid voltage phase angle, coordinate transformation is performed on the d-axis voltage modulation amount and the q-axis voltage modulation amount to obtain the initial three-phase modulation amount.

[0036] In some implementations of the first aspect, determining the d-axis voltage modulation amount according to the d-axis current reference value includes:

[0037] Obtaining the grid voltage feedforward coefficient, as well as the actual measured value of the d-axis current and the actual measured value of the d-axis voltage of the inverter;

[0038] Integrate the difference between the d-axis current reference value and the d-axis current measured value to obtain the d-axis current intermediate value;

[0039] The product of the measured value of the d-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the d-axis current are determined as the d-axis voltage modulation amount;

[0040] and / or, determining the q-axis voltage modulation amount according to the q-axis current reference value, including:

[0041] Obtaining the grid voltage feedforward coefficient, as well as the actual measured values ​​of the q-axis current and q-axis voltage of the inverter;

[0042] Integrate the difference between the q-axis current reference value and the q-axis current measured value to obtain the q-axis current intermediate value;

[0043] The product of the measured value of the q-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the q-axis current are determined as the q-axis voltage modulation amount.

[0044] Based on the same inventive concept, in a second aspect, an embodiment of the present application further provides an impedance reshaping device for an inverter, comprising:

[0045] An acquisition module, used for acquiring a first filter capacitor voltage and an initial three-phase modulation amount of the inverter;

[0046] A calculation module, connected to the acquisition module, for performing a differential operation on the voltage of the first filter capacitor to obtain a differential voltage;

[0047] A first determination module, connected to the operation module, for determining a virtual resistance voltage of the inverter according to the differential voltage;

[0048] The second determination module is connected to the first determination module and is used to determine the target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount. The target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter.

[0049] In some embodiments of the second aspect, the device further comprises:

[0050] A filtering module, used for performing low-pass filtering on the first filter capacitor voltage to obtain a second filter capacitor voltage;

[0051] Operation module, specifically used for:

[0052] A differential operation is performed on the second filter capacitor voltage to obtain a differential voltage.

[0053] In some implementations of the second aspect, the filtering module is specifically used to:

[0054] Performing low-pass filtering on the first filter capacitor voltage using a first-order Butterworth low-pass filter to obtain a second filter capacitor voltage;

[0055] Among them, the transfer function of the first-order Butterworth low-pass filter satisfies formula (1);

[0056] Formula (1) includes:

[0057]

[0058] Among them, G LPF(s) represents the transfer function of the first-order Butterworth low-pass filter; T LPF represents the time constant of the first-order Butterworth low-pass filter; s represents the frequency operation factor.

[0059] Based on the same inventive concept, in a third aspect, an embodiment of the present application further provides an inverter, comprising an inverter body and the impedance reshaping device of any one of the second aspects.

[0060] According to the impedance reshaping method, device and inverter of the inverter provided in the embodiment of the present application, a differential voltage is obtained by performing a differential operation on the first filter capacitor voltage, and then a virtual resistance voltage is determined according to the differential voltage. Then, a target pulse width modulation signal is determined according to the virtual resistance voltage and the initial three-phase modulation amount. The target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter. That is to say, compared with the methods of modifying the phase-locked loop structure, grid voltage weighted feedforward and grid impedance online measurement in the prior art, the embodiment of the present application introduces a differential link in the grid voltage feedforward path, which is equivalent to inserting a virtual resistor in series in the output filter capacitor branch, thereby reshaping the output impedance of the inverter so that the impedance characteristics of the inverter match the impedance characteristics of the grid. Without increasing the control complexity too much, it can improve the stability of the multi-inverter drag system, which is beneficial to increase the number of parallel groups of the drag system in the aging test plant and speed up the product delivery speed; compared with some active damping methods in the prior art, the embodiment of the present application can flexibly select the frequency range of the virtual resistor, and has almost no effect on the output voltage of the inverter, and there is no need to add additional voltage and current sensors; compared with the prior art parameter adjustment methods such as reducing the phase-locked loop bandwidth and reducing the grid voltage feedforward coefficient, the embodiment of the present application can directly reshape the output impedance of the inverter, which can more effectively improve the stability of the multi-inverter drag system. BRIEF DESCRIPTION OF THE DRAWINGS

[0061] Figure 1 A schematic diagram showing a structure of a towing system in the prior art is shown;

[0062] Figure 2 A schematic diagram showing a flow chart of an impedance reshaping method for an inverter provided in an embodiment of the present application;

[0063] Figure 3 A control structure block diagram of an impedance reshaping method for an inverter provided in an embodiment of the present application is shown;

[0064] Figure 4 A schematic diagram showing the voltage and current at the grid connection point in the prior art;

[0065] Figure 5 A schematic diagram showing output voltage and current of an inverter in the prior art;

[0066] Figure 6 A schematic diagram showing the inverter phase voltage and bus voltage in the prior art;

[0067] Figure 7 A schematic diagram showing the voltage and current of the grid connection point provided in an embodiment of the present application;

[0068] Figure 8 A schematic diagram showing the output voltage and current of the inverter provided in an embodiment of the present application;

[0069] Fig. 9 A schematic diagram showing the inverter phase voltage and bus voltage provided in an embodiment of the present application;

[0070] Fig.10 A schematic structural diagram of an impedance reshaping device for an inverter provided in an embodiment of the present application is shown;

[0071] Fig.11 A schematic structural diagram of an inverter provided in an embodiment of the present application is shown. DETAILED DESCRIPTION

[0072] In order to enable those skilled in the art to better understand the technical solution of the present application, the present application is further described in detail below with reference to the accompanying drawings and embodiments.

[0073] The features and exemplary embodiments of various aspects of the present application will be described in detail below. In order to make the purpose, technical solutions and advantages of the present application clearer, the present application will be further described in detail below in conjunction with the accompanying drawings and specific embodiments. It should be understood that the specific embodiments described herein are only configured to explain the present application and are not configured to limit the present application. For those skilled in the art, the present application can be implemented without the need for some of these specific details. The following description of the embodiments is only to provide a better understanding of the present application by illustrating the examples of the present application.

[0074] It should be noted that, in this article, relational terms such as first and second, etc. are only used to distinguish one entity or operation from another entity or operation, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Moreover, the terms "include", "comprise" or any other variants thereof are intended to cover non-exclusive inclusion, so that a process, method, article or device including a series of elements includes not only those elements, but also other elements not explicitly listed, or also includes elements inherent to such process, method, article or device. In the absence of further restrictions, the elements defined by the statement "include..." do not exclude the presence of other identical elements in the process, method, article or device including the elements.

[0075] It should be understood that the term "and / or" used in this article is only a description of the association relationship of associated objects, indicating that there can be three relationships. For example, A and / or B can represent: A exists alone, A and B exist at the same time, and B exists alone. In addition, the character " / " in this article generally indicates that the associated objects before and after are in an "or" relationship.

[0076] In order to better understand the impedance reshaping method of the inverter provided in the embodiment of the present application, the towing system and its related contents involved in the embodiment of the present application are introduced below.

[0077] like Figure 1 As shown, each group of towing systems 1 includes an inverter 11, a rectifier 12 and a first transformer 13. The inverter 11 corresponds to the rectifier 12 and the first transformer 13 one by one. The first transformer 11 is a transformer on the towing system side. For each group of towing systems 1, its corresponding inverter 11 and rectifier 12 can be connected through a DC bus. The rectifier 12 is connected to the grid connection point PCC through the inverter 11 and the first transformer 13 in turn; the DC buses of different groups of towing systems 1 are not connected. The rectifier 12 can be used to power the inverter 11.

[0078] The resistance of the line on the inverter 11 side is R i , the inductance is L i ; The resistance of the circuit on the rectifier 12 side is R r , the inductance is L r ; The resistance of the grid side line is R g , the inductance is L g The second transformer 21 is a transformer on the grid side. For example, the voltage corresponding to the first transformer 13 is 800V / 400V; the voltage corresponding to the second transformer 21 is 400V / 10kV.

[0079] Assume that the short-circuit ratio of the SCR on the inverter side is i , the short-circuit ratio of the rectifier side is SCR r , and the short-circuit ratio of the grid side is SCR g , then the impedance Z of the inverter side line i , the impedance Z of the rectifier side line r and the impedance Z of the grid-side route g satisfy:

[0080]

[0081] Among them, V g is the grid voltage, and P is the rated power. Assuming that the line resistance is 10% of the line inductance, the expressions of line resistance and inductance at each location satisfy:

[0082]

[0083] Among them, f g is the grid frequency.

[0084] It should be noted that the application scenario of the above-mentioned towing system 1 may include a microgrid, wherein the microgrid includes a small power generation and distribution system composed of a distributed power source, an energy storage device, an energy conversion device, a load, a monitoring and protection device, and the like.

[0085] The following describes an impedance reshaping method for an inverter provided in an embodiment of the present application.

[0086] Example 1

[0087] Figure 2 A schematic flow chart of an impedance reshaping method for an inverter provided in an embodiment of the present application is shown.

[0088] The impedance reshaping method of the inverter provided in the embodiment of the present application can be executed by a controller in the inverter, an impedance reshaping device of the inverter, etc. The following takes the impedance reshaping method of the inverter executed by the controller in the inverter as an example for description.

[0089] like Figure 2 As shown, the impedance reshaping method of the inverter provided in the embodiment of the present application may include steps S210 to S240.

[0090] S210, obtaining a first filter capacitor voltage and an initial three-phase modulation amount of the inverter.

[0091] S220 , performing a differential operation on the first filter capacitor voltage to obtain a differential voltage.

[0092] S230: Determine a virtual resistance voltage of the inverter according to the differential voltage.

[0093] S240, determining a target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount, wherein the target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter.

[0094] According to the impedance reshaping method of the inverter provided in the embodiment of the present application, a differential voltage is obtained by performing a differential operation on the first filter capacitor voltage, and then a virtual resistance voltage is determined based on the differential voltage. Then, a target pulse width modulation signal is determined based on the virtual resistance voltage and the initial three-phase modulation amount. The target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter. That is to say, compared with the methods of modifying the phase-locked loop structure, grid voltage weighted feedforward and grid impedance online measurement in the prior art, the embodiment of the present application introduces a differential link in the grid voltage feedforward path, which is equivalent to inserting a virtual resistor in series in the output filter capacitor branch, thereby reshaping the output impedance of the inverter so that the impedance characteristics of the inverter match the impedance characteristics of the grid. Without increasing the control complexity too much, it can improve the stability of the multi-inverter drag system, which is beneficial to increase the number of parallel groups of the drag system in the aging test plant and speed up the product delivery speed; compared with some active damping methods in the prior art, the embodiment of the present application can flexibly select the frequency range of the virtual resistor, and has almost no effect on the output voltage of the inverter, and there is no need to add additional voltage and current sensors; compared with the prior art parameter adjustment methods such as reducing the phase-locked loop bandwidth and reducing the grid voltage feedforward coefficient, the embodiment of the present application can directly reshape the output impedance of the inverter, which can more effectively improve the stability of the multi-inverter drag system.

[0095] The specific implementation methods of the above steps are introduced below.

[0096] In step S210, the inverter may be any inverter that needs to perform impedance reshaping. For example, the inverter may be an inverter in a microgrid traction system, which is used to convert electrical energy from a battery such as a solar panel or a DC power source into an AC power source.

[0097] The first filter capacitor voltage may be a filter capacitor voltage of the inverter that has not been preprocessed, that is, an initial filter capacitor voltage of the inverter.

[0098] The initial three-phase modulation amount may be the three-phase modulation amount obtained in the prior art, that is, the three-phase modulation amount obtained when the differential link and the low-pass filter link provided in the embodiment of the present application are not introduced. The initial three-phase modulation amount may include the initial A-phase modulation amount, the initial B-phase modulation amount, and the initial C-phase modulation amount.

[0099] As an example, the first filter capacitor voltage may be obtained by measurement.

[0100] As another example, the first filter capacitor voltage is pre-stored in the controller in the inverter for subsequent direct call.

[0101] In some implementations, obtaining the initial three-phase modulation amount may include:

[0102] Obtaining a DC voltage reference value, a DC voltage measured value, a q-axis current reference value, and a grid voltage phase angle of the inverter;

[0103] Integrate the difference between the DC voltage measured value and the DC voltage reference value to obtain the d-axis current reference value;

[0104] Determine the d-axis voltage modulation amount according to the d-axis current reference value;

[0105] Determine the q-axis voltage modulation amount according to the q-axis current reference value;

[0106] Based on the grid voltage phase angle, coordinate transformation is performed on the d-axis voltage modulation amount and the q-axis voltage modulation amount to obtain the initial three-phase modulation amount.

[0107] The DC voltage reference value may be a reference value of the DC voltage of the inverter.

[0108] The DC voltage measured value may be an actual measured value of the DC voltage of the inverter.

[0109] The q-axis current reference value may be a reference value of the current of the q-axis of the inverter in the dq coordinate system.

[0110] The grid voltage phase angle may be a phase angle of the grid voltage.

[0111] As an example, the DC voltage reference value, the DC voltage measured value, the q-axis current reference value and the grid voltage phase angle may be pre-stored in the controller of the inverter for subsequent direct call.

[0112] As another example, the measured value of the DC voltage may be obtained by measuring a voltage measuring device that is communicatively connected to the controller of the inverter.

[0113] For example, Figure 3 As shown, the measured DC voltage value v dc With DC voltage reference value V dc The difference v dc -V dc Through the transfer function H v (s) is integrated to obtain the d-axis current reference value i dr Among them, the transfer function H v (s) can satisfy H v (s) = K p_v +K i_v / s,K p_v is the first proportional coefficient of the PI regulator, K i_v is the first integral coefficient of the PI regulator, and s is the operation factor.

[0114] In some examples, determining the d-axis voltage modulation amount according to the d-axis current reference value may include:

[0115] Obtaining the grid voltage feedforward coefficient, as well as the actual measured value of the d-axis current and the actual measured value of the d-axis voltage of the inverter;

[0116] Integrate the difference between the d-axis current reference value and the d-axis current measured value to obtain the d-axis current intermediate value;

[0117] The product of the measured value of the d-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the d-axis current are determined as the d-axis voltage modulation amount.

[0118] The grid voltage feedforward coefficient is the grid voltage feedforward coefficient.

[0119] The d-axis current measured value is the actual measured value of the inverter's d-axis current in the dq coordinate system.

[0120] The measured value of the d-axis voltage is the actual measured value of the voltage of the d-axis of the inverter in the dq coordinate system.

[0121] As an example, the grid voltage feedforward coefficient, the d-axis current measured value, and the d-axis voltage measured value may be stored in the controller of the inverter for subsequent direct call.

[0122] As another example, the actual d-axis current value may be measured by a current measuring device that is communicatively connected to the controller of the inverter; and the actual d-axis voltage value may be measured by a voltage measuring device that is communicatively connected to the controller of the inverter.

[0123] For example, Figure 3 As shown, the d-axis current reference value i dr The measured value of d-axis current i d The difference i dr -i d Through the transfer function H i (s) is integrated to obtain the intermediate value of the d-axis current. Then the measured value of the d-axis voltage v d and grid voltage feedforward coefficient K pf The product of v d ×K pf , and the sum of the intermediate value of the d-axis current is determined as the d-axis voltage modulation. Among them, the transfer function H i (s) can satisfy: H i (s) = K p_i +K i_i / s,K p_i is the second proportional coefficient of the PI regulator, K i_i is the second integral coefficient of the PI regulator, and s is the operation factor.

[0124] In some examples, determining the q-axis voltage modulation amount according to the q-axis current reference value may include:

[0125] Obtaining the grid voltage feedforward coefficient, as well as the actual measured values ​​of the q-axis current and q-axis voltage of the inverter;

[0126] Integrate the difference between the q-axis current reference value and the q-axis current measured value to obtain the q-axis current intermediate value;

[0127] The product of the measured value of the q-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the q-axis current are determined as the q-axis voltage modulation amount.

[0128] The q-axis current measured value is the actual measured value of the current of the q-axis of the inverter in the dq coordinate system.

[0129] The q-axis voltage measured value is the actual measured value of the voltage on the q-axis of the inverter in the dq coordinate system.

[0130] The value of the grid voltage feedforward coefficient is greater than or equal to 0 and less than or equal to 1. Exemplarily, the value of the grid voltage feedforward coefficient may be 0.41, 0.5, 0.63, etc.

[0131] As an example, the actual measured value of the q-axis current and the actual measured value of the q-axis voltage may be stored in the controller of the inverter for subsequent direct call.

[0132] As another example, the actual q-axis current value may be measured by a current measuring device that is communicatively connected to the controller of the inverter; and the actual q-axis voltage value may be measured by a voltage measuring device that is communicatively connected to the controller of the inverter.

[0133] For example, Figure 3 As shown, the q-axis current reference value i qr The measured value of q-axis current i q The difference i qr -i q Through the transfer function H i (s) is integrated to obtain the intermediate value of the q-axis current. Then the measured value of the q-axis voltage v q and grid voltage feedforward coefficient K pf The product of v q ×K pf , and the sum of the intermediate value of the q-axis current is determined as the q-axis voltage modulation amount.

[0134] In the embodiment of the present application, the output of the DC voltage outer loop is used as the d-axis given value of the positive-sequence current inner loop, and the q-axis current given value is artificially set to avoid static instability problems caused by PCC voltage drop.

[0135] For example, Figure 3 As shown, the grid voltage phase angle θ cThe d-axis voltage modulation amount and the q-axis voltage modulation amount are input into the conversion module dq / abc of the dq coordinate and the abc coordinate to obtain the initial three-phase modulation amount.

[0136] In step S220, after obtaining the first filter capacitor voltage and the initial three-phase modulation amount of the inverter, the controller in the inverter may further perform a differential operation on the first filter capacitor voltage to obtain a differential voltage.

[0137] In some implementations, before performing a differential operation on the first filter capacitor voltage to obtain a differential voltage, the method further includes:

[0138] Performing low-pass filtering on the first filter capacitor voltage to obtain a second filter capacitor voltage;

[0139] Performing a differential operation on the voltage of the first filter capacitor to obtain a differential voltage includes:

[0140] A differential operation is performed on the second filter capacitor voltage to obtain a differential voltage.

[0141] In this embodiment, by low-pass filtering the first filter capacitor voltage, the influence of high-frequency components can be reduced or avoided, preventing high-frequency oscillation in the tow system, and limiting the frequency range of the grid voltage differential feedforward effect.

[0142] The second filter capacitor voltage may be a filter capacitor voltage obtained by performing low-pass filtering on the first filter capacitor voltage.

[0143] In some examples, low-pass filtering the first filter capacitor voltage to obtain the second filter capacitor voltage includes:

[0144] Performing low-pass filtering on the first filter capacitor voltage using a first-order Butterworth low-pass filter to obtain a second filter capacitor voltage;

[0145] Among them, the transfer function of the first-order Butterworth low-pass filter satisfies formula (1);

[0146] Formula (1) includes:

[0147]

[0148] Among them, G LPF (s) represents the transfer function of the first-order Butterworth low-pass filter; T LPF represents the time constant of the first-order Butterworth low-pass filter; s represents the frequency operation factor.

[0149] In this implementation, by using a first-order Butterworth low-pass filter to perform low-pass filtering on the first filter capacitor voltage, the phase lag introduced by the low-pass filtering link can be reduced, which is simple and effective.

[0150] The time constant determines the cutoff frequency of the first-order Butterworth low-pass filter. And the time constant satisfies: Among them, w c is the cut-off angular frequency (rad / s), f c is the cut-off frequency (Hz).

[0151] For example, the cut-off frequency f c is 100Hz, and the corresponding time constant is T LPF =1 / (2×π×100Hz)=1.59×10 -3 s. The above values ​​are only used for examples and are not used to limit the present application. The cut-off frequency f c It can be greater than 100Hz or less than 100Hz.

[0152] In other implementations, other low-pass filters besides the first-order Butterworth low-pass filter may also be used, which is not limited here.

[0153] In step S230, after the controller of the inverter performs a differential operation on the first filter capacitor voltage to obtain a differential voltage, it can also determine the virtual resistance voltage of the inverter according to the differential voltage.

[0154] The virtual resistor voltage can be used to reshape the impedance of the inverter.

[0155] In some implementations, determining a virtual resistance voltage of the inverter according to the differential voltage includes:

[0156] Obtain the filter capacitor capacitance and virtual resistor resistance of the inverter;

[0157] Substitute the filter capacitor value, differential voltage and virtual resistor value into formula (2) to calculate and obtain the virtual resistor voltage of the inverter;

[0158] Formula (2) includes:

[0159]

[0160] Among them, G df (s) represents the virtual resistance voltage; C f Indicates the capacitance of the filter capacitor; Represents differential voltage; R v Indicates the virtual resistance value.

[0161] The capacitance of the filter capacitor may be the capacitance of the filter capacitor of the inverter.

[0162] The virtual resistance value may be the resistance value of the virtual resistance of the inverter.

[0163] Exemplarily, the capacitance of the filter capacitor and the resistance of the virtual resistor may be stored in the controller of the inverter, and may be directly called later.

[0164] That is, the product of the differential voltage and the capacitance of the filter capacitor is determined as the filter capacitor current; and the product of the filter capacitor current and the resistance of the virtual resistor is determined as the virtual resistor voltage.

[0165] In an embodiment of the present application, a differential link is introduced into the grid voltage feedforward path, which is equivalent to inserting a virtual resistor in series in the output filter capacitor branch, thereby reshaping the inverter output impedance and improving the stability of the multi-inverter towing system. Among them, the grid voltage proportional feedforward path is located in the positive sequence current loop in the dq coordinate system; and the grid voltage differential feedforward path is located in the abc coordinate system, thereby reducing or avoiding the influence of the phase-locked loop on the grid voltage differential feedforward path. Among them, the differential link may include the extraction of the filter capacitor current and the virtual impedance voltage.

[0166] In other embodiments, the method of determining the virtual resistance voltage of the inverter may include: obtaining the output filter capacitor current; and taking the product of the filter capacitor current and the virtual resistance value as the virtual resistance voltage. That is, the output filter capacitor current can be directly sampled in the hardware circuit design, and the virtual resistance voltage can be obtained by multiplying the filter capacitor current by the virtual resistance value in the control loop, that is, G df (s) = i f ×R v , where i f is the filter capacitor current, R v is the virtual resistance value.

[0167] In step S240, after determining the virtual resistance voltage of the inverter according to the differential voltage, the controller of the inverter may also determine the target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount.

[0168] In some implementations, determining a target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount may include:

[0169] The virtual resistance voltage and the initial three-phase modulation amount are superimposed to obtain the target three-phase modulation amount;

[0170] The target three-phase modulation amount is pulse-width modulated to obtain a target pulse-width modulation signal.

[0171] For example, the virtual resistor voltage is V r The initial three-phase modulation includes the initial A-phase modulation V a , initial B phase modulation V b and the initial C phase modulation V c The target three-phase modulation includes the target A phase modulation ma 、Target B phase modulation m b and the target C phase modulation m c , then m a =V a +V r ;m b =V b +V r ;m b =V b +V r .

[0172] For example, Figure 3 As shown, the transfer function G LPF The virtual resistance voltage output by (s) and the initial three-phase modulation output by the conversion module dq / abc are superimposed to obtain the target three-phase modulation m abc .

[0173] In the simulation test of the embodiment of the present application, SCR i =2.75, SCR r =2.75, SCR g =3, V g =400V, P=330kW, f g =50Hz, virtual resistance value R v =13Ω as an example, the current multiplication method is used to simulate the scenario of multiple groups of drag systems in parallel. One group of drag systems is added every 1s until the system begins to have low-frequency oscillation, and the number of parallel groups with critical stability of the system can be obtained.

[0174] It should be noted that the above values ​​are only used as examples and are not used to limit this application. The virtual resistance value can be reasonably selected according to different types of inverters. If it is too small, it may not produce obvious effects. If it is too large, it may cause the control loop to not work properly. According to formula (2), it can be seen that the final applied virtual resistance voltage is related to the first filter capacitor voltage v abc And the filter capacitor value C f Therefore, the virtual resistor value needs to be appropriately adjusted according to parameters such as the voltage and current levels and filter capacitor values ​​of different models.

[0175] The simulation results in the prior art are as follows Figures 4 to 6 As shown in Figure 2, the simulation results without using grid voltage differential feedforward are as follows: Figures 4 to 6 As shown. Among them, Figures 4 to 6 The horizontal axis in the table is time, and the unit is s; Figure 4 The vertical axis of the upper middle figure is the grid connection point voltage, in V; Figure 4 The vertical axis of the lower middle figure is the grid-connected point current, in A; Figure 5The vertical axis of the upper middle figure is the inverter output voltage, in V; Figure 5 The vertical axis of the lower figure is the inverter output current, in A; Figure 6 The vertical axis of the upper middle figure is the inverter phase voltage (PV voltage), in V; Figure 6 The vertical axis in the lower middle figure is the inverter bus voltage, in V.

[0176] like Figures 4 to 6 As shown in the figure, when 6 groups of towing systems are connected in parallel, the system can operate stably. When 7 groups of towing systems are connected in parallel, the voltage and current at the grid connection point have a small low-frequency oscillation, and the oscillation amplitude is fixed; the inverter output voltage and current have only a very small low-frequency oscillation, and the inverter PV voltage and bus voltage also have a certain amplitude of low-frequency oscillation. It can be seen that when the grid voltage differential feedforward is not used, the number of parallel groups for critical stability of the system is 7.

[0177] The simulation results of the impedance reshaping method of the inverter provided by the embodiment of the present application are as follows: Figures 7 to 9 As shown in Figure 2, the simulation results using grid voltage differential feedforward are as follows: Figures 7 to 9 As shown. Among them, Figures 7 to 9 The horizontal axis in the table is time, and the unit is s; Figure 7 The vertical axis of the upper middle figure is the grid connection point voltage, in V; Figure 7 The vertical axis of the lower middle figure is the grid-connected point current, in A; Figure 8 The vertical axis of the upper middle figure is the inverter output voltage, in V; Figure 8 The vertical axis of the lower figure is the inverter output current, in A; Fig. 9 The vertical axis of the upper middle figure is the inverter phase voltage (PV voltage), in V; Fig. 9 The vertical axis in the lower middle figure is the inverter bus voltage, in V.

[0178] like Figures 7 to 9 As shown, when 11 groups of towing systems are connected in parallel, the system can operate stably. When 12 groups of towing systems are connected in parallel, the voltage and current at the grid connection point have small low-frequency oscillations, and the oscillation amplitude gradually increases; the inverter output voltage and current have only very small low-frequency oscillations, and the inverter PV voltage and bus voltage also have low-frequency oscillations of a certain amplitude. It can be seen that when the grid voltage differential feedforward is adopted, the number of parallel groups for critical stability of the system is 12 groups. Compared with not adopting the grid voltage differential feedforward, the number of parallel groups has increased by 5 groups, which verifies the effectiveness of the impedance reshaping method of the inverter proposed in the embodiment of the present application.

[0179] Compared with the prior art, the embodiments of the present application have at least the following beneficial effects:

[0180] Compared with the existing low-frequency oscillation suppression methods, the main advantages of the method proposed in the embodiment of the present application are the simplicity and flexibility of the control method. Compared with methods such as modifying the phase-locked loop structure, grid voltage weighted feedforward, and grid impedance online measurement, the embodiment of the present application can greatly improve the stability of the system without increasing the control complexity too much; compared with the partial active damping method, the embodiment of the present application can flexibly select the frequency range of the virtual resistance, and has almost no effect on the output voltage of the inverter, and there is no need to add additional voltage and current sensors; compared with the parameter adjustment methods such as reducing the phase-locked loop bandwidth and reducing the grid voltage feedforward coefficient, the embodiment of the present application directly reshapes the output impedance of the inverter, which can more effectively improve the system stability. This method directly reshapes the output impedance of the inverter, greatly improving the stability of the system without increasing the control complexity too much, thereby increasing the number of parallel groups of the towing system of the aging test plant and speeding up the product delivery speed.

[0181] Based on the same inventive concept, an embodiment of the present application further provides an impedance reshaping device for an inverter. The impedance reshaping device for an inverter provided in an embodiment of the present application is described below in conjunction with the accompanying drawings.

[0182] Example 2

[0183] Fig.10 A structural schematic diagram of an impedance reshaping device for an inverter provided in an embodiment of the present application is shown.

[0184] like Fig.10 As shown, the impedance reshaping device 1000 of the inverter provided in the embodiment of the present application may include an acquisition module 1010 , a calculation module 1020 , a first determination module 1030 and a second determination module 1040 .

[0185] The acquisition module 1010 is used to acquire the first filter capacitor voltage and the initial three-phase modulation amount of the inverter;

[0186] The operation module 1020 is connected to the acquisition module 1010 and is used to perform a differential operation on the first filter capacitor voltage to obtain a differential voltage;

[0187] A first determination module 1030, connected to the operation module 1020, is used to determine the virtual resistance voltage of the inverter according to the differential voltage;

[0188] The second determination module 1040 is connected to the first determination module 1030, and is used to determine the target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount. The target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter.

[0189] According to the impedance reshaping device of the inverter provided in the embodiment of the present application, a differential voltage is obtained by performing a differential operation on the first filter capacitor voltage, and then a virtual resistance voltage is determined based on the differential voltage. Then, a target pulse width modulation signal is determined based on the virtual resistance voltage and the initial three-phase modulation amount. The target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter. That is to say, compared with the methods of modifying the phase-locked loop structure, grid voltage weighted feedforward and grid impedance online measurement in the prior art, the embodiment of the present application introduces a differential link in the grid voltage feedforward path, which is equivalent to inserting a virtual resistor in series in the output filter capacitor branch, thereby reshaping the output impedance of the inverter so that the impedance characteristics of the inverter match the impedance characteristics of the grid. Without increasing the control complexity too much, it can improve the stability of the multi-inverter drag system, which is beneficial to increase the number of parallel groups of the drag system in the aging test plant and speed up the product delivery speed; compared with some active damping methods in the prior art, the embodiment of the present application can flexibly select the frequency range of the virtual resistor, and has almost no effect on the output voltage of the inverter, and there is no need to add additional voltage and current sensors; compared with the prior art parameter adjustment methods such as reducing the phase-locked loop bandwidth and reducing the grid voltage feedforward coefficient, the embodiment of the present application can directly reshape the output impedance of the inverter, which can more effectively improve the stability of the multi-inverter drag system.

[0190] In some embodiments, the impedance reshaping device 1000 may further include:

[0191] A filtering module, used for performing low-pass filtering on the first filter capacitor voltage to obtain a second filter capacitor voltage;

[0192] Operation module, specifically used for:

[0193] A differential operation is performed on the second filter capacitor voltage to obtain a differential voltage.

[0194] In some implementations, the filtering module may be specifically used to:

[0195] Performing low-pass filtering on the first filter capacitor voltage using a first-order Butterworth low-pass filter to obtain a second filter capacitor voltage;

[0196] Among them, the transfer function of the first-order Butterworth low-pass filter satisfies formula (1);

[0197] Formula (1) includes:

[0198]

[0199] Among them, G LPF (s) represents the transfer function of the first-order Butterworth low-pass filter; T LPFrepresents the time constant of the first-order Butterworth low-pass filter; s represents the frequency operation factor.

[0200] In some implementations, the first determining module 1030 may be specifically configured to:

[0201] According to the differential voltage, the virtual resistance voltage of the inverter is determined, including:

[0202] Obtain the filter capacitor capacitance and virtual resistor resistance of the inverter;

[0203] Substitute the filter capacitor value, differential voltage and virtual resistor value into formula (2) to calculate and obtain the virtual resistor voltage of the inverter;

[0204] Formula (2) includes:

[0205]

[0206] Among them, G df (s) represents the virtual resistance voltage; C f Indicates the capacitance of the filter capacitor; Represents differential voltage; R v Indicates the virtual resistance value.

[0207] In some implementations, the second determining module 1040 may be specifically configured to:

[0208] The virtual resistance voltage and the initial three-phase modulation amount are superimposed to obtain the target three-phase modulation amount;

[0209] The target three-phase modulation amount is pulse-width modulated to obtain a target pulse-width modulated signal.

[0210] In some implementations, the acquisition module 1010 may be specifically used to:

[0211] Obtaining a DC voltage reference value, a DC voltage measured value, a q-axis current reference value, and a grid voltage phase angle of the inverter;

[0212] Integrate the difference between the DC voltage measured value and the DC voltage reference value to obtain the d-axis current reference value;

[0213] Determine the d-axis voltage modulation amount according to the d-axis current reference value;

[0214] Determine the q-axis voltage modulation amount according to the q-axis current reference value;

[0215] Based on the grid voltage phase angle, coordinate transformation is performed on the d-axis voltage modulation amount and the q-axis voltage modulation amount to obtain the initial three-phase modulation amount.

[0216] In some implementations, the acquisition module 1010 may be specifically used to:

[0217] Obtaining the grid voltage feedforward coefficient, as well as the actual measured value of the d-axis current and the actual measured value of the d-axis voltage of the inverter;

[0218] Integrate the difference between the d-axis current reference value and the d-axis current measured value to obtain the d-axis current intermediate value;

[0219] The product of the measured value of the d-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the d-axis current are determined as the d-axis voltage modulation amount;

[0220] And / or, the acquisition module 1010 may be specifically used for:

[0221] Obtaining the grid voltage feedforward coefficient, as well as the actual measured values ​​of the q-axis current and q-axis voltage of the inverter;

[0222] Integrate the difference between the q-axis current reference value and the q-axis current measured value to obtain the q-axis current intermediate value;

[0223] The product of the measured value of the q-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the q-axis current are determined as the q-axis voltage modulation amount.

[0224] The impedance reshaping device of the inverter provided in the embodiment of the present application can be used to execute the impedance reshaping method of the inverter, that is, it has the beneficial effects and implementation methods of the impedance reshaping method of the inverter provided in Example 1 of the present application. For details, please refer to the specific description of the impedance reshaping method of the inverter in the above Example 1, and this embodiment will not be repeated here.

[0225] Based on the same inventive concept, an embodiment of the present application further provides an inverter. The inverter provided in the embodiment of the present application is described below in conjunction with the accompanying drawings.

[0226] Example 3

[0227] like Fig.11 As shown, the embodiment of the present application further provides an inverter 100, which may include an inverter body 2000 and the impedance reshaping device 1000 of any one of Embodiments 2.

[0228] The inverter provided in the embodiment of the present application includes the impedance reshaping device of any one of Embodiment 2, that is, it has the beneficial effects and implementation methods of the impedance reshaping device provided in Embodiment 2 of the present application. For details, please refer to the specific description of the impedance reshaping device in the above Embodiment 2, which will not be repeated in this embodiment.

[0229] It is to be understood that the above embodiments are merely exemplary embodiments used to illustrate the principles of the present application, but the present application is not limited thereto. For those skilled in the art, various modifications and improvements can be made without departing from the spirit and substance of the present application, and these modifications and improvements are also considered to be within the scope of protection of the present application.

Claims

1. An impedance reshaping method for an inverter, characterized in that: include: Acquiring a first filter capacitor voltage and an initial three-phase modulation amount of the inverter; Performing a differential operation on the first filter capacitor voltage to obtain a differential voltage; Determining a virtual resistance voltage of the inverter according to the differential voltage; A target pulse width modulation signal is determined according to the virtual resistance voltage and the initial three-phase modulation amount, and the target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter.

2. The method according to claim 1, characterized in that Before performing a differential operation on the first filter capacitor voltage to obtain a differential voltage, the method further includes: Performing low-pass filtering on the first filter capacitor voltage to obtain a second filter capacitor voltage; The performing a differential operation on the first filter capacitor voltage to obtain a differential voltage includes: A differential operation is performed on the second filter capacitor voltage to obtain a differential voltage.

3. The method according to claim 2, characterized in that The low-pass filtering the first filter capacitor voltage to obtain the second filter capacitor voltage includes: Using a first-order Butterworth low-pass filter to low-pass filter the first filter capacitor voltage to obtain a second filter capacitor voltage; Among them, the transfer function of the first-order Butterworth low-pass filter satisfies formula (1); The formula (1) includes: Among them, G LPF (s) represents the transfer function of the first-order Butterworth low-pass filter; T LPF represents the time constant of the first-order Butterworth low-pass filter; s represents the frequency operation factor.

4. The method according to claim 1, characterized in that: Determining the virtual resistance voltage of the inverter according to the differential voltage includes: Obtain the filter capacitor capacitance and virtual resistor resistance of the inverter; Substituting the capacitance of the filter capacitor, the differential voltage and the resistance of the virtual resistor into formula (2) for calculation, the virtual resistor voltage of the inverter is obtained; The formula (2) includes: Among them, G df (s) represents the virtual resistance voltage; C f Indicates the capacitance of the filter capacitor; Represents differential voltage; R v Indicates the virtual resistance value.

5. The method according to claim 1, characterized in that The step of determining a target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount comprises: Superimposing the virtual resistance voltage and the initial three-phase modulation amount to obtain a target three-phase modulation amount; The target three-phase modulation amount is pulse-width modulated to obtain a target pulse-width modulation signal.

6. The method according to claim 1, characterized in that Get the initial three-phase modulation quantity, including: Obtaining a DC voltage reference value, a DC voltage measured value, a q-axis current reference value, and a grid voltage phase angle of the inverter; Integrate the difference between the DC voltage measured value and the DC voltage reference value to obtain a d-axis current reference value; Determining a d-axis voltage modulation amount according to the d-axis current reference value; Determining a q-axis voltage modulation amount according to the q-axis current reference value; Based on the grid voltage phase angle, coordinate transformation is performed on the d-axis voltage modulation amount and the q-axis voltage modulation amount to obtain an initial three-phase modulation amount.

7. The method according to claim 6, characterized in that The step of determining the d-axis voltage modulation amount according to the d-axis current reference value comprises: Obtaining the grid voltage feedforward coefficient, as well as the actual measured value of the d-axis current and the actual measured value of the d-axis voltage of the inverter; Integrate the difference between the d-axis current reference value and the d-axis current measured value to obtain a d-axis current intermediate value; The product of the measured value of the d-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the d-axis current are determined as the d-axis voltage modulation amount; And / or, determining the q-axis voltage modulation amount according to the q-axis current reference value includes: Obtaining the grid voltage feedforward coefficient, as well as the actual measured values ​​of the q-axis current and q-axis voltage of the inverter; Integrate the difference between the q-axis current reference value and the q-axis current measured value to obtain a q-axis current intermediate value; The product of the actual measured value of the q-axis voltage and the grid voltage feedforward coefficient and the sum of the intermediate value of the q-axis current are determined as the q-axis voltage modulation amount.

8. An impedance reshaping device for an inverter, characterized in that: include: An acquisition module, used for acquiring a first filter capacitor voltage and an initial three-phase modulation amount of the inverter; A calculation module, connected to the acquisition module, and configured to perform a differential operation on the first filter capacitor voltage to obtain a differential voltage; A first determination module, connected to the calculation module, for determining a virtual resistance voltage of the inverter according to the differential voltage; The second determination module is connected to the first determination module and is used to determine a target pulse width modulation signal according to the virtual resistance voltage and the initial three-phase modulation amount. The target pulse width modulation signal is used to control the on and off of the three-phase converter bridge to achieve impedance reshaping of the inverter.

9. The device according to claim 8, characterized in that The device also includes: A filtering module, used for performing low-pass filtering on the first filter capacitor voltage to obtain a second filter capacitor voltage; The operation module is specifically used for: A differential operation is performed on the second filter capacitor voltage to obtain a differential voltage.

10. The device according to claim 9, characterized in that The filtering module is specifically used for: Using a first-order Butterworth low-pass filter to low-pass filter the first filter capacitor voltage to obtain a second filter capacitor voltage; Among them, the transfer function of the first-order Butterworth low-pass filter satisfies formula (1); The formula (1) includes: Among them, G LPF (s) represents the transfer function of the first-order Butterworth low-pass filter; T LPF represents the time constant of the first-order Butterworth low-pass filter; s represents the frequency operation factor.

11. An inverter, comprising an inverter body, characterized in that: It also includes the impedance reshaping device according to any one of claims 8 to 10.

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