Control method of two-way wireless charging system based on T-type three levels

Abstract

The invention provides a control method of a two-way wireless charging system based on T-type three levels. The control method is used for solving the problems that in an existing control method of the two-way wireless charging system, redundant switch modes are multiple, and control is complex. The method comprises the following specific steps: setting a direct-current voltage reference value, subtracting the direct-current voltage reference value from an acquired direct-current voltage value, obtaining a current reference value through a PI regulator, and acquiring a three-phase current value and a three-phase voltage value at a power grid side; performing adc / dq conversion on the three-phase voltage value and the current value to respectively output a disturbance variable and a current output variable, and inputting the shaft disturbance variable into a phase-locked loop device to perform phase detection; inputting the current output variable and the current reference value into a current loop decoupling device for decoupling, and outputting a voltage reference value; and carrying out adc / dq reverse conversion on the voltage reference value, outputting a three-phase voltage reference value and carrying out SVPWM (Space Vector Pulse Width Modulation). According to the control method, switching between the charging mode and the grid-connected mode is achieved, and the grid-connected current harmonic wave in the grid-connected mode meets the grid-connected requirement.

Description

Technical Field

[0001] The present invention relates to the field of wireless power transmission, and in particular to a control method for a T-type three-level bidirectional wireless charging system. Background Art

[0002] Traditional wireless power transmission (WPT) technology mainly focuses on one-way energy transfer, but with the diversification of application scenarios, bidirectional wireless power transmission (BWPT) technology has become particularly important because it can achieve flexible energy exchange between devices. In recent years, WPT technology has developed rapidly, and its application has expanded to many fields such as electric vehicles (EVs), medical equipment and underwater equipment.

[0003] As the number of EVs grows rapidly, the demand for charging increases, which poses new challenges to the power grid, which may cause the distribution network to be overloaded and even affect the stability of the power grid. EVs grid-connected technology supports both power supply from the grid to EVs (G2V) and power supply from EVs to the grid (V2G) by realizing two-way wireless power flow between the grid and EVs. This technology enables on-board batteries to provide auxiliary power during peak grid load periods and absorb excess power during valley periods, thereby improving the convenience of charging, reducing the burden on the grid, improving the quality of the grid, and enhancing the stability of the grid.

[0004] A typical BWPT system includes an on-board battery, a bidirectional converter, a resonant network, a coupling device, a bidirectional converter, and a power grid. In the case of bidirectional power transmission between the power grid and the DC battery energy storage system, the power grid is usually regarded as the primary side of the system, and the DC battery energy storage system is the secondary side. When the system performs forward power transmission, energy flows from the grid to the on-board battery; in the reverse transmission mode, energy flows back from the on-board battery to the grid.

[0005] The bidirectional converter connecting the grid and the resonant network is a key component of the system, responsible for the bidirectional flow of energy and grid-connected operation. Given its direct connection to the grid, it must meet strict harmonic and synchronization standards. The three-level circuit is an ideal choice due to its low harmonic content, high-quality output voltage waveform and low electromagnetic interference. In the document “Fault-tolerant control of three-level NPC grid-connected inverter based on SVPWM compensation optimization”, an optimized compensation type low common-mode voltage fault-tolerant control strategy based on space vector pulse width modulation is proposed, which can realize the stable and reliable operation of the grid-connected system after the single-phase bridge arm failure of the three-level NPC grid-connected inverter. The NPC circuit has the advantage of low switch tube loss, but its loss distribution is uneven; in the document “Circulation suppression method of ANPC type three-level grid-connected inverter based on new VLMZVM”, an active clamping type (ANPC) three-level circuit is adopted to establish a zero-sequence circulating current model of multi-machine ANPC type three-level grid-connected inverter. By detecting the zero-sequence circulating current state, the virtual large vector or actual large vector synthesis method is selected in real time to suppress the zero-sequence circulating current problem. The switch tube of the ANPC circuit has low withstand voltage and low loss, but it has many switching devices and redundant switching modes, and the control is relatively complex. Summary of the invention

[0006] The purpose of the present invention is to provide a control method for a bidirectional wireless charging system based on a T-type three-level circuit, so as to solve the technical problem of complex multi-control of redundant switch modes in the existing bidirectional wireless charging system control method.

[0007] A control method based on a T-type three-level bidirectional wireless charging system, the bidirectional wireless charging system comprising a bidirectional wireless charging circuit and a T-type three-level bidirectional AC-DC converter, the control method comprising a T-type three-level bidirectional AC-DC converter control method, the specific steps being:

[0008] S1.1 sets the DC voltage reference value, which is subtracted from the collected DC voltage value and passed through the PI regulator to obtain the d-axis current reference value, and collects the three-phase current value and three-phase voltage value on the grid side;

[0009] S1.2 performs adc / dq conversion on the three-phase voltage values ​​to output d-axis disturbance variables and q-axis disturbance variables, inputs the d-axis and q-axis disturbance variables into a phase-locked loop device for phase detection, and outputs the actual voltage vector angle;

[0010] S1.3 performs adc / dq conversion on the three-phase current values ​​to output d-axis current output variables and q-axis current output variables, inputs the d-axis and q-axis current output variables and the d-axis and q-axis current reference values ​​into the current loop decoupling device for decoupling, and outputs d-axis voltage reference values ​​and q-axis voltage reference values;

[0011] S1.4 performs adc / dq inverse conversion on the d-axis and q-axis voltage reference values ​​and outputs three-phase voltage reference values;

[0012] S1.5 inputs the three-phase voltage reference value into the SVPWM modulation module and obtains the driving signal through calculation.

[0013] Optionally, the specific steps in step S1.2 are:

[0014] S1.2.1 performs adc / dq conversion on the three-phase voltage values ​​to output d-axis disturbance variables and q-axis disturbance variables;

[0015] S1.2.2 Subtract the q-axis disturbance variable from the q-axis disturbance variable reference value and input the result into the PI regulator, and perform an integral operation on the output of the PI regulator;

[0016] S1.2.3 introduces the parameter 2π and performs MOD operation on the result of the integral operation in step S1.2.2 to output the actual voltage vector angle.

[0017] Optionally, the specific steps of step S1.3 are:

[0018] S1.3.1 performs adc / dq conversion on the three-phase current values ​​to output d-axis current output variables and q-axis current output variables;

[0019] S1.3.2 subtracts the d-axis and q-axis current output variables from the d-axis and q-axis current reference values ​​respectively and passes through the PI regulator;

[0020] S1.3.3 introduces d-axis and q-axis disturbance variables and d-axis and q-axis voltage feedforward quantities to eliminate the coupling quantity of adc / dq conversion, and outputs d-axis voltage reference value and q-axis voltage reference value.

[0021] Optionally, the adc / dq conversion includes Clark transform and Park transform.

[0022] Optionally, the phase angles required for the Park transformation in step S1.2 and the Park inverse transformation in step S1.4 are both output by a phase-locked loop device.

[0023] Optionally, the control method further includes a bidirectional wireless charging system control method, and the specific steps are:

[0024] S2.1 Based on Kirchhoff’s voltage law, write the circuit equations for the primary and secondary sides of the bidirectional wireless charging system;

[0025] S2.2 Calculate the resonant current of the primary and secondary sides of the bidirectional wireless charging system when the system is in a full resonant state based on the reactance of the primary and secondary resonant networks of the bidirectional wireless charging system;

[0026] S2.3 According to the resonant current of the primary and secondary sides of the bidirectional wireless charging system and the symmetrical phase shift modulation strategy, a functional relationship between the system transmission power and the external phase shift angle of the system and the internal phase shift angle of the primary and secondary sides of the system is constructed;

[0027] S2.4 controls the system output power by adjusting the inner phase shift angle and the outer phase shift angle parameters on both sides of the system.

[0028] Optionally, the primary side of the bidirectional wireless charging system includes a DC power supply, a first inverter, a primary side compensation network and a primary side coil connected in sequence;

[0029] The secondary side of the bidirectional wireless charging system includes a secondary side coil, a secondary side compensation network and a second inverter which are connected in sequence, and the other end of the second inverter is connected to a T-type three-level bidirectional AC-DC converter.

[0030] Optionally, the primary compensation network and the secondary compensation network are SS network topology structures, the primary compensation network includes a primary compensation capacitor, and the secondary compensation network includes a secondary compensation capacitor.

[0031] Optionally, the specific method in step S2.2 is:

[0032] S2.2.1 Write the primary reactance according to the system frequency, the capacitance value of the primary compensation capacitor and the self-inductance value of the primary coil, and write the secondary reactance according to the system frequency, the capacitance value of the secondary compensation capacitor and the self-inductance value of the secondary coil;

[0033] S2.2.2 When the primary reactance and the secondary reactance are both zero and the system is in a fully resonant state, a functional relationship is established between the primary resonant current and the primary voltage, the system frequency and the mutual inductance of the primary and secondary coils, and a functional relationship is established between the secondary resonant current and the secondary voltage, the system frequency and the mutual inductance of the primary and secondary coils.

[0034] Optionally, the specific steps of constructing the functional relationship between the system transmission power and the inner phase shift angle and the outer phase shift angle on both sides of the system in step S2.3 are:

[0035] S2.3.1 Analyze the functional relationship between the primary resonant current and the secondary resonant current. It can be seen that the primary resonant current lags the secondary excitation voltage by 90°, and the secondary resonant current also lags the primary excitation voltage by 90°.

[0036] S2.3.2 Construct the functional relationship between the system transmission power and the effective value of the fundamental component of the primary side excitation voltage, the effective value of the fundamental component of the secondary side excitation voltage and the external shift phase angle of the system;

[0037] S2.3.3 Construct the functional relationship between the effective value of the fundamental component of the primary excitation voltage and the phase shift angle within the primary side, and the functional relationship between the effective value of the fundamental component of the secondary excitation voltage and the phase shift angle within the secondary side when the symmetrical phase shift modulation strategy is adopted;

[0038] S2.3.4 constructs the functional relationship between the system transmission power and the system external phase shift angle and the system primary and secondary internal phase shift angle through the functional relationship constructed in steps S2.3.2 and S2.3.3.

[0039] Due to the adoption of the above technical solution, the present invention has the following advantages:

[0040] The control method of the present application realizes the switching between charging and grid-connected modes, and the grid-connected current harmonics in the grid-connected mode meet the grid-connected requirements. The T-type three-level bidirectional AC-DC converter adopted in the present application has fewer switching devices and low conduction loss.

[0041] Other advantages, objectives and features of the present invention will be described in the following description to some extent, and to some extent, will be obvious to those skilled in the art based on the following examination and study, or can be taught from the practice of the present invention. The objectives and other advantages of the present invention can be realized and obtained through the following description. BRIEF DESCRIPTION OF THE DRAWINGS

[0042] The accompanying drawings of the present invention are as follows.

[0043] Figure 1 The circuit diagram of the T-type three-level bidirectional wireless charging system of the present invention is shown in FIG.

[0044] Figure 2 1 is a circuit diagram of a T-type three-level bidirectional AC-DC converter according to the present invention.

[0045] Figure 3 The figure is a flow chart of a control method of a T-type three-level bidirectional AC-DC converter according to the present invention.

[0046] Figure 4 It is the control structure principle diagram of SSRF-SPLL of the present invention.

[0047] Figure 5 This is a schematic diagram of the current loop decoupling control principle of the present invention.

[0048] Figure 6 This is the SSRF-SPLL vector diagram of the present invention.

[0049] Figure 7 It is a control equivalent circuit diagram of the three-phase T-type three-level AC-DC converter of the present invention.

[0050] Figure 8 This is a diagram showing the voltage and current phase tracking effect of a single-phase power grid in the charging mode of the present invention.

[0051] Fig. 9 This is the voltage and current diagram of the primary and secondary sides of the wireless charging system in the charging mode of the present invention.

[0052] Fig.10 This is a bus voltage waveform diagram of the charging mode of the present invention.

[0053] Fig.11 This is the output waveform diagram of the current loop in the charging mode of the present invention.

[0054] Fig.12 This is the SVPWM modulation output waveform diagram of the charging mode of the present invention.

[0055] Fig.13 This is a harmonic analysis diagram of the grid-side current in the charging mode of the present invention.

[0056] Fig.14 This is a diagram showing the voltage and current phase tracking effect of a single-phase power grid in the grid-connected mode of the present invention.

[0057] Fig.15 This is the voltage and current diagram of the primary and secondary sides of the grid-connected mode wireless charging system of the present invention.

[0058] Fig.16 This is a bus voltage waveform diagram of the grid-connected mode of the present invention.

[0059] Fig.17 This is the output waveform diagram of the current loop in the grid-connected mode of the present invention.

[0060] Fig.18 This is the SVPWM modulation output waveform diagram of the grid-connected mode of the present invention.

[0061] Fig.19 This is a harmonic analysis diagram of the grid-side current in the grid-connected mode of the present invention. DETAILED DESCRIPTION

[0062] The present invention will be further described below in conjunction with the accompanying drawings and embodiments.

[0063] Embodiment 1:

[0064] like Figure 1 A T-type three-level bidirectional wireless charging system is shown, comprising a bidirectional wireless charging circuit and a T-type three-level bidirectional AC-DC converter. Figure 1 As shown, the bidirectional wireless charging circuit includes a primary circuit and a secondary circuit, and the primary circuit includes a DC power supply V connected in sequence. P , primary inverter, primary compensation network and primary coil L P The secondary circuit includes a secondary coil L connected in sequence S , a secondary side compensation network and a secondary side inverter, wherein the output end of the secondary side inverter is connected to one end of a T-type three-level bidirectional AC-DC converter.

[0065] In this embodiment, the primary inverter includes four switch tubes S1-S4, and the primary compensation network includes a primary compensation capacitor C PThe secondary inverter includes 4 switch tubes T1-T4, and the secondary compensation network includes the secondary compensation capacitor C S , secondary coil L S and the secondary coil L S Constitute a coupling mechanism.

[0066] like Figure 1 and Figure 2 As shown, the T-type three-level bidirectional AC-DC converter is a three-phase circuit, each phase circuit includes a bridge arm subcircuit and an intermediate clamp subcircuit, and the three-phase circuits are defined as phase a, phase b and phase c. The bridge arm subcircuit of phase a includes a switch tube S arranged in series in the same direction. A1 And switch tube S A4 The intermediate clamping subcircuit of phase a includes S A2 And switch tube S A3 The bridge arm subcircuit of phase a intersects with the middle clamp subcircuit at point A. The bridge arm subcircuit of phase b includes switch tubes S arranged in series in the same direction. B1 And switch tube S B4 The intermediate clamping subcircuit of phase b includes S B2 And switch tube S B3 The bridge arm subcircuit of phase b intersects with the middle clamp subcircuit at point B. The bridge arm subcircuit of phase c includes switch tubes S arranged in series in the same direction. C1 And switch tube S C4 The intermediate clamping subcircuit of phase a includes S C2 And switch tube S C3 ; The bridge arm sub-circuit of phase c intersects with the middle clamp sub-circuit at point C.

[0067] like Figure 1 and Figure 2 As shown, the T-type three-level bidirectional AC-DC converter also includes a bus capacitor C d1 and C d2 , bus capacitance C d1 and C d2 After being connected in series, it is connected in parallel with the three-phase bridge arm sub-circuit and the middle clamp sub-circuit. The other end of each three-phase bridge arm sub-circuit and the middle clamp sub-circuit is provided with an LCL filter circuit. The LCL filter circuit includes filter inductors L1 and L2, and filter capacitors C f The other ends of the three LCL filter circuits are connected to the three-phase e a 、e b and e c connect.

[0068] Embodiment 2:

[0069] A control method for a T-type three-level bidirectional wireless charging system is used to control the T-type three-level bidirectional wireless charging system described in Example 1, and the control method includes controlling a T-type three-level bidirectional AC-DC converter and controlling a bidirectional wireless charging circuit.

[0070] like Figure 3 As shown, the T-type three-level bidirectional AC-DC converter control method has the following specific steps:

[0071] S1.1 sets the DC voltage reference value, which is subtracted from the collected DC voltage value and passed through the PI regulator to obtain the d-axis current reference value, and collects the three-phase current value and three-phase voltage value on the grid side;

[0072] In this embodiment, the DC voltage reference value is The collected DC voltage value is u dc , the d-axis current reference value is

[0073] S1.2 As Figure 4 As shown, the three-phase voltage value is converted into ADC / DQ to output the d-axis disturbance variable and the q-axis disturbance variable, the d-axis and q-axis disturbance variables are input into the phase-locked loop device for phase detection, and the actual voltage vector angle is output; the specific steps are as follows:

[0074] S1.2.1 performs adc / dq conversion on the three-phase voltage values ​​to output d-axis disturbance variables and q-axis disturbance variables;

[0075] S1.2.2 Subtract the q-axis disturbance variable from the q-axis disturbance variable reference value and input the result into the PI regulator, and perform an integral operation on the output of the PI regulator;

[0076] S1.2.3 introduces the parameter 2π and performs MOD operation on the result of the integral operation in step S1.2.2 to output the actual voltage vector angle.

[0077] In this embodiment, adc / dq conversion includes Clark transformation and Park transformation. The three-phase voltages of the AC power grid are u ga 、u gb and u gc , the d-axis disturbance variable and the q-axis disturbance variable are u gd and u gq .

[0078] In this embodiment, the function of the phase-locked loop device is to lock the phase of the measured signal. In the grid-connected inverter control, it is necessary to dynamically obtain the phase information of the grid voltage in order to facilitate the control of the grid-connected active and reactive power (current). Therefore, it is necessary to use a phase-locked loop device to phase-lock the grid voltage phase. There are different classification methods for phase-locked loop devices, such as hardware phase-locked loop, software phase-locked loop, open-loop phase-locked loop, closed-loop phase-locked loop, single-phase phase-locked loop, three-phase phase-locked loop, etc. This application adopts a three-phase software phase-locked loop, and uses the synchronous locking of the rotating coordinate system to realize the control of the three-phase phase-locked loop single synchronous coordinate system software phase-locked loop (SSRF-SPLL).

[0079] like Figure 5 As shown in the figure, when the power grid is balanced, the actual voltage vector U in the two-phase stationary αβ coordinate system and the synchronous dq coordinate system is S and the phase-locked loop output voltage vector U SPLL The location of Figure 6 The angle of the actual voltage vector is θ, and the angle of the voltage vector output by the phase-locked loop device is θ′.

[0080] When the phase-locked loop is in precise phase locking, the voltage vector U SPLL The actual voltage vector U S Should be completely overlapped, that is, θ = θ′. At the moment of grid voltage phase mutation, vector U SPLL with U S The positions will be staggered, and appropriate closed-loop control must be adopted to make the output of the phase-locked loop device satisfy θ = θ'. The grid voltage undergoes Clark transformation and Park transformation, where the angle required for Park transformation is provided by the phase angle of the phase-locked loop output, θ' is the phase angle of the phase-locked loop output, and u gd and u gq is the vector quantity U SPLL The output of the PI controller can be used as the angular frequency ω, the integrator output ωt, and the MOD modulus obtains the phase value between 0 and 2π. When θ≠θ′, u gq ≠0, under the action of PI controller, u gd will tend to 0, so that θ≠θ′ and successful phase locking is achieved.

[0081] S1.3 performs adc / dq conversion on the three-phase current values ​​to output d-axis current output variables and q-axis current output variables, inputs the d-axis and q-axis current output variables and the d-axis and q-axis current reference values ​​into the current loop decoupling device for decoupling, and outputs the d-axis voltage reference value and the q-axis voltage reference value; the specific steps are:

[0082] S1.3.1 performs adc / dq conversion on the three-phase current values ​​to output d-axis current output variables and q-axis current output variables;

[0083] S1.3.2 subtracts the d-axis and q-axis current output variables from the d-axis and q-axis current reference values ​​respectively and passes through the PI regulator;

[0084] S1.3.3 introduces d-axis and q-axis disturbance variables and d-axis and q-axis voltage feedforward quantities to eliminate the coupling quantity of adc / dq conversion, and outputs d-axis voltage reference value and q-axis voltage reference value.

[0085] In this embodiment, if Figure 6 As shown, the adc / dq conversion includes Clark transformation and Park transformation, and the d-axis current output variable and the q-axis current output variable are i d and i q , the d-axis and q-axis current reference values ​​are and The d-axis and q-axis voltage feedforward values ​​are ω0L, and the d-axis voltage reference value and the q-axis voltage reference value are respectively and

[0086] In this embodiment, since the inductively coupled power transmission system is based on a resonant network of the SS topology, it outputs a constant current source and becomes a voltage source after passing through the bus capacitor. However, if the current provided by the current source does not match the current of the input inverter, the capacitor will be fed or charged, causing the voltage on the capacitor to change. Therefore, the voltage on the bus capacitor must be kept constant through control, so that the current provided by the current source is equal to the current on the input inverter side.

[0087] For the convenience of research, the d-axis of the synchronous rotating coordinate system can be made consistent with the direction of the grid voltage vector, and the value is the modulus of the grid voltage vector, which is zero. This processing method is called grid voltage vector orientation. The dual-loop control of the grid-connected inverter based on voltage vector orientation adopts a dual-loop control structure of a DC voltage outer loop and an active and reactive current inner loop, in which the current loop adopts current control based on feedforward decoupling, and the voltage loop refers to the control of the bus voltage.

[0088] Based on the grid voltage vector orientation, the instantaneous active and reactive power of the system is:

[0089]

[0090] It can be seen from the power expression that when there is no fluctuation in the grid voltage, the instantaneous active and reactive power of the grid-connected inverter is only related to the dq-axis component i of the grid-connected inverter current. d and i q The active power on the DC side of the grid-connected inverter is equal to I dc U dc , ignoring the inverter loss: I dc U dc =3 / 2ugd i d , when the grid voltage remains unchanged, the DC side voltage of the inverter U dc and grid-connected active power p and i d Proportional.

[0091] In this embodiment, for a three-phase T-type three-level AC-DC converter, the inverter is taken as an example for analysis. The control difference between rectification and inverter is only the direction of the inverter output current. For control, the control method of the three-level is the same as that of the two-level, so the circuit model diagram is as follows: Figure 7 As shown, the main circuit consists of an L filter, a three-phase full-bridge converter, and a bus capacitor. Figure 7 The middle is an IGBT module with an anti-parallel diode; U dc is the DC bus voltage; i k (k=a,b,c) is the inductor current; U k , Ug k (k=a, b, c) are the voltage on the inverter bridge arm side and the grid voltage respectively; L represents the filter inductance. When connected to the grid, L or LCL filters are usually used. When controlling, LCL filters are usually equivalent to L filters for modeling and control.

[0092] According to Kirchhoff's law, the inverter phase a loop equation can be obtained:

[0093]

[0094] Use S a To represent the switching function of phase a, then u a =u aN +u a0 =S a *1 / 2u aN +u a0 The mathematical model based on the abc three-phase stationary coordinate system 1 has a clear physical meaning, but the AC side of the model is all time-varying AC quantities. Although it is easy to understand, it is not conducive to the design of the control system. Therefore, the mathematical model in the synchronous rotating dq coordinate system is usually obtained through the rotating coordinate transformation to study the control of the inverter. The voltage equation in the three-phase stationary coordinate system is transformed to the rotating coordinate system:

[0095]

[0096] In the formula, u d and u q are the d-axis and q-axis components of the inverter output voltage, respectively. d and i q are the d-axis and q-axis components of the inverter output current, u gd and u gqThe d and q axis components of the grid voltage, ω0 is the angular frequency of the grid voltage,

[0097] In this embodiment, the rotating coordinate system rotates synchronously with the angular frequency of the grid voltage. The rotation 3 / 2 transformation introduces strong coupling between the d-axis and q-axis of the system. In addition to being affected by the control quantity, the system dq-axis output current is also affected by the coupling voltage, the output voltage, and the disturbance. If these disturbances are included in the control system, their influence can be eliminated. Since the disturbance coupling term is decoupled by feedforward compensation, it is a feedforward decoupling control. Thus, the current loop control strategy based on feedforward decoupling can be obtained. Figure 6 As shown, a capacitor voltage feedforward is introduced into the inner loop of the inductor current to compensate the inductor current.

[0098] S1.4 performs adc / dq inverse conversion on the d-axis and q-axis voltage reference values ​​and outputs three-phase voltage reference values.

[0099] In this embodiment, the adc / dq inverse conversion includes Clark inverse transformation and Park inverse transformation, and the phase angles required for Park inverse transformation are output by the phase-locked loop device. The three-phase voltage reference values ​​are respectively and

[0100] S1.5 inputs the three-phase voltage reference value into the SVPWM modulation module and obtains the driving signal through calculation.

[0101] In this embodiment, the SVPWM modulation module calculates the drive signal S a , S b , S c They are used to control the three phases of a T-type three-level bidirectional AC-DC converter respectively.

[0102] The S2 control method also includes a bidirectional wireless charging system control method, and the specific steps are:

[0103] S2.1 Based on Kirchhoff’s voltage law, write the circuit equations for the primary and secondary sides of the bidirectional wireless charging system;

[0104] In this embodiment, when the operating frequency of the BIPT system is the resonant frequency f, the system is in a full resonant state, f r It can be expressed as:

[0105]

[0106] Among them, ω r is the resonant angular frequency of the system.

[0107] The circuit model of the SS-BIPT system is analyzed by analytical method. According to Kirchhoff's voltage law, the circuit equations of the primary and secondary sides of the system can be written as:

[0108]

[0109] in, u p 、u s 、i p 、i s The phase expression form of .

[0110] S2.2 According to the reactance of the primary and secondary resonant networks of the bidirectional wireless charging system, the resonant current of the primary and secondary sides of the bidirectional wireless charging system is calculated when the system is in a full resonant state; the specific method is:

[0111] S2.2.1 Write the primary reactance according to the system frequency, the capacitance value of the primary compensation capacitor and the self-inductance value of the primary coil, and write the secondary reactance according to the system frequency, the capacitance value of the secondary compensation capacitor and the self-inductance value of the secondary coil;

[0112] S2.2.2 When the primary reactance and the secondary reactance are both zero and the system is in a fully resonant state, a functional relationship is established between the primary resonant current and the primary voltage, the system frequency and the mutual inductance of the primary and secondary coils, and a functional relationship is established between the secondary resonant current and the secondary voltage, the system frequency and the mutual inductance of the primary and secondary coils.

[0113] In this embodiment, the reactance X of the primary and secondary resonant networks of the system is p , X s They are:

[0114]

[0115] When the system is in full resonance, X p =X s = 0. At the same time, due to the equivalent loss resistance r of the resonant network p and r s is very small, so solving the equation yields:

[0116]

[0117] From the above formula, we can know that the primary resonant current i in the SS-BIPT system is p Lagging secondary excitation voltage u s 90°, secondary resonant current i s It also lags behind the primary excitation voltage u p 90°.

[0118] S2.3 According to the resonant current of the primary and secondary sides of the bidirectional wireless charging system and the symmetrical phase shift modulation strategy, a functional relationship between the system transmission power and the system external phase shift angle and the system primary and secondary side internal phase shift angle is constructed; the specific steps are:

[0119] S2.3.1 Analyze the functional relationship between the primary resonant current and the secondary resonant current. It can be seen that the primary resonant current lags the secondary excitation voltage by 90°, and the secondary resonant current also lags the primary excitation voltage by 90°.

[0120] S2.3.2 Construct the functional relationship between the system transmission power and the effective value of the fundamental component of the primary side excitation voltage, the effective value of the fundamental component of the secondary side excitation voltage and the external shift phase angle of the system;

[0121] S2.3.3 Construct the functional relationship between the effective value of the fundamental component of the primary excitation voltage and the phase shift angle within the primary side, and the functional relationship between the effective value of the fundamental component of the secondary excitation voltage and the phase shift angle within the secondary side when the symmetrical phase shift modulation strategy is adopted;

[0122] S2.3.4 constructs the functional relationship between the system transmission power and the system external phase shift angle and the system primary and secondary internal phase shift angle through the functional relationship constructed in steps S2.3.2 and S2.3.3.

[0123] In this embodiment, the transmission power P of the above analysis system can be derived as follows:

[0124]

[0125] Among them, u pf 、u sf They are square wave voltage u p 、u s The fundamental component, |u pf |、|u sf | are the effective values ​​of the two respectively, δ is the external phase shift angle of the system, that is, the secondary side square wave voltage u s Leading primary square wave voltage u p The phase angle of .

[0126] When the symmetrical phase shift modulation strategy is adopted, |u pf |、|u sf |It can be derived as:

[0127]

[0128] Among them, β p is the internal phase shift angle of the primary side of the system, that is, the square wave voltage u p The pulse width angle, β s is the internal phase shift angle of the secondary side of the system, that is, the square wave voltage u s The pulse width angle.

[0129] Therefore, the transmission power P of the system can be further expressed as:

[0130]

[0131] S2.4 controls the system output power by adjusting the inner phase shift angle and the outer phase shift angle parameters on both sides of the system.

[0132] In this embodiment, it can be seen from the above analysis that the internal phase shift angle β on both sides of the system can be adjusted. p , β s And the external phase shift angle δ is used to adjust the transmission power of the system. When the external phase shift angle δ∈(0°, 180°), sinδ is positive, the transmission power P is positive, and the energy is transmitted from the primary side of the system to the secondary side; when the external phase shift angle δ∈(-180°, 0°), sinδ is negative, the transmission power P is negative, and the energy is transmitted from the secondary side of the system to the primary side. In this way, the positive and negative value of δ determines the direction of energy flow in the BIPT system. When β p =β s When |δ|=0 and |δ|=90°, the system transmission power is maximum.

[0133] S3 simulation verification: passed Figure 1 Build simulation models, Figure 1 On this basis, an LC filter circuit is added between the primary high-frequency inverter and the DC power supply and between the secondary high-frequency inverter and the T-type three-level bidirectional AC-DC converter. The LC filter circuit is connected in parallel with the primary high-frequency inverter or the secondary high-frequency inverter. The T-type three-level bidirectional AC-DC converter model for the wireless charging and discharging device is built using MATLAB / simulink software, and the grid-side AC-DC converter adopts a three-phase T-type three-level converter. The vehicle battery terminal voltage is set to 700V, and the bus voltage reference values ​​of the charging mode and the grid-connected mode are 820V and 630V, respectively.

[0134] S3.1 simulates the charging mode and verifies the waveform of the above control strategy. Figure 8 The waveform of the voltage and current on the a-phase grid side in charging mode is shown in Figure 1. The grid phase is set to 60°. Figure 8 It can be seen that when the power is turned on, the current first fluctuates, and then stabilizes at a phase difference of 180° with the grid voltage. At this time, the grid charges the on-board battery. Fig. 9 The voltage and current of the primary and secondary sides of the wireless charging system are displayed. At this time, the secondary side voltage phase lags the primary side voltage phase by 90°, so the secondary side transmits electrical energy to the primary side.

[0135] Then look at the waveform output by the control strategy mentioned above. Fig.10 The figure shows the bus voltage under the control of the bus voltage loop. The reference value is set to 820V. It can be seen that the bus voltage stabilizes at 820V after the initial fluctuation when powered on. Fig.11 u is the output of the current loop d and u qAnd multiply the voltage waveform by the bridge arm gain, and then perform the inverse coordinate transformation to perform SVPWM modulation. Fig.12 is the output waveform after SVPWM modulation. When SPWM modulation is used, the waveform is a sine wave, while the output after SVPWM modulation is a ram horn wave. Fig.13 This is the data obtained after FFT harmonic analysis of the grid-side current. It can be seen from the figure that the harmonic of the grid-side current is 1.06%, which is a very small value.

[0136] S3.2 simulates the grid-connected mode and verifies the waveform of the control strategy. Fig.14 The waveform of the voltage and current on the grid side of phase a in the grid-connected mode is shown in Figure 1. The phase of the grid is set to 60°. Fig.14 It can be seen that when the power is turned on, the current first fluctuates, and then stabilizes at a phase difference of 0° with the grid voltage. At this time, the on-board battery energy is fed back to the grid. Fig.15 The voltage and current of the primary and secondary sides of the wireless charging system are displayed. At this time, the secondary side voltage phase leads the primary side voltage phase by 90°, so the primary side transmits electrical energy to the secondary side.

[0137] Then look at the waveform output by the control strategy mentioned above. Fig.16 The figure shows the bus voltage under the control of the bus voltage loop. The reference value is set to 630V. It can be seen that the bus voltage stabilizes at 630V after the initial fluctuation at power-on. Fig.17 u is the output of the current loop d and u q And multiply the voltage waveform by the bridge arm gain, and then perform the inverse coordinate transformation to perform SVPWM modulation. Fig.18 This is the output waveform after SVPWM modulation. The modulation wave at this time is somewhat different from the modulation wave in the charging mode and is more inclined to the saddle wave. This is because the modulation ratio increases. Fig.19 This is the data obtained after FFT harmonic analysis of the grid-side current. It can be seen from the figure that the harmonics of the grid-side current is 3.53% at this time, which meets the national standard requirement that the harmonics of the grid-connected circuit should be less than 5%.

[0138] In summary, this application studies the three-phase T-type three-level AC-DC converter for bidirectional wireless charging and discharging devices, and analyzes the control strategy for the entire wireless charging and discharging device, including the SVPWM modulation, current loop control, bus voltage loop control, and phase-locked loop control of the three-phase T-type three-level AC-DC converter and the primary-secondary switching control strategy of the wireless charging system. The waveforms of the above control strategies are analyzed and verified through simulation. The system can switch between charging mode and grid-connected mode well, and the grid-connected current harmonics in the grid-connected mode meet the grid-connected requirements.

[0139] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention rather than to limit it. Although the present invention has been described in detail with reference to the above embodiments, ordinary technicians in the relevant field should understand that the specific implementation methods of the present invention can still be modified or replaced by equivalents. Any modification or equivalent replacement that does not depart from the spirit and scope of the present invention should be covered within the scope of protection of the claims of the present invention.

Claims

1. A control method based on a T-type three-level bidirectional wireless charging system, wherein the bidirectional wireless charging system comprises a bidirectional wireless charging circuit and a T-type three-level bidirectional AC-DC converter, characterized in that: The control method includes a T-type three-level bidirectional AC-DC converter control method, and the specific steps are: S1.1 sets the DC voltage reference value, which is subtracted from the collected DC voltage value and passed through the PI regulator to obtain the d-axis current reference value, and collects the three-phase current value and three-phase voltage value on the grid side; S1.2 performs adc / dq conversion on the voltage value to output d-axis disturbance variables and q-axis disturbance variables, inputs the d-axis and q-axis disturbance variables into a phase-locked loop device for phase detection, and outputs the actual voltage vector angle; S1.3 performs adc / dq conversion on the three-phase current values ​​to output d-axis current output variables and q-axis current output variables, inputs the d-axis and q-axis current output variables and the d-axis and q-axis current reference values ​​into the current loop decoupling device for decoupling, and outputs d-axis voltage reference values ​​and q-axis voltage reference values; S1.4 performs adc / dq inverse conversion on the d-axis and q-axis voltage reference values ​​and outputs three-phase voltage reference values; S1.5 inputs the three-phase voltage reference value into the SVPWM modulation module and obtains the driving signal through calculation.

2. The control method of a T-type three-level bidirectional wireless charging system according to claim 1, characterized in that: The specific steps in step S1.2 are: S1.2.1 performs adc / dq conversion on the three-phase voltage values ​​to output d-axis disturbance variables and q-axis disturbance variables; S1.2.2 Subtract the q-axis disturbance variable from the q-axis disturbance variable reference value and input the result into the PI regulator, and perform an integral operation on the output of the PI regulator; S1.2.3 introduces the parameter 2π and performs MOD operation on the result of the integral operation in step S1.2.2 to output the actual voltage vector angle.

3. The control method of a T-type three-level bidirectional wireless charging system according to claim 1, characterized in that: The specific steps of step S1.3 are: S1.3.1 performs adc / dq conversion on the three-phase current values ​​to output d-axis current output variables and q-axis current output variables; S1.3.2 subtracts the d-axis and q-axis current output variables from the d-axis and q-axis current reference values ​​respectively and passes through the PI regulator; S1.3.3 introduces d-axis and q-axis disturbance variables and d-axis and q-axis voltage feedforward quantities to eliminate the coupling quantity of adc / dq conversion, and outputs d-axis voltage reference value and q-axis voltage reference value.

4. A control method based on a T-type three-level bidirectional wireless charging system according to any one of claims 1 to 3, characterized in that: The adc / dq conversion includes Clark transformation and Park transformation.

5. The control method of a T-type three-level bidirectional wireless charging system according to claim 4, characterized in that: The phase angles required for the Park transformation in step S1.2 and the Park inverse transformation in step S1.4 are both output by the phase-locked loop device.

6. The control method of a T-type three-level bidirectional wireless charging system according to claim 1, characterized in that: The control method also includes a bidirectional wireless charging system control method, and the specific steps are: S2.1 Based on Kirchhoff’s voltage law, write the circuit equations for the primary and secondary sides of the bidirectional wireless charging system; S2.2 Calculate the resonant current of the primary and secondary sides of the bidirectional wireless charging system when the system is in a full resonant state based on the reactance of the primary and secondary resonant networks of the bidirectional wireless charging system; S2.3 According to the resonant current of the primary and secondary sides of the bidirectional wireless charging system and the symmetrical phase shift modulation strategy, a functional relationship between the system transmission power and the system external phase shift angle and the system primary and secondary side internal phase shift angle is constructed; S2.4 controls the system output power by adjusting the inner phase shift angle and the outer phase shift angle parameters on both sides of the system.

7. The control method of a T-type three-level bidirectional wireless charging system according to claim 1, characterized in that: The primary side of the bidirectional wireless charging system includes a DC power supply, a first inverter, a primary side compensation network and a primary side coil connected in sequence; The secondary side of the bidirectional wireless charging system includes a secondary side coil, a secondary side compensation network and a second inverter which are connected in sequence, and the other end of the second inverter is connected to a T-type three-level bidirectional AC-DC converter.

8. The control method of a T-type three-level bidirectional wireless charging system according to claim 7, characterized in that: The primary compensation network and the secondary compensation network are SS network topology structures, the primary compensation network includes a primary compensation capacitor, and the secondary compensation network includes a secondary compensation capacitor.

9. The control method of a T-type three-level bidirectional wireless charging system according to claim 8, characterized in that: The specific method in step S2.2 is: S2.2.1 Write the primary reactance according to the system frequency, the capacitance value of the primary compensation capacitor and the self-inductance value of the primary coil, and write the secondary reactance according to the system frequency, the capacitance value of the secondary compensation capacitor and the self-inductance value of the secondary coil; S2.2.2 When the primary reactance and the secondary reactance are both zero and the system is in a fully resonant state, a functional relationship is established between the primary resonant current and the primary voltage, the system frequency and the mutual inductance of the primary and secondary coils, and a functional relationship is established between the secondary resonant current and the secondary voltage, the system frequency and the mutual inductance of the primary and secondary coils.

10. The control method of a T-type three-level bidirectional wireless charging system according to claim 6, characterized in that: The specific steps of constructing the functional relationship between the system transmission power and the inner phase shift angle and the outer phase shift angle on both sides of the system in step S2.3 are: S2.3.1 Analyze the functional relationship between the primary resonant current and the secondary resonant current. It can be seen that the primary resonant current lags the secondary excitation voltage by 90°, and the secondary resonant current also lags the primary excitation voltage by 90°. S2.3.2 Construct the functional relationship between the system transmission power and the effective value of the fundamental component of the primary side excitation voltage, the effective value of the fundamental component of the secondary side excitation voltage and the external shift phase angle of the system; S2.3.3 Construct the functional relationship between the effective value of the fundamental component of the primary excitation voltage and the phase shift angle within the primary side, and the functional relationship between the effective value of the fundamental component of the secondary excitation voltage and the phase shift angle within the secondary side when the symmetrical phase shift modulation strategy is adopted; S2.3.4 constructs the functional relationship between the system transmission power and the system external phase shift angle and the system primary and secondary internal phase shift angle through the functional relationship constructed in steps S2.3.2 and S2.3.3.

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