Bidirectional PCS converter power balance control method and device

Abstract

The present invention provides a power balance control method and device for a bidirectional PCS converter, the method comprising: dynamically compensating a PWM carrier counter according to the time difference between the interrupt timestamp sampled by an ADC and the count value of a PWM timer to obtain an optimal sampling moment of a PWM cycle; calculating a system resonant frequency according to an LCL filter network parameter, and dynamically modulating a switch delay time of a power tube according to its ratio to a switching frequency to obtain a switch delay voltage value; establishing a compensation channel according to the system resonant frequency, taking the switch delay voltage value as a feedforward compensation amount, and weightedly combining to obtain a PWM modulation signal; performing bandwidth limitation processing on a power calculation value at the optimal sampling moment according to the resonant frequency to obtain a power command value, and adjusting the PWM modulation signal accordingly to obtain a control amount of the converter charging and discharging power. The method eliminates the sampling timing deviation through dynamic compensation of a PWM carrier counter, thereby solving the system resonance problem of a bidirectional PCS converter.

Description

Technical Field

[0001] The present invention relates to the technical field of packaging and sintering, and in particular to a power balance control method and device for a bidirectional PCS converter. Background Art

[0002] The DSP-based bidirectional PCS converter is usually composed of a three-phase full-bridge topology, a DC bus capacitor, an LCL filter network and a DSP digital controller. Its core function is to realize the bidirectional energy flow between the energy storage system and the AC power grid. In power balance control, the DSP realizes dynamic adjustment through a dual closed-loop strategy of the current inner loop and the power outer loop. In the existing scheme, the PWM carrier signal is generated by the DSP timer, and the current and voltage sampling is usually based on the ADC trigger at a fixed time interval. The two have microsecond timing deviations due to factors such as hardware signal path delay and interrupt response jitter. Especially at low switching frequencies (such as 2-4kHz), such deviations will cause the sampled value to fail to accurately reflect the current average value within the PWM cycle, thereby causing the phase margin of the current loop to decrease. When the converter operates in a light load or bidirectional power frequent switching mode, the uncompensated timing error will couple with the resonant characteristics of the LCL filter to excite subharmonic oscillations, which is manifested as an increase in the low-frequency ripple of the DC bus voltage and a deterioration in the THD of the grid-connected current, affecting the reliability of the system operation. Summary of the invention

[0003] The main purpose of the present invention is to solve the technical problem that in a bidirectional PCS converter based on DSP, there is a microsecond deviation between the PWM carrier signal and the ADC sampling timing, which leads to coupling with the resonant characteristics of the LCL filter at a low switching frequency and causes subharmonic oscillation;

[0004] A first aspect of the present invention provides a bidirectional PCS converter power balance control method, the bidirectional PCS converter power balance control method comprising:

[0005] Dynamically compensating the comparison value and the period value of the PWM carrier counter according to the time difference between the sampling interrupt timestamp of the ADC sampling of the bidirectional PCS converter and the count value of the PWM timer to obtain the optimal sampling time of the PWM period;

[0006] The system resonant frequency is calculated according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and the switching delay time of the IGBT power tube in the bidirectional PCS converter is dynamically modulated according to the ratio of the system resonant frequency to the current switching frequency to obtain a compensated switching delay voltage value;

[0007] A resonance compensation channel is established according to the system resonance frequency, and the compensated switch delay voltage value is used as a feedforward compensation amount, and a PWM modulation signal is obtained through weighted combination processing;

[0008] The real-time power value obtained by performing power calculation at the optimal sampling time of the PWM period according to the system resonant frequency is subjected to bandwidth limitation processing to obtain a power command value, and the PWM modulation signal is adjusted according to the power command value to obtain a control quantity for controlling the charging and discharging power of the bidirectional PCS converter.

[0009] Optionally, in a first implementation of the first aspect of the present invention, dynamically compensating the comparison value and the period value of the PWM carrier counter according to the time difference between the sampling interrupt timestamp of the ADC sampling of the bidirectional PCS converter and the PWM timer count value to obtain the optimal sampling time of the PWM period includes:

[0010] Calculating an initial time difference between a sampling interruption timestamp of ADC sampling performed by the bidirectional PCS converter and a PWM timer count value;

[0011] Performing moving average filtering on the initial time difference to obtain an average time deviation, and performing linear compensation calculation on the period value of the PWM carrier counter according to the average time deviation to obtain a compensated period value;

[0012] Substitute the compensated period value into the phase compensation equation to obtain a compensation coefficient, and correct the comparison value of the PWM carrier counter according to the compensation coefficient to obtain the optimal sampling time of the PWM period.

[0013] Optionally, in a second implementation of the first aspect of the present invention, the system resonant frequency is calculated according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and the switching delay time of the IGBT power tube in the bidirectional PCS converter is dynamically modulated according to the ratio of the system resonant frequency to the current switching frequency, and the compensated switching delay voltage value is obtained, including:

[0014] Constructing a system state equation matrix according to the inductance value and the capacitance value of the LCL filter network in the bidirectional PCS converter to obtain a characteristic polynomial;

[0015] Solving the characteristic polynomial for an eigenvalue to obtain a system resonant angular frequency, and dividing the system resonant angular frequency by 2π and performing a ratio calculation on the current switching frequency to obtain a frequency ratio;

[0016] Performing piecewise function mapping on the reference switch delay time of the IGBT power tube in the bidirectional PCS converter according to the frequency ratio to obtain the modulated switch delay time;

[0017] The modulated switch delay time is multiplied by the DC bus voltage of the bidirectional PCS converter to obtain the compensated switch delay voltage value.

[0018] Optionally, in a third implementation of the first aspect of the present invention, establishing a resonant compensation channel according to the system resonant frequency, and using the compensated switch delay voltage value as a feedforward compensation amount, and obtaining a PWM modulation signal through weighted combination processing includes:

[0019] Determine the center frequency of the notch filter according to the system resonant frequency to obtain the notch filter transfer function, and perform bilinear transformation on the notch filter transfer function to obtain discrete domain compensator parameters;

[0020] Constructing a resonant compensation channel according to the discrete domain compensator parameters to obtain a compensation output, and weightedly superimposing the compensation output and the compensated switch delay voltage value to obtain a combined compensation amount;

[0021] The combined compensation amount is superimposed on the reference modulation wave of the bidirectional PCS converter to perform phase advance compensation operation to obtain the PWM modulation signal.

[0022] Optionally, in a fourth implementation of the first aspect of the present invention, determining the notch filter center frequency according to the system resonant frequency to obtain the notch filter transfer function, and performing a bilinear transformation on the notch filter transfer function to obtain the discrete domain compensator parameters includes:

[0023] Calculating a frequency ratio according to the ratio of the system resonant frequency to the switching frequency, performing interval judgment processing on the frequency ratio, and obtaining an adjustment coefficient of the center frequency of the notch filter;

[0024] Performing a product operation on the adjustment coefficient and the system resonant frequency, and constructing a second-order notch filter equation according to the operation result to obtain a notch filter transfer function;

[0025] Mapping the notch filter transfer function from the s domain to the z domain according to a bilinear transformation rule to obtain a z domain differential equation;

[0026] Discretization calculation is performed according to the numerator and denominator polynomial coefficients of the z-domain differential equation to obtain discrete domain compensator parameters.

[0027] Optionally, in a fifth implementation of the first aspect of the present invention, the real-time power value obtained by performing power calculation at the optimal sampling moment of the PWM period according to the system resonant frequency is subjected to bandwidth limitation processing to obtain the power command value, which includes:

[0028] Collecting instantaneous values ​​of voltage and current according to the optimal sampling time of the PWM cycle to obtain sampling data, and performing orthogonal decomposition operation on the sampling data to obtain active and reactive components;

[0029] Performing power calculation according to the active and reactive components to obtain a real-time power value, designing a bandpass filter bandwidth according to the system resonant frequency, performing filtering operation on the real-time power value to obtain a filtered power value;

[0030] The filtered power value is compared with a power reference value to obtain a power deviation; and the power deviation is subjected to slope limitation and amplitude limitation to obtain the power command value.

[0031] Optionally, in a sixth implementation of the first aspect of the present invention, adjusting the PWM modulation signal according to the power command value to obtain a control amount for controlling the charging and discharging power of the bidirectional PCS converter includes:

[0032] Convert the power command value into a command component in a d-axis and q-axis coordinate system to obtain a dq-axis power command value;

[0033] Performing a difference operation on the dq axis power command value and the current dq axis current value to obtain a dual axis current error;

[0034] Performing a proportional integral operation on the dual-axis current error to obtain a basic voltage vector, and performing a product operation on the basic voltage vector and a modulation depth of a PWM modulation signal to obtain a modulation compensation signal;

[0035] The modulation compensation signal is subjected to space vector transformation to obtain a three-phase control quantity, and a PWM drive signal with switch delay compensation is generated according to the three-phase control quantity to obtain the control quantity.

[0036] A second aspect of the present invention provides a bidirectional PCS converter power balance control device, the bidirectional PCS converter power balance control device comprising:

[0037] A timing compensation module, used for dynamically compensating the comparison value and the period value of the PWM carrier counter according to the time difference between the sampling interruption timestamp of the ADC sampling of the bidirectional PCS converter and the count value of the PWM timer, so as to obtain the optimal sampling time of the PWM period;

[0038] A switch delay modulation module is used to calculate the system resonant frequency according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and dynamically modulate the switch delay time of the IGBT power tube in the bidirectional PCS converter according to the ratio of the system resonant frequency to the current switching frequency to obtain a compensated switch delay voltage value;

[0039] A resonance suppression module, used to establish a resonance compensation channel according to the system resonance frequency, and use the compensated switch delay voltage value as a feedforward compensation amount to obtain a PWM modulation signal through weighted combination processing;

[0040] The power control module is used to perform bandwidth limiting processing on the real-time power value obtained by power calculation at the optimal sampling time of the PWM period according to the system resonant frequency, obtain a power command value, and adjust the PWM modulation signal according to the power command value to obtain a control amount for controlling the charging and discharging power of the bidirectional PCS converter.

[0041] The above-mentioned bidirectional PCS converter power balance control method and device utilizes the time difference between the interrupt timestamp sampled by ADC and the count value of the PWM timer to dynamically compensate the PWM carrier counter to obtain the optimal sampling time of the PWM cycle; calculates the system resonant frequency according to the LCL filter network parameters, and dynamically modulates the switch delay time of the power tube according to its ratio to the switching frequency to obtain the switch delay voltage value; establishes a compensation channel according to the system resonant frequency, uses the switch delay voltage value as the feedforward compensation amount, and weightedly combines to obtain the PWM modulation signal; performs bandwidth limitation processing on the power calculation value at the optimal sampling time according to the resonant frequency to obtain the power command value, and adjusts the PWM modulation signal accordingly to obtain the control amount of the converter charging and discharging power. This method eliminates the sampling timing deviation through dynamic compensation of the PWM carrier counter, thereby solving the system resonance problem of the bidirectional PCS converter.

[0042] Other features and advantages of the present invention will be described in the following description, and partly become apparent from the description, or understood by practicing the present invention. The purpose and other advantages of the present invention are realized and obtained by the structures particularly pointed out in the description, claims and drawings.

[0043] In order to make the above-mentioned objects, features and advantages of the present invention more obvious and easy to understand, preferred embodiments are given below and described in detail with reference to the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS

[0044] Figure 1 A schematic diagram of a first embodiment of a bidirectional PCS converter power balance control method according to an embodiment of the present invention;

[0045] Figure 2 Schematic diagram of an embodiment of a bidirectional PCS converter power balance control device in an embodiment of the present invention. DETAILED DESCRIPTION

[0046] In order to make the purpose, technical solution and advantages of the embodiments of the present invention clearer, the technical solution of the present invention will be clearly and completely described below in conjunction with the accompanying drawings. Obviously, the described embodiments are part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without creative work are within the scope of protection of the present invention.

[0047] The terms "including" and "having" and any variations thereof mentioned in the embodiments of the present invention are intended to cover non-exclusive inclusions. For example, a process, method, apparatus, product or device end including a series of steps or units is not limited to the listed steps or units, but may optionally include other steps or units that are not listed, or may optionally include other steps or units that are inherent to these processes, methods, products or device ends.

[0048] To facilitate understanding of this embodiment, a bidirectional PCS converter power balance control method disclosed in an embodiment of the present invention is first described in detail. Figure 1 As shown, the method comprises the following steps:

[0049] 101. Perform dynamic compensation processing on the comparison value and period value of the PWM carrier counter according to the time difference between the sampling interrupt timestamp of the ADC sampling of the bidirectional PCS converter and the count value of the PWM timer to obtain the optimal sampling time of the PWM period;

[0050] In one embodiment of the present invention, the dynamic compensation processing of the comparison value and the period value of the PWM carrier counter according to the time difference between the sampling interrupt timestamp of ADC sampling of the bidirectional PCS converter and the PWM timer count value to obtain the optimal sampling time of the PWM period includes: calculating the initial time difference between the sampling interrupt timestamp of ADC sampling of the bidirectional PCS converter and the PWM timer count value; performing moving average filtering processing on the initial time difference to obtain an average time deviation, and performing linear compensation calculation on the period value of the PWM carrier counter according to the average time deviation to obtain a compensated period value; substituting the compensated period value into the phase compensation equation to obtain a compensation coefficient, and correcting the comparison value of the PWM carrier counter according to the compensation coefficient to obtain the optimal sampling time of the PWM period.

[0051] Specifically, in the power balance control of the bidirectional PCS converter, the time difference between the ADC sampling interrupt timestamp and the PWM timer count value directly affects the accuracy of current sampling. First, the PWM timer count value when the ADC sampling interrupt is triggered is captured in real time by the high-precision timing module of the DSP, and the difference is calculated with the preset PWM period value to obtain the initial time difference. For example, when the PWM timer adopts the triangular carrier mode and the period is set to 500μs, if the count value read at the moment of the ADC sampling interrupt trigger is 480μs, the initial time difference is 20μs. This difference reflects the offset of the sampling time relative to the ideal position of the PWM carrier (such as the bottom or peak of the triangular wave), and its physical meaning is to quantify the phase misalignment between the sampling signal and the actual state of the PWM waveform. Due to factors such as the interrupt response delay of the DSP, the ADC conversion time, and the propagation delay of the hardware signal path, the initial time difference will change dynamically with the system operation state, and its influence needs to be eliminated through subsequent processing.

[0052] After the initial time difference is calculated, it needs to be processed by moving average filtering to suppress random interference. Moving average filtering maintains a fixed-length historical data queue (for example, storing the initial time difference of the last 10 cycles), takes the arithmetic mean of all data in the queue, and generates an average time deviation. For example, if the initial time difference values ​​stored in the queue are 18μs, 22μs, 19μs, etc., the average value is 20μs. The purpose of filtering is to eliminate outliers caused by noise or instantaneous disturbances in a single sampling, while retaining the long-term trend of the time difference. This process relies on the real-time data storage and computing capabilities of the DSP, and realizes efficient updating of historical data through the circular queue data structure. The generation of average time deviation ensures the stability of subsequent compensation operations and avoids frequent jumps in PWM parameters caused by instantaneous fluctuations.

[0053] Based on the average time deviation, the period value of the PWM carrier counter is linearly compensated. In the specific implementation, the average time deviation is proportionally superimposed with the current PWM period value to generate the compensated period value. For example, if the current PWM period is 500μs, the average time deviation is 20μs, and the compensation coefficient k is set to 0.8, the compensated period value is 500μs+0.8×20μs=516μs. The compensation coefficient here needs to be pre-calibrated according to the system switching frequency and hardware characteristics. The core idea is to gradually reduce the deviation between the sampling time and the ideal position by adjusting the PWM period value. The essence of linear compensation is to dynamically correct the time base value of the PWM carrier so that the ADC sampling window gradually converges to the target phase within multiple cycles. This operation requires real-time updating of the period register in the DSP's PWM module to ensure that the carrier waveform of the next cycle can reflect the compensation result.

[0054] The compensated period value needs to be further substituted into the phase compensation equation to calculate the compensation coefficient used to correct the comparison value. The design of the phase compensation equation is based on the geometric characteristics of the PWM carrier. For example, in the triangular carrier mode, the ideal sampling time is usually at the bottom of the carrier (the count value is 0) or the peak (the count value is the maximum value of the cycle). Assuming that the current cycle value is 516μs, and the ideal sampling time should be at the bottom of the valley, the compensation coefficient α can be calculated by the cycle deviation ratio, that is, α=(compensated period value-original period value) / original period value. For example, α=(516μs-500μs) / 500μs=0.032. This coefficient reflects the weight of the influence of the period adjustment on the phase of the comparison value. Its purpose is to convert the compensation of the period dimension into a direct correction of the duty cycle control amount.

[0055] Finally, the comparison value of the PWM carrier counter is dynamically corrected using the compensation coefficient. The comparison value determines the duty cycle of the PWM waveform, and its correction needs to be combined with the current working state and the target phase. For example, if the original comparison value is 250μs (corresponding to a 50% duty cycle) and the compensation coefficient α is 0.032, the corrected comparison value is 250μs×(1+α)=258μs. This correction advances or delays the rising or falling edge of the PWM waveform, thereby indirectly adjusting the phase position of the ADC sampling moment. The corrected comparison value is output to the power device through the PWM module of the DSP to ensure that the sampling moment is accurately aligned to the carrier valley or peak. At this point, the current value captured by the ADC can accurately reflect the average value within the PWM cycle, eliminating harmonic distortion or power calculation errors caused by timing deviations.

[0056] 102. Calculate the system resonant frequency according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and dynamically modulate the switching delay time of the IGBT power tube in the bidirectional PCS converter according to the ratio of the system resonant frequency to the current switching frequency to obtain a compensated switching delay voltage value;

[0057] In one embodiment of the present invention, the system resonant frequency is calculated according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and the switch delay time of the IGBT power tube in the bidirectional PCS converter is dynamically modulated according to the ratio of the system resonant frequency to the current switching frequency to obtain the compensated switch delay voltage value, including: constructing a system state equation matrix according to the inductance and capacitance values ​​of the LCL filter network in the bidirectional PCS converter to obtain a characteristic polynomial; solving the characteristic polynomial for the characteristic value to obtain the system resonant angular frequency, and dividing the system resonant angular frequency by 2π and calculating the ratio with the current switching frequency to obtain a frequency ratio; performing piecewise function mapping on the reference switch delay time of the IGBT power tube in the bidirectional PCS converter according to the frequency ratio to obtain the modulated switch delay time; multiplying the modulated switch delay time by the DC bus voltage of the bidirectional PCS converter to obtain the compensated switch delay voltage value.

[0058] Specifically, in a bidirectional PCS converter, the inductance and capacitance parameters of the LCL filter network directly affect the distribution characteristics of the system resonant frequency, and the interaction between the resonant frequency and the switching frequency determines the dynamic compensation requirements of the power device switching delay. First, based on the inductance and capacitance values ​​of the LCL filter network, a state equation matrix describing the dynamic behavior of the system is constructed. The establishment of the state equation requires the clarification of the state variables of the system. Usually, the inductor current and capacitor voltage are selected as the core variables. Combined with Kirchhoff's voltage law and current law, a differential equation group describing the relationship between the inductor voltage and the capacitor current is derived. By converting the differential equation group into a matrix form, a state matrix is ​​obtained, and its dimension is determined by the number of system state variables. The construction process of the state matrix is ​​essentially to abstract the nonlinear dynamic characteristics of the physical circuit into a linear time-invariant model, providing a mathematical framework for the subsequent analysis of the system stability and resonant characteristics. The establishment of the state equation must strictly follow the circuit topology and parameter constraints, such as the voltage drop of the inductor series resistor and the continuity conditions of the capacitor charging and discharging, to ensure that the model can accurately reflect the dynamic response of the actual system.

[0059] After the state equation matrix is ​​generated, the resonant angular frequency of the system needs to be extracted through eigenvalue analysis. The solution of the eigenvalue depends on the numerical calculation method. The characteristic equation is solved through iteration or matrix decomposition technology to obtain the pole distribution of the system. For LCL filters, the eigenvalue is usually expressed as a conjugate complex number pair, the real part of which characterizes the damping characteristics of the system, and the imaginary part corresponds to the resonant angular frequency. The physical meaning of the resonant angular frequency is to describe the free oscillation frequency of the system in the undamped state, and its numerical value is determined by the inverse square root of the product of the inductance and the capacitance. After converting the resonant angular frequency to the resonant frequency in Hertz, the ratio between it and the current switching frequency is further calculated to quantify the relative position of the resonant frequency in the switching frequency band. The evaluation of the frequency ratio is a key indicator for judging the stability of the system. The higher the ratio, the closer the resonance point is to the integer multiple frequency band of the switching frequency, and the easier it is for the system to excite the resonance phenomenon due to the high-frequency harmonics of the switching action, resulting in current distortion or voltage oscillation.

[0060] Based on the result of the frequency ratio, the reference switch delay time of the IGBT power tube is dynamically modulated. The modulation strategy adopts a piecewise function mapping mechanism, and defines different delay time adjustment rules according to the range of the frequency ratio. The design of the piecewise function needs to be combined with experimental data and stability criteria, and is usually divided into low-risk, medium-risk and high-risk intervals, corresponding to different delay increment coefficients. For example, in the low-risk interval, the delay time maintains the reference value to maintain the switching efficiency; in the medium-risk interval, the delay time increases linearly to suppress the accumulation of resonant energy; in the high-risk interval, the delay time increases significantly according to the nonlinear law to force damping oscillation. The realization of the modulation process depends on the real-time computing capability of the digital signal processor. The delay parameters are dynamically adjusted by table lookup or online calculation to ensure that the timing characteristics of the switching action can adapt to changes in system operating conditions. The dynamic adjustment of the delay time is essentially to reduce the edge rate of the switching device and reduce the excitation of the high-frequency harmonic component on the resonance point, thereby improving the stability of the system in a wide frequency domain.

[0061] The modulated switching delay time needs to be further converted into an equivalent voltage compensation value to offset the influence of the delay effect on the power control accuracy. The compensation value is calculated by multiplying the delay time with the DC bus voltage. Its physical meaning is to quantify the equivalent voltage loss caused by the time deviation of the switching action. The introduction of the voltage compensation value needs to be embedded in the modulation wave generation link, and the voltage error caused by the delay is pre-compensated in the PWM signal by feedforward superposition or duty cycle correction. For example, in space vector modulation, the compensation voltage can be converted into the action time adjustment amount of the voltage vector, and the fundamental component of the output waveform is ensured to be consistent with the reference value by correcting the timing distribution of the vector switching point. The realization of the compensation process needs to work in coordination with the current loop control algorithm to avoid the conflict between feedforward compensation and feedback control. At the same time, the dynamic limit of the compensation amount needs to be considered to prevent the modulation wave distortion caused by over-compensation.

[0062] 103. Establish a resonance compensation channel according to the system resonance frequency, and use the compensated switch delay voltage value as a feedforward compensation amount to obtain a PWM modulation signal through weighted combination processing;

[0063] In one embodiment of the present invention, a resonant compensation channel is established according to the system resonant frequency, and the compensated switch delay voltage value is used as a feedforward compensation amount, and a PWM modulation signal is obtained by weighted combination processing, including: determining the notch filter center frequency according to the system resonant frequency to obtain the notch filter transfer function, and performing a bilinear transformation on the notch filter transfer function to obtain discrete domain compensator parameters; constructing a resonant compensation channel according to the discrete domain compensator parameters to obtain a compensation output, and weightedly superimposing the compensation output and the compensated switch delay voltage value to obtain a combined compensation amount; and performing a phase lead compensation operation on the combined compensation amount superimposed on the reference modulation wave of the bidirectional PCS converter to obtain the PWM modulation signal.

[0064] Specifically, in the power balance control of the bidirectional PCS converter, the precise compensation of the system resonant frequency and the feedforward superposition of the switch delay voltage are the core links to improve the dynamic performance. First, the center frequency of the notch filter is determined according to the system resonant frequency. Its design goal is to form a deep attenuation near the resonant frequency and suppress the resonant current component caused by the LCL filter. The construction of the notch filter transfer function needs to be combined with the mathematical model of the second-order band-stop filter, and its center frequency is strictly aligned with the system resonant frequency. For example, if the system resonant frequency is 500Hz, the center frequency of the notch filter is set to 500Hz, and the poles and zeros of the transfer function need to be symmetrically configured around this frequency to ensure the maximum attenuation depth at the resonant frequency. The parameterization process of the transfer function needs to consider the trade-off between quality factor and bandwidth. The higher the quality factor, the narrower the notch filter stopband and the stronger the selective suppression ability of the resonant frequency, but it may introduce the risk of phase distortion.

[0065] After the notch filter transfer function is determined, it needs to be converted from the continuous domain (s domain) to the discrete domain (z domain) through bilinear transformation to meet the computing requirements of the digital controller. The essence of bilinear transformation is to map the differential equation in the s domain to the difference equation in the z domain. The core is to ensure that the frequency response after discretization is consistent with the continuous domain through frequency pre-distortion compensation. The sampling period parameter needs to be introduced during the transformation process. For example, when the control period is 100μs, the sampling frequency is 10kHz. At this time, the coefficients of the transfer function need to be normalized to ensure that the positions of the poles and zeros after discretization match the design goals. The generation of discrete domain compensator parameters includes the calculation of the numerator polynomial coefficients and the denominator polynomial coefficients. For example, for a second-order notch filter, its difference equation can be expressed as a linear combination of the current output and the historical input and output. The realization of discrete parameters depends on the real-time computing capability of the DSP, and the calculation accuracy and efficiency are ensured through floating-point operations or fixed-point number optimization.

[0066] The resonant compensation channel is constructed based on the discrete domain compensator parameters. Its essence is to pass the current or voltage signal through a digital filter, extract the disturbance component near the resonant frequency and generate a reverse compensation signal. For example, in the current loop control, the inductor current signal is input into the resonant compensation channel, and the compensation output is obtained after filtering by a notch filter. The output contains harmonic components near the resonant frequency, but its phase is opposite to the original signal. The amplitude and phase characteristics of the compensation output need to be calibrated experimentally to ensure that it can effectively offset the system resonant energy. At the same time, the compensated switch delay voltage value is used as the feedforward compensation amount, and its physical meaning is to offset the voltage error caused by the delay of the switching device. The generation of the feedforward compensation amount needs to be combined with the DC bus voltage and the delay time of the dynamic modulation. For example, the longer the delay time and the higher the bus voltage, the greater the absolute value of the feedforward compensation amount.

[0067] The resonant compensation output and the feedforward compensation are weighted and superimposed to generate a combined compensation. The weight coefficient needs to be dynamically adjusted according to the system operating conditions. For example, under light load or high resonance risk conditions, the weight of the resonant compensation is increased to strengthen oscillation suppression; under heavy load or stable conditions, the weight of the feedforward compensation is increased to optimize the dynamic response. The weighted superposition is achieved through the DSP multiplication and accumulation instructions. For example, the resonant compensation output is multiplied by the coefficient α, and the feedforward compensation is multiplied by the coefficient β. The sum of the two is obtained to obtain the combined compensation. The adaptive adjustment of the coefficient can be based on the output of the resonant energy observer or the power direction detection module. For example, when the harmonic energy near the resonant frequency is detected to exceed the threshold, the proportion of α is automatically increased.

[0068] The combined compensation is added to the reference modulation wave of the bidirectional PCS converter, and a phase lead compensation operation is performed. The reference modulation wave is usually generated by the output of the current loop or the power loop, and its essence is the desired voltage reference signal. After the combined compensation is added, the amplitude and phase characteristics of the modulation wave are corrected to achieve resonance suppression and delay compensation at the same time. The purpose of phase lead compensation is to offset the phase lag introduced by factors such as filtering and calculation delay in the control loop. For example, before PWM is generated, the modulation wave signal is corrected for the lead angle to ensure that the timing of the switching action is strictly synchronized with the reference signal. The calculation of the phase compensation amount needs to be combined with the open-loop frequency characteristics of the system, such as determining the compensation angle through Bode plot analysis, or estimating it based on the equivalent phase difference of the delay time.

[0069] Finally, the phase-compensated modulated wave signal is converted into a PWM drive signal through space vector modulation (SVPWM) or carrier comparison method. For example, in SVPWM, the compensated modulated wave is decomposed into basic voltage vectors and their action time, and the PWM module of the DSP generates a drive pulse with dead zone compensation to control the on and off of the IGBT power tube. The duty cycle and phase of the PWM signal strictly follow the compensated modulated wave, thereby achieving coordinated optimization of resonance suppression, delay compensation and dynamic response at the power device level.

[0070] Furthermore, the method of determining the center frequency of the notch filter according to the system resonant frequency to obtain the notch filter transfer function, and performing a bilinear transformation on the notch filter transfer function to obtain discrete domain compensator parameters includes: calculating the frequency ratio according to the ratio of the system resonant frequency to the switching frequency, performing interval judgment processing on the frequency ratio, and obtaining an adjustment coefficient of the center frequency of the notch filter; performing a product operation on the adjustment coefficient and the system resonant frequency, and constructing a second-order notch filter equation according to the operation result to obtain the notch filter transfer function; mapping the notch filter transfer function from the s domain to the z domain according to the bilinear transformation rule to obtain a z domain differential equation; and performing discretization calculation on the numerator and denominator polynomial coefficients of the z domain differential equation to obtain discrete domain compensator parameters.

[0071] Specifically, in the resonance suppression control of the bidirectional PCS converter, the dynamic calibration and discretization of the center frequency of the notch filter are the core links to ensure the effectiveness of the resonance compensation. First, the frequency ratio is calculated according to the ratio of the system resonant frequency to the switching frequency. Its physical meaning is to quantify the relative position relationship between the resonant frequency point and the switching frequency band. For example, when the system resonant frequency is 500Hz and the switching frequency is 4kHz, the frequency ratio is 0.125. The interval judgment of the frequency ratio divides the system operating conditions into multiple modes through a preset threshold: when the frequency ratio is less than 0.1, the resonant frequency is judged to be far away from the switching frequency band, and the center frequency of the notch filter maintains the nominal value; when the frequency ratio is between 0.1 and 0.3, the resonant frequency is close to the low-order harmonics of the switching frequency band, and a linear adjustment coefficient needs to be introduced to dynamically raise the center frequency; when the frequency ratio is greater than 0.3, the system is in a high-frequency resonance risk area, and an exponential adjustment coefficient is used to significantly shift the center frequency to avoid switching harmonic excitation. The design of the interval judgment and adjustment coefficient is intended to avoid the overlap of the center frequency of the notch filter and the switching harmonic frequency, thereby preventing the notch filter from excessively attenuating the useful signal or missing the key resonant components.

[0072] Based on the adjusted center frequency, a second-order notch filter transfer function is constructed. The core function of the notch filter is to form a stopband near a specific frequency point to suppress the resonant energy. The design of the transfer function needs to be combined with the zero-pole configuration, with the zero located at the target frequency to achieve deep attenuation and the pole located in the left half plane to maintain system stability. For example, when the adjusted center frequency is 550Hz, the stopband range of the notch filter needs to cover the fluctuation range near 550Hz, and the stopband width is controlled by the quality factor. The selection of the quality factor needs to balance the suppression effect and phase distortion. A higher quality factor can enhance frequency selectivity, but may introduce the risk of phase lag. The parameterization process of the transfer function must strictly follow the frequency domain response requirements to ensure the balance between the stopband attenuation depth and the passband flatness.

[0073] When converting the continuous domain transfer function to the discrete domain, the bilinear transformation method is used to avoid frequency aliasing. The bilinear transformation maps the differential equation in the s domain to the difference equation in the z domain through the pre-distortion compensation mechanism to ensure that the frequency response after discretization is consistent with the design target. The sampling period parameter needs to be introduced during the transformation process. For example, when the control period is 100μs, the sampling frequency is 10kHz. At this time, the coefficients of the transfer function need to be normalized. The discretized transfer function is expressed as a combination of the coefficients of the numerator and denominator polynomials. For example, the difference equation of the second-order notch filter can be expressed as the linear weighted sum of the current output and the historical input and output. The generation of discrete parameters relies on symbolic operations or numerical calculation tools to ensure that the pole and zero positions accurately match the continuous domain characteristics in the z plane.

[0074] The final form of the discrete domain compensator parameters is an executable differential equation, and its implementation depends on the real-time computing capability of the digital signal processor. For example, a circular buffer is configured in the DSP to store historical input and output data, and the iterative calculation of the differential equation is completed through multiplication and accumulation instructions. The dynamic update mechanism of the compensator parameters allows the coefficients to be adjusted in real time according to the change of the frequency ratio, thereby adapting to the dynamic fluctuations of the system working conditions. For example, when the frequency ratio increases from 0.1 to 0.2, the adjustment coefficient increases linearly, the center frequency of the notch filter shifts accordingly, and the discrete domain coefficients are updated synchronously to maintain the stopband characteristics. This process is implemented through table lookup or online calculation to ensure the real-time and stability of the control system.

[0075] 104. Perform bandwidth limitation processing on the real-time power value obtained by performing power calculation at the optimal sampling time of the PWM period according to the system resonant frequency to obtain a power command value, and adjust the PWM modulation signal according to the power command value to obtain a control quantity for controlling the charging and discharging power of the bidirectional PCS converter.

[0076] In one embodiment of the present invention, the real-time power value obtained by performing power calculation at the optimal sampling moment of the PWM period according to the system resonant frequency is subjected to bandwidth limitation processing to obtain the power command value, including: collecting instantaneous values ​​of voltage and current according to the optimal sampling moment of the PWM period to obtain sampling data, and performing orthogonal decomposition operation on the sampling data to obtain active and reactive components; performing power calculation according to the active and reactive components to obtain a real-time power value, and designing a bandpass filter bandwidth according to the system resonant frequency, performing filtering operation on the real-time power value to obtain a filtered power value; comparing the filtered power value with a power reference value to obtain a power deviation; performing slope limitation and amplitude limitation processing on the power deviation to obtain the power command value.

[0077] Specifically, in the power balance control of the bidirectional PCS converter, the voltage and current data acquisition and power calculation at the optimal sampling time of the PWM cycle are the basis for ensuring the control accuracy. First, at the bottom or peak moment of the PWM carrier (i.e., the optimal sampling moment), the instantaneous values ​​of the three-phase voltage and current are synchronously captured through the ADC module. For example, when the PWM cycle is 500μs, the sampling moment is strictly aligned to the midpoint of the carrier waveform to ensure that the collected voltage and current signals can accurately reflect the average value within a switching cycle and avoid high-frequency noise introduced by timing deviation. After the sampled data is filtered by hardware anti-aliasing, it is input into the data buffer of the DSP to provide the original signal for subsequent processing.

[0078] Perform orthogonal decomposition on the sampled data to extract active and reactive components. The implementation of orthogonal decomposition depends on the coordinate transformation algorithm. For example, the three-phase voltage and current are converted into components in the stationary coordinate system (αβ axis) through Clark transformation, and then rotated to the synchronous rotating coordinate system (dq axis) through Park transformation. In the dq coordinate system, the direct axis (d axis) component corresponds to active power, and the quadrature axis (q axis) component corresponds to reactive power. For example, if the d axis component of the sampled voltage is 220V, the q axis component is 0V, and the d axis component of the current is 10A, then the real-time active power is 220V×10A=2200W, and the reactive power is 0Var. The essence of orthogonal decomposition is to project the time domain signal to the direction of energy transmission, separate the active and passive components of power, and provide structured input for power calculation.

[0079] The core of calculating the real-time power value based on the active and reactive components is to combine the product of voltage and current with the integral operation. For the bidirectional PCS converter, the real-time power value includes instantaneous active power and instantaneous reactive power, which are directly obtained by the product of the voltage and current of the d-axis and q-axis, respectively. For example, the product of the d-axis voltage 220V and the d-axis current 10A is 2200W, and the product of the q-axis voltage 0V and the q-axis current 5A is 0Var. The calculation of the real-time power value needs to consider the continuity of the sampling interval, such as using the sliding window averaging method to eliminate instantaneous fluctuations and ensure the smoothness of the power data.

[0080] The bandwidth of the bandpass filter is designed according to the system resonant frequency, and the real-time power value is filtered. The design of the bandpass filter needs to take the resonant frequency as the center frequency, and the passband range covers the expected fluctuation range of the resonant frequency. For example, if the system resonant frequency is 500Hz, the passband can be set to 450Hz to 550Hz, and the stopband attenuation is not less than 40dB. The filter is implemented using a finite impulse response (FIR) or infinite impulse response (IIR) structure. For example, an IIR elliptical filter is used to take into account the steepness of the transition band and computational efficiency. The filtered power value retains the energy components near the resonant frequency, filters out high-frequency switching noise and low-frequency power frequency interference, and provides a pure input signal for power command generation.

[0081] The filtered power value is compared with the power reference value to generate a power deviation. The power reference value is usually set by the superior energy management system or the local control strategy. For example, the reference value is negative in the energy storage charging mode and positive in the discharge mode. The calculation of the deviation is achieved through subtraction. For example, if the filtered active power is 2100W and the reference value is 2000W, the deviation is +100W. The physical meaning of the deviation is to quantify the gap between the current power state and the target value. Its sign and amplitude directly determine the adjustment direction and strength of the controller.

[0082] The power deviation is slope-limited and amplitude-limited to generate the final power command value. Slope limitation prevents sudden changes in power commands from causing overcurrent or overvoltage by constraining the rate of change of the deviation. For example, if the maximum allowable change rate is set to 1000W / s, if the command value of the previous cycle is 2000W and the current deviation is +100W, the command value of this cycle will increase to a maximum of 2000W + 1000W×T (T is the control period). Amplitude limitation ensures that the power command value is within the hardware safety range by setting upper and lower thresholds, for example, the maximum charge and discharge power limit is ±3000W. The power command value that has been limited not only retains the dynamic adjustment capability, but also avoids system risks under extreme working conditions, providing stable input for the current loop or voltage loop.

[0083] Furthermore, the method of adjusting the PWM modulation signal according to the power command value to obtain a control quantity for controlling the charging and discharging power of the bidirectional PCS converter includes: converting the power command value into a command component in the d-axis and q-axis coordinate system to obtain a dq-axis power command value; performing a difference operation between the dq-axis power command value and the current dq-axis current value to obtain a dual-axis current error; performing a proportional-integral operation on the dual-axis current error to obtain a basic voltage vector, and multiplying the basic voltage vector by the modulation depth of the PWM modulation signal to obtain a modulation compensation signal; performing a space vector transformation on the modulation compensation signal to obtain a three-phase control quantity, and generating a PWM drive signal with switch delay compensation based on the three-phase control quantity to obtain the control quantity.

[0084] Specifically, in the power control of the bidirectional PCS converter, the conversion of the power command value to the PWM modulation signal is the core link to achieve precise regulation of the charging and discharging power. First, the power command value is converted into the command component in the d-axis and q-axis coordinate systems. Its essence is to map the target power to the reference input of the current loop through the power-current decoupling algorithm. For example, in the synchronous rotating coordinate system, the active power command corresponds to the d-axis current reference value , reactive power command corresponds to q-axis current reference value The conversion process needs to combine the phase information of the grid voltage and obtain the grid angular frequency through a phase-locked loop (PLL). The power command is decomposed into dq axes using the coordinate transformation matrix. For example, if the power command is 2000W active power and OVar reactive power, then ( is the d-axis voltage), and Keep it at zero.

[0085] The dual-axis current error is generated based on the difference between the dq axis current reference value and the actual current value. The actual current value is obtained by ADC sampling and coordinate transformation, such as the d-axis current and q-axis current The error is calculated by subtraction, for example and The physical meaning of the error is to quantify the degree of deviation of current tracking. Its sign and amplitude directly determine the adjustment direction and strength of the controller. If it is 10A lower than the reference value, , the controller needs to increase the output voltage to increase the current.

[0086] The proportional integral (PI) operation is performed on the dual-axis current error to generate the basic voltage vector. The parameters of the PI controller need to be adjusted according to the dynamic characteristics of the system, such as the proportional coefficient Determines response speed, integral coefficient Determines the steady-state accuracy. The calculation results are the d-axis voltage compensation and q-axis voltage compensation ,For example The amplitude and phase of the basic voltage vector reflect the voltage correction required to eliminate the current error, and its output needs to be adjusted in coordination with the modulation depth to avoid over-modulation or under-modulation. For example, when the modulation depth is 0.8, the voltage vector is scaled to ensure that its amplitude does not exceed 80% of the DC bus voltage.

[0087] The basic voltage vector is multiplied by the modulation depth to generate a modulation compensation signal. The dynamic adjustment of the modulation depth needs to be combined with the real-time value of the DC bus voltage. For example, when the bus voltage fluctuates, the output voltage stability is maintained through an adaptive algorithm. For example, if the DC bus voltage is 800V and the modulation depth is set to 0.9, the maximum output line voltage peak value is The generation of the modulation compensation signal needs to take into account the dead time effect and the nonlinear characteristics of the switching device, for example, by using pre-distortion compensation to offset the voltage loss caused by the dead time.

[0088] The modulation compensation signal is subjected to space vector transformation (SVPWM) to generate a three-phase control quantity. The core of space vector transformation is to convert the d-axis and q-axis voltage components into a three-phase modulation wave, and the process includes reference vector synthesis, sector judgment and action time calculation. For example, if the d-axis voltage is 220V and the q-axis voltage is 0V, the reference vector is located in the positive direction of the α-axis, and the corresponding three-phase modulation wave amplitude is distributed according to the sinusoidal law. The generation of the three-phase control quantity requires the carrier comparison method to compare the modulation wave with the triangular carrier to generate a PWMM pulse sequence with a specific duty cycle. Finally, a PWM drive signal with switch delay compensation is generated according to the three-phase control quantity. Switch delay compensation is implemented by a feedforward method, such as converting the delay time into an equivalent duty cycle correction amount and superimposing it on the original modulation wave. For example, if the switch delay is 2μs and the carrier period is 500μs, the duty cycle correction amount is , the corresponding voltage compensation is The compensated PWM signal is transmitted to the IGBT gate through the optocoupler isolation and drive circuit to control its inrush and shutdown timing, ensuring that the converter output power is consistent with the command value.

[0089] In this embodiment, the PWM carrier counter is dynamically compensated according to the time difference between the interrupt timestamp sampled by the ADC and the count value of the PWM timer to obtain the optimal sampling time of the PWM cycle; the system resonant frequency is calculated according to the LCL filter network parameters, and the switch delay time of the power tube is dynamically modulated according to its ratio to the switching frequency to obtain the switch delay voltage value; a compensation channel is established according to the system resonant frequency, and the switch delay voltage value is used as the feedforward compensation amount, and the PWM modulation signal is obtained by weighted combination; the power calculation value at the optimal sampling time is bandwidth-limited according to the resonant frequency to obtain the power command value, and the PWM modulation signal is adjusted accordingly to obtain the control amount of the converter charging and discharging power. This method eliminates the sampling timing deviation through dynamic compensation of the PWM carrier counter, thereby solving the system resonance problem of the bidirectional PCS converter.

[0090] The above describes the bidirectional PCS converter power balance control method in the embodiment of the present invention. The following describes the bidirectional PCS converter power balance control device in the embodiment of the present invention. Figure 2 , an embodiment of a bidirectional PCS converter power balance control device in an embodiment of the present invention includes:

[0091] A timing compensation module 201 is used to dynamically compensate the comparison value and period value of the PWM carrier counter according to the time difference between the sampling interruption timestamp of the ADC sampling of the bidirectional PCS converter and the count value of the PWM timer to obtain the optimal sampling time of the PWM period;

[0092] The switch delay modulation module 202 is used to calculate the system resonant frequency according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and dynamically modulate the switch delay time of the IGBT power tube in the bidirectional PCS converter according to the ratio of the system resonant frequency to the current switching frequency to obtain a compensated switch delay voltage value;

[0093] A resonance suppression module 203 is used to establish a resonance compensation channel according to the system resonance frequency, and use the compensated switch delay voltage value as a feedforward compensation amount to obtain a PWM modulation signal through weighted combination processing;

[0094] The power control module 204 is used to perform bandwidth limiting processing on the real-time power value obtained by power calculation at the optimal sampling time of the PWM period according to the system resonant frequency, obtain a power command value, and adjust the PWM modulation signal according to the power command value to obtain a control amount for controlling the charging and discharging power of the bidirectional PCS converter.

[0095] In the embodiment of the present invention, the bidirectional PCS converter power balance control device runs the above-mentioned bidirectional PCS converter power balance control method, and the bidirectional PCS converter power balance control device dynamically compensates the PWM carrier counter according to the time difference between the interrupt timestamp sampled by the ADC and the count value of the PWM timer to obtain the optimal sampling time of the PWM cycle; calculates the system resonant frequency according to the LCL filter network parameters, and dynamically modulates the switch delay time of the power tube according to its ratio to the switching frequency to obtain the switch delay voltage value; establishes a compensation channel according to the system resonant frequency, uses the switch delay voltage value as the feedforward compensation amount, and weightedly combines to obtain the PWM modulation signal; performs bandwidth limitation processing on the power calculation value at the optimal sampling time according to the resonant frequency to obtain the power command value, and adjusts the PWM modulation signal accordingly to obtain the control amount of the converter charging and discharging power. This method eliminates the sampling timing deviation through dynamic compensation of the PWM carrier counter, thereby solving the system resonance problem of the bidirectional PCS converter.

[0096] Those skilled in the art can clearly understand that, for the convenience and brevity of description, the specific working process of the above-described system, device, or unit can refer to the corresponding process in the aforementioned method embodiment and will not be repeated here.

[0097] If the integrated unit is implemented in the form of a software functional unit and sold or used as an independent product, it can be stored in a computer-readable storage medium. Based on this understanding, the technical solution of the present invention is essentially or the part that contributes to the prior art or the whole or part of the technical solution can be embodied in the form of a software product. The computer software product is stored in a storage medium, including several instructions to enable a computer device (which can be a personal computer, server, or network device, etc.) to perform all or part of the steps of the method described in each embodiment of the present invention. The aforementioned storage medium includes: U disk, mobile hard disk, read-only memory (ROM), random access memory (RAM), disk or optical disk and other media that can store program code.

[0098] As described above, the above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit the same. Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that the technical solutions described in the aforementioned embodiments may still be modified, or some of the technical features thereof may be replaced by equivalents. However, these modifications or replacements do not deviate the essence of the corresponding technical solutions from the spirit and scope of the technical solutions of the embodiments of the present invention.

Claims

1. A bidirectional PCS converter power balance control method, characterized in that: The bidirectional PCS converter power balance control method comprises: Dynamically compensating the comparison value and the period value of the PWM carrier counter according to the time difference between the sampling interrupt timestamp of the ADC sampling of the bidirectional PCS converter and the count value of the PWM timer to obtain the optimal sampling time of the PWM period; The system resonant frequency is calculated according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and the switching delay time of the IGBT power tube in the bidirectional PCS converter is dynamically modulated according to the ratio of the system resonant frequency to the current switching frequency to obtain a compensated switching delay voltage value; A resonance compensation channel is established according to the system resonance frequency, and the compensated switch delay voltage value is used as a feedforward compensation amount, and a PWM modulation signal is obtained through weighted combination processing; The real-time power value obtained by performing power calculation at the optimal sampling time of the PWM period according to the system resonant frequency is subjected to bandwidth limitation processing to obtain a power command value, and the PWM modulation signal is adjusted according to the power command value to obtain a control quantity for controlling the charging and discharging power of the bidirectional PCS converter.

2. The power balance control method for a bidirectional PCS converter according to claim 1, wherein the step of dynamically compensating a comparison value and a period value of a PWM carrier counter according to a time difference between a sampling interruption timestamp of ADC sampling of the bidirectional PCS converter and a count value of a PWM timer to obtain an optimal sampling time of a PWM period comprises: Calculating an initial time difference between a sampling interruption timestamp of ADC sampling performed by the bidirectional PCS converter and a PWM timer count value; Performing moving average filtering on the initial time difference to obtain an average time deviation, and performing linear compensation calculation on the period value of the PWM carrier counter according to the average time deviation to obtain a compensated period value; Substitute the compensated period value into the phase compensation equation to obtain a compensation coefficient, and correct the comparison value of the PWM carrier counter according to the compensation coefficient to obtain the optimal sampling time of the PWM period.

3. According to the power balance control method of the bidirectional PCS converter in claim 1, the system resonant frequency is calculated according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and the switching delay time of the IGBT power tube in the bidirectional PCS converter is dynamically modulated according to the ratio of the system resonant frequency to the current switching frequency, and the compensated switching delay voltage value is obtained, which includes: Constructing a system state equation matrix according to the inductance value and the capacitance value of the LCL filter network in the bidirectional PCS converter to obtain a characteristic polynomial; Solving the characteristic polynomial for an eigenvalue to obtain a system resonant angular frequency, and dividing the system resonant angular frequency by 2π and performing a ratio calculation on the current switching frequency to obtain a frequency ratio; Performing piecewise function mapping on the reference switch delay time of the IGBT power tube in the bidirectional PCS converter according to the frequency ratio to obtain the modulated switch delay time; The modulated switch delay time is multiplied by the DC bus voltage of the bidirectional PCS converter to obtain the compensated switch delay voltage value.

4. The bidirectional PCS converter power balance control method according to claim 1, wherein the resonant compensation channel is established according to the system resonant frequency, and the compensated switch delay voltage value is used as a feedforward compensation amount, and the PWM modulation signal is obtained by weighted combination processing, comprising: Determine the center frequency of the notch filter according to the system resonant frequency to obtain the notch filter transfer function, and perform bilinear transformation on the notch filter transfer function to obtain discrete domain compensator parameters; Constructing a resonant compensation channel according to the discrete domain compensator parameters to obtain a compensation output, and weightedly superimposing the compensation output and the compensated switch delay voltage value to obtain a combined compensation amount; The combined compensation amount is superimposed on the reference modulation wave of the bidirectional PCS converter to perform phase advance compensation operation to obtain the PWM modulation signal.

5. The bidirectional PCS converter power balance control method according to claim 4, wherein determining the notch filter center frequency according to the system resonant frequency to obtain the notch filter transfer function, and performing a bilinear transformation on the notch filter transfer function to obtain discrete domain compensator parameters comprises: Calculating a frequency ratio according to the ratio of the system resonant frequency to the switching frequency, performing interval judgment processing on the frequency ratio, and obtaining an adjustment coefficient of the center frequency of the notch filter; Performing a product operation on the adjustment coefficient and the system resonant frequency, and constructing a second-order notch filter equation according to the operation result to obtain a notch filter transfer function; Mapping the notch filter transfer function from the s domain to the z domain according to a bilinear transformation rule to obtain a z domain differential equation; Discretization calculation is performed according to the numerator and denominator polynomial coefficients of the z-domain differential equation to obtain discrete domain compensator parameters.

6. The bidirectional PCS converter power balance control method according to claim 1, wherein the real-time power value obtained by performing power calculation at the optimal sampling time of the PWM period according to the system resonant frequency is subjected to bandwidth limitation processing to obtain the power command value, comprising: Collecting instantaneous values ​​of voltage and current according to the optimal sampling time of the PWM cycle to obtain sampling data, and performing orthogonal decomposition operation on the sampling data to obtain active and reactive components; Performing power calculation according to the active and reactive components to obtain a real-time power value, designing a bandpass filter bandwidth according to the system resonant frequency, performing filtering operation on the real-time power value to obtain a filtered power value; The filtered power value is compared with a power reference value to obtain a power deviation; and the power deviation is subjected to slope limitation and amplitude limitation to obtain the power command value.

7. The power balance control method for a bidirectional PCS converter according to claim 1, wherein the step of adjusting the PWM modulation signal according to the power command value to obtain a control quantity for controlling the charging and discharging power of the bidirectional PCS converter comprises: Convert the power command value into a command component in a d-axis and q-axis coordinate system to obtain a dq-axis power command value; Performing a difference operation on the dq axis power command value and the current dq axis current value to obtain a dual axis current error; Performing a proportional integral operation on the dual-axis current error to obtain a basic voltage vector, and performing a product operation on the basic voltage vector and a modulation depth of a PWM modulation signal to obtain a modulation compensation signal; The modulation compensation signal is subjected to space vector transformation to obtain a three-phase control quantity, and a PWM drive signal with switch delay compensation is generated according to the three-phase control quantity to obtain the control quantity.

8. A bidirectional PCS converter power balance control device, characterized in that: The bidirectional PCS converter power balance control device comprises: A timing compensation module, used for dynamically compensating the comparison value and the period value of the PWM carrier counter according to the time difference between the sampling interruption timestamp of the ADC sampling of the bidirectional PCS converter and the count value of the PWM timer, so as to obtain the optimal sampling time of the PWM period; A switch delay modulation module is used to calculate the system resonant frequency according to the inductance and capacitance parameters of the LCL filter network in the bidirectional PCS converter, and dynamically modulate the switch delay time of the IGBT power tube in the bidirectional PCS converter according to the ratio of the system resonant frequency to the current switching frequency to obtain a compensated switch delay voltage value; A resonance suppression module, used to establish a resonance compensation channel according to the system resonance frequency, and use the compensated switch delay voltage value as a feedforward compensation amount to obtain a PWM modulation signal through weighted combination processing; The power control module is used to perform bandwidth limiting processing on the real-time power value obtained by power calculation at the optimal sampling time of the PWM period according to the system resonant frequency, obtain a power command value, and adjust the PWM modulation signal according to the power command value to obtain a control amount for controlling the charging and discharging power of the bidirectional PCS converter.

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