Control method of multi-active bridge transformer, electronic equipment and storage medium

Abstract

The embodiment of the invention provides a multi-active-bridge transformer control method, electronic equipment and a storage medium, and belongs to the technical field of controller design. The method comprises the following steps: observing actual output voltage through a linear extended state observer to obtain a first generalized total disturbance quantity estimated value, and observing the actual output voltage through a nonlinear extended state observer to obtain a second generalized total disturbance quantity estimated value. And obtaining a third generalized total disturbance quantity estimated value according to the first generalized total disturbance estimated value and the second generalized total disturbance estimated value, and performing error compensation according to the third generalized total disturbance quantity estimated value, a proportional controller, a critical gain and the reference voltage. According to the embodiment of the invention, the advantages of the linear extended state observer and the nonlinear extended state observer can be combined, the coupling influence between ports can be reduced, and real-time estimation and total disturbance compensation are realized to ensure the stability of the system.

Description

Technical Field

[0001] The present application belongs to the technical field of controller design, and in particular relates to a control method, electronic device and storage medium for a multi-active bridge transformer. Background Art

[0002] With the development of distributed renewable energy and DC loads, multi-port power electronic transformers are increasingly used in DC power distribution. Modular multi-active bridge transformers based on high-frequency AC busbars have attracted widespread attention due to their scalability, isolation, and independence between ports. Existing technologies mainly use a single linear expansion observer or a single nonlinear or state observer to adaptively control the current source submodule of the multi-active bridge transformer. However, as the number of ports in the multi-active bridge transformer increases (that is, the number of supported voltage source submodules and current source submodules), the existing state observer control strategy cannot solve the step phenomenon caused by multiple ports in the system, and the control system of the multi-active bridge transformer has poor disturbance resistance. Summary of the invention

[0003] In order to solve the above problems, the present application proposes a control method, electronic device and storage medium for a multi-active bridge transformer, which can more accurately control each current source sub-module in the multi-source bridge transformer in the scenario of multi-port coupling of the system, thereby improving the anti-disturbance capability of the control system of the multi-active bridge transformer.

[0004] To achieve the above-mentioned purpose, in a first aspect, an embodiment of the present application provides a control method for a multi-active bridge transformer, wherein the multi-active bridge transformer includes a plurality of circuit modules, wherein the plurality of circuit modules include a plurality of voltage source sub-modules and a plurality of current source sub-modules, wherein the voltage source sub-modules are connected to a primary winding of a high-frequency AC bus, and each of the current source sub-modules is connected to a secondary winding of a high-frequency AC bus; the control method is applied to each of the current source sub-modules, and the control method includes:

[0005] Acquiring a reference voltage of a current source submodule and an actual output voltage of the current source submodule;

[0006] The actual output voltage is observed by a linear extended state observer to obtain a first generalized total disturbance estimation value;

[0007] The actual output voltage is observed by a nonlinear extended state observer to obtain a second generalized total disturbance estimation value;

[0008] Obtaining a third generalized total disturbance estimate value according to the first generalized total disturbance estimate value and the second generalized total disturbance estimate value;

[0009] Generate a frequency change value according to the third generalized total disturbance estimation value, the proportional controller, the critical gain and the reference voltage;

[0010] A modulation control signal is generated according to the frequency change value and the nominal switching frequency, so as to perform voltage control on the current source submodule according to the modulation control signal.

[0011] In a second aspect, an electronic device provided according to an embodiment of the present application includes:

[0012] at least one processor;

[0013] at least one memory for storing at least one program;

[0014] When at least one of the programs is executed by at least one of the processors, the control method of the multi-active bridge transformer described in any one of the first aspects is implemented.

[0015] In a third aspect, a computer-readable storage medium is provided according to an embodiment of the application, storing computer-executable instructions, wherein the computer-executable instructions are used to execute the control method of the multi-active bridge transformer described in any one of the first aspects.

[0016] In summary, the above-mentioned embodiments of the present application estimate the actual output voltage through both the linear extended state observer and the nonlinear extended state observer, so that the nonlinear extended state observer estimates the disturbance estimation output by the linear extended state observer of the previous stage, thereby improving the tracking speed and estimation accuracy of the system for disturbances. Compared with the related art, the embodiments of the present application combine the advantages of the linear extended state observer and the nonlinear extended state observer, can reduce the coupling effect between ports, realize real-time estimation and compensation of total disturbances to ensure system stability. BRIEF DESCRIPTION OF THE DRAWINGS

[0017] Figure 1 A schematic structural diagram of an embodiment of a multi-active bridge transformer to which the control method for the multi-active bridge transformer provided in the present application is applied;

[0018] Figure 2 A schematic diagram of the control principle of the control system corresponding to each current source submodule of the multi-active bridge transformer provided in the present application;

[0019] Figure 3 A flow chart of a control method for a multi-active bridge transformer provided in the present application;

[0020] Figure 4 Bode diagram of the disturbance sensitivity transfer function of the control system applied to the control method of the multi-active bridge transformer provided in the present application (eN1=0.05);

[0021] Figure 5Bode diagram of noise sensitivity transfer function of different observer bandwidths of the control system applied to the control method of the multi-active bridge transformer provided in the present application;

[0022] Figure 6 A simulation diagram of the actual voltage output value of each current source submodule at the port when a small load disturbance occurs at the 220V port of the control system applied to the control method of the multi-active bridge transformer provided in the present application;

[0023] Figure 7 A simulation diagram of the actual output value of the port voltage of each current source submodule when the voltage fluctuation occurs at the 220V port of the control system applied to the control method of the multi-active bridge transformer provided in the present application;

[0024] Figure 8 A simulation diagram of the actual voltage output value of each current source submodule port when the load of the 220V port changes suddenly from 50% to 100% in the control system applied by the control method of the multi-active bridge transformer provided in the present application;

[0025] Fig. 9 A simulation diagram of the actual voltage output value of each current source submodule port when the load of the 48V port changes suddenly from 50% to 100% in the control system applied by the control method of the multi-active bridge transformer provided in the present application;

[0026] Fig.10 A schematic diagram of the hardware structure of the electronic device provided in this application. DETAILED DESCRIPTION

[0027] In order to make the purpose, technical solution and advantages of the present application more clearly understood, the present application is further described in detail below in conjunction with the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present application and are not used to limit the present application.

[0028] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as those commonly understood by those skilled in the art to which this application belongs. The terms used herein are only for the purpose of describing the embodiments of this application and are not intended to limit this application.

[0029] With the development of distributed renewable energy and DC loads, multi-port power electronic transformers are increasingly used in DC power distribution. Modular multi-active bridge (MAB) transformers based on high-frequency AC buses have attracted widespread attention due to their scalability, isolation, and independence between ports. However, the strong coupling between ports poses a challenge to dynamic and precise decoupling control under normal operation and fault conditions, and the control complexity increases significantly with the increase in the number of ports, requiring a more in-depth and detailed study of its control strategy. Most of the existing control strategies for multi-active bridge transformers belong to centralized control, that is, all measurements, digital control execution, and drive signal generation are concentrated in one processor. Although the centralized control architecture is more orderly, the computational burden, modularity, and scalability become major issues, and it is not conducive to hot-swap and high-reliability fault-through operation. Another solution is a decentralized control strategy, which distributes the computational burden of a central controller to several local controllers, thereby effectively improving the modularity, flexibility, and cost-effectiveness of the control system, but its technical difficulty lies in the control coordination and synchronization between multiple modules. In some scenarios, the decentralized control strategy uses additional communication links, such as EtherCAT and RS485, to achieve PWM synchronization, and requires the use of additional hardware. In other scenarios, although a decentralized control strategy for multi-active bridge transformers without communication is proposed to control the switching frequency of each sub-module to perform adaptive adjustment according to local information, realize the global synchronization of PWM without communication, and further improve modularity and flexibility. However, these adaptive control methods mainly solve the problem of power flow management, and do not adopt a robust decoupling mechanism to deal with step changes in the system, such as transient response deviation under load step changes. Therefore, it is still necessary to optimize the control strategy of multi-active bridge decentralized control to improve the system's ability to cope with step changes. Although a variety of control strategies have been proposed for step change scenarios, such as a multi-active bridge control strategy based on linear anti-disturbance control, specifically, by treating power / current coupling as a total disturbance term and eliminating it in the control loop by active compensation, the originally coupled multi-input and multi-output system is decomposed into an independent single-input and single-output system, achieving a control effect that is better than traditional feedforward decoupling and more independent active bridge module operation control. Another example is the use of sliding mode control (SMC) based on extended state observer (ESO) with traditional gain inversion matrix to completely eliminate the cross coupling in the three-active bridge converter. The combination of traditional decoupling method and SMC decoupling method enhances the robustness of the control system, and ESO can solve the chattering problem in SMC and improve the control performance.

[0030] However, the above control strategy is mainly based on centralized control, and the proposed decentralized control strategy is relatively simple, which cannot meet the requirements of stable and high-quality operation of multi-active bridge transformers under complex operating conditions. Based on this, the present application provides a control method, electronic device and storage medium for multi-active bridge transformers, which can more accurately control each current source submodule in the multi-source bridge transformer in the scenario of multi-port coupling of the system, and improve the anti-disturbance capability of the control system of the multi-active bridge transformer.

[0031] The control method provided in the present application is mainly used to independently control each current source submodule in a multi-active bridge transformer to achieve decentralized control of each current source submodule. The embodiment of the present application does not limit the number of current source submodules of the multi-active bridge transformer, and those skilled in the art can selectively set it according to actual needs.

[0032] The embodiment of the present application does not limit the control method of each voltage source submodule in the multi-active bridge transformer. The present application can be applied to the scenario where the number of submodules of the multi-active bridge transformer based on the high-frequency AC bus is greatly increased, and meet the requirements of stable and high-quality operation of the multi-active bridge transformer under complex operating conditions.

[0033] Reference Figure 1 As shown, taking the case where two current source submodules are provided, Figure 1 is a schematic diagram of the structure of an embodiment of a multi-active bridge transformer to which the control method of this embodiment is applied. Figure 1 As shown, the multi-active bridge transformer is a three-active bridge transformer, which includes a voltage source sub-module and two current source sub-modules; the voltage source sub-module includes a DC voltage source, an H-bridge inverter and a transformer; the current source sub-module includes a DC current source, an H-bridge inverter and a transformer, each voltage source sub-module is connected to the primary side winding of the high-frequency AC bus, and each current source sub-module is connected to the secondary side winding of the high-frequency AC bus.

[0034] Reference Figure 2 As shown, the embodiment of the present application provides a control system for a multi-active bridge transformer, the control system is set in one-to-one correspondence with the current source submodule of the multi-active bridge transformer, and the control system includes a nonlinear extended state observer, a linear extended state observer, a proportional gain device, a critical gain device, and a pulse width modulation device. The actual output voltage is observed by the linear extended state observer to obtain a first generalized total disturbance amount estimate; the actual output voltage is observed by the nonlinear extended state observer to obtain a second generalized total disturbance amount estimate; according to the third generalized total disturbance amount estimate, the proportional controller, the critical gain, and the reference voltage, a frequency change value is generated, and according to the frequency change value and the nominal switching frequency, a modulation control signal is generated to perform voltage control on the current source submodule according to the modulation control signal.

[0035] Figure 3 is a flow chart of the control method of the embodiment of the present application, referring to Figure 3 As shown, the control method of each current source submodule in the present application includes steps S101 to S107, which are specifically as follows:

[0036] Step S101, obtaining a reference voltage of a current source submodule and an actual output voltage of the current source submodule;

[0037] Step S102, observing the actual output voltage through a linear extended state observer to obtain a first generalized total disturbance estimation value;

[0038] Step S103, observing the actual output voltage through a nonlinear extended state observer to obtain a second generalized total disturbance estimation value;

[0039] Step S104, obtaining a third generalized total disturbance estimate value according to the first generalized total disturbance estimate value and the second generalized total disturbance estimate value;

[0040] Step S105, generating a frequency change value according to the third generalized total disturbance estimation value, the proportional controller, the critical gain and the reference voltage;

[0041] Step S106 , generating a modulation control signal according to the frequency change value and the nominal switching frequency, so as to perform voltage control on the current source submodule according to the modulation control signal.

[0042] Therefore, the actual output voltage is estimated by both the linear extended state observer and the nonlinear extended state observer, and the nonlinear extended state observer estimates the disturbance estimation of the linear extended state observer output at the previous stage, thereby improving the tracking speed and estimation accuracy of the system for disturbances. Compared with the related art, the embodiment of the present application combines the advantages of the linear extended state observer and the nonlinear extended state observer, can reduce the coupling effect between ports, and realize real-time estimation and compensation of total disturbances to ensure system stability.

[0043] In the embodiment of the present application, the linear extended state observer can adopt the linear extended state observer (LESO) used by the first-order LADRC to ensure that the system can operate stably under large disturbances. The nonlinear extended state observer in the embodiment of the present application can be used to estimate the disturbance estimation output by the previous linear extended state observer (LESO), improve the tracking speed and estimation accuracy of the system for disturbances, and thus improve the anti-disturbance ability of the system. When controlling with reference to steps S101 to S106, taking the 220V port 8kW50% to 100% load mutation, with 2 current source submodules as an example, relative to the traditional PI control, the control method of the present application based on the two current source submodules The adjustment time and transient voltage peak value are reduced by about 68.52%, 32.24% and 36.63%, 63.44% respectively compared with the PI control, and have faster adjustment time and stronger resistance to load step mutation.

[0044] In some embodiments, the linear extended state observer and the nonlinear extended state observer can estimate the uncertain disturbance in the time-frequency domain, and the third generalized total disturbance estimate includes the uncertain disturbance. When it is necessary to perform linear compensation of the uncertain disturbance for error compensation, it can be calculated directly based on the third generalized total disturbance estimate, the proportional controller, the critical gain and the reference voltage. For example, the third generalized total disturbance estimate includes the uncertain disturbance, that is, the estimated error of the disturbance by the nonlinear extended state observer. For example, f w Characterizes the uncertain disturbance, then when performing linear compensation, by offsetting f w Achieve accurate estimation of error compensation.

[0045] The proportional gain device is used to amplify the estimated value of the third generalized total disturbance to obtain an amplified error signal, and the critical gain is used to perform gain compensation on the amplified value. In some embodiments, the amplified voltage error and the third generalized total disturbance can be added together and then error compensated through the critical gain device to obtain a frequency change value.

[0046] In some embodiments, the state space expression (1) of the linear extended state observer and the state space expression (2) of the nonlinear extended state observer can be respectively expressed as follows:

[0047]

[0048] Wherein, z1 is the estimated value of the port voltage of the linear extended state observer linear current source submodule (that is, the estimated value of the output voltage), is the differential of z1, z2 is the first generalized total disturbance estimate of the linear extended state observer, is the differential of z2, β L1and β L2 is the observer gain of the linear extended state observer, e L1 is the estimated value of the port voltage of the current source submodule i and the actual output voltage v by the linear extended state observer i The error, b 0i represents the critical gain of the current source submodule i, is the phase shift angle of the current source submodule i, is the phase shift angle gain error.

[0049] Among them, s1 is the estimated value of the port voltage of the current source submodule i by the nonlinear extended state observer, s2 is the estimated value of the second generalized total disturbance by the nonlinear extended state observer, β N1 and β N2 is the observer gain of the nonlinear extended state observer, e N1 is the estimated value of the port voltage by the nonlinear extended state observer and the actual output voltage v i The error, b 0i is the critical gain of the current source submodule i, is the phase shift angle of the current source submodule i; fal(e N1 ,α2,δ), fal(e N1 ,α1,δ) are based on e N1 The nonlinear function of .

[0050] In some embodiments, based on the state space expression (1) and the state space expression (2), the control method further includes: to update.

[0051] In some embodiments, the reference voltage of the current source submodule is the voltage value when the current source submodule is expected to work normally, which can be set according to requirements. The actual output voltage is the output voltage of the current source submodule collected in real time by a voltage sensor or other collection device.

[0052] In some embodiments, generating a frequency change value according to the third generalized total disturbance estimation value, a proportional controller, a critical gain, and a reference voltage may include but is not limited to the following steps:

[0053] Obtaining a voltage error between a reference voltage of the current source submodule and an actual output voltage of the current source submodule;

[0054] The voltage error is amplified by a proportional controller;

[0055] According to the critical gain, the amplified voltage error and the third generalized total disturbance estimation value, the disturbance compensation is performed to obtain the frequency change value.

[0056] In some embodiments, the frequency change value is defined as Among them, f wi (t,v i ,ω) represents the total disturbance amount of the system, including the cross-coupling terms with the remaining submodules. The third generalized total disturbance amount estimate is an estimate of this amount. That is, in some embodiments, the frequency change value can be defined as Among them, u i is the error compensation value, b 0i is the critical gain of the current source submodule i, r i is the reference voltage of the current source submodule i, v i is the actual output voltage of the current source submodule i, s2 is the second generalized total disturbance estimate of the nonlinear extended state observer, z2 is the first generalized total disturbance estimate of the linear extended state observer, K p_i is the proportional gain of the proportional controller of the current source submodule i. In some embodiments, the proportional gain of the proportional controller can be obtained by performing simulation experiment data.

[0057] In some embodiments, the linear extended state observer and the nonlinear extended state observer of each current source submodule satisfy the following state space expression (3):

[0058]

[0059] Where z1 is the estimated value of the port voltage of the current source submodule i by the linear extended state observer, is the differential of z1, z2 is the first generalized total disturbance estimate of the linear extended state observer, is the differential of z2, s1 is the estimated value of the port voltage of the current source submodule i by the nonlinear extended state observer, is the differential of s1, s2 is the second generalized total disturbance estimate of the nonlinear extended state observer, is the differential of s2, β L1 and β L2 is the linear extended state observer gain, β N1 and β N2 is the nonlinear extended state observer gain, λ1(e N1 ) represents the nonlinear variable associated with the rate of change of the estimated value of the port voltage, corresponding to the e-based N1 The linear transformation law, λ2(e N1 ) represents the change in the estimated value of the second generalized total disturbance and e N1 The linear transformation law between P is the proportional gain of the proportional controller, r i is the reference voltage of the current source submodule i, v iis the actual output voltage of the current source submodule i.

[0060] It can be understood that the state space expression is determined based on the linear expanded state observer state space expression (the above state space expression (1)), the nonlinear expanded state observer state space expression (the above state space expression (2)) and the error compensation value to ensure that when the linear expanded state observer and the nonlinear expanded state observer observe the actual data value of the voltage, the accuracy of the error compensation meets the control requirements.

[0061] In some embodiments, the nonlinear function fal(e N1 ,α i ,δ) satisfies expression (4):

[0062]

[0063] Among them, e N1 is the error between the estimated value of the port voltage of the current source submodule by the nonlinear extended state observer and the actual output voltage, α i is the exponential power, δ is the length of the linear interval that can affect the filtering effect, and Sign() is the sign function. i , δ are the variable coefficients of the nonlinear function.

[0064] In some embodiments, a small signal model can be introduced to observe the frequency change of each current source submodule, the power change of the voltage source submodule, and the voltage change of the current source submodule. At this time, based on the small signal model, the relationship between the voltage change rate and the phase shift angle of each circuit module can be obtained as follows:

[0065]

[0066] b 0i is the critical gain of the current source submodule i, is the phase shift angle of the current source submodule i, is the actual output voltage differential of the current source submodule, f wi (t,v i ,ω) is the generalized total disturbance of the current source submodule i, v i is the actual output voltage of the current source submodule.

[0067] In some embodiments, in order to facilitate the performance calculation of the control system, the nonlinear function fal(e N1 ,α i ,δ) is linearized as shown in the following formula (6):

[0068]

[0069] According to (1), (2), (5) and (6), it can be deduced that the linear extended state observer and the nonlinear extended state observer of each current source submodule satisfy the state formula (3). At this time, the linear extended state observer and the nonlinear extended state observer can calculate f wi (t,v i ,ω) for real-time observation, then the error compensation is satisfied

[0070] Taking the case where two current source submodules are provided, namely current source submodule 2 and current source submodule 3, the stability, anti-disturbance sensitivity and noise sensitivity of the control system designed based on the state formula (3) are analyzed as follows:

[0071] (I) Stability analysis is as follows:

[0072] According to formula (5), the voltage change rate of current source submodule 2 and current source submodule 3 can be obtained as shown in the following formula (7):

[0073]

[0074] b 02 is the critical gain of the current source submodule 2, is the phase shift angle of current source submodule 2, is the actual output voltage differential of the current source submodule 2, f w2 (t,v2,ω) is the total disturbance of current source submodule 2, b 03 is the critical gain of the current source submodule 3, is the phase shift angle of the current source submodule 3, is the actual output voltage differential of the current source submodule 3, f w3 (t,v3,ω) is the total disturbance of current source submodule 3.

[0075] At this time, the observer disturbance estimation error can be obtained That is, the observer estimation error is as follows (8)

[0076]

[0077] in, is the observer disturbance estimation error, β N2 is the nonlinear extended state observer gain, λ2(e N1 ) represents the change in the estimated value of the second generalized total disturbance and e N1 The linear transformation law between L2 is the linear extended state observer gain, s is the complex variable in Laplace transform, v i (s) is the actual output voltage of current source submodule i, b 0iis the critical gain of the current source submodule i, is the phase shift angle of the current source submodule i; A and B satisfy the following formula (9):

[0078]

[0079] Where s is the complex variable in Laplace transform, β L1 and β L2 is the linear extended state observer gain, β N1 and β N2 is the nonlinear extended state observer gain, λ1(e N1 ) represents the nonlinear variable associated with the rate of change of the estimated value of the port voltage, corresponding to the e-based N1 The linear transformation law, λ2(e N1 ) characterizes the linear transformation law between the change in the estimate of the second generalized total disturbance and eN1.

[0080] Based on (8) to (9), we can get formula (10):

[0081]

[0082] Among them, β L1 is the linear extended state observer gain, β N1 is the nonlinear extended state observer gain, λ1(e N1 ) represents the nonlinear variable associated with the rate of change of the estimated value of the port voltage, corresponding to the e-based N1 The linear transformation law, λ2(e N1 ) represents the change in the estimated value of the second generalized total disturbance and e N1 The linear transformation law between them.

[0083] Therefore, when the selected observer gains are all greater than 0, that is, β L1 >0, β L2 >0, β N1 >0, β N2 >0. The linear extended state observer gain determined by the bandwidth method Therefore, in (10) That is, the roots of (9) are all distributed in the left half plane of s, so the control system is stable.

[0084] (II) Anti-disturbance sensitivity analysis is as follows:

[0085] According to (2), (5) and (6), we can deduce that the third generalized total disturbance (z2+s2) and the total disturbance f wi (t,v i ,ω) is as follows:

[0086]

[0087] Among them, β N2 is the nonlinear extended state observer gain, λ1(e N1 ) represents the nonlinear variable associated with the rate of change of the estimated value of the port voltage, corresponding to the e-based N1 The linear transformation law, λ2(e N1 ) represents the change in the estimated value of the second generalized total disturbance and e N1 The linear transformation law between L1 and β L2 is the linear extended state observer gain, and s is the complex variable in the Laplace transform.

[0088] Then, combining (7), (11) and The disturbance sensitivity function G can be obtained f_N (s) as formula (12):

[0089]

[0090] G f_N (s) is the disturbance sensitivity function, G Z_N (s) is the transfer function, s is the complex variable in Laplace transform, K p_i is the proportional gain of the proportional controller of the current source submodule i.

[0091] At this time, based on (12), simulation can be obtained: Figure 4 The Bode diagram of the disturbance sensitivity transfer function (eN1 = 0.05) is shown, where the phase and amplitude are coincident when the bandwidth ω is set to 100 and 1000. Figure 4 From the curves of , it can be seen that increasing the bandwidth ω of the extended state observer can reduce the amplitude of the disturbance sensitivity transfer function of the system using the cascade observer in the medium and low frequency bands, thereby improving the anti-disturbance capability. At the same time, under the same bandwidth, the amplitude of the anti-disturbance frequency characteristic of the cascaded linear-nonlinear extended state observer system is lower than that of the first-order linear extended state observer and the active disturbance rejection control based on the cascaded linear extended state observer, indicating that its anti-disturbance performance is stronger, verifying that the introduction of the nonlinear extended state observer system can improve the anti-disturbance performance of the system.

[0092] (III) Noise sensitivity analysis is as follows:

[0093] Since the observer bandwidth ω is also limited by the sensor measurement noise and sampling frequency, considering the influence of the measurement noise n, the error compensation is expressed as formula (13):

[0094]

[0095] Then the third generalized total disturbance can be expressed as formula (14):

[0096] z2+s2=G z_N (s)[f wi (s)+sn] (14);

[0097] Therefore, based on (7), (13) and (14), the measurement noise n and the actual output voltage v of the current source submodule can be obtained: i The transfer function between them is the noise sensitivity function G of the system. n (s) is shown in formula (15):

[0098]

[0099] Based on formula (15), Figure 5 The Bode plots of the noise sensitivity transfer function for different observer bandwidths are shown in Figure 2. Figure 5 , increasing the bandwidth of the two extended state observers will weaken the system's ability to suppress high-frequency noise, which is exactly the opposite of the change in anti-disturbance performance with bandwidth. Therefore, by reasonably selecting the bandwidth of the linear extended state observer and the nonlinear extended state observer, the system's ability to suppress noise can be improved and strong anti-disturbance performance can be maintained.

[0100] In summary, the design of the linear extended state observer and the nonlinear extended state observer based on (3) can meet the control requirements.

[0101] In some embodiments, the critical gain, β L1 , β L2 , β N1 , β N2 and K p The value of is determined by the following steps:

[0102] determining a critical gain initial value of the critical gain;

[0103] Based on the initial value of critical gain, β is determined by bandwidth method. L1 , β L2 The initial value of the linear observer gain and the determination of β by the bandwidth method N1 and β N2 The initial value of the nonlinear observer gain;

[0104] Based on the critical gain initial value, the initial values ​​of each linear observer gain and each nonlinear observer gain are optimized to determine β L1 , β L2 , β N1 and β N2 Parameter value of

[0105] Determining the variable coefficient of the nonlinear function according to the preset disturbance control condition;

[0106] Through trial and error and β L1 , β L2 , β N1 , β N2 The parameter value determines the value of the critical gain and the value of the proportional gain.

[0107] For example, assuming that the initial value of the critical gain is b0, the gain of the linear extended state observer is adjusted according to the bandwidth method, such as β L1 =2ω0, ω0 is the observer bandwidth. Similarly, β N1 , β N2 Then, in order to avoid oscillation of the system, we choose N2 For optimization, such as selecting β N1 =2ω 01 , At the same time, based on the fact that the bandwidth of the latter observer is greater than that of the former observer, it can more accurately estimate the disturbance estimation residual of the former output, and weigh the anti-disturbance performance and noise suppression performance of the system, the bandwidth of the former linear extended state observer is determined to be ω 0_1 =400,ω 0_2 =600, and the bandwidths of the subsequent nonlinear extended state observers are ω 1_1 =800,ω 1_2 =700. Then based on ω 0_1 =400,ω 0_2 =600,ω 1_1 =800,ω 1_2 =700 for simulation, L1 , β L2 , β N1 and β N2 The heuristic method is used to optimize and obtain β N1_1 =24ω 1_1 =19200, β N1_2 =6ω 1_2 =4200,

[0108] For the nonlinear function fal(e N1 ,α i ,δ), whose variable coefficients include α i and δ; α i The smaller the value of , the faster the tracking speed of the noise signal, but the filtering effect will be worse. The linear interval length δ is just the opposite. Increasing δ can improve the filtering effect, but it also increases the tracking delay. In some embodiments, α iThe value of δ is between 0 and 1; the value range of δ is: 0.01<δ<0.1. i There is no restriction on how α1=0.5, α2=0.3, and δ=0.01 can be selected based on the anti-disturbance control requirements of the DC transformer.

[0109] Exemplarily, the critical gain of the controller and the proportional gain Kp in the state error feedback control rate are determined, and according to the transient response characteristics expected to be achieved by the three-active bridge DC transformer, the critical gain and the proportional gain Kp are synchronously adjusted through trial and error to determine the critical gain and the proportional gain Kp.

[0110] In some embodiments, determining a critical gain initial value of a critical gain includes:

[0111] Obtaining a small signal model of a current source submodule and a circuit module port average current variation matrix, wherein the circuit module port average current variation matrix represents a relationship between a phase shift angle and a partial differential of each circuit module in the same multi-active bridge transformer;

[0112] Determine the theoretical critical gain based on the small signal model and the average current variation matrix of the circuit module port;

[0113] According to the preset total disturbance formula and theoretical critical gain, the initial value of the critical gain is determined. The total disturbance formula is used for observing the disturbance in the time-frequency domain by the linear extended state observer and the nonlinear extended state observer.

[0114] In some embodiments, the small signal model is shown in equation (16):

[0115]

[0116] in, is the change in output power of voltage source submodules 1 to k, is the change in the actual output voltage of the current source submodule k+1~n, is the actual output voltage of the voltage source submodules 1 to k, is the change in the switching frequency of the voltage source submodules 1 to k, is the change in the switching frequency of the current source submodules k+1~n, s is the complex variable in the Laplace transform, H K+1 ~H n is the relationship between the actual output voltage and actual output current of the current source submodules k+1~n, G kn It is the partial differential of the average current of circuit module k with respect to the phase shift angle of circuit module n.

[0117] In some embodiments, the small signal model can be derived and determined by the following steps:

[0118] (I) According to the switching frequency f of circuit module i si and the switching frequency f of circuit module j sj , determine the phase shift angle between circuit modules i and j Formula (17):

[0119]

[0120] is the phase shift angle between circuit module i and circuit module j, f si is the switching frequency of circuit module i, f sj is the switching frequency of circuit module j;

[0121] (ii) The expression (18) of the transmission power between each circuit module is as follows:

[0122]

[0123] P ij is the transmission power between each circuit module, n ij is the transformation ratio of the transformer between each circuit module, L ij is the equivalent inductance, v i is the actual output voltage of circuit module i, v j is the actual output voltage of circuit module j, is the phase shift angle between each circuit module; f is the switching frequency of the circuit module.

[0124] (III) Introduce the small signal disturbance of the phase shift angle of the corresponding circuit module to each port of TAB, and then calculate the partial differential G of the average current of each circuit module with respect to the phase shift angle of the circuit module at the steady-state operating point Q. ij , ignoring small signal disturbances and disturbances of the second order or higher, and performing linearization, and obtaining linearization parameters, the average current variation matrix of the circuit module port can be obtained to satisfy the following formula (19):

[0125]

[0126] is the change in the average current of circuit module i, is the change in phase shift angle of circuit module i, G ij It is the partial differential of the average current of circuit module i with respect to the phase shift angle of circuit module j.

[0127] (IV) According to the resistance load R connected to the circuit module i i , capacitor C connected to circuit module i iAnd the complex variable s in the Laplace transform, based on Kirchhoff's current law, the relationship between the actual output voltage and current of the circuit module is shown in equation (20):

[0128]

[0129] v i is the terminal voltage of circuit module i, I i is the port current of circuit module i, R i is the resistance load connected to circuit module i, s is the complex variable in Laplace transform, C i is the capacitance connected to submodule i.

[0130] In summary, based on (17), (19) and (20), we can obtain (16).

[0131] In some embodiments, the total disturbance satisfies equation (21):

[0132]

[0133] Among them, v i is the actual output voltage of circuit module i, C i is the capacitor connected to circuit module i, R i is the resistance load connected to circuit module i, is the phase shift angle of circuit module i, G ij is the partial differential of the average current of circuit module i relative to the phase shift angle of circuit module j, b i represents the theoretical critical gain of circuit module i, b 0i represents the critical gain of the current source submodule i.

[0134] In some embodiments, after generating the error compensation value, the method further comprises:

[0135] Obtaining the critical gain of the current source submodule;

[0136] According to the error compensation value and the critical gain of the current source submodule, the phase-shift angle gain errors of the linear extended state observer and the nonlinear extended state observer are updated respectively, wherein the phase-shift angle gain error of the linear extended state observer represents the relationship between the rate of change of the voltage estimate of the linear extended state observer, the first generalized total disturbance estimate and the voltage gain error; the phase-shift angle gain error of the nonlinear extended state observer represents the relationship between the rate of change of the voltage estimate of the nonlinear extended state observer, the first generalized total disturbance estimate and the voltage gain error.

[0137] The following is based on Figure 1The structure diagram of the multi-active bridge transformer shown in FIG. 1 is a process simulation of voltage control of the current source based on PI control, first-order LADRC and the above-mentioned method of the present application, wherein the parameters of the multi-active bridge transformer are shown in Tables 1 to 4 below, the controller parameters corresponding to the first-order LADRC are shown in Table 3, and the critical gain, β in the method of the present application are shown in Table 3. L1 , β L2 , β N1 , β N2 , K p And the variable coefficients of the nonlinear function are shown in Table 4 for simulation experiments.

[0138] Among them, Tables 1 to 4 are as follows:

[0139]

[0140]

[0141] Table 1

[0142] Among them, the rated voltages VdN1, VdN2, and VdN3 represent the rated voltages of the voltage source submodule, the current source submodule 2, and the current source submodule 3, respectively; the inductors L1, L2, and L3 represent the inductors of the voltage source submodule, the current source submodule 2, and the current source submodule 3, respectively. The capacitors C2 and C3 represent the capacitances of the current source submodule 2 and the current source submodule 3, respectively; and the rated loads R2 and R3 represent the rated loads of the current source submodule 2 and the current source submodule 3, respectively.

[0143] PI controller parameters <![CDATA[k p ]]> <![CDATA[k i ]]> Current source submodule 2 (220V) 15 3000 Current source submodule 3 (48V) 30 3000

[0144] Table 2

[0145] LADRC controller parameters <![CDATA[ω0]]> <![CDATA[ω c ]]> <![CDATA[b0]]> Current source submodule 1 (220V) 40000 240.79 1022.3 Current source submodule 2 (48V) 50000 150.17 720.57

[0146] Table 3

[0147] <![CDATA[K p ]]> <![CDATA[b0]]> <![CDATA[β L1 ]]> <![CDATA[β L2 ]]> <![CDATA[β N1 ]]> <![CDATA[β N2 ]]> <![CDATA[α1]]> <![CDATA[α2]]> δ 220V 10000 255.57 800 160000 19200 53333 0.5 0.3 0.01 48V 5500 72.057 1200 360000 4200 10208 0.5 0.3 0.01

[0148] Table 4

[0149] Among them, Table 3 and Table 4 record the parameter values ​​of each controller under 220V and 48V.

[0150] Therefore, the simulation test is performed based on Tables 1 to 4 as follows:

[0151] (I) Small load disturbance:

[0152] like Figure 6As shown in the simulation diagram of the actual output value of the port voltage of each current source submodule when a small load disturbance occurs at the 220V port, at 0.3s, a resistance load of 50Ω (where 50Ω simulates a small load disturbance occurring at the connection port of the TAB DC transformer during normal operation) is connected in parallel to the current source submodule 1 (220V), and the port resistance change of the current source submodule 1 is approximately -2.6983%. As shown in the figure, when facing a small load disturbance at the port, different control strategies (based on first-order LADRC, the control method of the present application, and decentralized PI) are used to control the actual output voltage of the two current source submodules. Figure 6 As shown, the quantitative performance comparison analysis is shown in Table 5:

[0153]

[0154]

[0155] Table 5

[0156] according to Figure 6 It can be seen from Table 5 that the steady-state response characteristics of the current source submodule using CL-NLADRC (that is, the control method of the present application) are close to PI control and first-order LADRC, the port voltage offset is about 0, and the port voltage ripple is relatively small. In terms of transient response characteristics, the two current source submodules use CL-NLADRC to deal with small load disturbances. The transient voltage peak is the smallest, the adjustment time is short, and it has a strong anti-disturbance ability. According to Table 5, it can be calculated that the transient voltage peak of CL-NLADRC is reduced by about 31.86%, 57.70% and 27.00%, 54.39% compared with PI control and first-order LADRC, respectively, and the adjustment time is 58.09%, 26.90% and 61.31%, 31.22% of PI control and first-order LADRC, respectively.

[0157] In summary, compared with PI control and first-order LADRC, the TAB DC transformer current source submodule using CL-NLADRC can also achieve basically no steady-state error and small output port voltage ripple. At the same time, CL-NLADRC has better transient response characteristics and can effectively suppress small load disturbances caused by step changes.

[0158] (ii) Voltage disturbance:

[0159] Figure 7 This is a simulation diagram of the actual output value of the port voltage of each current source submodule when the voltage fluctuation occurs at the port of the 220V current source submodule. Figure 7As shown, the voltage at the port of the current source submodule 1 is set to increase by 5V (the change is about 2.27%) at 0.3s, and then the port voltage is set to decrease by 5V at 0.45s, which is used to simulate the small voltage fluctuation at the connection port of the TAB DC transformer during normal operation. Figure 7 As shown in Figure 6, the dynamic waveforms of the output voltages of the two current source sub-modules under different control strategies of PI control, first-order LADRC and CL-NLADRC are compared and analyzed quantitatively.

[0160]

[0161] Table 6

[0162] analyze Figure 7 As shown in Table 6, when the voltage of the current source submodule 1 (220V) suddenly increases (0.3s), the port with voltage fluctuation adopts CL-NLADRC, PI control, and first-order LADRC, and the difference in steady-state and transient response characteristics is not large, and the adjustment time and the port voltage ripple in steady state are slightly larger than PI control. However, for another current source submodule 2 (48V) without voltage fluctuation, CL-NLADRC reduces the voltage sudden increase by 37.89% and 44.88% compared with PI and first-order LADRC, respectively. Similarly, when the port voltage suddenly decreases (0.45s), the steady-state and transient response characteristics of the port with voltage fluctuation are also relatively close when comparing different current source submodule control strategies, but for another port, CL-NLADRC reduces the voltage drop by 30.84% ​​and 40.26% compared with PI and first-order LADRC, respectively.

[0163] Compared with PI control and first-order LADRC control, the TAB DC transformer current source submodule adopts CL-NLADRC, although it cannot improve the transient and steady-state characteristics of the port where voltage fluctuation occurs, it can significantly reduce the output voltage fluctuation phenomenon caused by the coupling relationship between TAB ports at the port where no voltage fluctuation occurs.

[0164] (III) 50% to 100% load mutation:

[0165]

[0166] Table 7

[0167] Among them, Table 7 shows the performance comparison of the current source submodule under different control strategies when the 220V port has a 50% to 100% load mutation; Figure 8 This is a simulation diagram of the actual voltage output value of each current source submodule when the 220V port experiences a 50% to 100% load mutation. Figure 8It can be seen from Table 7 that when the current source submodule 3 experiences a sudden load change from 50% to 100%, the transient voltage peak value of the current source submodule 3 using CL-NLADRC is the smallest, the adjustment time is the shortest, and it has good transient response characteristics and anti-large load disturbance performance. According to Table 7, it can be calculated that the transient voltage peak value of CL-NLADRC is reduced by about 36.63%, 25.61%, 63.44%, 58.09% compared with PI control and first-order LADRC, respectively, and the voltage transient response time is 68.52%, 88.20%, 32.24%, 42.31% of that of PI control and first-order LADRC control, respectively.

[0168]

[0169] Table 8

[0170] Fig. 9 This is a simulation diagram of the actual voltage output value of each current source submodule when the load of the 48V port changes from 50% to 100%. Fig. 9 It can be seen from Table 8 that, similar to the case of a step mutation in the load of the 220V port, when the current source submodule 2 undergoes a 50% to 100% load mutation, the transient voltage peak value of CL-NLADRC is the smallest and the adjustment time is the shortest. It can be calculated from Table 8 that the transient voltage peak value of CL-NLADRC is reduced by about 52.61%, 38.35% and 32.59%, 26.82% compared with PI control and first-order LADRC, respectively, and the voltage transient response time is 54.96%, 65.93% and 44.90%, 60.47% of that of PI control and first-order LADRC control, respectively.

[0171] In summary, when the current source submodule of the TAB DC transformer undergoes a 50% to 100% load mutation, the current source submodule adopts the method based on this application for self-disturbance control, which has better transient response characteristics and stronger anti-disturbance capability than PI control and first-order LADRC. In addition, according to the analysis of Tables 7 and 8, the regulation ability of CL-NLADRC for non-load mutation ports is significantly better than PI control and traditional first-order LADRC, and the non-disturbed ports can quickly restore their normal working state, which is conducive to the distributed application of multi-active bridge DC transformers.

[0172] The embodiment of the present application also provides an electronic device, the electronic device includes a memory and a processor, the memory stores a computer program, and the processor implements the above-mentioned multi-active bridge transformer control method when executing the computer program. The electronic device can be any intelligent terminal including a tablet computer, a car computer, etc.

[0173] See also Fig.10 , Fig.10 The hardware structure of an electronic device of another embodiment is illustrated, and the electronic device includes:

[0174] The processor 901 may be implemented by a general-purpose CPU (Central Processing Unit), a microprocessor, an application-specific integrated circuit (Application Specific Integrated Circuit, ASIC), or one or more integrated circuits, and is used to execute relevant programs to implement the technical solutions provided in the embodiments of the present application;

[0175] The memory 902 can be implemented in the form of a read-only memory (ROM), a static storage device, a dynamic storage device, or a random access memory (RAM). The memory 902 can store an operating system and other application programs. When the technical solution provided in the embodiment of this specification is implemented by software or firmware, the relevant program code is stored in the memory 902, and the processor 901 calls and executes the multi-active bridge transformer control method of the embodiment of this application;

[0176] Input / output interface 903, used to implement information input and output;

[0177] Communication interface 904, used to realize communication interaction between the device and other devices, which can be realized by wired mode (such as USB, network cable, etc.) or wireless mode (such as mobile network, WIFI, Bluetooth, etc.);

[0178] A bus 905 that transmits information between various components of the device (e.g., the processor 901, the memory 902, the input / output interface 903, and the communication interface 904);

[0179] The processor 901 , the memory 902 , the input / output interface 903 and the communication interface 904 are connected to each other in communication within the device via a bus 905 .

[0180] In some embodiments, the embodiments of the present application further provide a computer-readable storage medium, which stores a computer program, and when the computer program is executed by a processor, the above-mentioned multi-active bridge transformer control method is implemented.

[0181] The memory, as a non-transient computer-readable storage medium, can be used to store non-transient software programs and non-transient computer executable programs. In addition, the memory may include a high-speed random access memory, and may also include a non-transient memory, such as at least one disk storage device, a flash memory device, or other non-transient solid-state storage device. In some embodiments, the memory may optionally include a memory remotely disposed relative to the processor, and these remote memories may be connected to the processor via a network. Examples of the above-mentioned network include, but are not limited to, the Internet, an intranet, a local area network, a mobile communication network, and combinations thereof.

[0182] The embodiments described in the embodiments of the present application are intended to more clearly illustrate the technical solutions of the embodiments of the present application and do not constitute a limitation on the technical solutions provided in the embodiments of the present application. Those skilled in the art will appreciate that with the evolution of technology and the emergence of new application scenarios, the technical solutions provided in the embodiments of the present application are also applicable to similar technical problems.

[0183] Those skilled in the art will appreciate that the technical solutions shown in the figures do not constitute a limitation on the embodiments of the present application, and may include more or fewer steps than shown in the figures, or a combination of certain steps, or different steps.

[0184] The device embodiments described above are merely illustrative, and the units described as separate components may or may not be physically separated, that is, they may be located in one place or distributed on multiple network units. Some or all of the modules may be selected according to actual needs to achieve the purpose of the solution of this embodiment.

[0185] Those skilled in the art will appreciate that all or some of the steps in the methods disclosed above, and the functional modules / units in the systems and devices may be implemented as software, firmware, hardware, or a suitable combination thereof.

[0186] The terms "first", "second", "third", "fourth", etc. (if any) in the specification of the present application and the above-mentioned drawings are used to distinguish similar objects, and are not necessarily used to describe a specific order or sequence. It should be understood that the data used in this way can be interchangeable where appropriate, so that the embodiments of the present application described herein can be implemented in an order other than those illustrated or described herein. In addition, the terms "including" and "having" and any of their variations are intended to cover non-exclusive inclusions, for example, a process, method, system, product or device comprising a series of steps or units is not necessarily limited to those steps or units clearly listed, but may include other steps or units that are not clearly listed or inherent to these processes, methods, products or devices.

[0187] It should be understood that in the present application, "at least one (item)" means one or more, and "plurality" means two or more. "And / or" is used to describe the association relationship of associated objects, indicating that three relationships may exist. For example, "A and / or B" can mean: only A exists, only B exists, and A and B exist at the same time, where A and B can be singular or plural. The character " / " generally indicates that the objects associated before and after are in an "or" relationship. "At least one of the following" or similar expressions refers to any combination of these items, including any combination of single or plural items. For example, at least one of a, b or c can mean: a, b, c, "a and b", "a and c", "b and c", or "a and b and c", where a, b, c can be single or multiple.

[0188] In the several embodiments provided in the present application, it should be understood that the disclosed devices and methods can be implemented in other ways. For example, the device embodiments described above are only schematic. For example, the division of the above units is only a logical function division. There may be other division methods in actual implementation, such as multiple units or components can be combined or integrated into another system, or some features can be ignored or not executed. Another point is that the mutual coupling or direct coupling or communication connection shown or discussed can be through some interfaces, indirect coupling or communication connection of devices or units, which can be electrical, mechanical or other forms.

[0189] The units described above as separate components may or may not be physically separated, and the components shown as units may or may not be physical units, that is, they may be located in one place or distributed on multiple network units. Some or all of the units may be selected according to actual needs to achieve the purpose of the solution of this embodiment.

[0190] In addition, each functional unit in each embodiment of the present application may be integrated into one processing unit, or each unit may exist physically separately, or two or more units may be integrated into one unit. The above-mentioned integrated unit may be implemented in the form of hardware or in the form of software functional units.

[0191] If the integrated unit is implemented in the form of a software functional unit and sold or used as an independent product, it can be stored in a computer-readable storage medium. Based on this understanding, the technical solution of the present application, or the part that contributes to the prior art, or all or part of the technical solution can be embodied in the form of a software product, and the computer software product is stored in a storage medium, including multiple instructions to enable a computer device (which can be a personal computer, server, or network device, etc.) to execute all or part of the steps of the methods of various embodiments of the present application. The aforementioned storage medium includes: U disk, mobile hard disk, read-only memory (Read-Only Memory, referred to as ROM), random access memory (Random Access Memory, referred to as RAM), disk or optical disk and other media that can store programs.

Claims

1. A control method for a multi-active bridge transformer, characterized in that: The multi-active bridge transformer includes a plurality of circuit modules, and the plurality of circuit modules are composed of a voltage source submodule and a plurality of current source submodules, wherein the voltage source submodule is connected to a primary winding of a high-frequency AC bus, and each of the current source submodules is connected to a secondary winding of the high-frequency AC bus; The control method is applied to each of the current source sub-modules, and the control method includes: Acquiring a reference voltage of a current source submodule and an actual output voltage of the current source submodule; The actual output voltage is observed by a linear extended state observer to obtain a first generalized total disturbance estimation value; The actual output voltage is observed by a nonlinear extended state observer to obtain a second generalized total disturbance estimation value; Obtaining a third generalized total disturbance estimate value according to the first generalized total disturbance estimate value and the second generalized total disturbance estimate value; Perform error compensation according to the third generalized total disturbance estimation value, the proportional controller, the critical gain and the reference voltage to obtain a frequency change value; A modulation control signal is generated according to the frequency change value and the nominal switching frequency, so as to perform voltage control on the current source submodule according to the modulation control signal.

2. The control method according to claim 1, characterized in that: The error compensation is performed according to the third generalized total disturbance estimation value, the proportional controller, the critical gain and the reference voltage, including: Acquire a voltage error between a reference voltage of the current source submodule and an actual output voltage of the current source submodule; amplifying the voltage error by the proportional controller; Disturbance compensation is performed according to the critical gain, the amplified voltage error and the third generalized total disturbance estimate.

3. The control method according to claim 1, characterized in that: The linear extended state observer and the nonlinear extended state observer of the same current source submodule satisfy the following state space expression: Where z1 is the estimated value of the port voltage of the current source submodule i by the linear extended state observer, is the differential of z1, z2 is the first generalized total disturbance estimate of the linear extended state observer, is the differential of z2, s1 is the estimated value of the port voltage of the current source submodule i by the nonlinear extended state observer, is the differential of s1, s2 is the second generalized total disturbance estimate of the nonlinear extended state observer, is the differential of s2, β L1 and β L2 is the observer gain of the linear extended state observer, β N1 and β N2 is the observer gain of the nonlinear extended state observer, λ1(e N1 ) represents the nonlinear variable associated with the rate of change of the estimated value of the port voltage, corresponding to the e-based N1 The linear transformation law, λ2(e N1 ) represents the change in the estimated value of the second generalized total disturbance and e N1 The linear transformation law between P is the proportional gain of the proportional controller, r i is the reference voltage of the current source submodule i, v i is the actual output voltage of the current source submodule i.

4. The control method according to claim 3, characterized in that: The critical gain, the β L1 , the β L2 , the β N1 , the β N2 And the K p The value of is determined by the following steps: determining a critical gain initial value of the critical gain; Based on the initial value of the critical gain, the β L1 , the β L2 The initial value of the linear observer gain and the β are determined according to the bandwidth method. N1 and the β N2 The initial value of the nonlinear observer gain; Based on the critical gain initial value, each of the linear observer gain initial values ​​and each of the nonlinear observer gain initial values ​​are parameter optimized to determine the β L1 , the β L2 , the β N1 and the β N2 Parameter value of Determining the variable coefficient of the nonlinear function according to a preset disturbance control condition; By trial and error and the L1 , the β L2 , the β N1 , the β N2 The parameter value determines the value of the critical gain and the value of the proportional gain.

5. The control method according to claim 4, characterized in that: The step of determining the initial value of the critical gain of the critical gain comprises: Obtaining a small signal model of a current source submodule and a circuit module port average current variation matrix, wherein the circuit module port average current variation matrix represents a relationship between a phase shift angle and a partial differential of each circuit module in the same multi-active bridge transformer; Determining a theoretical critical gain according to the small signal model and the average current variation matrix of the circuit module port; The initial value of the critical gain is determined according to a preset total disturbance formula and a theoretical critical gain, and the total disturbance formula is used for the linear extended state observer and the nonlinear extended state observer to observe the disturbance in the time-frequency domain.

6. The control method according to claim 5, characterized in that: The small signal model is as follows: in, is the change in output power of voltage source submodules 1 to k, is the change in the actual output voltage of the current source submodule k+1~n, is the actual output voltage of the voltage source submodules 1 to k, is the change in the switching frequency of the voltage source submodules 1 to k, is the change in the switching frequency of the current source submodules k+1~n, s is the complex variable in the Laplace transform, H k+1 ~H n is the relationship between the actual output voltage and actual output current of the current source submodules k+1~n, G kn It is the partial differential of the average current of circuit module k with respect to the phase shift angle of circuit module n.

7. The control method according to claim 5, characterized in that: The total disturbance satisfies the following formula: Among them, v i is the actual output voltage of circuit module i, C i is the capacitor connected to circuit module i, R i is the resistance load connected to circuit module i, is the phase shift angle of circuit module i, G ij is the partial differential of the average current of circuit module i relative to the phase shift angle of circuit module j, b i represents the theoretical critical gain of circuit module i, b 0i represents the critical gain of the current source submodule i.

8. The control method according to claim 1, characterized in that: After generating the frequency change value, the method further includes: Obtaining the critical gain of the current source submodule; According to the error value and the critical gain of the current source submodule, the phase-shift angle gain errors of the linear extended state observer and the nonlinear extended state observer are updated respectively, wherein the phase-shift angle gain error of the linear extended state observer characterizes the relationship between the rate of change of the voltage estimate of the linear extended state observer, the first generalized total disturbance estimate and the voltage gain error; the phase-shift angle gain error of the nonlinear extended state observer characterizes the relationship between the rate of change of the voltage estimate of the nonlinear extended state observer, the first generalized total disturbance estimate and the voltage gain error.

9. An electronic device, comprising: at least one processor; at least one memory for storing at least one program; When at least one of the programs is executed by at least one of the processors, the method according to any one of claims 1 to 8 is implemented.

10. A computer-readable storage medium storing computer-executable instructions, wherein the computer-executable instructions are used to execute the method according to any one of claims 1 to 8.

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