Nonlinear compensation circuit and method for signal acquisition probe

Abstract

The invention discloses a non-linear compensation circuit for a signal acquisition probe, which comprises a sampling resistor, and is characterized in that the sampling resistor and an equipotential input end arranged at one end of the sampling resistor are utilized, a secondary amplification method is utilized, and a sampling resistor R2 network is matched with differential input during primary amplification, so that a signal / noise path is separated; a dynamic ground potential compensation system is constructed before amplification input, noise is dynamically monitored and shielded, post-filtering is conducted on the noise, and meanwhile phase compensation is achieved through a filtering network; and finally, noise cancellation and main signal amplification are completed by using a next-stage amplifier. The circuit overcomes grounding noise interference caused by multiple distributed grounding points in the prior art, and meanwhile, due to the fact that dynamic potential compensation is added, high-sensitivity detection and noise compensation are achieved. The invention further provides a method based on the nonlinear compensation circuit.

Description

Technical Field

[0001] The present invention relates to the field of electrical signal detection, and in particular to a nonlinear compensation circuit and method for a signal acquisition probe. Background Art

[0002] Coaxial cable is a common carrier for high-frequency signal transmission. It generally includes a copper core, an inner insulating layer outside the copper core, a metal shielding layer wrapped around the insulating layer, and an outer insulating layer outside the metal shielding layer. Coaxial cable can transmit high-frequency signals, and the metal shielding layer can shield most external interference.

[0003] At present, when using coaxial cables, the metal shielding layer is grounded. Long coaxial cables need to have multiple grounding points. When the coaxial cables are grounded, current is easily generated, causing voltage drop, ground bounce, and ground interference, which eventually cause measurement errors of the coaxial cables. In addition, for the measurement system of coaxial cables, the measurement system often uses a DC-DC power supply to achieve low power consumption and small size.

[0004] When measuring a coaxial cable with multi-point grounding, the cable will produce a voltage drop due to the large current on the coaxial cable, and a large current will flow through the coaxial cable. This current is not constant, nor is it in an ideal linear relationship with the output signal. This interference is distributed interference formed by the shared ground terminal of multi-point grounding.

[0005] When you need to collect signals from coaxial cables with multiple grounding points, you need to remove these interferences.

[0006] Specifically, a coaxial cable requires multiple data relays from the signal transmitting end to the receiving end, and each data relay has a grounding terminal. When the coaxial signal is measured at the receiving end, distributed loop interference of multiple grounding terminals is naturally introduced.

[0007] In a coaxial cable multi-point grounding system (the number of grounding points n ≥ 2), the shielding layer forms a closed loop, and its equivalent circuit satisfies Kirchhoff's voltage law: in, , represents the distributed impedance of the shielding layer of the kth segment.

[0008] When there is a difference in ground potential between different grounding points: When , a circulation will occur: ,in, is the additional impedance at the remote ground path location. For example: when the length of the coaxial cable exceeds , (λ is the wavelength corresponding to the interference frequency), the circulating current of a 10-meter coaxial cable at 30MHz can reach 12%~18% of the signal current.

[0009] In addition, when the measurement system uses a DC-DC power supply, and the common DC-DC power supply is a Buck type, then, , that is, the inductor current pattern; where, is the switching frequency of the switching power supply; the high-frequency oscillation waveform of the switching node Contains fast edges, this fast edge , the voltage derivative of the fast edge is: , the inverse of this voltage is coupled to the shielding layer through parasitic capacitance, and a displacement current is formed between the shielding layer and the grounding terminal. The displacement current is: .

[0010] Assumptions: ,when When , 10mA The displacement current is sufficient to form parasitic modulation in the transmission of sensitive high-frequency signals. Due to the existence of parasitic modulation signals, if you want to restore the original modulated high-frequency signal during measurement, you must also remove the parasitic modulation signals. Finally, for coaxial cables, high-frequency dynamic currents will form a magnetic field around the shielding layer. The inductance per unit length of the shielding layer is: ; When multiple coaxial cables are transmitted close together, the current changes in the adjacent lines When , the induced electromotive force is: , is the mutual inductance; if the mutual inductance coefficient k on the two coaxial cables is 0.2, the parasitic inductance , then the coupled noise per centimeter is: , this level of noise is enough to affect the accuracy of signal transmission and detection. Summary of the invention

[0011] The purpose of the present invention is to provide a nonlinear compensation circuit for a signal acquisition probe, aiming to solve the technical problem of distributed grounding interference caused by multi-point grounding in the prior art.

[0012] The nonlinear compensation circuit for a signal acquisition probe provided by the present invention comprises: A detection terminal, the detection terminal is electrically connected to a coaxial cable, the coaxial cable comprises a first copper core and a second copper core, a first shielding layer is arranged around the first copper core, and a second shielding layer is arranged around the second copper core; the first copper core has a first resistor, the second copper core has a third resistor, the first shielding layer has a fifth resistor, and the second shielding layer has a seventh resistor; A sampling resistor, wherein a first end of the sampling resistor is connected to the first copper core, a second end of the sampling resistor is respectively connected to the second copper core, a first shielding layer and a second shielding layer, and the first shielding layer is grounded; Wherein, the first end of the sampling resistor is connected to the negative phase input end of the second amplifier through the first resistor and the second resistor in sequence; The second end of the sampling resistor is connected to the equipotential input terminal through a third resistor and a fifth resistor respectively, and the third resistor and the fifth resistor are connected in parallel; The equipotential input terminal is connected to the negative phase input terminal of the first amplifier through the sixth resistor; the equipotential input terminal is connected to the positive phase input terminal of the second amplifier through the fourth resistor, and is connected to the negative phase input terminal of the first amplifier through the sixth resistor; The sampling resistor is connected to the positive phase input terminal of the first amplifier through the seventh resistor; the output terminal of the first amplifier is connected to the negative phase input terminal of the second amplifier through the ninth resistor; the output terminal of the first amplifier is connected to the negative phase input terminal of the first amplifier through the eighth resistor, and capacitors are connected in parallel at both ends of the eighth resistor; the negative phase input terminal of the second amplifier is connected to the output terminal of the second amplifier through the tenth resistor; the output terminal of the second amplifier is connected to the output terminal of the probe through the eleventh resistor.

[0013] The circuit disclosed in the present invention achieves the goal through a three-stage architecture: 1- First stage (input conditioning): The sampling resistor R2 network cooperates with the differential input, that is, the fifth resistor RL3 and the seventh resistor RL4 enter the differential amplification of the first amplifier U12, thereby separating the signal / noise path.

[0014] 2- Second stage (common mode rejection): The first amplifier U12 builds a dynamic ground potential compensation system.

[0015] 3- The third stage (signal synthesis): The second amplifier U1 completes the noise cancellation and main signal amplification.

[0016] The present invention also discloses a nonlinear compensation method for a signal acquisition probe in the process of realizing the above circuit, the method comprising: Step S1: sampling the current between the first copper core and the second copper core and converting it into voltage; Step S2: sampling the noise signal between the first copper core and the first shielding layer; Step S3: setting an equipotential input terminal between the second copper core and the first shielding layer so that the signals on the second copper core and the first shielding layer form a dynamic potential balance point, so as to reduce the influence of displacement current caused by multi-point grounding on the coaxial cable; Step S4: differentially amplifying the noise between the first shielding layer and the second shielding layer, eliminating common mode interference and isolating the influence on the signal on the first copper core by differential amplification, and filtering the noise after differential amplification; Step S5: superimposing the filtered noise signal with the signal on the first copper core, and amplifying the signal with the gain of the second copper core after superposition and outputting the signal to the detection end; The filtered noise signal is used to perform phase compensation on the signal on the first copper core.

[0017] The method separates the signal / noise path, constructs a dynamic ground potential compensation system, completes noise cancellation and main signal amplification, and can also achieve phase compensation. BRIEF DESCRIPTION OF THE DRAWINGS

[0018] Figure 1 It is the circuit principle diagram of the present invention; Figure 2 It is a flow chart of the method of the present invention. DETAILED DESCRIPTION

[0019] The present invention will be further described and illustrated below in conjunction with specific embodiments and accompanying drawings: Please refer to Figure 1 The present invention discloses a nonlinear compensation circuit for a signal acquisition probe, the circuit comprising: a detection terminal current, the detection terminal current is electrically connected to a coaxial cable X, and the other end of the detection terminal current is connected to a detection device, such as an oscilloscope, a spectrum analyzer, etc., through a probe. The coaxial cable X comprises a first copper core x1 and a second copper core x2, a first shielding layer y1 is disposed on the periphery of the first copper core x1, and a second shielding layer y2 is disposed on the periphery of the second copper core x2; the first copper core x1 has a first resistor RL1, the second copper core x2 has a third resistor RL2, the first shielding layer y1 has a fifth resistor RL3, and the second shielding layer y2 has a seventh resistor RL4.

[0020] The detection terminal current includes a sampling resistor R2, a first end of the sampling resistor R2 is connected to the first copper core x1, a second end of the sampling resistor R2 is respectively connected to the second copper core x2, the first shielding layer y1 and the second shielding layer y2, and the first shielding layer y1 is grounded GND1.

[0021] Among them, the first end of the sampling resistor R2 is connected to the negative phase input terminal of the second amplifier U1 through the first resistor RL1 and the second resistor R8 in sequence; the second end of the sampling resistor R2 is connected to the equipotential input terminal GND through the third resistor RL2 and the fifth resistor RL3 respectively, and the third resistor RL2 and the fifth resistor RL3 are connected in parallel; the equipotential input terminal GND is connected to the negative phase input terminal of the first amplifier U12 through the sixth resistor R5; the equipotential input terminal GND is connected to the positive phase input terminal of the second amplifier U1 through the fourth resistor R65, and is connected to the positive phase input terminal of the first amplifier U12 through the sixth resistor R5. negative phase input terminal; the sampling resistor R2 is connected to the positive phase input terminal of the first amplifier U12 through the seventh resistor RL4; the output terminal of the first amplifier U12 is connected to the negative phase input terminal of the second amplifier U1 through the ninth resistor R64; the output terminal of the first amplifier U12 is connected to the negative phase input terminal of the first amplifier U12 through the eighth resistor R1, and the two ends of the eighth resistor R1 are connected in parallel with the capacitor C1; the negative phase input terminal of the second amplifier U1 is connected to the output terminal of the second amplifier U1 through the tenth resistor R10; the output terminal of the second amplifier U1 is connected to the output terminal of the probe through the eleventh resistor R6; Among them, the sampling resistor R2 is less than or equal to 1Ω, the resistance of the second resistor R8 is 249Ω, the fourth resistor R65 is 49.9Ω, the sixth resistor R5 is 1K, the eighth resistor R1 is 1K, the ninth resistor R64 is 10K, the tenth resistor R10 is 330Ω, and the eleventh resistor R6 is 49.9Ω; the capacitor C1 is 10pF.

[0022] The sampling resistor R2 is an alloy resistor to reduce the error caused by sampling temperature drift.

[0023] The equipotential input terminal GND is used to form a redundant current path with the third resistor (RL2), the fifth resistor (RL3), and the seventh resistor (RL4) connected in parallel, and to establish a common reference ground to discharge high-frequency noise on the second copper core x2 and the first shielding layer y1.

[0024] The negative phase input terminal of the first amplifier U12 is connected to the equipotential input terminal GND to destroy the circulating current loop on the first shielding layer y1 and / or the second shielding layer y2 on the coaxial cable X, thereby reducing the circulating current grounding interference.

[0025] Among them, the positive input end of the first amplifier U12 is connected to the second shielding layer y2, so as to detect the potential fluctuation caused by the displacement current on the second shielding layer y2 in real time through the seventh resistor RL4. The first amplifier U12 is used to amplify the potential fluctuation to the output end of the first amplifier U12, and utilize the filtering and feedback branch formed by the eighth resistor R1 and capacitor C at the output end of the first amplifier U12 to achieve noise cancellation and feedback gain maintenance, thereby reducing the influence of the front-end noise of the first amplifier U12.

[0026] The following is an explanation of the overall design principle, working process and component functions of this circuit.

[0027] 1. Overall circuit architecture and core functional requirements This circuit is designed for precise current detection of high-frequency coaxial cable X, and its core is to solve the following five technical problems: a-Distributed ground loop interference; b-interference coupling of displacement current in the shielding layer; c- High frequency impact of parasitic modulation effect; d- Waveform distortion caused by phase shift; e- Improve the acquisition accuracy of microvolt level signals.

[0028] This application achieves its goals through a three-level architecture: 1- First stage (input conditioning): The sampling resistor R2 network cooperates with the differential input, that is, the fifth resistor RL3 and the seventh resistor RL4 enter the differential amplification of the first amplifier U12, thereby separating the signal / noise path.

[0029] 2- Second stage (common mode rejection): The first amplifier U12 builds a dynamic ground potential compensation system.

[0030] 3- The third stage (signal synthesis): The second amplifier U1 completes the noise cancellation and main signal amplification.

[0031] 2. Analysis of key modules layer by layer 1. Detection terminal current and shielding network design Sampling resistor R2 (≤1Ω): The sampling resistor R2 is arranged between the first copper core x1 (the first copper core's own resistance is the first resistor RL1) and the second copper core x2 (the second copper core's own resistance is the third resistor RL2) to form a current-voltage conversion core.

[0032] Technical points: The impedance of the sampling resistor R2 below 1Ω ensures that the voltage drop is insufficient to cause temperature drift when measuring the cable current (assuming that only 1V loss occurs at full scale of 1A).

[0033] Multi-path reflux design: The second end of the sampling resistor R2 is connected in parallel with the second copper core x2 (corresponding to the third resistor RL2), the first shielding layer y1 (corresponding to the fifth resistor RL3), and the second shielding layer y2 (corresponding to the seventh resistor RL4) to form a redundant current path.

[0034] Shield layer processing mechanism: The first shielding layer y1 (GND1 is directly grounded): establishes the main reference ground plane and discharges high-frequency noise to the low-impedance ground. The second shielding layer y2 (through the seventh resistor RL4 → the positive input terminal of the first amplifier U12): dynamically monitors the displacement current, detects and compensates through the operational amplifier. Hybrid grounding strategy: The three-terminal parallel connection adopts star grounding and dynamic potential tracking, that is, when the second copper core x2, the first shielding layer y1 and the second shielding layer y2 are connected to the input terminal of the first amplifier U12, a three-point star connection is formed. The star grounding can destroy the ground loop structure of the traditional multi-layer shielding.

[0035] 2. Construction of equipotential input GND Implementation method: The parallel point of the second copper core x2 and the first shielding layer y1, that is, the equipotential input terminal GND, has an equivalent impedance , for example, when RL2=0.1Ω, RL3=0.05Ω, Z GND ≈0.033Ω. At this time, the role of the equipotential input terminal GND is to provide a reference point for the signal processing circuit; bypass the high-frequency noise component to the ground path; and actively correct the potential potential offset through the operational amplifier.

[0036] 3. Noise cancellation loop of the first amplifier U12 The positive input terminal of the first amplifier U12: samples the voltage drop generated by the displacement current through the seventh resistor RL4 (parasitic resistance of the second shielding layer y2). The negative input terminal of the first amplifier U12: obtains the voltage reference of the equipotential input terminal GND through the fifth resistor R5 (1 kΩ) on the first shielding layer y1.

[0037] The feedback network of the first amplifier U12 is composed of an eighth resistor R1 (1 kΩ) and a capacitor C1 (10 pF) connected in parallel to form a composite feedback, and the composite feedback includes: DC channel: The eighth resistor R1 keeps the closed-loop gain at 1 to ensure a stable operating point.

[0038] High-frequency filtering: A low-pass characteristic is formed at 159MHz (f=1 / (2πRC)) to filter out UHF band noise.

[0039] Phase compensation: to cope with the transmission delay of high-speed signals and prevent self-oscillation.

[0040] Core function: Real-time detection of the potential fluctuation (ΔV RL4); amplify the fluctuation signal and output it to the next stage for noise cancellation; create an active ground potential adjustment system to eliminate the phase shift of the equipotential input terminal GND itself.

[0041] 4. The main signal processing channel of the second amplifier U1 Signal input layer: a. Inverting end path, three steps of signal processing are performed: The first resistor RL1 on the first copper core x1 → voltage drop correction of the inherent impedance of the cable; connect the second resistor R8 (249Ω) on the first copper core x1 → set the voltage attenuation ratio (coordinated with R10=330Ω for amplitude stabilization); receive the compensation voltage output by the first amplifier U12 (via R64=10kΩ).

[0042] b. Forward path: The fourth resistor R65 (49.9Ω) and the third resistor RL2 on the second copper core x2 constitute a reference input, and 49.9Ω (quasi-50Ω) is selected to achieve high-frequency impedance matching and reduce standing wave reflection.

[0043] Calculation of the amplification parameters of the second amplifier U12: Closed loop gain A v = 1 + (R10) / (R8 + RL1), substituting the following parameters: R10=330Ω, R8=249Ω, assuming RL1=0.01Ω (high-quality copper core), we get A v ≈ 1 + 330 / (249+0.01) ≈ 2.326 times (basic magnification).

[0044] Cascade Noise Cancellation: The compensation signal output by the first amplifier U12 is injected into the inverting input terminal of the second amplifier U1 through the ninth resistor R64 (10 kΩ), and the corresponding cancellation coefficient K = (R10 / R64) / (1 + R10 / R64) ≈ 0.032, thereby achieving adaptive correction of amplitude and phase.

[0045] 5. Output interface design The eleventh resistor R6 (49.9Ω) forms the best match with the 50Ω input of the oscilloscope (SWR=1.002) → reflection coefficient Γ= (50-49.9) / (50+49.9)≈0.00099; it isolates the capacitive load between the op amp output and the probe input; it limits the short-circuit current and protects the front-end circuit.

[0046] Full channel bandwidth calculation: The dominant pole is determined by the gain-bandwidth product (GBP) of the second amplifier U1. Assuming that the second amplifier U1 uses ADA4817 (GBP=400MHz): The theoretical bandwidth BW of the second amplifier U1 = GBP / A v = 400 / 2.326≈172MHz. This bandwidth meets most high-frequency test requirements (such as USB3.0, 5Gbps signal analysis).

[0047] 3. Solutions to key technical issues Problem 1: Distributed ground loop interference The solution adopted is: a. Star-tree mixed grounding: the first shielding layer y1 is single-point grounded (GND1), and the second shielding layer y2 is modulated by the operational amplifier to form a dynamic potential balance.

[0048] b. Equipotential node absorption: The equipotential input terminal GND is connected in parallel with the current component of the third resistor RL2 of the second copper core x2 and the fifth resistor RL3 of the first shielding layer, and the ground potential difference is isolated through the high resistance of the sixth resistor R5 (1kΩ) to reduce ground interference.

[0049] Simulation verification: Assuming that there is 1mV / 100kHz interference between the ground terminal GND1 and the equipotential input terminal GND, if the CCMRR (common mode rejection ratio) of the first amplifier U12 reaches 80dB, it will produce 0.01μV residual at the output end, then the ground interference after secondary suppression in the second amplifier U1 stage is almost unmeasurable.

[0050] Problem 2: Displacement Current Interference Mechanism analysis: At high frequencies, the capacitive coupling effect of the shielding layer of coaxial cables (C Shield ) and the cable length produce displacement current: I dis = jω·C Shield ·V CM .

[0051] Suppression strategies: 1. Active monitoring compensation: The positive terminal of the first amplifier U12 monitors the voltage drop V of the seventh resistor RL4 on the second shielding layer x2 in real time. D = I dis ·RL4.

[0052] 2. Mirror injection: The output of the first amplifier U12 is injected into the inverting terminal of the second amplifier U1 through the ninth resistor R64, and vector-subtracted from the main signal (the signal from the first copper core x1 and the second copper core x2).

[0053] 3. Physical isolation: The second shielding layer y2 is not directly grounded, and absorbs current through the high impedance input of the first amplifier U12.

[0054] Problem 3: Parasitic Modulation An ultra-low inductance sampling resistor (R2≤1Ω) is used; capacitor C1 is selected to be 10pF, and capacitor C1 introduces a controllable pole in the feedback loop formed with the eighth resistor R1 to suppress radio frequency interference; the eleventh resistor R6 is selected to be 49.9Ω to achieve transmission matching and eliminate signal rebound. The transmission matching is adapted to the front-end impedance of the coaxial cable X of 50Ω.

[0055] Problem 4: Phase offset control Mechanism and compensation: Phase shift mainly comes from the distributed parameters of the resistor and capacitor components in the signal path, which is particularly significant at high frequencies. This circuit addresses this problem from three aspects: 1. Symmetrical input path design: Main signal path: first resistor RL1 → second resistor R8 → second amplifier U1 inverting terminal (total impedance of 249.01Ω) Compensation signal path: seventh resistor RL4 → first amplifier U12 → ninth resistor R64 → second amplifier U1 inverting terminal (total impedance = RL4 + R64) In this application, path delay balance is achieved through precise resistance matching (R8=249Ω, RL4≈equivalent resistance of C1 and R1+10kΩ of R64).

[0056] 2. Active phase compensation network: The capacitor C1 = 10 pF and the eighth resistor R1 = 1 kΩ are connected in parallel in the feedback loop of the first amplifier U12 to form a compensation pole.

[0057] The frequency calculation formula of the compensation pole is: Compensation frequency .

[0058] Effect: Introduce phase boost at 15.9MHz to offset the additional delay caused by distributed capacitance (such as 2pF / cm of PCB traces).

[0059] 3. Operational amplifier bandwidth matching (taking AD8065 as an example): The first amplifier U12 is configured as a voltage follower, and the bandwidth = gain bandwidth product (GBW) / gain = 145 MHz / 1 = 145 MHz.

[0060] The second amplifier U1 is configured with a gain of ≈2.3 times, and an equivalent bandwidth of 145MHz / 2.3≈63MHz. The phase shift superposition of the two-stage amplifier is controlled within ±0.1° through simulation verification, meeting the high-speed signal requirements.

[0061] Question 5: Achieving microvolt-level signal accuracy The core is guaranteed by four levels of technology: 1. TCR optimization of sampling resistor R2: The sampling resistor R2 is made of manganese-copper alloy (TCR≤50ppm / ℃). Assuming that the error is only 0.05% when ΔT=10℃, the temperature drift effect is minimized.

[0062] 2. Operational amplifier input bias current control: The selection criteria require that the input bias current is ≤1nA. For example, the ADA4528-1 operational amplifier is selected in this application; Voltage error caused by the operational amplifier: This error is 5 orders of magnitude lower than the 1V voltage drop of a 1Ω sampling resistor at a current of 1A. Such a voltage error can be ignored.

[0063] Common-mode rejection enhancement: The common mode rejection ratio (CMRR) of the first amplifier U12 is ≥100dB; the CMRR of the second amplifier U1 using differential input is ≥90dB; the total CMRR is: Such a suppression ratio can suppress 1V common mode noise to below 0.45μV, which has reached an extremely high level.

[0064] Thermal Noise Suppression Design: In this application, the total noise is calculated as: ; So the noise on the second resistor R8 is: ,achieve level.

[0065] When a low-noise op amp is selected with an input noise of 1.9nV / √Hz, the low-frequency noise is: , and also reach Therefore, the total input noise of the circuit is in the range of 8.7μV_rms to a peak-to-peak value of about 57μV. When the full scale is 1V, the signal-to-noise ratio SNR = 20log(1V / 57μV) ≈ 104dB.

[0066] In this application, the circuit uses a three-layer grounding mechanism to form a distributed noise absorption system: a) Main ground plane GND1: directly shares the earth and discharges high-frequency noise to a low-impedance path; b) Equipotential input terminal GND: as the signal processing reference; Equivalent circuit: a star network composed of the third resistor RL2 (second copper core), the fifth resistor RL3 (first shielding layer), and the seventh resistor RL4 (second shielding).

[0067] Potential stabilization mechanism: real-time monitoring and adjustment by the first amplifier U12 The second shielding layer RL4: through the active regulation of the first amplifier op amp, the displacement current Idis can be guided to the op amp input virtual ground to avoid contaminating the main signal ground.

[0068] For the first amplifier U12 in the present application: noise amplification suppression is achieved by input differential pair detection. (Reflects the shield layer current); negative end signal .

[0069] At this time, the closed-loop gain of the transfer function of the first amplifier U12 is set to 1 (due to the ), to achieve active reduction of shielding layer noise.

[0070] In addition, for the first amplifier U12, after the capacitor C1 is introduced, the phase margin of the open-loop gain curve at the unity gain intersection reaches 65°, which can prevent oscillation.

[0071] For the second amplifier U1 of the present application, the total current equation of the inverting input terminal of the second amplifier U1 is: The voltage at the non-inverting input terminal of the second amplifier U1 is: , and the third resistor RL2 forms a voltage divider, the third resistor RL2≈R65, at this time, the voltage at the non-inverting input terminal of the second amplifier U1 is half of the voltage across the sampling resistor R2. Therefore, the output voltage is , thus achieving vector cancellation of noise components.

[0072] In the example calculation, assuming RL1=0.01Ω, R8=249Ω, R10=330Ω, R64=10kΩ, the main signal path gain is: Compensation Path Gain: The total output expression is: when , partial cancellation of shielding noise is achieved.

[0073] In this application, capacitor C1 has different functions in different frequency bands, specifically: At low frequencies (f<1MHz): C1 impedance is much larger than R1 (1kΩ→10pF at 1MHz = 15.9kΩ), feedback is dominated by R1, and the op amp works in unity gain mode to ensure DC stability.

[0074] In the mid-frequency band (1-15MHz): the capacitive reactance of C1 gradually decreases and begins to form a parallel effect with R1. At this time, phase lead compensation is introduced to improve transient response (reduce ringing); At high frequencies (>15MHz), C1 becomes the primary feedback path, extending the bandwidth of U12 to the GBW limit, thereby suppressing the coupling of RF interference (such as 900MHz mobile phone signals).

[0075] In this application, the circuit component parameters are selected as follows: Calculate the key ratios based on the values ​​in the table above: Among them, the signal-to-noise ratio improvement factor is: →After conversion, a 32dB improvement was achieved. Among them, the ground loop isolation is: →Achieved -89dB coupling attenuation.

[0076] The circuit in this application systematically solves the core interference problem in coaxial cable X detection through multi-level feedback, common mode compensation and impedance matching technology.

[0077] Its beneficial effects include: 1. High common mode rejection ratio (CMRR): The dual op amp nested structure can theoretically achieve a CMRR of more than 120dB.

[0078] 2. Wideband response: Extended to hundreds of MHz through small capacitor compensation, suitable for high-speed signal sampling.

[0079] 3. Anti-ground loop design: Dynamic virtual ground technology eliminates the potential risks of multi-point grounding.

[0080] 4. Phase consistency guarantee: Symmetrical layout and compensation network ensure group delay.

[0081] This circuit is particularly suitable for scenarios with strict requirements on signal integrity, such as high-frequency communication cables (such as 5G base station feeders) and medical ultrasound probes. It can be further adapted to different cable specifications by adjusting parameters (such as gain and bandwidth).

[0082] See also Figure 1 and Figure 2 The present invention further discloses a nonlinear compensation method for a signal acquisition probe, the method is used to detect a high-frequency electrical signal on a coaxial cable X, the method comprising: Step S1: sampling the current between the first copper core x1 and the second copper core x2 and converting it into voltage; Step S2: sampling the noise signal between the first copper core x1 and the first shielding layer y1; Step S3: setting an equipotential input terminal between the second copper core x2 and the first shielding layer y1, so that the signals on the second copper core x2 and the first shielding layer y1 form a dynamic potential balance point, so as to reduce the influence of displacement current caused by multi-point grounding on the coaxial cable X; Step S4: differentially amplify the noise between the first shielding layer y1 and the second shielding layer y2, eliminate common mode interference by differential amplification, isolate the influence on the signal on the first copper core x1, and filter the noise after differential amplification; Step S5: superimposing the filtered noise signal with the signal on the first copper core x1, and amplifying the signal with the gain of the signal on the second copper core x2 after superposition, and then outputting to the detection end; The filtered noise signal is used to perform phase compensation on the signal on the first copper core x1.

[0083] The second copper core x2, the first shielding layer y1 and the second shielding layer y2 form a connection loop to the noise differential amplifier input terminal, thereby destroying the multi-point grounding structure of the coaxial cable X to avoid the distributed parameters from forming common mode noise.

[0084] Among them, during the noise differential amplification, it also includes real-time monitoring of the potential fluctuations caused by the noise on the first shielding layer y1 and the second shielding layer y2, and adjusting the potential bias point according to the monitoring results when superimposing the filtered noise signal with the signal on the first copper core x1, maintaining the common mode rejection ratio on the first shielding layer y1 and the second shielding layer y2, so as to achieve the purpose of noise suppression.

[0085] When the signal gain on the second copper core x2 is amplified, the filtered noise signal and the superimposed signal on the first copper core x1 are used as reference inputs to eliminate the noise.

[0086] Finally, it should be noted that the above embodiments are only used to illustrate the technical solution of the present invention, rather than to limit the scope of protection of the present invention. Although the present invention has been described in detail with reference to the preferred embodiments, those skilled in the art should understand that the technical solution of the present invention can be modified or replaced by equivalents without departing from the essence and scope of the technical solution of the present invention.

Claims

1. A nonlinear compensation circuit for a signal acquisition probe, characterized in that: The circuit comprises: A detection terminal, the detection terminal is electrically connected to a coaxial cable, the coaxial cable comprises a first copper core and a second copper core, a first shielding layer is arranged around the first copper core, and a second shielding layer is arranged around the second copper core; the first copper core has a first resistor, the second copper core has a third resistor, the first shielding layer has a fifth resistor, and the second shielding layer has a seventh resistor; A sampling resistor, wherein a first end of the sampling resistor is connected to the first copper core, a second end of the sampling resistor is respectively connected to the second copper core, a first shielding layer and a second shielding layer, and the first shielding layer is grounded; Wherein, the first end of the sampling resistor is connected to the negative phase input end of the second amplifier through the first resistor and the second resistor in sequence; The second end of the sampling resistor is connected to the equipotential input terminal through a third resistor and a fifth resistor respectively, and the third resistor and the fifth resistor are connected in parallel; The equipotential input terminal is connected to the negative phase input terminal of the first amplifier through the sixth resistor; the equipotential input terminal is connected to the positive phase input terminal of the second amplifier through the fourth resistor, and is connected to the negative phase input terminal of the first amplifier through the sixth resistor; The sampling resistor is connected to the positive phase input terminal of the first amplifier through the seventh resistor; the output terminal of the first amplifier is connected to the negative phase input terminal of the second amplifier through the ninth resistor; the output terminal of the first amplifier is connected to the negative phase input terminal of the first amplifier through the eighth resistor, and capacitors are connected in parallel at both ends of the eighth resistor; the negative phase input terminal of the second amplifier is connected to the output terminal of the second amplifier through the tenth resistor; the output terminal of the second amplifier is connected to the output terminal of the probe through the eleventh resistor.

2. The nonlinear compensation circuit for a signal acquisition probe according to claim 1, characterized in that: The sampling resistor is an alloy resistor to reduce the error caused by sampling temperature drift.

3. The nonlinear compensation circuit for a signal acquisition probe according to claim 1, characterized in that: The equipotential input terminal is used to form a redundant current path with the third resistor, the fifth resistor, and the seventh resistor connected in parallel, and to establish a common reference ground to discharge high-frequency noise on the second copper core and the first shielding layer.

4. The nonlinear compensation circuit for a signal acquisition probe according to claim 1, characterized in that: The negative phase input terminal of the first amplifier is connected to the equipotential input terminal to destroy the circulating current loop on the first shielding layer and / or the second shielding layer on the coaxial cable, thereby reducing the circulating current grounding interference.

5. The nonlinear compensation circuit for a signal acquisition probe according to claim 1, characterized in that: The positive input terminal of the first amplifier is connected to the second shielding layer so as to detect the potential fluctuation caused by the displacement current on the second shielding layer in real time through the seventh resistor. The first amplifier is used to amplify the potential fluctuation to the output terminal of the first amplifier, and utilize the filtering and feedback branch formed by the eighth resistor and capacitor at the output terminal of the first amplifier to achieve noise cancellation and feedback gain maintenance, thereby reducing the impact of the front-end noise of the first amplifier.

6. The nonlinear compensation circuit for a signal acquisition probe according to claim 1, characterized in that: The sampling resistor is less than or equal to 1Ω, the resistance of the second resistor is 249Ω, the fourth resistor is 49.9Ω, the sixth resistor is 1K, the eighth resistor is 1K, the ninth resistor is 10K, the tenth resistor is 330Ω, and the eleventh resistor is 49.9Ω; the capacitor is 10pF.

7. A nonlinear compensation method for a signal acquisition probe, characterized in that: The method is used to detect a high-frequency electrical signal on a coaxial cable, wherein the coaxial cable comprises a first copper core and a second copper core, wherein a first shielding layer is disposed on the periphery of the first copper core, and a second shielding layer is disposed on the periphery of the second copper core; The first copper core has a first resistance, the second copper core has a third resistance, the first shielding layer has a fifth resistance, and the second shielding layer has a seventh resistance; The method comprises: Step S1: sampling the current between the first copper core and the second copper core and converting it into voltage; Step S2: sampling the noise signal between the first copper core and the first shielding layer; Step S3: setting an equipotential input terminal between the second copper core and the first shielding layer so that the signals on the second copper core and the first shielding layer form a dynamic potential balance point, so as to reduce the influence of displacement current caused by multi-point grounding on the coaxial cable; Step S4: differentially amplifying the noise between the first shielding layer and the second shielding layer, eliminating common mode interference and isolating the influence on the signal on the first copper core by differential amplification, and filtering the noise after differential amplification; Step S5: superimposing the filtered noise signal with the signal on the first copper core, and amplifying the signal with the gain of the second copper core after superposition and outputting the signal to the detection end; The filtered noise signal is used to perform phase compensation on the signal on the first copper core.

8. The nonlinear compensation method for a signal acquisition probe according to claim 7, characterized in that: The second copper core, the first shielding layer, and the second shielding layer to the noise differential amplifier input end form a connection loop, thereby destroying the multi-point grounding structure of the coaxial cable to avoid the formation of common mode noise by distributed parameters.

9. The nonlinear compensation method for a signal acquisition probe according to claim 7, characterized in that: During noise differential amplification, it also includes real-time monitoring of the potential fluctuations caused by the noise on the first shielding layer and the second shielding layer, and adjusting the potential bias point according to the monitoring results when superimposing the filtered noise signal with the signal on the first copper core, maintaining the common mode rejection ratio on the first shielding layer and the second shielding layer to achieve the purpose of noise suppression.

10. The nonlinear compensation method for a signal acquisition probe according to claim 7, characterized in that: When the signal gain on the second copper core is amplified, the filtered noise signal and the superimposed signal on the first copper core are used as reference inputs, thereby eliminating the noise.

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