A kind of wired and wireless dual load high frequency system based on load impedance compression and parameter design method

Abstract

The application is a wired and wireless dual-load high-frequency system based on load impedance compression and a parameter design method. The application relates to the technical field of load impedance compression design, and proposes a load impedance compression mechanism, and increases an impedance conversion network to adjust the impedance value of a branch. From the perspective of impedance trajectory, the influence of the change of the coupling coefficient is compressed, the complex interlaced trajectory curve among multiple loads is decoupled, the independence of the load power is restored, and a theoretical basis is provided for the further packaging and expansibility design of the multiple load modules.

Description

Technical Field

[0001] The invention relates to the technical field of load impedance compression design, in particular to a wired and wireless dual-load high-frequency system based on load impedance compression and a parameter design method. Background Art

[0002] After entering the Industrial Scientific Medical Band (ISM), namely 6.78MHz, 13.56MHz and 27.12MHz, the rectifier input voltage and current no longer show zero phase angle characteristics, such as Figure 1 As shown. Due to the increase in the proportion of parasitic capacitance effect, the input current exceeds the input voltage, and the rectifier exhibits capacitive impedance characteristics, and produces nonlinear changes with power changes. Take the full-bridge rectifier as an example:

[0003]

[0004]

[0005] Among them, Q j is the junction charge stored in the diode junction capacitance. rec_FBR and X rec_FBR R and V are the real and imaginary input impedances of the full-bridge rectifier, respectively. o These correspond to the rectifier output load and voltage, respectively. In low-frequency systems, the load-independent characteristics of LCC / S (or double-sided LCC) derived from classic fundamental wave analysis will vary in high-frequency systems. Therefore, the present invention employs trajectory analysis from a more accurate impedance compression perspective.

[0006] Figure 2 The impedance trajectory changes of a single wired load system and a wired / wireless dual load system under traditional impedance compression design are demonstrated. Figure 3 Shows Figure 2 The output performance of the Class-E power amplifier (PA) used in the analysis. The yellow area is the high efficiency impedance range of the PA, with the PA output impedance Z pa Performing a load pull sweep, as shown by the red line, the PA efficiency increases as it moves toward the center. The purple curve is the PA power output contour line, and as shown by the green arrow, the power output gradually decreases. So, for the single wire load case, the load resistance R L After the rectifier, the capacitive input impedance Z rec ,like Figure 2 (a). Obviously, at this time Z rec It is not within the efficient range of PA, so an impedance matching network (IMN) is needed torec Convert to Z pa On one hand, the PA works efficiently, and on the other hand, the power level required by the system is regulated. However, when the number of loads begins to increase, as shown in Figure 2 (b), the impedance trajectory begins to become complex. Similarly, the load R L1 , R L2 after the rectifier presents a capacitance.

[0007] Figure 2 High-frequency system impedance trajectory changes; Figure 2 (a) single-point impedance compression design of a wired load system; Figure 2 (b) impedance compression design of a high-frequency wired / wireless dual load system; Figure 2 (c) impedance compression design of a high-frequency wired / wireless dual load system based on PA efficiency optimization;

[0008] The wireless branch needs to have the ability to move freely, so the impedance Z rec_1 (azure circles) after the coil coupling gets the reflected impedance Z ref_1 (red triangles) under the offset. It can be seen that the impedance Z ref_1 has a wide range of impedance changes, which will lead to a large range of changes in the synthesized impedance Z IMN (green circles). Under the same impedance compression design, Z pa may not be able to maintain the PA efficient work. Of course, some dynamic IMN can be used to re-optimize the impedance trajectory to ensure that Z pa falls into the high-efficiency region, as shown in Figure 2 (c). But this will increase the complexity of the system. More importantly, the impedance trajectory of the system is still relatively chaotic, and the power distribution of each load cannot be accurately obtained. There is power coupling between the loads, and the characteristics of each load are not independent. In this way, the requirement of load scalability cannot be further met. SUMMARY

[0009] Miniaturization is one of the main trends in the development of power supplies, and high frequency helps to improve the flexibility of the magnetic field space of wireless power transmission technology, making it more suitable for multi-load application scenarios. However, high frequency also increases the sensitivity of the system, causing power coupling problems in multi-load design. Therefore, the present application proposes a load impedance compression mechanism for wired / wireless dual load applications, which increases the impedance conversion network to adjust the impedance values of the branches. From the perspective of impedance trajectory, the influence of coupling coefficient change is compressed, the complex trajectory curve of the interweaving of multiple loads is decoupled, and the independence of load power is restored, providing a theoretical basis for further packaging and scalability design of multi-load modules.

[0010] The application provides a wired and wireless dual-load high-frequency system based on load impedance compression and a parameter design method.

[0011] A wired and wireless dual-load high-frequency system based on load impedance compression, the system comprising: a constant AC voltage source, a capacitor C tx , an inductor L tx , an inductor L rx , a capacitor C rx , a diode D1, a diode D2, a capacitor C r1 , a capacitor C r2 , an inductor L 01 , an inductor L 02 , a capacitor C0, a resistor R0, a capacitor C b1 , an inductor L b1 , a capacitor C b2 , an inductor L b2 , a diode D F1 , a diode D F2 , a capacitor C rF1 , a capacitor C rF2 , an inductor L 0F1 , an inductor L 0F2 , a capacitor C 0F , and a resistor R 0F .

[0012] One end of the constant AC voltage source is connected to one end of the capacitor C tx and the capacitor C b1 ; the other end of the capacitor C tx is connected to one end of the inductor L tx , the inductor L tx and the inductor L rx are mutually inductive; one end of the inductor L rx is connected to one end of the capacitor C rx , the other end of the capacitor C rx is connected to one end of the diode D1, the capacitor C r1 and the inductor L 01 , the other end of the inductor L 01 is connected to one end of the inductor L 02 , the capacitor C0 and the resistor R0, the other end of the capacitor C0 is connected to the other end of the resistor R0 and grounded; the other end of the diode D1 and the capacitor C r1 is respectively connected to one end of the diode D2 and the capacitor C r2 ; the other end of the diode D2 and the capacitor C r2 is respectively connected to the other end of the inductor L rx and the inductor L 02 .

[0013] The other end of the constant AC voltage source is connected to the inductor L tx and the inductor Lb1 The other end of the capacitor C b1 The other end of the inductor L b1 and capacitor C b2 One end of the capacitor C b2 The other end of the inductor L b2 One end of the inductor L b2 The other end of the diode D F1 , capacitor C rF1 and inductor L 0F1 One end of the inductor L 0F1 The other end of the inductor L 0F2 , capacitor C 0F and resistor R 0F One end of the capacitor C 0F The other end of the resistor R 0F The other end of the diode D F1 and capacitor C rF1 The other end of each diode is connected to F2 and capacitor C rF2 One end of the diode D F2 and capacitor C rF2 The other end of the inductor L b1 and inductor L 0F2 the other end.

[0014] Preferably, the load branch reflected impedance Z ref_1 and Z ref_2 Connect in parallel to reduce the synthetic impedance Z IMN , add an impedance conversion network in the wired branch to adjust the reflected impedance Z ref_2 , to achieve fixed power transmission of wired branches.

[0015] A synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression, the method specifically comprising:

[0016] For wireless branches, compensate for the reactive component on the secondary side to improve coil efficiency;

[0017] By using the load-pull function and taking the rectifier input impedance as a benchmark, the efficient impedance range of the coil under different coupling coefficients k is depicted;

[0018] Select the minimum coupling coefficient for full compensation and design the rectifier shunt capacitor to achieve the maximum efficiency of the coil under this coupling coefficient;

[0019] By analyzing and calculating the impedance values ​​under different coupling coefficients, the impedance trajectory is obtained;

[0020] Compare the impedance traces to see if the impedance values ​​under different coupling coefficients are within the high efficiency range of the coil, and adjust the value of the rectifier shunt capacitor.

[0021] When the stable voltage is generated, the wired load power will remain constant, keeping the impedance value unchanged; in the case of ignoring the line resistance, the synthesis impedance Z is determined IMN .

[0022] Preferably, for the wireless branch, the secondary reactive component is compensated to improve the coil efficiency, that is:

[0023]

[0024] When the output power of the wireless branch changes, its corresponding X rec_1 The complete compensation design is carried out by using the traditional method:

[0025]

[0026] At this time, the reflected impedance Z ref_1 is:

[0027]

[0028] The coil efficiency is expressed by the following formula:

[0029]

[0030] In the traditional method, the coil efficiency is simplified as:

[0031]

[0032] The peak efficiency is derived:

[0033]

[0034] The real impedance of the rectifier corresponding to the peak efficiency of the coil is:

[0035]

[0036] Using the load traction function, taking the rectifier input impedance Z rec_1 as the basis, the high-efficiency impedance range of the coil under different coupling coefficients k is drawn.

[0037] Preferably, for the wired load branch, the reflected impedance Z ref_2 is reduced by ICN, according to Kirchhoff's voltage / current law:

[0038]

[0039] Where u ab is the sinusoidal voltage source generated by the PA through the IMN, and further derivation is obtained:

[0040]

[0041] Preferably, the resonant element satisfies the following equation:

[0042]

[0043] The simplified result is obtained:

[0044]

[0045] When u ab After the stable voltage is generated, the wired load power P oF Will remain constant, keeping Z rec_2 Invariable.

[0046] Preferably, the reflected impedance eliminates the reactive component, moves left and right on the zero imaginary impedance horizontal axis, has parameter flexibility, and needs to approach Z IMN To the left on the horizontal axis to reduce the change range of the synthesized impedance Z ref_2 .

[0047] Preferably, in the case of ignoring the line resistance, Z IMN Is equal to:

[0048]

[0049] A computer readable storage medium having stored thereon a computer program, the program being executed by a processor to implement a synthesized impedance design method of a wired and wireless dual-load high-frequency system based on load impedance compression.

[0050] A computer device comprising a memory and a processor, the memory storing a computer program, and the processor implementing a synthesized impedance design method of a wired and wireless dual-load high-frequency system based on load impedance compression when executing the computer program.

[0051] The present application has the following beneficial effects:

[0052] Compared with the prior art, the present application has the following advantages:

[0053] Based on analyzing the design limitations of traditional multi-load impedance trajectories, the present application makes the multi-load trajectory design more clear from the perspective of trajectory optimization, thereby facilitating the realization of load scalability. The present application does not require additional costs and is simple to implement, and has a wide application prospect in the high-frequency field. BRIEF DESCRIPTION OF DRAWINGS

[0054] In order to more clearly illustrate the technical solutions in the specific embodiments of the present application or the prior art, the drawings needed to be used in the specific embodiments or prior art description will be briefly introduced. Obviously, the drawings in the following description are some embodiments of the present application, and other drawings can be obtained by those skilled in the art without creative effort on the basis of these drawings.

[0055] Figure 1 Performance comparison diagram for 100 kHz and 6.78 MHz rectifiers;

[0056] Figure 2 Impedance trajectory change diagram for high-frequency system;

[0057] Figure 3 Z pa Impedance-based E-class PA output performance diagram;

[0058] Figure 4 Configuration diagram for intermediate constant-voltage AC bus;

[0059] Figure 5 Conceptual diagram for load impedance compression;

[0060] Figure 6 Circuit configuration diagram;

[0061] Figure 7 Z rec_1 Impedance trajectory-based coil efficiency analysis diagram;

[0062] Figure 8 Equivalent analysis diagram for circuit;

[0063] Figure 9 Impedance change diagram for wired branch under different C r and ICN values;

[0064] Figure 10 Impedance trajectory change diagram for high-frequency system with wireless / wired dual load;

[0065] Figure 11 PA output constant-voltage source u ab schematic diagram through IMN;

[0066] Figure 12 Z pa Impedance trajectory design diagram;

[0067] Figure 13 6.78 MHz high-frequency system diagram with wired / wireless dual load;

[0068] Figure 14 Experimental prototype diagram;

[0069] Figure 15Waveform diagram when mutual inductance is 1.796uH and power output is 33.87W;

[0070] Figure 16 Waveform diagram when mutual inductance is 1.48uH and power output is 36.28W;

[0071] Figure 17 Waveform diagram when mutual inductance is 1.157uH and power output is 40.93W. DETAILED DESCRIPTION

[0072] The technical solutions of the present application will be described below in detail with reference to the drawings. Obviously, the described embodiments are only part of the embodiments of the present application, rather than all the embodiments. Based on the embodiments in the present application, all other embodiments obtained by those skilled in the art without creative work fall within the scope of the present application.

[0073] In the description of the present application, it should be noted that the terms "center", "upper", "lower", "left", "right", "vertical", "horizontal", "inner", "outer" and the like indicate the orientation or positional relationship shown in the drawings, and are only for the convenience of describing the present application and simplifying the description, and do not indicate or imply that the device or element referred to must have a particular orientation, be constructed and operated in a particular orientation, and therefore cannot be understood as a limitation on the present application. In addition, the terms "first", "second", "third" are only for descriptive purposes and cannot be understood as indicating or implying relative importance.

[0074] In the description of the present application, it should be noted that unless otherwise explicitly specified and limited, the terms "mounting", "connection", "connection" should be understood broadly, for example, it can be fixedly connected, or it can be detachably connected, or integrally connected; it can be mechanically connected, or it can be electrically connected; it can be directly connected, or it can be indirectly connected through an intermediate medium; it can be the communication inside two elements. For those skilled in the art, the specific meaning of the above terms in the present application can be understood according to the specific circumstances.

[0075] In addition, the technical features involved in the different embodiments of the present application described below can be combined with each other as long as they do not conflict with each other.

[0076] The present application will be described in detail below in combination with specific embodiments. Embodiment one:

[0078] According to Figures 1-17 As shown in the drawings, the specific optimization technical solution adopted by the present application to solve the above technical problems is: the present application relates to a wired and wireless dual-load high-frequency system based on load impedance compression and a parameter design method.

[0079] The present invention provides a wired and wireless dual-load high-frequency system based on load impedance compression, the system comprising: a constant AC voltage source, a capacitor C tx 、Inductor L tx 、Inductor L rx , capacitor C rx , diode D1, diode D2, capacitor C r1 , capacitor C r2 、Inductor L 01 、Inductor L 02 , capacitor C0, resistor R0, capacitor C b1 、Inductor L b1 , capacitor C b2 、Inductor L b2 , diode D F1 , diode D F2 , capacitor C rF1 , capacitor C rF2 、Inductor L 0F1 、Inductor L 0F2 , capacitor C 0F and resistor R 0F ;

[0080] One end of the constant AC voltage source is connected to a capacitor C tx and capacitor C b1 One end of the capacitor C tx The other end of the inductor L tx One end of the inductor L tx and inductor L rx Mutual inductance; inductance L rx One end of the capacitor C rx One end of the capacitor C rx The other end is connected to diode D1 and capacitor C r1 and inductor L 01 One end of the inductor L 01 The other end of the inductor L 02 , capacitor C0 and one end of resistor R0, the other end of capacitor C0 is connected to the other end of resistor R0 and grounded; diode D1 and capacitor C r1 The other end is connected to diode D2 and capacitor C r2 One end of the diode D2 and capacitor C r2 The other end of the inductor L rx and inductor L 02 the other end;

[0081] The other end of the constant AC voltage source is connected to the inductor L tx and inductor L b1 The other end of the capacitor C b1 The other end of the inductor Lb1 and one end of the capacitor C b2 , the other end of the capacitor C b2 is connected to one end of the inductor L b2 , the other end of the inductor L b2 is connected to the diode D F1 , the capacitor C rF1 and one end of the inductor L 0F1 , the other end of the inductor L 0F1 is connected to the inductor L 0F2 , the capacitor C 0F and one end of the resistor R 0F , the other end of the capacitor C 0F is connected to the other end of the resistor R 0F and is grounded; the other end of the diode D F1 and the capacitor C rF1 is connected to the diode D F2 and the capacitor C rF2 respectively, the other end of the diode D F2 and the capacitor C rF2 is connected to the inductor L b1 and the inductor L 0F2 respectively. Specific embodiment two:

[0083] The difference between the embodiment two and the embodiment one of the application is only that:

[0084] By connecting the load branch reflection impedance Z ref_1 and Z ref_2 in parallel, the synthetic impedance Z IMN is reduced, the impedance conversion network is added in the wired branch to adjust the reflection impedance Z ref_2 , and the fixed power transmission of the wired branch is realized. Specific embodiment three:

[0086] The difference between the embodiment three and the embodiment two of the application is only that:

[0087] For the wireless branch, the secondary side reactive component is compensated to improve the coil efficiency;

[0088] By using the load traction function, the high-efficiency impedance range of the coil under different coupling coefficients k is plotted based on the rectifier input impedance;

[0089] The minimum coupling coefficient is selected for complete compensation, and the rectifier parallel capacitor is designed to achieve the maximum efficiency of the coil under this coupling coefficient;

[0090] By analyzing and calculating the impedance values under different coupling coefficients, the impedance trajectory is obtained;

[0091] Compare the impedance trajectory, observe whether the impedance value is in the coil efficient range under different coupling coefficients, and adjust the value of the rectifier parallel capacitor;

[0092] When the stable voltage is generated, the power of the wired load will remain constant, keeping the impedance value unchanged; in the case of ignoring the line resistance, the synthesized impedance Z IMN . Specific embodiment four:

[0094] The difference between the embodiment four and the embodiment three of the application is only that:

[0095] For the wireless branch, the secondary side reactive component is compensated to improve the coil efficiency, that is:

[0096]

[0097] When the output power of the wireless branch changes, its corresponding X rec_1 The traditional method is used for complete compensation design:

[0098]

[0099] At this time, the reflected impedance Z ref_1 is:

[0100]

[0101] The coil efficiency is expressed by the following formula:

[0102]

[0103] In the traditional method, the coil efficiency is simplified as:

[0104]

[0105] The peak efficiency is derived as:

[0106]

[0107] The real part impedance of the rectifier corresponding to the peak efficiency of the coil is:

[0108]

[0109] Using the load traction function, taking the rectifier input impedance Z rec_1 As the reference, the high efficient impedance range of the coil under different coupling coefficients k is depicted. Specific embodiment five:

[0111] The difference between the embodiment five and the embodiment four of the application is only that:

[0112] For the wired load branch, the ICN is used to reduce the reflected impedance Zref_2 According to the Kirchhoff voltage / current law:

[0113]

[0114] Where u ab is the sinusoidal voltage source generated by the PA through the IMN, and further derivation gives:

[0115] Specific embodiment six:

[0117] The difference between the embodiment six and the embodiment five is only that:

[0118] Let the resonant element satisfy the following equation:

[0119]

[0120] The simplified result is obtained:

[0121]

[0122] When u ab is stable, the power P oF of the wired load will remain constant, and Z rec_2 will remain unchanged. Specific embodiment seven:

[0124] The difference between the embodiment seven and the embodiment six is only that:

[0125] The reflected impedance eliminates the reactive component and moves left and right on the zero imaginary impedance horizontal axis, and has parameter flexibility, in order to reduce the change range of the synthesized impedance Z IMN , Z ref_2 needs to be close to the left on the horizontal axis. Specific embodiment eight:

[0127] The difference between the embodiment eight and the embodiment seven is only that:

[0128] In the case of ignoring the line resistance, Z IMN is equal to:

[0129] Specific embodiment nine:

[0131] The difference between the embodiment nine and the embodiment eight is only that:

[0132] The application provides a computer readable storage medium, which stores a computer program, and the program is executed by a processor to realize a kind of based on load impedance compression wired and wireless dual load high frequency system's synthesized impedance design method. Specific embodiment ten:

[0134] The only difference between the tenth embodiment of the present invention and the ninth embodiment is that:

[0135] The present invention provides a computer device, comprising a memory and a processor, wherein the memory stores a computer program, and when the processor executes the computer program, a synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression is implemented. Specific embodiment eleven:

[0137] After entering the Industrial Scientific Medical Band (ISM), namely 6.78MHz, 13.56MHz and 27.12MHz, the rectifier input voltage and current no longer show zero phase angle characteristics, such as Figure 1 As shown. Due to the increase in the proportion of parasitic capacitance effect, the input current exceeds the input voltage, and the rectifier exhibits capacitive impedance characteristics, and produces nonlinear changes with power changes. Take the full-bridge rectifier as an example:

[0138]

[0139] Among them, Q j is the junction charge stored in the diode junction capacitance. rec_FBR and X rec_FBR R and V are the real and imaginary input impedances of the full-bridge rectifier, respectively. o These correspond to the rectifier output load and voltage, respectively. In low-frequency systems, the load-independent characteristics of LCC / S (or double-sided LCC) derived from classic fundamental wave analysis will vary in high-frequency systems. Therefore, the present invention employs trajectory analysis from a more accurate impedance compression perspective.

[0140] Figure 2 The impedance trajectory changes of a single wired load system and a wired / wireless dual load system under traditional impedance compression design are demonstrated. Figure 3 Shows Figure 2 The output performance of the Class-E power amplifier (PA) used in the analysis. The yellow area is the high efficiency impedance range of the PA, with the PA output impedance Z pa Performing a load pull sweep, as shown by the red line, the PA efficiency increases as it moves toward the center. The purple curve is the PA power output contour line, and as shown by the green arrow, the power output gradually decreases. So, for the single wire load case, the load resistance R L After the rectifier, it presents a capacitive input impedance Z rec ,like Figure 2 (a). Obviously, at this time Z recIt is not within the efficient range of PA, so an impedance matching network (IMN) is needed to rec Convert to Z pa On the one hand, it can meet the needs of PA to work efficiently, and on the other hand, it can adjust the power level required by the system. However, when the number of loads begins to increase, such as Figure 2 As shown in (b), the impedance trajectory begins to become complicated. Similarly, the load R L1 、R L2 (purple triangle) becomes capacitive after passing through the rectifier.

[0141] The wireless branch needs to be able to move freely, so Z rec_1 (Cyan circle) The reflected impedance Z under offset is obtained through coil coupling ref_1 (red triangle). It can be seen that Z ref_1 The impedance variation range is relatively wide, which will lead to the synthetic impedance Z IMN (Green circle) has a larger variation range. Under the same impedance compression design, Z pa It may not be possible to maintain PA high efficiency. Of course, some dynamic IMN can be used to re-optimize the impedance trajectory to ensure Z pa Falling into the high efficiency area, such as Figure 2 (c) shows this. However, this increases system complexity. More importantly, the system's impedance trajectory remains chaotic, making it impossible to accurately determine the power distribution of each load. Power coupling exists between loads, and the independent characteristics of each load are not achieved. Consequently, the requirements for load scalability cannot be met.

[0142] Miniaturization is one of the main trends in the development of power supplies, and high frequency helps to improve the flexibility of the magnetic field space of wireless power transmission technology, making it more suitable for multi-load application scenarios. However, high frequency will also increase the sensitivity of the system, causing power coupling problems in multi-load designs. To this end, the present invention proposes a load impedance compression mechanism for wired / wireless dual-load applications, adding an impedance conversion network to adjust the branch impedance value. From the perspective of impedance trajectory, the impact of the change in coupling coefficient is compressed, the complex and intertwined trajectory curves between multiple loads are decoupled, and the independence of load power is restored, providing a theoretical basis for further packaging and scalability design of multi-load modules.

[0143] Aiming at the problem of chaotic impedance trajectory of multiple loads under traditional impedance compression design, this paper proposes a load impedance compression mechanism. The basic idea is to build an intermediate constant voltage AC bus and divide the complex impedance trajectory design into two parts to solve it, such as Figure 4 shown.

[0144] In this way, compared with the power directly output by the PA, which is difficult to determine, the power obtained by the load through the secondary bus is actually easier to estimate. Therefore, in order to achieve the constant voltage effect of the PA output, it is necessary to significantly reduce the synthetic impedance Z IMN , which is beneficial to Z PA After all, the impedance compression capability of static IMN is limited. The design idea of ​​load impedance compression came into being, such as Figure 5 As shown. By setting the load branch reflected impedance Z ref_1 and Z ref_2 Connect in parallel to reduce the synthetic impedance Z IMN The reflection impedance of the general wireless load branch is affected by the change of the coupling coefficient, resulting in Z ref_1 Therefore, adding an impedance conversion network (ICN) in the wired branch can not only adjust the reflected impedance Z ref_2 , while achieving fixed power transmission of the wired branch. Figure 6 The present invention adopts a post-stage circuit structure that uses a full-wave rectifier, which is relatively common and can provide a good real impedance.

[0145] Next is the formula derivation, the rectifier input impedance is equal to:

[0146]

[0147] Among them, R rec and X rec They represent the real and imaginary parts of the full-wave rectifier input impedance respectively:

[0148]

[0149]

[0150]

[0151]

[0152] Here, D is the duty cycle of the diode, φ rec The above formula can also be regarded as a nonlinear function of load, power and parallel capacitance.

[0153]

[0154] For the wireless branch, it is necessary to compensate the secondary side reactive component as much as possible to improve the coil efficiency, that is:

[0155]

[0156] But as formula (8) shows, when the output power of the wireless branch changes, its corresponding X rec_1 will also change. Therefore, the above formula cannot guarantee that the secondary side is fully compensated under all coupling coefficients. The primary side compensation has no effect, and the traditional method can be used for full compensation design.

[0157]

[0158] At this time, the reflected impedance Z ref_1 is:

[0159]

[0160] Then the coil efficiency can be expressed as:

[0161]

[0162] In the traditional method, only the efficiency analysis of the coil in the full resonance state is usually considered, and at this time the coil efficiency can be simplified as:

[0163]

[0164] Further, the peak efficiency is derived:

[0165]

[0166] At this time, the real part impedance of the rectifier corresponding to the peak efficiency of the coil is:

[0167]

[0168] Formula (15) is usually used as a benchmark for the design of the rectifier parameters of the wireless branch. At the maximum power point, the real part of the input impedance of the rectifier is equal to But in this invention, since the wireless branch cannot be fully compensated under all deviations, a new design method is needed from the perspective of impedance trajectory. Similarly, the load pulling function is used. Here, the input impedance Z rec_1 of the rectifier is used as a benchmark to depict the high-efficiency impedance range of the coil under different coupling coefficients (k), as shown in Figure 7 .

[0169] As can be seen from formula (12) to formula (14), the increase of mutual inductance (M) helps to improve the efficiency of the coil. Therefore, first select the minimum coupling coefficient for full compensation, and design the rectifier parallel capacitor to achieve the maximum efficiency of the coil under this coupling coefficient. Then, by analyzing and calculating the Z rec_1 value under different coupling coefficients, the impedance trajectory of the pink diamond in Figure 7 can be obtained. Finally, by comparing the Z rec_1 impedance trajectory, the Zrec_1 Are they all within the coil's high efficiency range, so as to further adjust the rectifier parallel capacitor C r Therefore, for the wireless branch, C r The choice is relatively flexible.

[0170] For the wired load branch, the design idea is to reduce the reflected impedance Z through ICN. ref_2 . First, according to Kirchhoff's voltage / current law:

[0171]

[0172] Where u ab is the sinusoidal voltage source generated by PA through IMN. Further derivation yields:

[0173]

[0174] At this time, let the resonant element satisfy the following equation:

[0175]

[0176] The simplified result is:

[0177]

[0178] When u ab After the stable voltage is generated, the wired load power P oF will remain constant, keeping Z rec_2 On this basis, we can further ensure that formulas (18) and (19) are valid. This is actually a state interlocking process. Figure 9 Different C r The input impedance Z of the wired branch rectifier under the resonance value of ICN rec_2 and the reflected impedance Z ref_2 It can be seen that the reflected impedance eliminates the reactive component and can move left and right on the zero imaginary impedance horizontal axis, which has strong parameter flexibility. IMN The range of change is needed to make Z ref_2 On the horizontal axis, it approaches the left. But on the other hand, Z ref_2 The value should not be too small, otherwise it will increase the branch current I t_2 , resulting in additional losses. When the internal resistance of the line is ignored, Z IMN equal:

[0179]

[0180] Figure 10The final impedance locus of the wireless / wired dual load high frequency system and its corresponding impedance parameters are given. It can be seen that the wide range of reflection impedance Z ref_1 is caused by the coupling variation ref_2 Under the proper impedance compression of Z IMN , the wide range of Z IMN variation is greatly reduced. The real part of Z IMN changes only 7.25Ω. The three impedance loci are basically maintained near the zero imaginary impedance horizontal axis, which reduces the reactive loss of the system.

[0181] In addition, the extreme compression of Z pa will greatly reduce the design difficulty of the impedance compression network, and facilitate the planning of Z pa impedance locus. The IMN is placed after the classic class-E power amplifier, which plays the role of impedance compression. According to Figure 13 , the PA output impedance Z pa is further derived:

[0182]

[0183] At this time, the three parameter freedoms of the IMN (X T1 -X T3 ) need to be adjusted, so that Z pa impedance locus falls into the PA efficient interval, and at the same time, the power transmission direction meets the relationship between constant voltage output and load, that is, as the load increases, the power obtained should be reduced to maintain constant voltage.

[0184] Under such constraints, Figure 12 the locus of Z pa is given, whose real impedance changes little, and the imaginary part satisfies:

[0185]

[0186] Table 1 Circuit parameters of dual load WPT system

[0187]

[0188] In order to verify the correctness of the proposed theory, an experimental prototype is built, Figure 13 which shows the complete circuit of the proposed system. Table 1 gives the parameter values of each component. Figure 14 As a physical prototype, two electronic loads are used here to simulate the R o and R oFFor high frequency system, the actual circuit production and debugging is more cautious than low frequency kHz system. For example, when drawing the hardware circuit board, it is necessary to pay attention to avoid two different networks directly opposite to each other, avoid additional parasitic capacitance, after all, the commonly used capacitance value of high frequency system is pF level. At the same time, since high frequency system mostly uses air core coil to make resonant inductor, the inductor should be placed along the orthogonal magnetic field direction and leave enough distance to avoid unnecessary coupling between inductive elements. For parameter determination, high precision and high bandwidth impedance network analyzer is needed to measure multiple times and take average value to reduce device error. Finally, according to different circuit boards, actual welding components and circuit simulation, the parameters need to be corrected to obtain the results close to the theory.

[0189] So from the experimental results, when the mutual inductance changes from 1.157 μH to 1.796 μH, the wired branch load always outputs 30W, and the voltage maintains at about 20V, as shown in Figure 15 (c)- Figure 17 (c). Although the rectified waveform has certain asymmetry, it is still within the acceptable range. In a sense, this is also one of the unique working states of this wired / wireless dual load high frequency system. As for the wireless load branch, due to the secondary filtering effect of coil matching, this part of the rectified waveform is not affected by harmonics and maintains good symmetry. With the decrease of mutual inductance, its power output capability gradually increases. Figure 15 Corresponding to the total output of the system 33.87W, the total efficiency of the system is 85.8%. Figure 16 Corresponding to the total output of the system 36.28W, the total efficiency of the system is 88.5%. Figure 17 Corresponding to the total output of the system 40.93W, the total efficiency of the system is 87.3%. Such ideal efficiency is maintained, mainly due to the load compression principle, which reduces the numerical change range of the synthesized impedance Z IMN , so as to realize the high efficiency work of the power amplifier.

[0190] As shown in Figure 15 (a) to Figure 17 (a), it can be seen that under different coil coupling conditions, the PA maintains good soft switching state. Before the driving waveform reaches the threshold voltage of gallium nitride switch, the drain-source voltage of PA has almost dropped to 0 voltage. In addition, by observing the primary and secondary currents of the coil, it can be found that they show good sinusoidal characteristics, which proves that the sinusoidal nature of the intermediate bus voltage is good. The secondary current leads the primary current by nearly 90 degrees, which shows that the compensation network plays a corresponding role.

[0191] The above merely describes a preferred embodiment of the wired and wireless dual load high-frequency system and parameter design method based on load impedance compression, and the protection scope of the wired and wireless dual load high-frequency system and parameter design method based on load impedance compression is not limited to the above-described embodiment. Any technical solution under the same idea belongs to the protection scope of the present application. It should be noted that, for those skilled in the art, some improvements and changes without departing from the principles of the present application should also be considered as the protection scope of the present application.

Claims

1. A synthetic impedance design method for a wired and wireless dual-load high-frequency system based on load impedance compression, the method being implemented based on a wired and wireless dual-load high-frequency system based on load impedance compression, the system comprising: Constant AC voltage source, capacitor C tx 、Inductor L tx 、Inductor L rx , capacitor C rx , diode D1, diode D2, capacitor C r1 , capacitor C r2 、Inductor L 01 、Inductor L 02 , capacitor C0, resistor R0, capacitor C b1 、Inductor L b1 , capacitor C b2 、Inductor L b2 , diode D F1 , diode D F2 , capacitor C rF1 , capacitor C rF2 、Inductor L 0F1 、Inductor L 0F2 , capacitor C 0F and resistor R 0F ; One end of the constant AC voltage source is connected to a capacitor C tx and capacitor C b1 One end of the capacitor C tx The other end of the inductor L tx One end of the inductor L tx and inductor L rx Mutual inductance; inductance L rx One end of the capacitor C rx One end of the capacitor C rx The other end is connected to diode D1 and capacitor C r1 and inductor L 01 One end of the inductor L 01 The other end of the inductor L 02 , capacitor C0 and one end of resistor R0, the other end of capacitor C0 is connected to the other end of resistor R0 and grounded; diode D1 and capacitor C r1 The other end is connected to diode D2 and capacitor C r2 One end of the diode D2 and capacitor C r2 The other end of the inductor L rx and inductor L 02 the other end; The other end of the constant AC voltage source is connected to the inductor L tx and inductor L b1 The other end of the capacitor C b1 The other end of the inductor L b1 and capacitor C b2 One end of the capacitor C b2 The other end of the inductor L b2 One end of the inductor L b2 The other end of the diode D F1 , capacitor C rF1 and inductor L 0F1 One end of the inductor L 0F1 The other end of the inductor L 0F2 , capacitor C 0F and resistor R 0F One end of the capacitor C 0F The other end of the resistor R 0F The other end of the diode D F1 and capacitor C rF1 The other end of each diode is connected to F2 and capacitor C rF2 One end of the diode D F2 and capacitor C rF2 The other end of the inductor L b1 and inductor L 0F2 The other end of the; its characteristics are: For wireless branches, the reactive component on the secondary side is compensated to improve coil efficiency: When the output power of the wireless branch changes, its corresponding X rec_1 Using traditional methods for fully compensated design: At this time, the reflected impedance Z ref_1 for: The coil efficiency is expressed as follows: In the traditional method, the coil efficiency is simplified to: Derive the peak efficiency: The real impedance of the rectifier corresponding to the peak efficiency of the coil is: Use the load pull function to rectifier input impedance Z rec_1 As a benchmark, different coupling coefficients are plotted. k The efficient impedance range of the lower coil; For wired load branches, ICN is used to reduce reflected impedance. Z ref_2 , according to Kirchhoff's voltage / current law: in, u ab is the sinusoidal voltage source generated by PA through IMN, and further derivation yields: ； By using the load-pull function and taking the rectifier input impedance as a benchmark, the efficient impedance range of the coil under different coupling coefficients k is depicted; Select the minimum coupling coefficient for full compensation and design the rectifier shunt capacitor to achieve the maximum efficiency of the coil under this coupling coefficient; By analyzing and calculating the impedance values ​​under different coupling coefficients, the impedance trajectory is obtained; Compare the impedance traces to see if the impedance values ​​under different coupling coefficients are within the high efficiency range of the coil, and adjust the value of the rectifier shunt capacitor. When the stable voltage is generated, the wired load power will remain constant, keeping the impedance value unchanged; when ignoring the line internal resistance, determine the synthetic impedance Z IMN .

2. The method according to claim 1, wherein: By adding the reflected impedance of the load branch Z ref_1 and Z ref_2 Connect in parallel to reduce the combined impedance Z IMN , add an impedance conversion network in the wired branch to adjust the reflected impedance Z ref_2 , to achieve fixed power transmission of wired branches.

3. The method according to claim 1, wherein: Let the resonant element satisfy the following equation: The simplified result is: when u ab After the stable voltage is generated, the wired load power P oF will remain constant, maintain Z rec_2 The unchanged.

4. The method according to claim 3, wherein: The reflected impedance eliminates the reactive component and moves left and right on the zero imaginary impedance horizontal axis, with parameter flexibility to reduce the synthetic impedance. Z IMN The range of change requires Z ref_2 Approaching the left on the horizontal axis.

5. The method according to claim 4, wherein: When the internal resistance of the line is ignored, Z IMN equal: 。 6. A computer-readable storage medium having a computer program stored thereon, characterized in that: The program is executed by a processor to implement the method according to any one of claims 1 to 5.

7. A computer device comprising a memory and a processor, wherein the memory stores a computer program, wherein: When the processor executes the computer program, the method according to any one of claims 1 to 5 is implemented.

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