A three-phase frequency modulation control method and system suitable for SS compensation wireless power transmission system

Abstract

The application discloses a three-phase-shift frequency modulation control method and system suitable for SS compensation wireless power transmission systems and belongs to the field of wireless power transmission. The method comprises the following steps: by adjusting the inter-bridge outer phase shift angle delta and the switching frequency f in real time, minimum reactive ZVS of all switching tubes of the primary side inverter and the secondary side rectifier under system steady state is realized; by PI control on the inter-bridge inner phase shift angle gamma 2 of the rectifier bridge arm, closed loop adjustment on wide output voltage gain is realized; by system voltage gain deduction, sweep parameter and least square method fitting, the optimal efficiency working curve gamma 1 = f (gamma 2) is obtained, so as to realize real-time tracking on the highest efficiency working point of the system under various working conditions. The application can realize minimum reactive ZVS of all switching tubes of the system, system wide output voltage gain adjustment and efficiency optimization by fully and reasonably utilizing four control degrees of freedom of the phase shift angles gamma 1, gamma 2, delta and the switching frequency f, and the introduction of additional power hardware links is avoided.

Description

TECHNICAL FIELD

[0001] The present application belongs to the field of wireless power transfer, and more particularly, to a three-phase frequency-shift control method and control system suitable for SS compensation wireless power transfer (WPT) system. BACKGROUND

[0002] At present, the control strategy of WPT system mainly aims at the realization of zero voltage switch (ZVS) of the converter, wide range power regulation and efficiency optimization, etc. The specific control methods include frequency modulation, phase shift and pulse density modulation, etc.

[0003] In order to realize the ZVS of the primary inverter and the secondary rectifier, some scholars have proposed a double active frequency-shift control strategy which needs to obtain the phase of high-frequency current of the primary and secondary sides in real time. However, the accuracy of current phase detection is easily affected by noise, which makes it difficult to be applied to high-frequency resonant network. In order to solve the above problem, a dynamic ZVS angle control method based on uniform time delay compensation (UTDCM) can effectively improve the accuracy of phase detection. Since this method involves multiple closed loops, the cross coupling between each loop may cause instability at the system level. On the other hand, some scholars have realized the output power regulation and ZVS by using variable frequency control. However, the large range adjustment of switching frequency makes the system seriously detuned, which will lead to a significant reduction in system efficiency. Based on this, subsequent research further optimizes the combination of frequency and phase shift angle, so that the frequency deviation is minimized. However, the iterative calculation based on the model will greatly affect the dynamic performance of the system. In view of this, it is expected to develop a simple control method to realize ZVS and adjustable output voltage without any additional DC-DC converter, switching element or complex resonant network.

[0004] The methods for realizing the efficiency optimization of the WPT system can be mainly divided into three categories: constructing optimal load impedance, feedback control based on wireless communication and feedback control without wireless communication. A direct method based on constructing optimal load impedance is to use the duty cycle control of the DC-DC converter to power the actual load, and according to the type of the power converter, a duty cycle control algorithm can be found to adjust the input impedance of the DC-DC converter to make the load equal to the optimal load impedance value. For the feedback control method without wireless communication, the common method is to adjust the duty cycle of the transmitting inverter to obtain the minimum input power through the perturbation and observation (P&O) method for any output power, so as to automatically realize the maximum energy efficiency of the entire WPT system. However, the above two methods often need to add additional converters or auxiliary circuits, and the power circuit is complicated. In contrast, the feedback control method based on wireless communication can directly use the active rectifier bridge to accurately and stably track the optimal efficiency point without adding additional power hardware units.

[0005] In summary, the existing research on the control method of the wireless power transmission system has described various methods for ensuring the ZVS operation of the inverter, the power or gain control and the efficiency optimization. However, there is still a design challenge in how to fully utilize the control degrees of freedom without adding additional power hardware units, to realize ZVS while maintaining optimal efficiency in a wide load range. SUMMARY

[0006] In view of the defects and improvement needs of the prior art, the present application discloses a three-phase frequency modulation control method and system suitable for SS compensation wireless power transmission system, which aims to fully utilize the control degrees of freedom of the SS compensation double active bridge topology without adding any additional power hardware units, so that all power MOS tubes of the system work at the ZVS operating point with the minimum reactive power in the steady state, and simultaneously realize the wide range controllable output voltage and the dcdc efficiency optimization.

[0007] To achieve the above object, the application provides a three-phase-shift frequency modulation control method suitable for SS compensation wireless power transmission system, which is applied to an SS resonant compensation double active bridge circuit, the SS resonant compensation double active bridge circuit comprising a direct current voltage source, a phase-shift full-bridge inverter, an SS resonant compensation network, a magnetic coupling mechanism, a phase-shift full-bridge rectifier, a filter capacitor and a load resistor, the phase-shift full-bridge inverter comprising power MOS tubes S1-S4, the phase-shift full-bridge rectifier comprising power MOS tubes S5-S8, the magnetic coupling mechanism comprising a primary side transmitting coil and a secondary side receiving coil, and the SS resonant compensation network comprising a primary side capacitor C1 and a secondary side capacitor C2, the three-phase-shift frequency modulation control method comprising four independent control degrees of freedom: a switching frequency f, a phase angle δ of S5 leading S1 opening, a phase angle γ1 of S1 leading S4 opening and a phase angle γ2 of S5 leading S8 opening, and three control targets: tracking of minimum reactive ZVS working points of all power MOS tubes by adjusting f and δ, optimal dcdc efficiency by constraining the numerical relationship between γ1 and γ2, and wide range controllable output voltage gain by adjusting γ1 and γ2.

[0008] Further, the tracking of minimum reactive ZVS working points of all power MOS tubes by adjusting f and δ comprises:

[0009] analyzing phase conditions that voltage and current of all power MOS tubes should satisfy when the minimum reactive ZVS working points are achieved;

[0010] under the phase conditions, establishing a steady-state phasor model of the SS resonant compensation double active bridge circuit;

[0011] combining Kirchhoff's law, calculating circuit equations from the steady-state phasor model, and deducing phasor expressions of key state quantities of the circuit;

[0012] drawing a phasor diagram according to the phasor expressions and the phase conditions, obtaining constraint conditions that f and δ should satisfy when the minimum reactive ZVS working points are achieved according to geometric relationships in the phasor diagram, and thus tracking the minimum reactive ZVS working points under different working conditions by adjusting f and δ in real time.

[0013] Further, the optimal dcdc efficiency by constraining the numerical relationship between γ1 and γ2 comprises:

[0014] sweeping parameters of γ1 and γ2 in the range of 0-π, and obtaining system output voltage V dc and efficiency under different combinations of γ1 and γ2;

[0015] selecting a continuous and monotonic interval, and requiring δV dc / δγ1<0 and δV dc / δγ2<0;

[0016] Selecting the upper and lower limits of the output voltage and the incremental step size, the working curves between γ1 and γ2 under each output voltage are obtained respectively;

[0017] Selecting the combination of γ1 and γ2 with the optimal efficiency on the working curve under each output voltage and drawing a line, the working curve γ1=f(γ2) is obtained by linear least squares fitting, and the optimal dcdc efficiency is realized.

[0018] Further, the wide-range controllable output voltage gain is realized by adjusting γ1 and γ2, including: adjusting the output voltage V dc The deviation signal is input into the PI controller to output γ2, and then γ1 is obtained by γ1=f(γ2), so as to guide the generation of the PWM driving signal and drive the switching action, and the controllable output voltage gain is realized.

[0019] The application also provides a three-phase-shift frequency modulation control system suitable for SS compensation wireless power transmission system, comprising: a SS resonant compensation double active bridge circuit, a DSP chip and a control circuit; wherein the DSP chip is used for executing a three-phase-shift frequency modulation control method, and then receiving a sampling signal and issuing a control instruction through the control circuit such as isolation, signal amplification, filtering and driving, so as to drive the switching devices in the SS resonant compensation double active bridge circuit to complete power control. The three-phase-shift frequency modulation control method comprises four independent control degrees of freedom: the unified switching frequency f of all power switching devices of the primary side inverter and the secondary side rectifier, the phase angle δ of S5 leading S1 opening, the phase angle γ1 of S1 leading S4 opening and the phase angle γ2 of S5 leading S8 opening. By adjusting the above four control variables, three core control targets can be realized: the real-time solution and output of the minimum reactive ZVS working point of all power MOS tubes are realized by adjusting f and δ, the optimal dcdc efficiency is realized by constraining the numerical relationship between γ1 and γ2, and the wide-range controllable output voltage gain is realized by adjusting γ1 and γ2.

[0020] Further, the SS resonant compensation double active bridge circuit comprises a direct current voltage source, a phase-shifted full-bridge inverter, a SS resonant compensation network, a magnetic coupling mechanism, a phase-shifted full-bridge rectifier, a filter capacitor and a load resistor; wherein the phase-shifted full-bridge inverter comprises power MOS tubes S1-S4, each of which is provided with an anti-parallel body diode; the phase-shifted full-bridge rectifier comprises power MOS tubes S5-S8, each of which is provided with an anti-parallel body diode; the SS resonant compensation network comprises a primary side capacitor C1 and a secondary side capacitor C2; the magnetic coupling mechanism comprises a primary side transmitting coil and a secondary side receiving coil; the circuit topology model comprises a primary side self-inductance L1, a primary side line loss resistor R1, a secondary side self-inductance L2, a secondary side line loss resistor R2 and a mutual inductance M; the center frequency is f c , and there are:

[0021]

[0022] To achieve ZVS at minimum reactive power, the phase difference between the AC side voltage and current of the inverter and rectifier should be exactly equal to γ1 / 2 and -γ2 / 2. The system circuit topology is equivalent to Thevenin, and the KVL equation is written on this basis, and the following formula is obtained:

[0023]

[0024] wherein, are the fundamental phasors of the inverter output voltage and the rectifier input voltage (correspondingly, U1 and U2 are the fundamental amplitudes of the inverter output voltage and the rectifier input voltage), are the fundamental phasors of the inverter output current and the rectifier input current, wherein and are opposite to the reference direction, and are the same as the reference direction (correspondingly, I1 and I2 are the fundamental amplitudes of the inverter output current and the rectifier input current), ω = 2πf is the switching angular frequency of the system, and ω0 = 2πf c is the central angular frequency of the system; the meaning of k and its expression are as follows:

[0025]

[0026] Then,

[0027]

[0028] When the system works at the optimal ZVS working point, the following formula is obtained by ignoring the line loss resistance:

[0029]

[0030] wherein, i = 1, 2, are the power factor angles of the AB end and the ab end, respectively, and V in is the inverter DC bus voltage, V dc is the rectifier output voltage. According to the basic equation 1, the basic equation 2, and the optimal working point constraint, the system phasor diagram can be drawn. By using the sine theorem and the cosine theorem, the following formula is easily obtained:

[0031]

[0032] wherein, γ1 and γ2 can be obtained by the DSP chip receiving the calculation results of the previous stage, and U1 and U2 can be obtained by sampling V in and V dcAfter obtaining, the inductance is a known quantity. The unknown quantity is only k related to the frequency f and the phase shift angle δ, which can be actively adjusted to meet the constraint equation, so that the system tracks the optimal ZVS operating point.

[0033] Further, considering line loss, the present application derives the rectifier input (ab terminal) voltage amplitude and dcdc efficiency expression as follows:

[0034]

[0035] Solving the above equation group for all working points (γ2, γ1) in the range of {γ1, γ2|0<γ1<π, 0<γ2<π}, obtaining the system output voltage and efficiency matrix under different γ1 and γ2 combinations; eliminating the no solution interval, selecting the continuous monotonic interval (requiring δV dc / δγ1<0, δV dc / δγ2<0), finally obtaining the method feasible region; selecting the upper and lower limits of the output voltage and the incremental step, respectively obtaining the working curve between γ1 and γ2 under each output voltage; selecting the efficiency optimal working point (γ2, γ1) on each output voltage working curve and drawing it in the two-dimensional coordinate system, performing least squares linear fitting, and finally obtaining the optimal efficiency working curve γ1=f(γ2).

[0036] Further, the output voltage V dc is closed-loop controlled, and V in and V dc are obtained by sampling; the deviation signal is processed and input to the PI control method output γ2 instruction in the DSP chip, and then γ1 is obtained from γ1=f(γ2), and then input to the ZVS constraint link to solve the constraint equations 1 and 2 to obtain δ and f; the above four control variables are input to the PWM generation link to drive the switch action, completing the whole control process.

[0037] Overall, the above technical solutions conceived by the present application can achieve the following beneficial effects:

[0038] (1) Compared with the traditional voltage regulation method of the wireless power transmission system and the feedback control efficiency optimization method without wireless communication, the present application does not need to add extra power hardware link, fully utilizes the control freedom of the inverter and active rectifier, effectively reduces the system redundancy, and reduces the dynamic interference between the front and rear stages of the system.

[0039] (2) The present application adjusts the inverter phase shift angle γ1 and the rectifier phase shift angle γ2 to regulate the output voltage, which can widen the voltage regulation range and realize efficiency optimization compared with the single phase shift control method.

[0040] (3) The three-phase frequency modulation control method suitable for the SS compensation wireless power transmission system proposed in the application successfully realizes the three core goals of ZVS, voltage regulation and efficiency optimization in the same system without adding any additional hardware links, well deals with the common problems faced by the wireless power transmission system, and has strong comprehensive performance and application value. BRIEF DESCRIPTION OF DRAWINGS

[0041] Figure 1 The three-phase frequency modulation control method suitable for the SS compensation wireless power transmission system provided by the application is provided with a block diagram and a system schematic diagram;

[0042] Figure 2 The (a) SS compensation wireless power transmission system circuit topology and its (b) fundamental equivalent simplified topology provided by the application;

[0043] Figure 3 The flow direction conditions that i1 and i2 should meet when the topology described in the application realizes full tube ZVS and the corresponding u AB , i1, u ab , i2 should meet the phase conditions;

[0044] Figure 4 The corresponding phasor diagrams (take 0<δ<90° as an example) of (a) basic equation 1 and (b) basic equation 2 of the system described in the application when realizing ZVS of all power MOS tubes under minimum reactive power;

[0045] Figure 5 The (a) gain and (b) efficiency distribution diagrams of all feasible working points of the system described in the embodiment of the application in the range of {γ1, γ2 | 0<γ1<π, 0<γ2<π};

[0046] Figure 6 The (a) all optimal efficiency working points (γ1, γ2) under each gain and the optimal efficiency working curve γ1=f(γ2) obtained by least square fitting of the system described in the embodiment of the application, and (b) the η-P curve of all working points on the curve;

[0047] Figure 7 The simulation schematic diagram of the load voltage closed-loop regulation of the system described in the embodiment of the application;

[0048] Figure 8 The voltage and current waveforms of the inverter and the rectifier on the AC side when the DC output voltage V dc of the system described in the embodiment of the application is controlled to (a) 45V and (b) 10V. DETAILED DESCRIPTION

[0049] In order to make the objects, technical solutions and advantages of the present application clearer, the present application will be further described in detail below with reference to the drawings and embodiments. It should be understood that the specific embodiments described herein are only used to explain the present application and should not be used to limit the present application. In addition, the technical features involved in the various embodiments of the present application described below can be combined with each other as long as they do not conflict with each other.

[0050] In the present application, the terms "first", "second", etc. (if any) in the present application and the drawings are used to distinguish similar objects, and do not necessarily describe a specific order or sequence.

[0051] Embodiment:

[0052] Referring to Figure 1 , the embodiment provides a three-phase frequency modulation control system suitable for SS compensation wireless power transmission system, which is composed of an SS resonant compensation double active bridge circuit topology, a three-phase frequency modulation control method and a control circuit. Among them, the SS resonant compensation double active bridge circuit topology is responsible for realizing power conversion, the control method is responsible for waveform and power control, and the control circuit is responsible for hardware implementation of the control method. The basic parameters used in the embodiment are shown in Table 1.

[0053] Table 1

[0054] Parameter Value Parameter Value Resonant frequency f c (kHz) 100 Primary side inductance L1 (μH) 40 Secondary side inductance L2 (μH) 40 Mutual Inductance M (μH) 10 Primary capacitor C1 (nF) 63.3 Secondary capacitor C2 (nF) 63.3 Primary line loss resistance R1 (Ω) 0.135 Secondary wire loss resistance R2 (Ω) 0.135 Load Resistance R (Ω) 5 [Source voltage V in (V)]]> 78.54V

[0055] Firstly, the equivalent modeling of the system under steady state is needed. Specifically, due to the existence of the resonant compensation unit, the left part of port AB and the right part of port ab can be respectively subjected to fundamental wave equivalence and Thevenin equivalence, and finally the simplified topology is obtained as shown in (b) of Figure 2 . The phasor method is used to replace the time domain quantities in the system with phasors.

[0056] Further, the zero-voltage turn-on condition of each MOS tube on the topology shown in (a) of Figure 2 is analyzed. The basic principle is that the current i1 or i2 can make the antiparallel diode conductive within the dead time after the opposite side tube is turned off, so as to realize the voltage clamping of the switch tube and the turn-on with the end voltage close to 0. Based on this idea, the flow direction conditions that i1 and i2 should satisfy to realize the zero-voltage turn-on of S1-S8 tubes (with the current reference direction in (a) of Figure 2 as positive) are shown in Table 2.

[0057] Table 2

[0058] Switch Tube No. [i1 flow direction] Switch Tube No. [i2 flow direction] S1 - S5 + S2 + S6 - S3 + S7 - S4 - S8 +

[0059] Further, to meet the current flow conditions described above while minimizing the reactive circulating current, the power factor angle of the inverter output or the rectifier input should be as small as possible. Therefore, if there is a working point that minimizes the power factor angle of the inverter and the rectifier at the same time, it is called the minimum reactive ZVS working point, and the waveform at this time is shown in Figure 3 .

[0060] Further, in combination with the constraint conditions shown in Figure 3 , the phasor diagrams of the basic equations 1 and 2 can be obtained as shown in Figure 4 (a) and (b) (for example, 0<δ<90°). By using the sine theorem, the cosine theorem and classifying discussion, the ZVS constraint equation set with universality can be obtained. This equation set strictly limits the given δ and f required for the system to achieve minimum reactive ZVS under any feasible γ1, γ2. For example, if the upper-level instruction received is γ1=γ2=0, then after calculation, δ=0.5π, f=100kHz.

[0061] Further, for all working points (γ2, γ1) in the range of {γ1, γ2|0<γ1<π, 0<γ2<π}, the system output voltage and efficiency matrix under different combinations of γ1 and γ2 are solved by the equation set; the no-solution interval is eliminated, and the continuous monotonic interval (requiring δV dc / δγ1<0, δV dc / δγ2<0) is selected, and finally the method feasible region is obtained, as shown in Figure 5 .

[0062] The working curve between γ1 and γ2 under each output voltage is obtained; the working point (γ2, γ1) with the optimal efficiency on each output voltage working curve is selected and plotted in a two-dimensional coordinate system, and the least squares linear fitting is performed, and finally the optimal efficiency working curve γ1=f(γ2) is obtained, as shown in Figure 6 (a). The η-P curve corresponding to all working points on the working curve is shown in Figure 6 (b), and when the output efficiency reaches 15% of the maximum output efficiency, the efficiency can reach more than 90%.

[0063] When the system is working, the voltage sampling + low-pass filtering module can obtain the DC side voltage of the inverter and the rectifier in real time. The deviation signal obtained by subtracting the voltage reference value from the collected output voltage is processed and input into the PI control method in the DSP chip to obtain the γ2 instruction. The corresponding γ1 is obtained after the γ2 is input into the optimal efficiency curve γ1=f(γ2). The γ1, γ2 and the collected input voltage V in are input into the ZVS constraint equation to solve δ and f; the above four control variables are input into the PWM generation link to drive the switch action, and finally the whole control process is completed. Among them, the voltage closed-loop effect is as follows:Figure 7 As shown, the system can complete the voltage closed-loop regulation of 1V-49V in about 50ms of adjustment time; meanwhile, the minimum reactive ZVS of all switching tubes can be realized in the steady state working condition of different output voltages, and the AC side voltage and current waveforms of the inverter and rectifier are as shown in Figure 8 .

[0064] Those skilled in the art can easily understand that the above description is only a preferred embodiment of the present application, and is not intended to limit the present application, and any modification, equivalent replacement and improvement made within the spirit and principle of the present application shall be included in the protection scope of the present application.

Claims

1. A three-phase-shift frequency modulation control method suitable for SS-compensated wireless power transmission systems, applied to an SS-resonant compensation dual active bridge circuit, wherein the SS-resonant compensation dual active bridge circuit includes a DC voltage source, a phase-shifted full-bridge inverter, an SS-resonant compensation network, a magnetic coupling mechanism, a phase-shifted full-bridge rectifier, a filter capacitor, and a load resistor; the phase-shifted full-bridge inverter includes power MOSFETs. S 1 ~ S 4. The phase-shifted full-bridge rectifier includes power MOSFETs. S 5 ~ S 8; The magnetic coupling mechanism includes a primary-side transmitting coil and a secondary-side receiving coil; The SS resonant compensation network includes a primary-side capacitor C1 and a secondary-side capacitor C2; characterized in that... The three-phase-shift frequency modulation control method includes: By adjusting f and δ To achieve minimum reactive power ZVS operating point tracking for all power MOSFETs, including: Analyze the phase conditions that the voltage and current of all power MOSFETs must satisfy when operating at the minimum reactive ZVS point; Under the aforementioned phase conditions, a steady-state phasor model of the SS resonant compensation dual active bridge circuit is established; Combining Kirchhoff's laws, the circuit equations are solved using the steady-state phasor model, and the phasor expressions for the key state quantities of the circuit are derived. Phasor diagrams are drawn based on phasor expressions and phase conditions. The operating point at minimum reactive power ZVS is obtained based on the geometric relationships within the phasor diagrams. f and δ The constraints that must be met, thereby enabling real-time adjustments. f and δ To achieve tracking of the minimum reactive ZVS operating point; where, f For switching frequency, δ for S 5 Advanced S 1. Phase angle of activation γ 1 is S 1. Advanced S 4. Phase angle of activation γ 2 is S 5 Advanced S 8. Phase angle at which it is activated; Through constraints γ 1 and γ The numerical relationship between 2 achieves optimal DC-DC efficiency; By adjusting γ 1 and γ 2. Achieve controllable output voltage gain; in, f For switching frequency, δ for S 5 Advanced S 1. Phase angle of activation γ 1 is S 1. Advanced S 4. Phase angle of activation γ 2 is S 5 Advanced S 8. Phase angle at which it is activated.

2. The three-phase-shift frequency modulation control method as described in claim 1, characterized in that, The constraint γ 1 and γ The numerical relationship between 2 achieves optimal DC-DC efficiency, including: right γ 1 and γ 2. Perform a parameter scan within the range of 0 to π to obtain different... γ 1 and γ System output voltage under 2 combinations V dc and efficiency; Select a continuous monotonic interval, and require... V dc / γ 1<0, V dc / γ 2<0; Select the upper and lower limits of the output voltage and the increment step size, and obtain the output voltage at each lower limit. γ 1 and γ Working curves between 2; Select the one with the highest efficiency on the operating curve for each output voltage. γ 1 and γ The two curves are combined and plotted as a line, then subjected to least squares linear fitting to obtain the final working curve. γ 1= f ( γ 2) Achieve optimal DC-DC efficiency.

3. The three-phase-shift frequency modulation control method as described in claim 2, characterized in that, The adjustment γ 1 and γ 2. Achieve controllable output voltage gain, including: controlling the output voltage V dc To perform closed-loop control, the deviation signal is input to the PI controller output. γ 2, then by γ 1= f ( γ 2) Obtain γ 1. This guides the generation of PWM drive signals and drives the switching action, thereby achieving controllable output voltage gain.

4. A three-phase-shift frequency modulation control system suitable for SS-compensated wireless power transmission systems, characterized in that, include: SS resonant compensation dual active bridge circuit, DSP chip, control circuit; The SS resonant compensation dual active bridge circuit includes a DC voltage source, a phase-shifted full-bridge inverter, an SS resonant compensation network, a magnetic coupling mechanism, a phase-shifted full-bridge rectifier, a filter capacitor, and a load resistor. The phase-shifted full-bridge inverter includes power MOSFETs. S 1 ~ S 4. The phase-shifted full-bridge rectifier includes power MOSFETs. S 5 ~ S 8; The magnetic coupling mechanism includes a primary-side transmitting coil and a secondary-side receiving coil; The SS resonant compensation network includes a primary-side capacitor. C 1 and secondary capacitor C 2; The DSP chip is used to execute the three-phase-shift frequency modulation control method. It receives sampling signals and issues control commands through the control circuit to drive the switching devices in the SS resonant compensation dual active bridge circuit to complete power control. The three-phase-shift frequency modulation control method includes: adjusting... f and δ Achieve minimum reactive power ZVS operating point tracking for all power MOSFETs, through constraints. γ 1 and γ The numerical relationship between 2 achieves optimal DC-DC efficiency by adjusting γ 1 and γ 2. Achieve controllable output voltage gain; among which, f For switching frequency, δ for S 5 Advanced S 1. Phase angle of activation γ 1 is S 1. Advanced S 4. Phase angle of activation γ 2 is S 5 Advanced S 8. Phase angle at which the circuit is open; by adjusting f and δ To achieve minimum reactive power ZVS operating point tracking for all power MOSFETs, including: Analyze the phase conditions that the voltage and current of all power MOSFETs must satisfy when operating at the minimum reactive ZVS point; Under the aforementioned phase conditions, a steady-state phasor model of the SS resonant compensation dual active bridge circuit is established; Combining Kirchhoff's laws, the circuit equations are solved using the steady-state phasor model, and the phasor expressions for the key state quantities of the circuit are derived. Phasor diagrams are drawn based on phasor expressions and phase conditions. The operating point at minimum reactive power ZVS is obtained based on the geometric relationships within the phasor diagrams. f and δ The constraints that must be met, thereby enabling real-time adjustments. f and δ To achieve tracking of the minimum reactive ZVS operating point; where, f For switching frequency, δ for S 5 Advanced S 1. Phase angle of activation γ 1 is S 1. Advanced S 4. Phase angle of activation γ 2 is S 5 Advanced S 8. Phase angle at which it is activated.

5. The three-phase-shift frequency modulation control system as described in claim 4, characterized in that, Power MOSFET S 1 ~ S 4. Each of them has a body diode, and the cathode and anode of each body diode are connected to the drain and source of the corresponding power MOSFET, respectively. S The source of 1 and S The drains of the two terminals are connected, and their common terminal A is the first AC output terminal. S The drain of 1 is connected to the positive terminal of the DC voltage source. S The source of 2 is connected to the negative terminal of the DC voltage source. S 1. S 2 form a bridge arm; S 3 source pole and S The drains of the four electrodes are connected, and their common terminal B is the second AC output terminal. S The drain of pin 3 is connected to the positive terminal of the DC power supply. S The source of 4 is connected to the negative terminal of the DC power supply. S 3. S 4 form a bridge arm; Power MOSFET S 5 ~ S Each of the 8 transistors has a body diode, and the cathode and anode of each body diode are connected to the drain and source of the corresponding MOSFET, respectively. S 5 source pole and S The drains of the 6 terminals are connected, and their common terminal a is the first AC input terminal. S 5's drain and filter capacitor C R The positive terminal and one end of the load resistor are connected together. S 6 source and filter capacitor C R The negative terminal is connected to the other end of the load resistor. S 5. S 6 form a bridge arm; S 7 source pole and S The drains of the 8 terminals are connected, and their common terminal b is the second AC input terminal. S 7's drain and filter capacitor C R The positive terminal and one end of the load resistor are connected together. S 8 source and filter capacitor C R The negative terminal is connected to the other end of the load resistor. S 7. S 8 form a bridge arm; The primary capacitor C The first terminal of capacitor 1 is connected to the first AC output terminal A, and the second terminal is connected to one end of the primary-side transmitting coil; the secondary-side capacitor... C 2. The first end is connected to one end of the secondary receiving coil, and the second end is connected to the first AC input terminal a.

6. The three-phase-shift frequency modulation control system as described in claim 4, characterized in that, The constraint γ 1 and γ The numerical relationship between 2 achieves optimal DC-DC efficiency, including: right γ 1 and γ 2. Perform a parameter scan within the range of 0 to π to obtain different... γ 1 and γ System output voltage under 2 combinations V dc and efficiency; Select a continuous monotonic interval, and require... V dc / γ 1<0, V dc / γ 2<0; Select the upper and lower limits of the output voltage and the increment step size, and obtain the output voltage at each lower limit. γ 1 and γ Working curves between 2; Select the one with the highest efficiency on the operating curve for each output voltage. γ 1 and γ The two curves are combined and plotted as a line, then subjected to least squares linear fitting to obtain the final working curve. γ 1= f ( γ 2) Achieve optimal DC-DC efficiency.

7. The three-phase-shift frequency modulation control system as described in claim 6, characterized in that, The adjustment γ 1 and γ 2. Achieve controllable output voltage gain, including: controlling the output voltage V dc To perform closed-loop control, the deviation signal is input to the PI controller output. γ 2, then by γ 1= f ( γ 2) Obtain γ 1. This guides the generation of PWM drive signals and drives the switching action, thereby achieving controllable output voltage gain.

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