Dual circularly polarized filtering antenna based on magnetoelectric dipole

Abstract

The invention belongs to the technical field of antennas, and particularly relates to a dual circularly polarized filtering antenna based on a magnetoelectric dipole, which comprises a dielectric substrate, a 3dB coupler printed on the lower surface of the dielectric substrate, a metal floor printed on the upper surface of the dielectric substrate, and two inverted L-shaped branches arranged on the metal floor in a crossed manner; the two inverted L-shaped branches are crossed to form four quadrants, the four quadrants are internally provided with crossed magnetoelectric dipoles, and the bottom ends of the inverted L-shaped branches are connected with the 3dB coupler. By printing the 3dB coupler feed network on the back of the dielectric substrate, the reconfigurable characteristic of dual circular polarization is realized, the circular polarization axial ratio bandwidth is wide, the introduction of a high-frequency filtering zero point is realized under the condition of not adding an additional filtering structure, the complexity of the antenna is effectively reduced, and the use efficiency of the antenna is improved.

Description

Technical Field

[0001] The present invention belongs to the technical field of antennas, and particularly relates to a dual-circularly polarized filtering antenna based on magnetoelectric dipoles. Background Art

[0002] In long-distance communication scenarios such as satellite communication, it is difficult to achieve linear polarization alignment in the signal transmission and reception systems. The use of circularly polarized antennas can effectively achieve the effective reception of various linear polarization waves; when electromagnetic waves are transmitted over long distances, they will encounter many obstacles, which will cause scattering, reflection, and diffraction of the electromagnetic waves. The receiving antenna will receive signals from different propagation paths, which will cause signal distortion. After the circularly polarized wave is reflected by an obstacle, the rotation direction of the waveform will be reversed. The direct wave and the scattered wave have opposite rotation directions and will cancel each other out; when electromagnetic waves pass through the ionosphere, they will encounter a strong magnetic field. This magnetic field will cause the electromagnetic waves to rotate. If it is a linearly polarized wave, the rotation will cause signal loss. However, if it is a circularly polarized wave, the rotation will not affect the signal; in mobile communication scenarios, in order to achieve communication of various signals within a limited frequency band, an antenna with filtering characteristics is required to suppress signals outside the operating frequency band and ensure that the signal is not interfered by other devices. To generate a circularly polarized wave, two linearly polarized waves with orthogonal directions, equal amplitudes, and a phase difference of 90° are required. According to the feeding method, circularly polarized antennas can be divided into those realized by single-point feeding and multi-point feeding. Among them, the single-feed method is to add some perturbation elements to the antenna, such as slotting, chamfering, loading parasitic elements, etc. These perturbation elements can change the surface current distribution to generate two orthogonal degenerate modes. The amplitudes of the two modes are equal and the phase difference is 90°. The two are combined to radiate a circularly polarized wave. However, the circularly polarized bandwidth excited by the perturbation elements is often narrow and cannot meet the broadband requirements. The other multi-feed method usually feeds multiple antenna elements with a feeding network. The phase difference between the ports is adjusted by adjusting the length of the microstrip line so that the phase difference between the antenna elements is 90°. The radiation modes of multiple antenna elements are combined to synthesize a circularly polarized wave. However, the use of the feeding network and multiple antenna elements increases the antenna aperture. Common circularly polarized antennas often lack filtering capabilities. To improve the suppression ability of out-of-band signals, an additional filter component needs to be cascaded, which introduces additional insertion loss and production cost to the communication system.

[0003] Designing a circularly polarized antenna requires considering many aspects. Existing circularly polarized antennas often achieve a narrow circularly polarized bandwidth by adding perturbation elements to the antenna, or applying orthogonal signals to multiple antenna elements using a feeding network. These practices usually have some disadvantages, such as narrow bandwidth, large antenna aperture, complex structure, etc.

[0004] Examples of existing technical solutions: T-shaped circularly polarized crossed dipole antenna In 2021, Heesu Wang et al. from Ajou University in Suwon, South Korea proposed a size-reduced circularly polarized cross-dipole antenna. This antenna consists of two printed cross-dipole arms and a pair of quarter-wavelength printed loops for phase shifting, which is used to generate circularly polarized radiation. The T-shaped cross-dipole antenna is similar to the traditional cross-dipole antenna, but the difference is that half of the cross-dipole is rotated by 90°. This rotation reduces the area coverage of the dipole by nearly half, thus forming a compact antenna whose CP radiation performance is similar to that of the traditional cross-dipole antenna, achieving a 3 dB axial ratio bandwidth of 12.8% (2.78 - 3.16 GHz). Although the circular polarization bandwidth is wider than that of the traditional microstrip patch type, the communication system has an increasing demand for the broadband of the antenna, and there is still room for improvement in this bandwidth.

[0005] Differential-fed broadband circularly polarized antenna Tu Zhi-Hong et al. from South China University of Technology proposed a differential-fed circularly polarized antenna with common-mode suppression and unidirectional radiation wide axial ratio bandwidth in 2018. First, the proposed antenna uses two cross-dipoles and two ring delay line phase shifters to achieve circular polarization. Second, parasitic patches are used to broaden the axial ratio bandwidth. Third, the cross-dipoles of the proposed antenna can be effectively excited under differential-mode operation, but not under common-mode operation; thus the antenna shows good common-mode suppression. The simulation results show that the proposed antenna has a 3 dB axial ratio bandwidth of 31% (1.74 - 2.38 GHz) at the circular polarization center frequency of 2.06 GHz and a wide -10 dB impedance bandwidth of 60.5% (1.50 - 2.80 GHz) at the center frequency of 2.15 GHz.

[0006] High-gain circularly polarized slot patch antenna Hao Si-Si et al. from Northwestern Polytechnical University proposed a high-gain circularly polarized slot patch antenna in 2020. The proposed antenna consists of a slotted radiation patch and a ground plane, and the structure is as simple as Figure 2-3 shown. Slots are etched on the radiation patch to radiate circular polarization. The proposed slotted patch antenna can stably radiate left-handed circular polarization in the range of 3.25 to 3.43 GHz, achieving a 3 dB circular polarization bandwidth of 5.4%.

[0007] Yang Wan-Jun et al. from South China University of Technology proposed a single-layer low-profile circularly polarized filtering patch antenna for wireless local area networks. Its circularly polarized radiation characteristics are composed of a feeding loop, four L-shaped radiators, and a square ground plane. To integrate the filtering function, four narrow open stubs and four shorting pins are introduced into the patch, and two U-shaped slots are etched in the feeding loop. The circularly polarized bandwidth of the antenna is 4.1%, the peak gain is 8.3 dBic, there is obvious sideband selectivity at the edge of the cutoff bandwidth, and the out-of-band radiation suppression level exceeds 20 dB.

[0008] Therefore, the existing technologies have technical problems such as low circularly polarized bandwidth, the need for an additional feeding network, only having unidirectional circular polarization, and lacking filtering characteristics. Summary of the Invention

[0009] The purpose of the present invention is to provide a dual-circularly polarized filtering antenna based on magnetoelectric dipoles to solve the technical problems existing in the prior art, such as low circularly polarized bandwidth, the need for an additional feeding network, only having unidirectional circular polarization, and lacking filtering characteristics.

[0010] To achieve the above object, the present invention adopts the following technical solutions: A dual-circularly polarized filtering antenna based on magnetoelectric dipoles includes a dielectric substrate. A 3dB coupler is printed on the lower surface of the dielectric substrate, and a metal floor is printed on the upper surface of the dielectric substrate. Two Γ-shaped stubs are cross-arranged on the metal floor; the two Γ-shaped stubs cross to form four quadrants, and cross magnetoelectric dipoles are arranged in all four quadrants. The bottom end of the Γ-shaped stub is connected to the 3dB coupler.

[0011] Preferably, each of the two Γ-shaped stubs includes a first stub, a second stub, and a third stub. The second stub is vertically connected to the first stub and the third stub at both ends to form a Γ shape, and the heights of the first stubs of the two Γ-shaped stubs are different.

[0012] Preferably, fixing holes are opened at the positions of the Γ-shaped stubs on the dielectric substrate. A fixing stub adapted to the fixing holes is provided at the lowermost part of the first stub, and the bottom end of the fixing stub is connected to the microstrip line of the 3dB coupler on the lower surface of the dielectric substrate.

[0013] Preferably, the dielectric substrate is made of FR4 material.

[0014] Preferably, the metal floor is a copper metal floor.

[0015] Preferably, the cross magnetoelectric dipole includes two magnetic arms and one electric arm. The magnetic arms are vertically fixed on the metal floor and are connected at an angle of 90 degrees, and the electric arm is arranged on the two magnetic arms.

[0016] Preferably, a cut groove is opened at the connection position of the electric arm and the magnetic arm.

[0017] Preferably, the 3dB coupler includes a Port1 port, a Port2 port, a Port3 port, and a Port4 port; the Port3 port and the Port4 port are respectively connected to two Γ-shaped stubs; When the Port1 port is used as the coupler input port, the Port2 port is the isolation port, the Port3 port is the coupling port, and the Port4 port is the through port. At this time, the output phase of the through port leads the phase of the coupling port by 90°; When the Port2 port is used as the coupler input port, the Port1 port is the isolation port, the Port3 port is the through port, and the Port4 port is the coupling port.

[0018] Preferably, the theoretical value of the equivalent circuit of the 3dB coupler is calculated by the following formula:

[0019]

[0020]

[0021]

[0022] In the formula, is the characteristic impedance of the circuit; is the impedance of the microstrip line between the Port1 port, the Port2 port, the Port3 port, or the Port4 port and the three-line coupled line; is the impedance of the microstrip line between the Port2 port and the Port4 port or between the Port1 port and the Port3 port; is the impedance of the three-line coupled line; is the power distribution coefficient of the coupler; and are respectively the odd-mode impedance and the even-mode impedance of the three-line coupled line.

[0023] Preferably, the cross-magnetoelectric dipole is made of a metal copper plate.

[0024] Compared with the prior art, the present invention has the following beneficial effects: The integrated feeding structure realizes the reconfigurable characteristic of dual circular polarization. For example, the several antennas listed in the second part only have one circular polarization radiation mode and do not have the circular polarization reconfigurable characteristic; Antenna 2 uses dual-port differential feeding and does not integrate the design of the differential feeding network. In the actual use process, an additional feeding network is required to apply equal-amplitude and 90° phase-difference excitation signals to the two ports of the antenna.

[0025] No additional filtering structure is added. The first three antennas listed in the second part do not have filtering characteristics. Antenna four introduces filtering zeros by adding open stubs and short pins.

[0026] To address the above-mentioned problems, the antenna proposed by the present invention realizes the reconfigurable characteristic of dual circular polarization by printing a 3dB coupler feeding network on the back of the dielectric substrate. It has a wide circular polarization axial ratio bandwidth and introduces high-frequency filtering zeros without adding an additional filtering structure, effectively reducing the complexity of the antenna and improving the antenna usage efficiency. Brief Description of the Drawings

[0027] To more clearly illustrate the technical solutions of the embodiments of the present invention, the following will briefly introduce the drawings required for the embodiments. It should be understood that the following drawings only show some embodiments of the present invention and should not be regarded as limiting the scope. For those of ordinary skill in the art, without creative efforts, other related drawings can also be obtained based on these drawings.

[0028] Figure 1 Schematic diagram of the antenna structure of the present invention; Figure 2 Side view of the Γ-shaped stub of the present invention; Figure 3 Top view of the 3dB coupler of the present invention; Figure 4 Equivalent circuit diagram of the 3dB coupler of the present invention; Figure 5 Schematic diagram of the magnetic dipole arm structure of the present invention; Figure 6 Curve graph of the input impedance Z11 of the present invention; Figure 7 Side view of the magnetic dipole arm and schematic diagram of the stepped impedance resonator of the present invention; where (a) is the side view of the magnetic dipole arm and (b) is the schematic diagram of the stepped impedance resonator Figure 8 Echo loss and realized gain graph of the LHCP mode antenna of the present invention; Figure 9 Axial ratio curve of the LHCP mode antenna of the present invention; Figure 10 Radiation pattern of the LHCP mode antenna of the present invention at different frequency points; where (a) is the frequency point of 1.94 GHz, (b) is the frequency point of 2.65 GHz, (c) is the frequency point of 2.9 GHz, and (d) is the frequency point of 3.42 GHz; Figure 11 Echo loss and realized gain curves of the RHCP mode antenna of the present invention; Figure 12Axial ratio curve of the RHCP mode antenna of the present invention; Figure 13 Radiation pattern of the RHCP mode antenna of the present invention at different frequency points; wherein, (a) is at 2.6 GHz and (b) is at 3.32 GHz.

[0029] Wherein: 1 - crossed magnetoelectric dipole; 101 - electric arm; 102 - magnetic arm; 2 - Γ-shaped stub; 201 - first stub; 2011 - fixed stub; 202 - second stub; 203 - third stub; 3 - 3dB coupler; 4 - dielectric substrate; 5 - slot; 6 - copper metal floor. Detailed implementation manners

[0030] To make the objectives, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings in the embodiments of the present invention. Apparently, the described embodiments are some but not all of the embodiments of the present invention. Usually, the components of the embodiments of the present invention described and illustrated herein can be arranged and designed in various different configurations.

[0031] Therefore, the following detailed description of the embodiments of the present invention provided in the drawings is not intended to limit the scope of the claimed present invention, but merely represents selected embodiments of the present invention. All other embodiments obtained by those of ordinary skill in the art based on the embodiments of the present invention without creative efforts fall within the scope of protection of the present invention.

[0032] It should be noted that: similar reference numerals and letters denote similar items in the following drawings. Therefore, once an item is defined in one drawing, it does not need to be further defined and explained in subsequent drawings.

[0033] In the description of the embodiments of the present invention, it should be noted that if terms such as "upper", "lower", "horizontal", "inner", etc. are used to indicate the orientation or positional relationship, it is based on the orientation or positional relationship shown in the drawings, or the orientation or positional relationship in which the product of the present invention is usually placed during use. It is only for the convenience of describing the present invention and simplifying the description, rather than indicating or implying that the device or element referred to must have a specific orientation, be constructed and operated in a specific orientation, and therefore should not be construed as a limitation of the present invention. In addition, terms such as "first", "second", etc. are only used for distinguishing descriptions and cannot be understood as indicating or implying relative importance.

[0034] In addition, if the term "horizontal" appears, it does not mean that the component is required to be absolutely horizontal, but it can be slightly inclined. For example, "horizontal" only means that its direction is more horizontal relative to "vertical", and does not mean that the structure must be completely horizontal, but it can be slightly inclined.

[0035] In the description of the embodiments of the present invention, it should also be noted that unless otherwise clearly specified and limited, if the terms "set", "installed", "connected", and "coupled" are used, they should be understood in a broad sense. For example, it can be a fixed connection, a detachable connection, or an integral connection; it can be a mechanical connection or an electrical connection; it can be directly connected or indirectly connected through an intermediate medium, and it can be the communication inside two components. For those of ordinary skill in the art, the specific meanings of the above terms in the present invention can be understood according to specific situations.

[0036] The present invention will be further described in detail below with reference to the accompanying drawings: The present invention discloses a dual-circularly polarized filtering antenna based on magnetoelectric dipoles. Refer to Figure 1 , which includes a dielectric substrate 4. A 3 dB coupler 3 is printed on the lower surface of the dielectric substrate 4. A metal floor is printed on the upper surface of the dielectric substrate 4. Two Γ-shaped branches 2 are cross-arranged on the metal floor; the two Γ-shaped branches 2 cross to form four quadrants, and cross magnetoelectric dipoles 1 are arranged in all four quadrants. The bottom ends of the Γ-shaped branches 2 are connected to the 3 dB coupler 3.

[0037] In some embodiments, refer to Figure 1 , Figure 2 , the two Γ-shaped branches 2 both include a first branch 201, a second branch 202, and a third branch 203. The two ends of the second branch 202 are respectively vertically connected to the first branch 201 and the third branch 202 to form a Γ shape. The heights of the first branches 201 of the two Γ-shaped branches 2 are different.

[0038] In some embodiments, refer to Figure 1 , Figure 2 , fixing holes are opened at the positions of the Γ-shaped branches 2 on the dielectric substrate 4. A fixing branch 2011 adapted to the fixing holes is arranged at the lowermost part of the first branch 201, and the bottom end of the fixing branch 2011 is connected to the microstrip line of the 3 dB coupler 3 on the lower surface of the dielectric substrate 4.

[0039] In some embodiments, the dielectric substrate 4 is made of FR4 material.

[0040] In some embodiments, the metal floor is a copper metal floor.

[0041] In some embodiments, refer to Figure 5 , the cross magnetoelectric dipole 1 includes two magnetic arms 102 and one electric arm 101. The magnetic arms 102 are perpendicularly fixed on the metal floor and are connected at an angle of 90 degrees. The electric arm 101 is arranged on the two magnetic arms 102.

[0042] In some embodiments, refer to Figure 5, a cut groove 5 is provided at the connection between the electric arm 101 and the magnetic arm 102.

[0043] In some embodiments, referring to Figure 3 , the 3dB coupler 3 includes a Port1 port, a Port2 port, a Port3 port, and a Port4 port; the Port3 port and the Port4 port are respectively connected to two Γ-shaped stubs 2; When the Port1 port is used as the coupler input port, the Port2 port is the isolation port, the Port3 port is the coupling port, and the Port4 port is the through port. At this time, the output phase of the through port leads the phase of the coupling port by 90°; When the Port2 port is used as the coupler input port, the Port1 port is the isolation port, the Port3 port is the through port, and the Port4 port is the coupling port.

[0044] In some embodiments, referring to Figure 4 , which is the equivalent circuit diagram of the 3dB coupler. The theoretical values of the equivalent circuit of the 3dB coupler 3 are calculated by the following formula:

[0045]

[0046]

[0047]

[0048] In the formula, is the characteristic impedance of the circuit; is the impedance of the microstrip line between the Port1 port, the Port2 port, the Port3 port, or the Port4 port and the three-wire coupling line; is the impedance of the microstrip line between the Port2 port and the Port4 port or between the Port1 port and the Port3 port; is the impedance of the three-wire coupling line; is the power distribution coefficient of the coupler; and are respectively the odd-mode impedance and the even-mode impedance of the three-wire coupling line.

[0049] In some embodiments, the crossed magnetoelectric dipole 1 is made of a metal copper plate.

[0050] In some embodiments, the present invention discloses a dual-circularly polarized filtering antenna based on a magnetoelectric dipole, which is composed of a crossed magnetoelectric dipole 1, Γ-shaped stubs 2, and a 3dB coupler 3 printed on the back of a dielectric substrate. Its antenna structure is as shown in Figure 1As shown. The proposed antenna uses an FR4 material with a thickness of 0.8 mm as the bottom dielectric substrate 4, and a copper metal floor 6 is printed on the upper surface of the dielectric substrate 4. Four crossed magnetoelectric dipoles 1 are arranged above the copper metal floor 6. The width W1 of the crossed magnetoelectric dipole 1 is 35 mm, and the height H is 20 mm. It is made of a metal copper plate with a thickness of 0.5 mm to ensure the stability of the structure. A slot 5 with a length of 18 mm and a width of 0.5 mm is made at the connection of the electric arm 101 and the magnetic arm 102 of each crossed magnetoelectric dipole 1.

[0051] In some embodiments, two Γ-shaped branches 2 located in the middle of the four crossed magnetoelectric dipoles 1 are crossed and arranged for coupling feeding of the crossed magnetoelectric dipoles 1. Figure 2 is a side view of the Γ-shaped branch 2. The Γ-shaped branch is also made of a copper plate with a thickness of 0.5 mm. The entire branch can be divided into three cuboids with lengths of 20 mm, 10 mm, and 12 mm respectively and a width of 2.4 mm. To avoid interference between the two branches, the height of one of the branches is reduced by 1.5 mm. The protruding part at the bottom of the branch with a width of 1.6 mm and a height of 0.8 mm is consistent with the size of the reserved hole on the substrate. The protruding part is snapped into the reserved space to fix the branch on the substrate and is connected to the coupler microstrip line below the substrate.

[0052] Figure 3 is a top view of the 3dB coupler printed on the back of the substrate. This coupler is an improved design based on a branch-line coupler. A short-circuit coupling line is used in the middle parallel-coupled three-line structure to improve the coupling performance. At the same time, the parallel-coupled three-line structure provides a higher coupling coefficient and symmetric transmission phase, improving the bandwidth and performance. Figure 4 is the equivalent circuit diagram of the 3dB coupler. Using the following equations, the theoretical values of the equivalent circuit can be calculated:

[0053] where is the characteristic impedance of the circuit, is the power distribution coefficient of the coupler, , and are the odd-mode impedance and even-mode impedance of the three-line coupling line respectively. When Port 1 is connected to the excitation and Port 2 is connected to the matching load, Port 1 serves as the input port of the coupler, Port 2 is the isolation port, Port 3 is the coupling port, and Port 4 is the through port. At this time, the output phase of the through port leads the phase of the coupling port by 90°; conversely, when Port 2 is used as the input port, Port 1 is the isolation port, Port 3 is the through port, and Port 4 is the coupling port. Port 3 and Port 4 are connected to the Γ-shaped branch. To ensure a constant phase difference of the output signals, the lengths of the microstrip lines connecting Port 3 and Port 4 should be kept as equal as possible.

[0054] Figure 6 It is a curve graph of the input impedance Z11 of the proposed antenna. As can be seen from the figure, at 1.5 GHz, the input resistance of the antenna is very high and the input reactance is zero, which indicates that at this frequency, the antenna has a serious impedance mismatch and is in an open circuit state, resulting in a filtering zero point at this location. Analyzing its principle, the dipole is regarded as a lossy transmission line, and the calculation formula for its input impedance is as follows:

[0055] In the formula, is the average characteristic impedance of the dipole, is the length of the dipole, is the width of the dipole, is the equivalent attenuation constant of the dipole, R is the radiation resistance, is the phase constant of the dipole, , is the effective dielectric constant of the dipole. Then the magnetic arm is regarded as a parallel plate transmission line, and its characteristic impedance is , where is the free space wave impedance, and d and W respectively represent the spacing and width of the parallel plate transmission line. Finally, the input impedance of the transformed dipole along the parallel plate transmission line is calculated by using the impedance transformation formula. For a transmission line terminated by an arbitrary load , we can get:

[0056] Based on the above formula, the theoretically calculated input impedance can be obtained, and it also shows that by changing the physical dimensions of the magnetic dipole and the electric dipole, the position of the zero point can be adjusted, indicating that this zero point is a controllable zero point.

[0057] As Figure 7 shown, making a slot with a length of Lc1 and a width of Wc1 on the side of the dipole arm can be equivalent to a second-order SIR composed of three transmission lines with different impedances and electrical lengths. From bottom to top, the characteristic impedance and electrical length of each section of the transmission line are and . Taking the first-order SIR as an example, the calculation formula for the reflection coefficient: , the impedance ratio , when K≠1, the transmitted signal is reflected, and the stronger the reflection, the stronger the suppression level of the generated filtering zero point. The electrical length θ of each section of the transmission line also affects the frequency position of the transmission zero point. Therefore, by adjusting the physical dimensions of each section of the transmission line, the depth and position of the transmission zero point can be adjusted. For the antenna proposed in this application, the frequency position of the filtering zero point and the out-of-band suppression level can be adjusted by changing the dimensions of the magnetoelectric dipole and the length and width of the slot.

[0058] When the antenna is excited by Port1, the antenna operates in the left-hand circular polarization (LHCP) mode. At this time, the return loss and realized gain of the antenna are as Figure 8 shown. The impedance bandwidth of the antenna (S11 < -10 dB) is 73.9% (1.822 - 3.96 GHz). The peak gain is 9.1 dBi, the suppression level of the filtering zero point at low frequencies is greater than 37 dB, and the suppression level of the filtering zero point at high frequencies is greater than 20 dB. Figure 9 is the axial ratio curve of the antenna in the LHCP mode. At this time, the 3 dB axial ratio bandwidth is 64.8% (1.868 - 3.661 GHz). The overlapping operating bandwidth is 64.8% (1.868 - 3.661 GHz).

[0059] Select 4 axial ratio minimum points (1.94 GHz, 2.65 GHz, 2.9 GHz, 3.42 GHz) to observe the radiation pattern of the antenna. The results are as Figure 10 shown. The maximum radiation direction of the proposed antenna is stably maintained in the positive Z direction. In the maximum radiation direction, the difference between the left-hand circular polarization and the right-hand circular polarization is always greater than 20 dB.

[0060] When the Port2 port is used for excitation, the antenna operates in the right-hand circular polarization (RHCP) mode. At this time, the return loss and realized gain curves of the antenna are as Figure 11 shown. The impedance bandwidth of the antenna (S22 < -10 dB) is 75.2% (1.791 - 3.949 GHz). The peak gain is 9.1 dBi, the suppression level of the filtering zero point at low frequencies is greater than 35 dB, and the suppression level of the filtering zero point at high frequencies is greater than 20 dB. Figure 12 is the axial ratio curve of the antenna in the RHCP mode. At this time, the 3 dB axial ratio bandwidth is 46.9% (2.282 - 3.680 GHz). The overlapping operating bandwidth is 46.9% (2.282 - 3.680 GHz).

[0061] Select 2 axial ratio minimum points (2.6 GHz, 3.32 GHz) to observe the radiation pattern of the antenna. The results are as Figure 13 shown. The maximum radiation direction of the proposed antenna is stably maintained in the positive Z direction. In the maximum radiation direction, the difference between the right-hand circular polarization and the left-hand circular polarization is always greater than 20 dB.

[0062] The above are only the preferred embodiments of the present invention and are not used to limit the present invention. For those skilled in the art, the present invention can have various changes and modifications. Any modification, equivalent replacement, improvement, etc. made within the spirit and principle of the present invention shall be included in the protection scope of the present invention.

Claims

1. A dual circular polarization filtering antenna based on magnetoelectric dipole, characterized in that: The invention comprises a dielectric substrate (4), a 3dB coupler (3) is printed on the lower surface of the dielectric substrate (4), a metal floor is printed on the upper surface of the dielectric substrate (4), and two Γ-shaped branches (2) are cross-arranged on the metal floor; the two Γ-shaped branches (2) cross-form four quadrants, each of the four quadrants is provided with a cross magnetoelectric dipole (1), and the bottom end of the Γ-shaped branch (2) is connected to the 3dB coupler (3).

2. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 1, characterized in that: The two Γ-shaped branches (2) each comprise a first branch (201), a second branch (202) and a third branch (203); two ends of the second branch (202) are respectively vertically connected to the first branch (201) and the third branch (202) to form a Γ shape; the first branches (201) of the two Γ-shaped branches (2) have different heights.

3. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 2, characterized in that: A fixing hole is provided at the position of the Γ-shaped branch node (2) on the dielectric substrate (4); a fixing branch node (2011) adapted to the fixing hole is provided at the bottom of the first branch node (201); and the bottom end of the fixing branch node (2011) is connected to the microstrip line of the 3dB coupler (3) on the lower surface of the dielectric substrate (4).

4. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 1, characterized in that: The dielectric substrate (4) is made of FR4 material.

5. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 1, characterized in that: The metal floor is a copper metal floor.

6. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 1, characterized in that: The cross magnetoelectric dipole (1) comprises two magnetic arms (102) and an electric arm (101); the magnetic arms (102) are vertically fixed to a metal floor and are connected at an angle of 90 degrees; and the electric arm (101) is arranged on the two magnetic arms (102).

7. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 6, characterized in that: A groove (5) is provided at the connection between the electric arm (101) and the magnetic arm (102).

8. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 1, characterized in that: The 3dB coupler (3) comprises a Port 1 port, a Port 2 port, a Port 3 port and a Port 4 port; the Port 3 port and the Port 4 port are respectively connected to two Γ-shaped branches (2); When Port1 is used as the coupler input port, Port2 is an isolation port, Port3 is a coupling port, and Port4 is a through port. At this time, the output phase of the through port leads the phase of the coupling port by 90°. When Port2 is used as the coupler input port, Port1 is an isolation port, Port3 is a through port, and Port4 is a coupling port.

9. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 8, characterized in that: The theoretical value of the equivalent circuit of the 3dB coupler (3) is calculated by the following formula: In the formula, is the characteristic impedance of the circuit; is the impedance of the microstrip line between Port1, Port2, Port3 or Port4 and the three-wire coupled line; is the impedance of the microstrip line between Port2 and Port4 or between Port1 and Port3; is the impedance of the three-wire coupled line; is the power distribution coefficient of the coupler; and They are the odd-mode impedance and even-mode impedance of the three-wire coupled line respectively.

10. The dual circular polarization filtering antenna based on magnetoelectric dipole according to claim 1, characterized in that: The crossed magnetoelectric dipole (1) is made of a metal copper plate.

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