Pulse width modulation method for open-winding permanent magnet synchronous motor with bisymmetric phase voltage arrangement

Abstract

The invention discloses a pulse width modulation method for an open-winding permanent magnet synchronous motor with bisymmetric phase voltage arrangement, and the method comprises the steps: obtaining the three-phase current of the open-winding permanent magnet synchronous motor, and obtaining the dq-axis current according to the three-phase current; obtaining a dq-axis reference voltage according to the dq-axis current; obtaining alpha-beta-axis reference voltage according to the dq-axis reference voltage, obtaining ABC three-phase reference voltage through coordinate transformation, obtaining expected conduction time difference of two bridge arm switching devices in each phase of the inverter according to the ABC three-phase reference voltage through a pulse width modulation method of double-symmetric phase voltage arrangement, and obtaining the difference of the conduction time of the two bridge arm switching devices in each phase of the inverter according to the difference of the conduction time of the two bridge arm switching devices. And the switching state switching time of the upper bridge arm switching device of each phase in the first inverter and the second inverter is obtained, and the switching state switching time is compared with the PWM triangular wave to obtain a switching signal of an inverter driving motor. The current quality of the open-winding permanent magnet synchronous motor can be optimized, the operation efficiency of the open-winding permanent magnet synchronous motor is improved, and the method is simple and easy to implement.

Description

Technical Field

[0001] The present invention belongs to the technical field of motor drive and control, and in particular relates to a pulse width modulation method for an open-winding permanent magnet synchronous motor with a dual-symmetrical phase voltage arrangement. Background Art

[0002] Permanent magnet synchronous motor (PMSM) has become a research focus due to its high efficiency, superior power density, fast dynamic response and other characteristics. The development of high-performance PMSM and its drive control technology has important scientific, economic and social value in promoting the development of related technologies. An important trend in the development of PMSM is the development towards high speed. The high speed of PMSM helps to improve the power density and torque density of the motor, thereby reducing the size and weight of the motor and reducing the consumption of raw materials. In order to broaden the speed regulation range of the drive system, open-winding permanent magnet synchronous motors have begun to be widely studied.

[0003] Since the open-winding permanent magnet synchronous motor needs to control two inverters at the same time, the pulse width modulation (PWM) method of the open-winding motor is relatively complicated, which further increases the requirements of the PWM method on the control chip. At present, some methods have been used to simplify PWM, but at the same time, they increase current harmonics and reduce motor efficiency. Therefore, it is necessary to study simple and efficient PWM methods for open-winding permanent magnet synchronous motors. At present, the PWM methods for open-winding permanent magnet synchronous motors can be roughly divided into the following three categories:

[0004] (1) Space Vector Pulse Width Modulation (SVPWM)

[0005] This type of method divides the three-phase voltage vector of the AC motor into a hexagonal space and selects a suitable space vector for modulation. SVPWM makes the voltage output smoother and more symmetrical, thereby significantly reducing the high-order harmonics in the current and making the motor more efficient. However, this type of PWM method has a large amount of calculation and requires a high calculation speed of the controller.

[0006] (2) Sine wave pulse width modulation (SPWM)

[0007] This type of method transforms the dq0 three-axis reference voltage coordinate of the inverter to the ABC three-phase stationary coordinate system, and modulates according to the reference voltage of the ABC three-phase stationary coordinate system, which can save complex vector synthesis calculations and achieve the purpose of simplifying the PWM algorithm. However, the current harmonic content of the existing SPWM method is high, and the motor efficiency is not as high as the SVPWM method.

[0008] (3) Model Predictive Pulse Width Modulation (MPPWM)

[0009] This type of method brings all voltage vectors into the prediction equation, and brings the predicted value into the cost function, and selects the voltage vector with the minimum cost function to act on the motor. This type of method has a simple principle and strong flexibility. However, this method relies heavily on motor parameters, and considering the large number of voltages in the dual-inverter system of the open-winding motor, this method will greatly increase the burden on the controller. Summary of the Invention

[0010] The object of the present invention is to provide a pulse width modulation method for an open-winding permanent magnet synchronous motor with double-symmetric phase voltage arrangement, which is applicable to an open-winding permanent magnet synchronous motor, is efficient, simple and easy to implement, and has low requirements for the hardware system.

[0011] In order to achieve the above object, the solution of the present invention is:

[0012] A pulse width modulation method for an open-winding permanent magnet synchronous motor with double-symmetric phase voltage arrangement, including,

[0013] Obtain the three-phase current of the open-winding permanent magnet synchronous motor, and obtain the dq-axis current according to the three-phase current;

[0014] Obtain the dq-axis reference voltage according to the dq-axis current;

[0015] Obtain the αβ-axis reference voltage according to the dq-axis reference voltage:

[0016] Obtain the ABC three-phase reference voltage according to the αβ-axis reference voltage:

[0017] Obtain the expected conduction time difference of the two bridge arm switching devices in each phase of the inverter according to the ABC three-phase reference voltage:

[0018] Obtain the switching state switching moments of the upper bridge arm switching devices in each phase of the first inverter and the second inverter according to the expected conduction time difference of the two bridge arm switching devices in each phase of the inverter;

[0019] Compare the switching state switching moments of the upper bridge arm switching devices in each phase of the first inverter and the second inverter with the PWM triangular wave to obtain the switching signal for the inverter to drive the motor.

[0020] Among them, obtaining the three-phase current of the open-winding permanent magnet synchronous motor and obtaining the dq-axis reference voltage according to the three-phase current includes,

[0021] Obtain the three-phase current of the open-winding permanent magnet synchronous motor and obtain the dq-axis current;

[0022] Obtain the dq-axis reference voltage according to the dq-axis current.

[0023] Among them, obtaining the three-phase current of the open-winding permanent magnet synchronous motor and obtaining the dq-axis current includes,

[0024] Obtain the phase A current \(i_{A}\) of the open - winding permanent - magnet synchronous motor A 、phase B current \(i_{B}\) B 、and phase C current \(i_{C}\); C ;

[0025] Calculate the d - axis current \(i_{d}\) d 、q - axis current \(i_{q}\) q and zero - sequence current \(i_{0}\) according to the following formula,

[0026]

[0027] where \(\theta\) is the rotor position angle.

[0028] Among them, according to the dq - axis currents, obtain the dq - axis reference voltages, including,

[0029] Calculate the d - axis reference voltage \(u_{d}\) dref 、q - axis reference voltage \(u_{q}\) qref and 0 - axis reference voltage \(u_{0}\) according to the following formula, 0ref ,

[0030]

[0031] where \(i_{d}^{*}\) dref is the d - axis reference current of the open - winding permanent - magnet synchronous motor, \(i_{q}^{*}\) qref is the q - axis reference current of the open - winding permanent - magnet synchronous motor, \(i_{0}^{*}\) 0ref is the zero - sequence reference current of the open - winding permanent - magnet synchronous motor, \(G_{i}\) PI (s) is the proportional - integral controller of the current loop, \(G_{r}\) PR (s) is the proportional - resonant controller of the current loop; \(i_{d}\) d is the d - axis current, \(i_{q}\) q is the q - axis current, and \(i_{0}\) is the zero - sequence current.

[0032] Among them, according to the dq - axis reference voltages of the open - winding permanent - magnet synchronous motor, obtain the \(\alpha\beta\) - axis reference voltages, including,

[0033] Calculate the \(\alpha\) - axis reference voltage \(u_{\alpha}\) αref and \(\beta\) - axis reference voltage \(u_{\beta}\) according to the following formula, βref ,

[0034]

[0035] where \(u_{d}\) dref is the d - axis reference voltage of the open - winding permanent - magnet synchronous motor, \(u_{q}\) qref is the q - axis reference voltage of the open - winding permanent - magnet synchronous motor, and \(\theta\) is the rotor position angle.

[0036] Among them, according to the \(\alpha\beta\) - axis reference voltages, obtain the ABC - phase reference voltages, including,

[0037] The reference voltage u of phase A is calculated according to the following formula Aref , the reference voltage u of phase B Bref and the reference voltage u of phase C C re f,

[0038]

[0039] wherein, u αref is the reference voltage of the α-axis, u βref is the reference voltage of the β-axis, u 0ref is the reference voltage of the 0-axis.

[0040] Among them, according to the reference voltages of the three phases of ABC, the expected conduction time differences of the switching devices of the two arms in each phase of the inverter are obtained, including

[0041] The expected conduction time difference Δt of the switching devices of the two arms in phase A of the inverter is calculated according to the following formula Aref , the expected conduction time difference Δt of the switching devices of the two arms in phase B Bref , the expected conduction time difference Δt of the switching devices of the two arms in phase C Cref ,

[0042]

[0043] wherein, U DC is the DC bus voltage, T s is the switching period of the inverter switching device, t A1 is the conduction time of the upper-bridge arm switching device in phase A of the first inverter, t A2 is the conduction time of the upper-bridge arm switching device in phase A of the second inverter, t B1 is the conduction time of the upper-bridge arm switching device in phase B of the first inverter, t B2 is the conduction time of the upper-bridge arm switching device in phase B of the second inverter, t C1 is the conduction time of the upper-bridge arm switching device in phase C of the first inverter, t C2 is the conduction time of the upper-bridge arm switching device in phase C of the second inverter; u Aref is the reference voltage of phase A, u Bref is the reference voltage of phase B, u Cref is the reference voltage of phase C.

[0044] Among them, according to the expected conduction time differences of the switching devices of the two arms in each phase of the inverter, the switching state transition moments of the upper-bridge arm switching devices in each phase of the first inverter and the second inverter are obtained, including

[0045] The switching state transition moment T of the upper-bridge arm switching device in phase A of the first inverter is calculated according to the following formulaA1 The switching time T of the switching device of the upper arm of phase A in the second inverter A2 The switching time T of the switching device of the upper arm of phase B in the first inverter B1 The switching time T of the switching device of the upper arm of phase B in the second inverter B2 The switching time T of the switching device of the upper arm of phase C in the first inverter C1 The switching time T of the switching device of the upper arm of phase C in the second inverter C2 ,

[0046]

[0047] where Δt Aref is the expected conduction time difference between the switching devices of the two arms of phase A of the inverter, and Δt Bref is the expected conduction time difference between the switching devices of the two arms of phase B of the inverter, and Δt Cref is the expected conduction time difference between the switching devices of the two arms of phase C of the inverter.

[0048] After adopting the above scheme, the present invention calculates the dq0-axis reference voltage through the field-oriented control method, obtains the ABC three-phase reference voltage through coordinate transformation, and then uses the pulse width modulation method with double-symmetric phase voltage arrangement to optimize the current quality of the open-winding permanent magnet synchronous motor and improve the operation efficiency of the open-winding permanent magnet synchronous motor, which is simple and easy to implement.

[0049] Compared with the prior art, the beneficial effects of the present invention are as follows:

[0050] (1) Compared with the existing space vector pulse width modulation method, the present invention can achieve the same high-quality current as SVPWM, and the calculation amount is greatly reduced, which can reduce the requirements for the control chip.

[0051] (2) Compared with the existing sine wave pulse width modulation method, the present invention further reduces the current ripple according to the principle of double-symmetric phase voltage arrangement, effectively improving the operation efficiency of the motor.

[0052] (3) Compared with the existing model predictive pulse width modulation method, the present invention does not require accurate motor parameters, has stronger robustness, and the calculation amount is greatly reduced. Brief Description of the Drawings

[0053] Figure 1 is a schematic diagram of the control structure of the present invention;

[0054] Figure 2 is a flowchart of the present invention;

[0055] Figure 3It is a schematic diagram of the switching state and waveform of the pulse width modulation method for an open - winding permanent magnet synchronous motor with double - symmetric phase voltage arrangement;

[0056] Among them, (a) is the schematic diagram of the switching state and waveform corresponding to when the phase reference voltage is greater than 0, and (b) is the schematic diagram of the switching state and waveform corresponding to when the phase reference voltage is less than 0;

[0057] Figure 4 It is a comparison chart of experimental results;

[0058] Among them, (a) is the experimental result of traditional space vector pulse width modulation, (b) is the experimental result of the present invention, and (c) is the experimental result of traditional sine wave pulse width modulation;

[0059] Figure 5 It is a comparison chart between the method of the present invention and the traditional space pulse width modulation method. Detailed implementation mode

[0060] The technical solutions and beneficial effects of the present invention will be described in detail below in conjunction with the accompanying drawings.

[0061] The drive system of the embodiment of the present invention includes: a DC voltage source, an inverter circuit, an open - winding permanent magnet synchronous motor, a drive circuit, a current sampling circuit, and a digital control chip. Among them, the DC voltage source provides the DC bus voltage for the inverter circuit, and the current sampling circuit measures the three - phase current of the motor. The connection method of the open - winding permanent magnet synchronous motor and the inverter is as Figure 1 shown. Two sets of traditional three - phase inverters 1 and 2 are respectively connected to one end of the three - phase windings of the open - winding permanent magnet synchronous motor (OW - PMSM), and the two sets of three - phase inverters are powered by the same DC voltage source.

[0062] In this embodiment, the parameters of the open - winding permanent magnet synchronous motor are: DC bus voltage U DC = 100V, number of pole pairs p n = 4, stator phase resistance R s = 1Ω, direct - axis inductance L d = 3mH, quadrature - axis inductance L q = 3mH, zero - axis inductance L0 = 3mH, fundamental permanent - magnet flux linkage ψ f = 0.137Wb, third - harmonic permanent - magnet flux linkage ψ 3f = 0.0062Wb. The specific experimental conditions are: rated speed 1000r / min, switching frequency 5kHz, load torque 3.65Nm, and phase - current amplitude 4A.

[0063] As Figure 1 and Figure 2 shown, the specific steps included in the embodiment are as follows:

[0064] The first step: First, collect the A - phase current i of the motorA and the B-phase current \(i_{B}\) B and the C-phase current \(i_{C}\) C as well as the rotor position angle \(\theta\), the d-axis current \(i_{d}\) d and the q-axis current \(i_{q}\) q and the zero-sequence current \(i_{0}\) of the motor are obtained through coordinate transformation, as shown in Equation (1):

[0065]

[0066] Then, the d-axis reference voltage \(u_{d}\) dref and the q-axis reference voltage \(u_{q}\) qref and the 0-axis reference voltage \(u_{0}\) 0ref of the open-winding permanent magnet synchronous motor are calculated by the current loop controller through the field-oriented control method, as shown in Equation (2):

[0067]

[0068] where \(i_{d}^{*}\) dref is the d-axis reference current of the motor, \(i_{q}^{*}\) qref is the q-axis reference current of the motor, \(i_{0}^{*}\) 0ref is the zero-sequence reference current of the motor, \(G_{PI}(s)\) PI is the proportional-integral controller of the current loop, and \(G_{PR}(s)\) PR is the proportional-resonant controller of the current loop;

[0069] Step 2: According to the d-axis reference voltage \(u_{d}\) dref and the q-axis reference voltage \(u_{q}\) qref as well as the rotor position angle \(\theta\), the \(\alpha\)-axis reference voltage \(u_{\alpha}\) αref and the \(\beta\)-axis reference voltage \(u_{\beta}\) βref are calculated through Equation (3):

[0070]

[0071] Step 3: According to the \(\alpha\)-axis reference voltage \(u_{\alpha}\) αref and the \(\beta\)-axis reference voltage \(u_{\beta}\) βref as well as the 0-axis reference voltage \(u_{0}\) 0ref the A-phase reference voltage \(u_{A}\) Aref and the B-phase reference voltage \(u_{B}\) Bref and the C-phase reference voltage \(u_{C}\) Cref are calculated through Equation (4):

[0072]

[0073] Step 4: According to the A-phase reference voltage \(u_{A}\) Aref and the B-phase reference voltage \(u_{B}\) Bref and the C-phase reference voltage \(u_{C}\) Cref and the DC bus voltage \(U_{dc}\) DCand the switching period T of the inverter switching device s , the difference in the expected conduction time Δt of the switching devices of the two arms of the A-phase of the inverter is calculated by Equation (5) Aref , the difference in the expected conduction time Δt of the switching devices of the two arms of the B-phase of the inverter Bref , the difference in the expected conduction time Δt of the switching devices of the two arms of the C-phase of the inverter Cref :

[0074]

[0075] where t A1 is the conduction time of the upper-arm switching device of the A-phase in Inverter 1, t A2 is the conduction time of the upper-arm switching device of the A-phase in Inverter 2, t B1 is the conduction time of the upper-arm switching device of the B-phase in Inverter 1, t B2 is the conduction time of the upper-arm switching device of the B-phase in Inverter 2, t C1 is the conduction time of the upper-arm switching device of the C-phase in Inverter 1, t C2 is the conduction time of the upper-arm switching device of the C-phase in Inverter 2.

[0076] Step 5: The difference in the expected conduction time Δt of the switching devices of the two arms of the A-phase of the inverter Aref , the difference in the expected conduction time Δt of the switching devices of the two arms of the B-phase of the inverter Bref , the difference in the expected conduction time Δt of the switching devices of the two arms of the C-phase of the inverter Cref , the switching state transition moment T A1 of the upper-arm switching device (S A1 ) of the A-phase in Inverter 1 is calculated by Equation (6), the switching state transition moment T A2 of the upper-arm switching device (S A2 ) of the A-phase in Inverter 2, the switching state transition moment T B1 of the upper-arm switching device (S B1 ) of the B-phase in Inverter 1, the switching state transition moment T B2 of the upper-arm switching device (S B2 ) of the B-phase in Inverter 2, the switching state transition moment T C1 ) of the upper-arm switching device (S C1 ) of the C-phase in Inverter 1, the switching state transition moment T C2 ) of the upper-arm switching device (S C2 of the C-phase in Inverter 2:

[0077]

[0078] Step 6: Compare the switching time T of the switching device of the upper arm of phase A in the inverter 1 A1 , the switching time T of the switching device of the upper arm of phase A in the inverter 2 A2 , the switching time T of the switching device of the upper arm of phase B in the inverter 1 B1 , the switching time T of the switching device of the upper arm of phase B in the inverter 2 B2 , the switching time T of the switching device of the upper arm of phase C in the inverter 1 C1 , the switching time T of the switching device of the upper arm of phase C in the inverter 2 C2 with the PWM triangular wave to obtain the switching signal. Each switching time corresponds to controlling one phase of the bridge arm, that is, the switching time T of the switching device of the upper arm of phase A in the inverter 1 A1 corresponds to controlling the switching device S of the upper arm of phase A in the inverter 1 A1 ; the switching time T of the switching device of the upper arm of phase A in the inverter 2 A2 corresponds to controlling the switching device S of the upper arm of phase A in the inverter 2 A2 ; and so on. The schematic diagram of comparing the switching time with the PWM triangular wave to obtain the switching signal is as shown in Figure 3 , taking Figure 3 (a) as an example, when the switching time T A1 is greater than or equal to the value of the PWM triangular wave, the output of the switching device S of the upper arm of phase A in the inverter 1 A1 is low level, that is, the switching device S of the upper arm of phase A in the inverter 1 A1 is turned off; when the switching time T A1 is less than the value of the PWM triangular wave, the output of the switching device S of the upper arm of phase A in the inverter 1 A1 is high level, that is, the switching device S of the upper arm of phase A in the inverter 1 A1 is turned on; the same applies to the other devices. The switching devices of the same bridge arm of the two inverters all adopt the complementary conduction mode, that is, when the switching device S A1 is turned on, the switching device S of the same bridge arm A3 is turned off; when the switching device S A1 is turned off, the switching device S of the same bridge arm A3 is turned on. Therefore, the switching times T A1 , T B1 , T C1 are used to drive and control the switching devices S of the inverter 1 A1 and S A3 , S B1 and S B3 , S C1 and S C3; Similarly, the switch state switching moments T A2 、T B2 、T C2 for driving and controlling the switching devices S of the inverter 2 A2 and S A4 、S B2 and S B4 、S C2 and S C4 . According to the above steps, the twelve switching devices of the two inverters can be controlled, and then the motor can be driven.

[0079] The present invention performs pulse width modulation according to the principle of double symmetric phase voltage arrangement, optimizes the current ripple of the open winding permanent magnet synchronous motor, and can significantly reduce the computational complexity of the control algorithm while improving the motor efficiency. In the experiment, the bus voltage is 100V, and the reference speed is set to 1000r / min. The experimental results are as Figure 4 shown, and the algorithm comparison results are as Figure 5 shown. The experimental results and the algorithm comparison results show that the present invention can achieve the same high-quality current as the traditional space vector pulse width modulation method, while reducing the algorithm running time by 35% and the chip memory occupancy by 76.5%.

[0080] In summary, the technical solution of the present invention first calculates the dq0-axis reference voltage through the field-oriented control method; obtains the αβ-axis reference voltage by coordinate transformation; obtains the ABC three-phase reference voltage by coordinate transformation; then uses the pulse width modulation method of double symmetric phase voltage arrangement to optimize the current quality of the open winding permanent magnet synchronous motor and improve the operation efficiency of the open winding permanent magnet synchronous motor.

[0081] Those skilled in the art should understand that the embodiments of the present invention can be provided as methods, systems, or computer program products. Therefore, the present invention can take the form of a complete hardware embodiment, a complete software embodiment, or an embodiment combining software and hardware aspects. Moreover, the present invention can take the form of a computer program product implemented on one or more computer-usable storage media (including but not limited to disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code. The solutions in the embodiments of the present invention can be implemented in various computer languages, for example, object-oriented programming languages such as Java and interpreted scripting languages such as JavaScript.

[0082] The present invention is described with reference to the flowcharts and / or block diagrams of methods, apparatus (systems), and computer program products according to embodiments of the present invention. It should be understood that each flow and / or block in the flowchart and / or block diagram, and combinations of flows and / or blocks in the flowchart and / or block diagram can be implemented by computer program instructions. These computer program instructions can be provided to the processor of a general purpose computer, special purpose computer, embedded processor, or other programmable data processing device to produce a machine such that the instructions executed by the processor of the computer or other programmable data processing device generate means for implementing the functions specified in one flow Figure 1 one flow or more flows and / or blocks Figure 1 or means for implementing the functions specified in one block or more blocks.

[0083] These computer program instructions can also be stored in a computer-readable memory that can direct a computer or other programmable data processing device to work in a specific manner, such that the instructions stored in the computer-readable memory produce a manufacture including instruction means that implement the functions specified in one flow Figure 1 one flow or more flows and / or blocks Figure 1 or means for implementing the functions specified in one block or more blocks.

[0084] These computer program instructions can also be loaded onto a computer or other programmable data processing device, such that a series of operation steps are executed on the computer or other programmable device to produce a computer-implemented process, so that the instructions executed on the computer or other programmable device provide steps for implementing the functions specified in one flow Figure 1 one flow or more flows and / or blocks Figure 1 or means for implementing the functions specified in one block or more blocks.

[0085] Although the preferred embodiments of the present invention have been described, those skilled in the art can make additional changes and modifications once they learn the basic creative concept. Therefore, the appended claims are intended to be construed to include the preferred embodiments as well as all changes and modifications that fall within the scope of the present invention.

[0086] Obviously, those skilled in the art can make various changes and variations to the present invention without departing from the spirit and scope of the present invention. Thus, if these modifications and variations of the present invention fall within the scope of the claims of the present invention and their equivalent technologies, the present invention is also intended to include these modifications and variations.

Claims

1. A pulse width modulation method for an open-winding permanent magnet synchronous motor with a dual-symmetrical phase voltage arrangement, characterized in that: include, Obtain the three-phase current of the open-winding permanent magnet synchronous motor, and obtain the dq axis current according to the three-phase current; According to the dq axis current, the dq axis reference voltage is obtained; According to the dq axis reference voltage, the αβ axis reference voltage is obtained: According to the αβ axis reference voltage, the ABC three-phase reference voltage is obtained: According to the ABC three-phase reference voltage, the expected conduction time difference of the two bridge arm switching devices in each phase of the inverter is obtained: According to the expected conduction time difference of the two bridge arm switch devices in each phase of the inverter, the switch state switching time of the upper bridge arm switch devices of each phase in the first inverter and the second inverter is obtained; The switching state switching time of the upper bridge arm switch devices of each phase in the first inverter and the second inverter is compared with the PWM triangle wave to obtain the switching signal of the inverter driving the motor.

2. The method according to claim 1, characterized in that: Get the three-phase current of the open-winding permanent magnet synchronous motor, and get the dq axis reference voltage according to the three-phase current, including: Obtain the three-phase current of the open-winding permanent magnet synchronous motor to obtain the dq axis current; According to the dq axis current, the dq axis reference voltage is obtained.

3. The method according to claim 2, characterized in that: Get the three-phase current of the open-winding permanent magnet synchronous motor and get the dq axis current, including, Get the A phase current i of the open winding permanent magnet synchronous motor A 、B phase current i B 、C phase current i C ; The d-axis current i is calculated according to the following formula d , q-axis current i q and zero sequence current i0, Where θ is the rotor position angle.

4. The method according to claim 2, characterized in that: According to the dq axis current, the dq axis reference voltage is obtained, including, The d-axis reference voltage u is calculated according to the following formula dref , q-axis reference voltage u qref and 0 axis reference voltage u 0ref , Among them, i dref is the d-axis reference current of the open-winding permanent magnet synchronous motor, i qref is the q-axis reference current of the open-winding permanent magnet synchronous motor, i 0ref is the zero-sequence reference current of the open-winding permanent magnet synchronous motor, G PI (s) is the proportional-integral controller of the current loop, G PR (s) is the proportional resonant controller of the current loop; i d is the d-axis current, i q is the q-axis current, and i0 is the zero-sequence current.

5. The method according to claim 1, characterized in that: According to the dq axis reference voltage of the open-winding permanent magnet synchronous motor, the αβ axis reference voltage is obtained, including: The α-axis reference voltage u is calculated according to the following formula αref and β-axis reference voltage u βref , Among them, u dref is the d-axis reference voltage of the open-winding permanent magnet synchronous motor, u qref is the q-axis reference voltage of the open-winding permanent magnet synchronous motor, and θ is the rotor position angle.

6. The method according to claim 1, characterized in that: According to the αβ axis reference voltage, the ABC phase reference voltage is obtained, including, The A phase reference voltage u is calculated according to the following formula Aref , B phase reference voltage u Bref and C phase reference voltage u C re f, Among them, u αref is the α-axis reference voltage, u βref is the β-axis reference voltage, u 0ref is the 0-axis reference voltage.

7. The method according to claim 1, characterized in that: According to the ABC three-phase reference voltage, the expected conduction time difference of the two bridge arm switching devices in each phase of the inverter is obtained, including: The expected conduction time difference Δt of the two bridge arm switching devices of inverter phase A is calculated according to the following formula: Aref , the expected conduction time difference Δt of the two bridge arm switching devices of phase B Bref , the expected conduction time difference Δt of the two bridge arm switching devices of phase C Cref , Among them, U DC is the DC bus voltage, T s is the switching period of the inverter switching device, t A1 is the conduction time of the upper bridge arm switch device of phase A in the first inverter, t A2 is the conduction time of the upper bridge arm switch device of phase A in the second inverter, t B1 is the conduction time of the upper bridge arm switch device of phase B in the first inverter, t B2 is the conduction time of the upper bridge arm switch device of phase B in the second inverter, t C1 is the conduction time of the upper bridge arm switch device of phase C in the first inverter, t C2 is the conduction time of the upper bridge arm switch device of phase C in the second inverter; u Aref is the reference voltage of phase A, u Bref is the reference voltage of phase B, u Cref is the C phase reference voltage.

8. The method according to claim 1, characterized in that: According to the expected conduction time difference between the two bridge arm switch devices in each phase of the inverter, the switch state switching time of the upper bridge arm switch devices in each phase of the first inverter and the second inverter is obtained, including: The switching state switching time T of the upper bridge arm switch device of phase A in the first inverter is calculated according to the following formula: A1 , the switching state switching time T of the A-phase upper bridge arm switch device in the second inverter A2 , the switching state switching time T of the B-phase upper bridge arm switch device in the first inverter B1 , the switching state switching time T of the B-phase upper bridge arm switch device in the second inverter B2 , the switching state switching time T of the C-phase upper bridge arm switch device in the first inverter C1 , the switching state switching time T of the C-phase upper bridge arm switch device in the second inverter C2 , Among them, Δt Aref is the expected conduction time difference between the two bridge arm switching devices of inverter phase A, Δt Bref is the expected conduction time difference between the two bridge arm switching devices of inverter phase B, Δt Cref is the expected conduction time difference of the two bridge arm switching devices of the inverter C phase.

Images