OFDM radar communication integration method based on airspace sidelobe control

Abstract

The invention discloses an OFDM (Orthogonal Frequency Division Multiplexing) radar communication integration method based on airspace sidelobe control, which comprises the following steps of: designing a group of weight vectors and ensuring that main lobe amplitudes of an antenna transmitting directional diagram are consistent and sidelobes in a specific direction have different levels; in a pulse repetition period, each OFDM subcarrier selects a corresponding weight vector according to carried communication information, and transmits the information in a sidelobe amplitude modulation mode; a communication receiving end separates different subcarrier signals and demodulates communication information through matched filtering and threshold judgment; and a radar receiving end performs beam forming, separates echo signals and performs complex gain compensation, and extracts echo delay and Doppler frequency information after matched filtering. According to the invention, by using the OFDM technology and the sidelobe modulation technology, stable high-speed data transmission is realized on the premise of ensuring the main lobe detection performance of the radar.

Description

Technical Field

[0001] The present invention belongs to the field of radar communication technology, and particularly relates to an OFDM radar-communication integration technology based on spatial sidelobe control. Background Art

[0002] Radar-communication integration is a technology that integrates radar detection and communication functions into one system. By realizing the dual functions of target detection and data transmission on the same platform, the resource utilization efficiency can be improved, and the spectrum occupancy can be reduced. To achieve radar-communication integration, a key issue lies in signal design and research on processing methods. Specifically, it is necessary to design a new type of multiplexing waveform that can meet the requirements of radar target detection, effectively carry communication information, and design corresponding signal processing methods and communication information modulation and demodulation strategies.

[0003] The airspace dimension integrated signal design utilizes the directivity of the antenna beam to direct radar and communication signals in different directions in the airspace, enabling independent transmission and effectively reducing interference between the two. This method does not require additional spectrum resources, can significantly improve system performance, and is applicable to scenarios where the directions of radar and communication signals are different. The integrated signal design based on sidelobe control is a scheme to achieve airspace dimension integrated signals. The basic idea is to use the main lobe of the antenna beam for radar detection and perception and use the sidelobes to transmit information. The literature "Euzière J, Guinvarc'h R, Lesturgie M, et al. Dual function radar communication Time modulated array[C]. 2014 International Radar Conference, 2014: 1-4." proposed a radar-communication integration method based on information embedding in a time-modulated array. By changing the phases of the transmitting array elements of different pulses, different sidelobe levels are received by the communication receiver in the set direction, representing different communication symbols. Due to the highly non-linear optimization problem, it is difficult to design multiple transmitting beam patterns with the same main lobe; the literature "Hassanien A, Amin M G, Zhang Y D, et al. Dual-Function Radar-Communications: Information Embedding Using Sidelobe Control and Waveform Diversity[J]. IEEE Transactions on Signal Processing, 2016, 64(8): 2168-2181." proposed a radar-communication integration method based on sidelobe control and waveform diversity. This method controls the sidelobe level of the transmitting direction pattern by optimizing the transmitting weight vector and introduces orthogonal waveforms to improve the communication rate and reduce the bit error rate. The communication receiver judges the symbols through matched filtering and level detection. The disadvantage of this method is that it is difficult to design absolutely orthogonal waveforms, and the non-orthogonality between waveforms may cause interference, thereby affecting the information transmission quality; therefore, the number of orthogonal waveforms should not be too large, the pulse repetition interval is much larger than the pulse period, and the improvement degree of the communication rate is also limited to a certain extent. For the case where the communication direction cannot be accurately estimated, the literature " T, Koivunen V. Improved Beamforming Encoding for Joint Radar and Communication[C]. 2022 IEEE International Conference on Acoustics, Speech and Signal Processing (ICASSP), 2022: 4923-4927. "proposed an improved method to better control the transmit pattern and reduce the bit error rate by optimizing the covariance matrix of the transmit waveform."

[0004] Orthogonal Frequency Division Multiplexing (OFDM) is a modulation technique that divides the spectrum into multiple narrow subcarriers and transmits them orthogonally; by allocating data to multiple orthogonal subcarriers, it effectively resists multipath fading and significantly improves the robustness of the system; since multiple subcarriers can transmit data in parallel, the data transmission rate is greatly increased; in addition, the different subcarriers are closely arranged, making full use of the spectrum resources. These characteristics enable OFDM to have strong anti-interference ability and efficient transmission performance in complex wireless environments and are suitable for high-data-rate communication scenarios. Summary of the Invention

[0005] The problem to be solved by the present invention is that in the scenario where the radar target and the communication user are located in different directions, aiming at the problem of low communication rate existing in the current integrated radar and communication signal with sidelobe amplitude modulation, the present invention provides an integrated radar and communication method with a higher sidelobe communication rate on the premise of ensuring that the main lobe detection performance of the radar is not affected.

[0006] The technical solution adopted by the present invention to solve the above technical problems is an OFDM integrated radar and communication method based on spatial sidelobe control, which is characterized by including the following steps:

[0007] During a pulse repetition period, the integrated signal transmitter selects the corresponding weight vector from the weight vector group according to the communication information of the current time slot to adjust the amplitude and phase of each transmitting antenna in the transmitting antenna array, thereby completing the amplitude modulation of the sidelobes of each OFDM subcarrier and transmitting the OFDM signal. After the transmission is completed, it pauses for a period of time for preamble transmission and then enters the next pulse repetition period; each weight vector in the weight vector group can ensure that the main lobe amplitude of the antenna radiation pattern is consistent, and the sidelobes in specific directions have different sidelobe levels, and the communication information is transmitted through the high and low of the sidelobe levels; the number of OFDM symbols transmitted in a pulse repetition period is equal to the number of bits of the phase encoding signal on each OFDM subcarrier;

[0008] The communication receiving end down-converts the received OFDM signal to a baseband signal, and then separates the received signals of each sub-carrier from the baseband signal for matched filtering. The communication information is demodulated by performing threshold decision on the amplitude of the sub-carrier after matched filtering.

[0009] The radar receiving end performs receiving beamforming on the OFDM signals received by each receiving antenna in the receiving array, and then separates the target echo signals on different sub-carriers. For each sub-carrier, after complex gain compensation, matched filtering is performed, and the range information and velocity information of the target are obtained based on the sub-carrier after matched filtering.

[0010] The present invention combines OFDM with a radar-communication integrated signal with sidelobe control, and gives the corresponding signal processing method, and uses the orthogonal sub-carriers of the OFDM signal to achieve parallel transmission. Combining the advantages of multi-sub-carrier parallel transmission of OFDM with the sidelobe control technology of the array antenna effectively improves the communication rate and anti-interference ability of the system, and at the same time ensures the normal operation of the radar detection function, achieving a balance between radar detection performance and communication performance.

[0011] The beneficial effects of the present invention are as follows: By transmitting information simultaneously on multiple sub-carriers, the communication rate is significantly improved; using the orthogonal characteristics of OFDM technology, the interference between sub-carriers is avoided, ensuring the stable transmission of communication signals; since the radar and communication directions are different, the main lobe detection performance of the radar is not affected, and the accuracy of radar detection can be maintained while ensuring high-efficiency communication. BRIEF DESCRIPTION OF THE DRAWINGS

[0012] Figure 1 is a schematic flow chart of the method of the present invention;

[0013] Figure 2 is the transmit power direction diagram when the communication sidelobe level is 4;

[0014] Figure 3 is the BER simulation diagram under different communication sidelobe levels;

[0015] Figure 4 is the output result diagram of the radar after matched filtering within a single pulse;

[0016] Figure 5 is the result diagram after performing IFFT processing between different sub-carriers within a single pulse;

[0017] Figure 6 is the result diagram after performing FFT processing between different pulses. DETAILED DESCRIPTION OF THE EMBODIMENTS

[0018] An OFDM radar-communication integration method based on spatial sidelobe control, as Figure 1 shown, includes the following steps:

[0019] Step 1: Design a set of weight vectors. Different weight vectors ensure that the main lobe amplitude of the antenna radiation pattern is consistent, and the side lobes in specific directions have different side lobe levels. Use the high and low levels of the side lobe levels to transmit information;

[0020] Specifically, the method for designing the transmit weight vector is as follows: Assume that the radar operates in a certain spatial sector Θ, and take Θ as the main lobe area. The design goal is to minimize the difference between the desired radiation pattern and the actual transmit radiation pattern within the radar operating area; at the same time, the following conditions need to be met: the side lobe area level does not exceed a certain value ε; the level of the communication direction θ c is equal to a specific side lobe level SLL, and the high and low levels of the side lobe level represent different communication information; this design problem can be modeled as:

[0021]

[0022] where, represents minimizing the maximum deviation between the actual radiation pattern and the desired radiation pattern in all possible directions θ ∈ Θ under the given constraint conditions s.t. by adjusting the beamforming vector u n , u n is the nth weight vector in the weight vector group, that is, the nth transmit beamforming weight vector, G d (θ) is the desired transmit beam radiation pattern, Θ is the main lobe area where the radar operates, is the side lobe area, a(θ) is the transmit steering vector in the direction θ, ε is the level threshold, Δ n is the nth specific side lobe level corresponding to the communication direction θ c , and (·) H represents conjugate transpose.

[0023] Step 2: The integrated signal transmitter transmits multiple orthogonal OFDM subcarriers within one pulse repetition period. Each OFDM subcarrier selects a corresponding weight vector according to the carried communication information, and acts on the transmit signal of the subcarrier to transmit the communication information in the form of side lobe amplitude modulation; specifically, the integrated signal transmitter selects a weight vector according to the communication information to adjust the baseband signal to form a subcarrier. After K subcarriers are modulated, they are processed by the inverse discrete Fourier transform IDFT or the inverse fast Fourier transform IFFT, and then through digital-to-analog conversion and up-conversion to form a radio frequency OFDM signal;

[0024] Specifically, consider a radar-communication integrated system that is equipped with N t transmit antennas, and there are N rA receiving antenna, the communication user is a single user and is equipped with a single receiving antenna; assuming that the OFDM signal has K subcarriers, and a phase-encoded signal is carried on each subcarrier. Since the information is modulated by the sidelobe amplitude in this scheme, the phase-encoded signal is only used for subsequent matched filtering processing to improve the signal-to-noise ratio and is not used for transmitting information. The same encoded signal can be used between different subcarriers. There are M OFDM symbols in one time slot (slot) of the signal, and M is equal to the number of bits of the phase-encoded signal. The subcarrier spacing is Δf = 1 / T s , T s is the duration of one OFDM symbol, and the cyclic prefix is not considered. After transmitting M OFDM symbols, the integrated signal will pause for a period of time T off , T off is the preamble reservation time. The integrated signal is transmitted in pulses, and the time of one time slot is equal to the time T of the pulse repetition period r . M OFDM symbols are transmitted within one pulse period, which can be called multi-symbol pulse OFDM radar. However, different from general pulse radars, the T of the present invention off is very short, and the time T off except for transmitting the OFDM signal within one pulse repetition period is used to transmit the preamble. The radar antenna is separated for transmitting and receiving, and it can be considered that the radar operates in a continuous wave regime in the time domain.

[0025] Let the transmit matrix be represents the transmit data on the k-th subcarrier, which is a phase-encoded signal with a length of M; the transmit beamforming weight vector of the k-th subcarrier obtained from step 1 is Applying the weight vector to the transmit signal of this subcarrier, the transmit signal of the m-th symbol on the k-th subcarrier is obtained as:

[0026]

[0027] where, where, g k,m (l) represents the sampling point l of the transmit signal of the m-th symbol on the k-th subcarrier, (·) * represents conjugation, ψ k,m (l) is the complex envelope of the OFDM subcarrier signal, x k,m is the m-th OFDM symbol on the k-th subcarrier in the transmit matrix X, l is the sampling point, k = 1,..., K, K represents the number of OFDM signal subcarriers, m = 1,..., M, M represents the number of OFDM symbols transmitted in one time slot; is the transmit signal on all transmit antennas; calculate the far-field synthesized signal of the antenna in a certain direction θ:

[0028]

[0029] where y m,θ (l) is the composite signal of the m-th symbol in a certain direction θ, is the transmit steering vector; is the radiation pattern of the antenna, and different weight vectors will form specific sidelobe levels in the communication direction.

[0030] Step 3: The communication receiving end performs analog-to-digital conversion A / D and down-conversion on the received RF OFDM signal to obtain a baseband signal. The baseband signal is processed by discrete Fourier transform DFT or fast Fourier transform FFT to separate the received signals on different subcarriers; the received data on each subcarrier is matched-filtered with the transmitted data; the amplitude after the matched filtering is compared with a preset threshold value to demodulate the communication information;

[0031] Specifically, assuming that the communication user is located at the direction θ c and the user has only a single receiving antenna, the signal received by the antenna is down-converted and analog-to-digital converted to obtain:

[0032]

[0033] where is the received signal after down-conversion and analog-to-digital conversion, α is the channel coefficient, which represents the fading experienced by the signal. Without considering the multipath effect, it is assumed that the channel coefficient remains unchanged within a pulse; n c (l) is zero-mean additive white Gaussian noise. The received signal is processed by DFT / FFT, and the output results are arranged to obtain the communication receiving matrix Z c = [z c0 , z c1 , …, z c(K-1) T , where is the received data on a subcarrier, and η c is the output of the noise processed by DFT / FFT.

[0034] Since the phase coding sequences on different subcarriers are known, the received data on a subcarrier is cross-correlated with the transmitted phase coding sequence to implement signal matched filtering, and the following is obtained:

[0035]

[0036] where E c (k, n) is the result of the received data on the k-th subcarrier being matched-filtered with the transmitted phase coding sequence, n is the sampling point after the matched filtering, is the autocorrelation function of the phase coding sequence of the k-th subcarrier, and ψ c (n) is the output of the noise matched filtering;​ is a fixed value; when the channel coefficient α is known, the amplitude after matched filtering depends only on the antenna pattern By adjusting the transmit weight vector u k , the propagation level in the communication direction can be changed, and the signal amplitude after matched filtering at the receiving end is compared with a preset threshold value, thereby demodulating the communication information.

[0037] Step 4: The radar receiving end performs analog-to-digital conversion A / D and down-conversion on the OFDM signals received by different antennas to obtain baseband signals, performs receiving beamforming and DFT / FFT processing on the baseband signals, and separates the target echo signals on different subcarriers; for each subcarrier, complex gain compensation is performed, and the received data is matched-filtered with the transmitted data; the data after matched filtering of each subcarrier is subjected to IDFT / IFFT processing to extract echo delay information; DFT / FFT processing is performed on the received data between different pulses to extract the Doppler frequency information of the echo.

[0038] Specifically, assume that the target is located at direction θ r , the relative distance between the target and the radar is R, the relative speed is v, and the number of receiving antennas at the radar receiving end is N r , the antennas are aligned with the target direction, and the radar blind area is not considered; calculate the signal in the p-th pulse and the m-th OFDM symbol received by the radar:

[0039]

[0040] where is the signal in the p-th pulse and the m-th OFDM symbol received by the radar at direction θ r , β is a coefficient related to target scattering, is the receiving steering vector, is zero-mean additive white Gaussian noise, is the baseband echo signal considering the Doppler frequency and delay; considering that the signal is narrowband, the baseband echo signal is expressed as:

[0041]

[0042] where exp(-j 2πkΔfτ) represents the phase term caused by delay, τ is the target echo delay; exp(j2πmT s f d ) represents the phase term caused by the Doppler frequency of the m-th OFDM symbol; exp(j2πpT r f d ) represents the phase term caused by the Doppler frequency of the p-th pulse; exp(j2πf d lT s( / K) represents the phase term caused by the Doppler frequency at the l-th sampling point. Since the sampling frequency is much higher than the Doppler frequency, this term can be ignored;

[0043] Perform receive beamforming and DFT / FFT processing on the received signal in Equation (6) in the θ r ' direction to obtain:

[0044]

[0045] where r m,p (k) is the signal of the k-th subcarrier on the p-th pulse and the m-th OFDM symbol separated, and η r (k) is the result after noise DFT, is the receive beamforming vector in the θ r ' direction; since the transmit right vectors selected for different subcarriers are different, the phases of will also be different. To ensure coherent accumulation in the subsequent process, complex gain compensation is performed:

[0046]

[0047] where z m,p (k) is the output after compensation, is a constant. When the beamforming direction is equal to the target direction θ r , is completely compensated; arrange the compensated result to obtain the radar receive three-dimensional matrix Z r (k,m,p), where k, m, and p represent the row index, column index, and layer index respectively, corresponding to the k-th subcarrier, the m-th OFDM symbol, and the p-th pulse; perform matched filtering on the data of each row:

[0048]

[0049] where E c (k,n,p) is the result after matched filtering of the k-th row data sequence, n is the sampling point after matched filtering, corresponding to the autocorrelation function of the phase coding sequence. Due to the existence of the Doppler frequency phase term exp(j2πmT s f d ), the autocorrelation function will undergo a certain deformation; ψ r (k,n,p) is the output after noise matched filtering. The role of matched filtering is to determine whether there is a target echo signal. If the amplitude of the matched filtering output signal is low, it means there is no target echo signal and no subsequent steps are required. After matched filtering, E c(k,n,p) generates a peak along the n-axis. Since there is a phase difference of exp(-j2πkΔfτ) between different subcarriers k of the signal, perform IDFT / IFFT on the time-delay terms of each column of E c (k,n,p) to obtain:

[0050]

[0051] where E c (i,n,p) is the result after IDFT / IFFT transformation, i is the sampling point after transformation. When i / K = Δfτ, the exponents cancel each other out, the phase difference is compensated, the signal achieves coherent integration, and the signal E c (i,n,p) generates a peak in the i-n plane. Calculate the echo delay of the signal based on the i coordinate of the peak position, and then determine the target distance. Extract the K×1-dimensional vector corresponding to the column where the peak is located, and combine the vectors of different layers into a K×P-dimensional matrix E' c (i,p). Since there is a phase difference of exp(j2πpT r f d ) between different layers, similarly, perform DFT / FFT processing on the matrix along the p-axis to compensate for the phase difference:

[0052]

[0053] where E' c '(i,u) is the result after DFT / FFT transformation. When u / P = T r f d , the exponents cancel each other out. Calculate the Doppler frequency of the echo based on the u coordinate of the peak position, and then determine the relative velocity of the target.

[0054] Embodiment

[0055] The present invention will be further described in detail below with reference to the accompanying drawings and examples. Set the number of transmitting antennas of the integrated system to N t = 16, the number of receiving antennas of the radar to N r = 16, and the interval between antennas is half a wavelength; the subcarriers of the OFDM signal adopt 16-bit P4 code signals, the number of subcarriers is 256, and the P4 code signals on each subcarrier are the same; the duration of one OFDM symbol is 10 microseconds, the subcarrier interval is 100 kHz, each pulse contains 16 OFDM symbols, and the pulse width of the corresponding signal is 160 microseconds; the pulse repetition period of the signal is 200 microseconds, T off is 40 microseconds, and the pulse repetition frequency is 5 kHz; the carrier frequency f c is 5 GHz.

[0056] Consider a scenario where the direction of the target is different from that of the communication user, and the radar beam points to θ r = 30°, and the main lobe region is [20°, 40°]; the direction where the communication user is located is θ c = -30°. Assume that the transmitter and receiver are completely synchronized, and the cyclic prefix is ignored. Use MATLAB simulation to simulate transmitting 10 4 pulses to the communication user; the channel is modeled as a random variable with a constant magnitude and uniformly distributed random phase, and the multipath effect is not considered. The channel coefficient remains unchanged within each pulse. The target detected by the radar is located in the 32° direction, with a relative distance of 1000 m from the integrated system and a relative speed of 35 m / s. The integrated system transmits a detection signal to the target and generates an echo. The target scattering coefficient is set to a fixed value, the echo signal-to-noise ratio is 0 dB, and the receiver accumulates 256 echo pulses.

[0057] Figure 2 is the relationship diagram between the normalized transmission power of the four weight vectors and the angle when ε = 0.1, and the communication sidelobe levels Δ1 = 0.01, Δ2 = 0.04, Δ3 = 0.07, and Δ4 = 0.1. It can be seen that the patterns obtained by different weight vectors are basically similar in the main lobe region, ensuring the radar detection performance; in the communication direction, the patterns have different sidelobe powers, representing different communication information.

[0058] Figure 3 is the curve of the bit error rate of the integrated signal varying with the signal-to-noise ratio when the number of sidelobe levels is 2, 4, 8, and 16. It can be seen that the fewer the number of sidelobe levels, the lower the bit error rate; as the signal-to-noise ratio increases, the bit error rate gradually decreases; when the number of sidelobe levels is 4, that is, the number of communication bits embedded in the sidelobe is 2 bits, the bit error rate of the present invention drops below 10 -5 at a signal-to-noise ratio of 11 dB, and the corresponding communication rate at this time is 2.56 Mbps. The results show that the present invention can maintain a high communication quality at a low bit error rate and achieve a high communication rate.

[0059] Figure 4 is the output result when the radar received signal within one pulse undergoes subcarrier separation, receive beamforming, complex gain compensation, and is matched filtered with the transmitted signal. The subcarrier generates a peak at the center of the matched filtering sample points.

[0060] Figure 5 is the output result after performing IFFT processing between different subcarriers. When the distance sampling point i = 171, the signals between different subcarriers achieve coherent accumulation, and the output signal generates a peak.

[0061] Figure 6It is the output result after extracting the column where the peak is located and performing FFT processing between different pulses. The signal achieves coherent accumulation between different pulses, generating a peak in the range-Doppler plane. After calculation, the measured range value corresponding to the peak is 1001.953 m, and the measured velocity value is 35.742 m / s.

[0062] The above simulation experiments verify the effectiveness and reliability of the present invention.

[0063] In summary, a signal design method for integrated radar and communication based on spatial sidelobe control. This method realizes the effective integration of radar detection and communication signals by utilizing the subcarriers and sidelobe modulation technology of OFDM signals; it can achieve stable high-speed data transmission under the condition of low bit error rate, and can ensure that the radar system has high detection accuracy, achieving the balance between communication performance and radar detection performance.

Claims

1. An OFDM radar communication integration method based on spatial sidelobe control, characterized in that: The following steps are involved: The integrated signal transmitter selects the corresponding weight vector from the selection weight vector group to adjust the amplitude and phase of each transmitting antenna in the transmitting antenna array according to the communication information of the current time slot within a pulse repetition period, thereby completing the amplitude modulation of the side lobe of each OFDM subcarrier and sending the OFDM signal. After the transmission is completed, it pauses for a period of time for the preamble code transmission and then enters the next pulse repetition period; each weight vector in the weight vector group can ensure that the main lobe amplitude of the antenna transmission pattern is consistent, and the side lobe in a specific direction has different side lobe levels, and the communication information is transmitted by the high and low side lobe levels; The number of OFDM symbols sent in one pulse repetition period is equal to the number of phase coding signal bits on each OFDM subcarrier; The communication receiving end down-converts the received OFDM signal to a baseband signal, separates each subcarrier receiving signal from the baseband signal for matched filtering, and demodulates the communication information by thresholding the subcarrier amplitude after matched filtering. The radar receiving end performs receiving beamforming on the OFDM signals received by each receiving antenna in the receiving array, and then separates the target echo signals on different subcarriers; For each subcarrier, complex gain compensation is performed and then matched filtering is performed, and then the distance information and speed information of the target are obtained based on the subcarrier after matched filtering.

2. The method according to claim 1, characterized in that: When the weight vector acts on the kth OFDM subcarrier to perform sidelobe amplitude modulation, the transmitted signal of the mth symbol is: Among them, g k,m (l) represents the transmitted signal of the mth symbol on the kth subcarrier, l represents the sampling point, u k represents the beamforming weight vector for the kth subcarrier selection, (·)* represents conjugation, and x k,m Represents the mth symbol data on the kth subcarrier, k = 1, ..., K, K represents the number of OFDM signal subcarriers, m = 1, ..., M, M represents the number of OFDM symbols sent in one time slot.

3. The method according to claim 1, characterized in that: In order to avoid the phase difference of the target direction antenna diagram caused by the different transmission weight vectors selected by the integrated signal transmitter for different subcarriers and ensure the coherent accumulation of matched filtering, the radar receiver performs complex gain compensation on the received signal of each subcarrier before matched filtering. The specific method is as follows: Among them, z m,p (k) is the value of r m,p (k) The signal after complex gain compensation, r m,p (k) is the separated signal of the kth subcarrier on the pth pulse and the mth OFDM symbol; u k is the transmit beamforming weight vector of the kth subcarrier, a(θ r ') is the direction θ r 'The launch guidance vector, θ r ' is the beamforming direction, (·) H represents the conjugate transpose.

4. The method according to claim 1, characterized in that: The specific method of obtaining the distance information and speed information of the target based on the subcarrier after matched filtering is as follows: extracting the echo delay information from the subcarriers after matched filtering, and extracting the Doppler frequency information of the echo from the pulses in the subcarrier after matched filtering; finally, obtaining the distance information of the target based on the echo delay information, and obtaining the speed information of the target based on the Doppler frequency information of the echo.

5. The method according to claim 1, characterized in that: When the integrated signal transmitter performs OFDM signal modulation, the phase coding signal on each OFDM subcarrier is not used to transmit information, and the same phase coding signal is used.

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