Fitting MTPV depth field weakening control method based on lead angle in combination with negative q-axis current compensation

Abstract

The invention discloses a fitting MTPV depth field weakening control method based on combination of a lead angle and negative q-axis current compensation. The method comprises the following steps: firstly, analyzing a field weakening control model of the permanent magnet synchronous motor on the basis of vector control, dividing field weakening control into three areas, pointing out a range suitable for depth field weakening, and designing a novel stator current synthesis vector limiting track control scheme suitable for depth field weakening by demonstrating the property of a maximum torque-voltage ratio curve; after a novel negative q-axis current compensation amount gain is designed, a traditional control system is improved, and the whole fitting MTPV deep field weakening control based on the combination of the lead angle and the negative q-axis current compensation is completed. According to the method, the calculation complexity of a control algorithm is effectively reduced, the motor can be subjected to deep field weakening under the condition that a table look-up method is not used, the risk that a current loop is saturated and out of control is effectively reduced, and the motor has good operation performance in a high-speed domain.

Description

Technical Field

[0001] The present invention relates to the technical field of motor drive control, and in particular to a method for deep field-weakening control of an interior permanent magnet synchronous motor. Background Art

[0002] Permanent magnet synchronous motors have been widely used in the industrial field due to their small size, high performance, and excellent speed regulation performance, especially in applications where the motor system is required to provide high-precision operation over a wide speed range and ensure sufficient torque output. Interior permanent magnet synchronous motors have a stronger torque output ability due to their reluctance torque, and thus can meet the requirements of high power and high torque. Generally, within the rated speed range, in order to ensure minimum copper loss and achieve optimal efficiency, the maximum torque per ampere control strategy is adopted. However, once the output voltage reaches the maximum output voltage of the inverter, the magnetic field cannot be further adjusted, resulting in the motor being unable to generate sufficient torque. In addition, when the motor operates in the high-speed region, if the control strategy is improper, reverse torque may be generated, leading to motor out of control. Therefore, it is of great significance to adopt a field-weakening control strategy to ensure that the motor can provide sufficient torque during high-speed operation.

[0003] Traditional field-weakening control generally adopts voltage closed-loop feedback control, but this control method cannot achieve deep field-weakening. As the motor speed increases, the stator current synthesis vector will run downward along the current limit circle until it gets out of control. When the characteristic current point is in an infinite speed-up system within the current limit circle, it is necessary to switch to a deep field-weakening control method to further increase the speed. Deep field-weakening can generally be divided into two methods. One is the look-up table method that obtains the optimal control commands under different working conditions through finite element analysis or off-line experiments. However, the look-up table method depends on pre-collected experimental data, has a high storage cost, and is not portable. The other is to limit the amplitude at the characteristic current point and replace the negative increase of the d-axis current by negatively compensating the q-axis current. This method has a relatively simple control method, but cannot make full use of the maximum torque.

[0004] Researching a new deep field-weakening control system can improve the speed regulation range and operating performance of the motor, while taking into account reducing the control cost, reducing the risk of system out of control, and improving the portability of the control system. Summary of the Invention

[0005] Aiming at the deficiencies in the prior art, the present invention provides a fitting MTPV deep field-weakening control method based on the leading angle combined with negative q-axis current compensation. It reduces the structural complexity of the control system, simplifies the calculation difficulty of the control system, improves the dynamic response speed of the drive system, reduces the risk of system out of control, and enables the motor to have better torque output ability in the high-speed region.

[0006] To achieve the above object, the present invention adopts the following technical solutions:

[0007] In a first aspect, the present invention proposes a fitting MTPV deep field-weakening control method based on the leading angle combined with negative q-axis current compensation, and the method includes the following steps:

[0008] S1: Analyze the problem of establishing the field-weakening control of a permanent magnet synchronous motor based on vector control.

[0009] S2: Design a new stator current synthesis vector control trajectory scheme by demonstrating the properties of the maximum torque per voltage ratio curve.

[0010] S3: Design a new compensation gain scheme based on the demonstrated properties of the maximum torque per voltage ratio curve.

[0011] S4: Design a new field-weakening system and design a new stator current synthesis vector control trajectory operation scheme for the motor in the deep field-weakening region.

[0012] Further, in step S1, the permanent magnet synchronous motor model under field-weakening control is:

[0013] First, the stator voltage equation of the IPMSM in the d-q coordinate system is:

[0014]

[0015] where u d and u q are the d-axis and q-axis stator voltages, i d and i q are the d-axis and q-axis stator currents, L d and L q are the d-axis and q-axis stator inductances, ω e is the rotor electrical angular velocity, R s is the stator winding resistance, ψ d and ψ q are the d-axis and q-axis stator fluxes, and the flux equation components are:

[0016]

[0017] where ψ f is the permanent magnet flux, and the electromagnetic torque equation is thus obtained as:

[0018] T e = 1.5n p (ψ d i q - ψ q i d )

[0019] where T e is the electromagnetic torque, and n p is the number of pole pairs of the motor.

[0020] Due to the relatively small stator resistance of the permanent magnet motor, when the motor operates in the field-weakening region, the voltage drop generated by the resistance part can be ignored because the motor speed is relatively high. And when the motor is in a steady state, the differential term within a unit cycle of the current loop can be regarded as zero. Therefore, the steady-state voltage equation of the IPMSM can be expressed as:

[0021]

[0022] When the IPMSM operates in the self-control mode, its operating state is restricted by the characteristics of the motor itself and the output capacity of the inverter. This restriction includes two aspects: current and voltage. Among them, the maximum value of the current limit is set as I max , and the voltage amplitude limit is set as U max . These two limit parameters are related to the motor parameters and the DC bus voltage, and can be expressed by the following equations:

[0023]

[0024] In the formula, i s is the stator current synthesis vector, and u s is the voltage synthesis vector. From the above formula, in the d-q axis plane, the circle formed by the stator current synthesis vector with the origin as the center is the current limit circle, and the ellipse formed by the voltage synthesis vector with as the center is the voltage limit ellipse.

[0025] When the motor is below the base speed, in order to ensure the minimum copper loss and achieve the best efficiency, the maximum torque per ampere (MTPA) control strategy is adopted. According to the Lagrange extreme value theorem, by introducing the motor parameter formula, the MTPA curve equation below the base speed is obtained:

[0026]

[0027] When the stator current synthesis vector i s moves along the MTPA curve to the current limit circle, it cannot continue to increase the speed along the MTPA curve. At this time, the voltage closed-loop feedback control can be used to make i s be restricted by both the current limit circle and the voltage limit ellipse to operate, so that the motor can continue to increase the speed. If the characteristic current point is located inside the current limit circle, there is a deep field-weakening region, and the stator current synthesis vector i s can move along a new trajectory to make the motor continue to increase the speed.

[0028] If the motor has a deep field-weakening region, limiting along the maximum torque per voltage (MTPV) trajectory can obtain the maximum torque. According to the Lagrange extreme value theorem and introducing the motor parameter formula, the MTPV curve equation is obtained:

[0029]

[0030] However, the MTPV curve equation is relatively complex when controlling the current trajectory, and the MTPV curve can be simplified.

[0031] Define the convex machine ratio Characteristic current The MTPV curve equation can be changed to:

[0032]

[0033] Further let It can be obtained that:

[0034]

[0035] Where m and n are fixed values composed of motor parameters. Analyzing the above formula, the hyperbolic property of the MTPV curve can be obtained. In Figure 3 The center of the hyperbola is (m, 0), and the slopes of the two asymptotes are It can be obtained that an asymptote L of the MTPV curve passing through the second quadrant is:

[0036]

[0037] The asymptote intersects the current limit circle at a point Q in the second quadrant. The abscissa i of point Q dQ is:

[0038]

[0039] The ordinate i of point Q qQ is:

[0040]

[0041] The intersection of the MTPV curve and the current limit circle is defined as point P. The abscissa i of point P dP is:

[0042]

[0043] In the d-q axis voltage plane, if u d and u qIf both are saturated, both current regulators fail, which is called complete out-of-control. The slope of the line connecting the saturation out-of-control point in the second quadrant and the origin is -1. When converted to the current plane, the saturation out-of-control line is a straight line with a slope of -1 passing through the characteristic current point. The intersection of this line and the current limit circle is defined as R, and this line is defined as L RT It turns at point P and does not reach the saturation out-of-control line. In this ideal state, the motor should continue to increase its speed along the MTPV curve. A general method is to avoid complex calculations. The intersection of the line perpendicular to the d-axis passing through the characteristic current point and the current limit circle is defined as N, and L NT is the common operating trajectory of the stator current synthesis vector.

[0044] From the coordinates of point Q and the coordinates of the characteristic current point T a straight line L can be obtained QT , and the slope k n of the straight line is:

[0045]

[0046] The abscissa of the characteristic current point In the d-q axis current plane, it can be obtained that:

[0047] i dP <i dQ <T

[0048] According to the geometric properties of the hyperbola and the straight line, the coordinates of point Q are located above and to the right of point P. Since point P does not reach the saturation out-of-control line, point Q is still within the limit out-of-control range. Then it is easy to obtain:

[0049] k n <-1

[0050] It is easy to obtain that the straight line intersects the MTPV curve at the characteristic current point and a certain point in the second quadrant, and the straight line is close to the MTPV curve. The equation of the straight line can be substituted into the motor formula for pre-calculation to obtain a fixed slope value, avoiding the real-time calculation of the MTPV curve formula. L QT The equation of the straight line is:

[0051] i q =k n (i d -T)

[0052] L QT is the straight line (LMTPV) fitted by the MTPV curve. However, the pre-calculation of this slope still has a certain degree of complexity, which will be further simplified below.

[0053] The torque formula of the motor can be expressed as:

[0054] T e =1.5n p i q ((Ld -L q )i d +ψ f )

[0055] On the d-q axis current plane, the property of the constant torque curve is an inverse proportional function. For the interior permanent magnet synchronous motor L d <L q , where L d -L q has a relatively small absolute value, ψ f has a relatively large value, the magnitude of the torque is mainly affected by the parameter i q . And when the motor operates in the field-weakening region, the running trend of the stator current synthesis vector i s is that i q decreases, i d increases negatively, and the decrease of i q has a greater reduction effect on the magnitude of the torque T e , while the reverse increase of i d has a relatively small increasing effect on the magnitude of the torque T e .

[0056] Then it can be easily inferred that when field-weakening is carried out along the current limit circle and after passing the straight line perpendicular to the characteristic current point, without losing control, the greater the negative increase of i d , the more beneficial it is to output a high torque. Define the slope of the straight line L MT passing through the characteristic current point as k l . If this straight line is taken between L QT and L NT , the closer it is to L QT , the greater the maximum output torque that can be achieved on this trajectory, and the better the performance in the high-speed section. At this time, the range of k l is:

[0057] k l <k n <-1

[0058] Since the slope k l is negative, the smaller k l is taken, until -∞, the closer the straight line L MT is to the L NT selected by the traditional method. The greater k l is taken, the closer it is to -1, and the closer it is to the saturation out-of-control line L RT .

[0059] In the traditional method of negative q-axis current increment compensation, limiting is carried out at i dT . The Δi d generated by the trend of i d increasing negatively under field-weakening control is compensated by the gain k m to generate Δi q, so that i q decreases to continue to increase the speed. k m is:

[0060]

[0061] k m Essentially, the MTPV curve takes i q as the curve slope of the variable. The purpose of this method is to use the change law of MTPV as the compensation amount gain to dynamically reduce i q to compensate for the limited i d , and obtain a current trajectory consistent with the voltage ellipse contraction trend.

[0062] From the above analysis, the property of MTPV is a hyperbola, and it can be obtained that it approaches the asymptote slope at the negative infinity of the d-axis and approaches the slope of the line perpendicular to the d-axis at . Take k m absolute value k' m , then it can be obtained that during the field-weakening control process as the speed increases, i q decreases, the gain becomes larger and larger, and i q is prone to fluctuations in the deep field-weakening area, which is extremely likely to cause the current loop to saturate and get out of control.

[0063] In fact, the control range of deep field-weakening is always limited within the voltage limit ellipse by the voltage closed-loop feedback. Figure 4 In, define a new compensation amount gain k n , similarly, take k n absolute value k' n . A smaller k' n can relieve the pressure of the current loop integration and reduce the possibility of the current loop getting out of control, but too small a k' n will result in a limited speed increase range. It can be set that taking k' n as an example. At this time, limited by the voltage closed-loop feedback compensation amount, Δi is at most I d , then Δi max is at most I q / ρ. If the Q point defined above is taken as the turning point, the Q point is geometrically close to the intersection point P of the MTPV curve and the current limit circle. At the turning point, i max after the reduction of Δi qQ can approach or cross the d-axis, and the speed increase range is sufficient. Limited by the performance of the motor body, a motor with an infinite speed increase system cannot actually increase the speed infinitely in practice. If the demand for the speed increase amount is not large, a smaller fixed value k' q can be selected. n, further reducing the possibility of the current loop getting out of control, that is, reducing the difficulty of PI tuning for the current loop and the voltage closed-loop feedback.

[0064] Based on the traditional voltage closed-loop feedback negative q-axis current increment compensation method, an advanced angle field weakening method is adopted. The control framework is basically the same, which simplifies the calculation burden and speeds up the response speed. Taking the MTPV fitting straight line L described above QT as an example, select i dQ as the d-axis current limit point, and the compensation gain is taken as k n . Add a MAX comparison module. After compensation, corresponds to a point id* on the MTPV fitting straight line L QT . Through the limitation of the comparison module, the current trajectory will be limited to the MTPV fitting straight line and its right side. In Figure 5 , that is, it changes from point G to point F, so that the limit operating trajectory of the stator current synthesis vector i s runs along the current limit circle to point Q and then switches to running along the straight line L QT . The operation process is always restricted by the voltage closed-loop feedback and the current limit circle. The new field weakening limit trajectory will change from the traditional method of A→B→C→D→V→Q to A→B→C→D→E→F. The new method has a larger speed regulation range of the stator current, a larger maximum output torque under the same voltage in the deep field weakening region, and stronger load-carrying capacity in the high-speed section. Using LMTPV to replace the traditional MTPV curve simplifies the calculation amount.

[0065] Adopting the above method can construct a fitting MTPV deep field weakening control system based on the advanced angle negative q-axis current compensation.

[0066] The beneficial effects of the present invention are:

[0067] 1. Provide a new type of field weakening system, which can enable the motor to perform deep field weakening without using the look-up table method.

[0068] 2. Propose the property that the MTPV curve is a hyperbola, and design a MTPV fitting straight line with a small calculation amount.

[0069] 3. Based on the traditional negative q-axis current increment compensation method, propose a new fixed compensation gain to avoid real-time calculation of the gain.

[0070] 4. The proposed new type of field weakening method has good portability, and the above method can be used in segmented field weakening systems, advanced angle field weakening systems, and composite field weakening systems. Description of the Drawings

[0071] Figure 1 is a flowchart of a fitting MTPV deep field weakening control method based on the advanced angle negative q-axis current compensation according to an embodiment of the present invention.

[0072] Figure 2 It is the structural diagram of a fitting MTPV deep field-weakening control method based on leading angle negative q-axis current compensation according to an embodiment of the present invention.

[0073] Figure 3 It is a schematic diagram of the fitting MTPV straight line in the d-q axis coordinate plane according to an embodiment of the present invention.

[0074] Figure 4 It is a schematic diagram of the compensation gain of the traditional method and the new method in the d-q axis coordinate plane according to an embodiment of the present invention.

[0075] Figure 5 It is a schematic diagram of the current limit trajectory of the new field-weakening system in the d-q axis coordinate plane according to an embodiment of the present invention.

[0076] Figure 6 It is the motor simulation result diagram of the new field-weakening system with a given ramp speed according to an embodiment of the present invention Specific implementation scheme

[0077] The present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments.

[0078] Embodiment 1

[0079] Figure 1 It is the flow chart of a fitting MTPV deep field-weakening control method based on leading angle negative q-axis current compensation according to an embodiment of the present invention. Figure 2 It is the structural diagram of a fitting MTPV deep field-weakening control method based on leading angle negative q-axis current compensation. This embodiment proposes a fitting MTPV deep field-weakening control method based on the combination of leading angle and negative q-axis current compensation. The method includes the following steps:

[0080] S1: Analyze the problem of establishing the field-weakening control of the permanent magnet synchronous motor based on vector control.

[0081] S2: Design a new stator current synthesis vector control trajectory scheme by demonstrating the properties of the maximum torque voltage ratio curve.

[0082] S3: Design a new compensation amount gain scheme by the demonstrated properties of the maximum torque voltage ratio curve.

[0083] S4: Design a new field-weakening system and design the operation scheme of the new stator current synthesis vector control trajectory of the motor in the deep field-weakening region.

[0084] I. Mathematical model of permanent magnet synchronous motor under field-weakening control

[0085] Furthermore, in step S1, the permanent magnet synchronous motor model under weak magnetic control is as follows:

[0086] First, the stator voltage equation of the IPMSM in the d-q coordinate system is:

[0087]

[0088] In the formula, u d , u q are the stator voltages on the d and q axes, i d , i q are the stator currents on the d and q axes, L d , L q are the stator inductances on the d and q axes, ω e is the rotor electrical angular velocity, R s is the stator winding resistance, ψ d , ψ q are the stator flux linkages on the d and q axes. Among them, the components of the flux linkage equation are:

[0089]

[0090] In the formula, ψ f is the permanent magnet flux linkage. From this, the electromagnetic torque equation is obtained as:

[0091] T e =1.5n p (ψ d i q -ψ q i d )

[0092] In the formula, T e is the electromagnetic torque, n p is the number of pole pairs of the motor.

[0093] Since the stator resistance of the permanent magnet motor is small, when the motor operates in the weak magnetic region, due to the high motor speed, the voltage drop generated by the resistance part can be ignored. And when the motor is in a steady state, the differential term within a unit period of the current loop can be regarded as zero. Therefore, the steady-state voltage equation of the IPMSM can be expressed as:

[0094]

[0095] When the IPMSM operates in the self-control mode, its operating state is restricted by the motor's own characteristics and the output capacity of the inverter. This restriction includes two aspects: current and voltage. Among them, the maximum value of the current limit is set as I max , and the voltage amplitude limit is set as U max . These two limit parameters are related to the motor parameters and the DC bus voltage, and can be expressed by the following equations:

[0096]

[0097] In the formula, i s is the resultant vector of the stator current, and u s is the resultant vector of the voltage. From the above formula, in the d-q axis plane, the circle formed by the resultant vector of the stator current with the origin as the center is the current limit circle, and the ellipse formed by the resultant vector of the voltage with as the center is the voltage limit ellipse.

[0098] When the motor is below the base speed, in order to ensure the minimum copper loss and achieve the best efficiency, the maximum torque per ampere (MTPA) control strategy is adopted. According to the Lagrange extreme value theorem, the motor parameter formula is introduced to obtain the MTPA curve equation below the base speed:

[0099]

[0100] When the resultant vector i of the stator current s moves along the MTPA curve to the current limit circle, it cannot continue to increase the speed along the MTPA curve. At this time, the voltage closed-loop feedback control can be used to make i s operate under the current limit circle and the voltage limit ellipse at the same time to make the motor continue to increase the speed. If the characteristic current point is located inside the current limit circle, there is a deep flux-weakening region, and the resultant vector i of the stator current s can run along a new trajectory to make the motor continue to increase the speed.

[0101] If the motor has a deep flux-weakening region, the maximum torque can be obtained by limiting the amplitude along the maximum torque per voltage (MTPV) trajectory. According to the Lagrange extreme value theorem, the motor parameter formula is introduced to obtain the MTPV curve equation:

[0102]

[0103] However, the MTPV curve equation is relatively complex when controlling the current trajectory, and the MTPV curve can be simplified.

[0104] II. Properties of the MTPV Curve and Design of the Fitting Line

[0105] Define the convex machine ratio Characteristic current The MTPV curve equation can be changed to:

[0106]

[0107] Further let It can be obtained that:

[0108]

[0109] where m and n are fixed values composed of motor parameters. Analyzing the above formula, the hyperbolic property of the MTPV curve can be obtained. In Figure 3 , the center point of the hyperbola is (m, 0), and the slopes of the two asymptotes are It can be obtained that an asymptote L of the MTPV curve passing through the second quadrant is:

[0110]

[0111] The asymptote intersects the current limit circle at a point Q in the second quadrant. The abscissa i of point Q dQ is:

[0112]

[0113] The ordinate i of point Q qQ is:

[0114]

[0115] The intersection point of the MTPV curve and the current limit circle is defined as P. Then the abscissa i of point P dP is:

[0116]

[0117] In the d-q axis voltage plane, if u d and u q are both saturated, then both current regulators fail, which is called total out-of-control. The slope of the line connecting the saturated out-of-control point in the second quadrant and the origin is -1. Converting to the current plane, the saturated out-of-control line is a straight line with a slope of -1 passing through the characteristic current point. The intersection point of this line and the current limit circle is defined as R. Then this line is defined as L RT . It turns at point P and has not reached the saturated out-of-control line. At this time, in the ideal state, the motor should continue to accelerate along the MTPV curve. Generally, to avoid complex calculations, the intersection point of the line perpendicular to the d-axis passing through the characteristic current point and the current limit circle is defined as N. L NT is the common operating trajectory of the stator current synthesis vector.

[0118] From the coordinates of point Q and the coordinates of the characteristic current point T a straight line L QT can be obtained. The slope k n of the straight line is:

[0119]

[0120] The abscissa of the characteristic current point can be obtained in the d-q axis current plane as:

[0121] idP <i dQ <T

[0122] According to the geometric properties of the hyperbola and the straight line, the coordinates of point Q are located above and to the right of point P. Since point P has not reached the saturation out-of-control line, point Q is still within the limit out-of-control range. Then it is easy to obtain:

[0123] k n <-1

[0124] It is easy to obtain that the straight line intersects the MTPV curve at the characteristic current point and a certain point in the second quadrant, and the straight line is close to the MTPV curve. The equation of the straight line can be substituted into the motor formula for pre-calculation to obtain a fixed slope value, avoiding the real-time calculation of the MTPV curve formula. L QT The equation of the straight line is:

[0125] i q =k n (i d -T)

[0126] L QT That is the straight line (LMTPV) fitted by the MTPV curve. However, the pre-calculation of this slope still has a certain complexity, which will be further simplified below.

[0127] The torque formula of the motor can be expressed as:

[0128] T e =1.5n p i q ((L d -L q )i d +ψ f )

[0129] On the d-q axis current plane, the property of the constant torque curve is an inverse proportional function. For the interior permanent magnet synchronous motor L d <L q , where L d -L q has a relatively small absolute value, and ψ f has a relatively large value. The magnitude of the torque is mainly affected by the parameter i q . When the motor is operating in the field-weakening region, the running trend of the stator current synthesis vector i s is that i q decreases, i d increases negatively, and the decrease of i q has a greater reduction effect on the magnitude of the torque T e , while the reverse increase of i d has a relatively small increase effect on the magnitude of the torque T e .

[0130] It is easy to infer that when weak magnetic field weakening is carried out along the current limit circle, after passing the straight line perpendicular to the characteristic current point, without losing control, the greater the negative increase of i d is, the more beneficial it is to output high torque. Define the straight line L MT passing through the characteristic current point with a slope of k l . If this straight line is taken between L QT and L NT , the closer it is to L QT , the greater the maximum output torque that can be achieved on this trajectory, and the better the performance in the high-speed section. At this time, the range of k l is:

[0131] k l < k n < -1

[0132] Since the slope k l is negative, the smaller k l is taken, until -∞, the straight line L MT is closer to the L NT selected by the traditional method. The larger k l is taken, the closer it is to -1, and the closer it is to the saturation out-of-control line L RT .

[0133] III. Fixed-value gain compensation of the new weak magnetic system

[0134] In the traditional negative q-axis current increment compensation method, amplitude limiting is carried out at i dT . The Δi d generated by the negative increase trend of i d under weak magnetic control is compensated by the gain k m to generate Δi q , so that i q is reduced to continue to increase the speed. k m is:

[0135]

[0136] k m is essentially the curve slope of the MTPV curve with i q as the variable. The purpose of this method is to use the change law of MTPV as the compensation amount gain to dynamically reduce i q to compensate for the amplitude-limited i d , and obtain a current trajectory consistent with the voltage ellipse contraction trend.

[0137] From the above analysis, the property that MTPV is a hyperbola can be obtained, and it can be seen that it approaches the asymptote slope at the negative infinity of the d-axis, and approaches the slope of the straight line perpendicular to the d-axis at . Take the absolute value k' m of k m, then it can be obtained during the field-weakening control process As the speed increases, i q decreases, the gain becomes larger and larger, and i q is prone to fluctuations in the deep field-weakening region, which is extremely likely to cause the current loop to saturate and get out of control.

[0138] Actually, the control range of deep field-weakening is always limited within the voltage limit ellipse by the voltage closed-loop feedback. Figure 4 In it, a new compensation gain k n is defined. Similarly, take k n absolute value k' n . A smaller k' n can relieve the pressure of the current loop integration and reduce the possibility of the current loop getting out of control. However, too small a k' n will cause the speed increase range to be limited. It can be set that Taking k' n as an example, at this time, limited by the voltage closed-loop feedback compensation amount, Δi is at most I d , then Δi max is at most I q / ρ. If the Q point defined above is taken as the turning point, the Q point is geometrically close to the intersection point P of the MTPV curve and the current limit circle. At the turning point, i max After the reduction of Δi qQ , it can approach or cross the d-axis, and the speed increase range is sufficient. Limited by the performance of the motor body, a motor with an infinite speed increase system cannot actually increase the speed infinitely in practice. If the demand for the speed increase amount is not large, a smaller fixed value k' q can be selected to further reduce the possibility of the current loop getting out of control, that is, reduce the difficulty of PI tuning of the current loop and the voltage closed-loop feedback. n

[0139] IV. Control Scheme of the New Field-weakening System

[0140] Based on the traditional voltage closed-loop feedback negative q-axis current increment compensation method, an advanced angle field-weakening method is adopted. The control framework is basically the same, which simplifies the calculation burden and speeds up the response speed. Taking the MTPV fitting line L QT described above as an example, select i dQ as the d-axis current limit point, the compensation gain is k n , a new MAX comparison module is added. After compensation, corresponds to a point id* on the MTPV fitting line L QT . Limited by the comparison module, the current trajectory will be limited to the MTPV fitting line and its right side. In Figure 5 , that is, it changes from point G to point F, so that the limit operation trajectory of the stator current synthesis vector i s runs along the current limit circle to point Q and then switches to run along the line LQT It runs, and the running process is always restricted by voltage closed-loop feedback and current limit circle. The new field-weakening limit trajectory will change from A→B→C→D→V→Q of the traditional method to A→B→C→D→E→F. The new method has a larger speed regulation range of stator current, a larger maximum output torque at the same voltage in the deep field-weakening region, a stronger load-carrying capacity in the high-speed section, and uses LMTPV to replace the traditional MTPV curve, simplifying the calculation amount.

[0141] In matlab / simulink, the motor is started with a ramp speed, and the final speed is 7500 r / min. The rated speed of the motor is 3600 r / min, and the torque is set to 20 N·m. Below the rated speed, the motor is controlled in the MTPA mode, and then controlled by voltage closed-loop feedback. After the d-axis current reaches the switching point of the deep field-weakening region, the stator current synthesis vector i s runs along the MTPV fitting line, and the overall field-weakening trajectory switches smoothly, and there is no current saturation out-of-control situation during the whole process. The simulation experiment shows that the algorithm proposed by the present invention not only greatly simplifies the parameter calculation process during the deep field-weakening process, but also expands the deep field-weakening speed regulation range, effectively avoiding motor out-of-control and making the motor more reliable and stable during operation.

[0142] The present invention has good portability in the deep field-weakening region. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art can still modify the technical solutions recorded in the foregoing embodiments, or perform equivalent substitution on some of the technical features. Any modification, equivalent substitution, improvement, etc. made within the spirit and principle of the present invention shall be included within the protection scope of the present invention.

Claims

1. A fitting MTPV deep magnetic weakening control method based on lead angle combined with negative q-axis current compensation, the method comprising the following steps: S1: The problem of establishing weak magnetic field control of permanent magnet synchronous motor based on vector control is analyzed. S2: Design a new stator current synthetic vector control trajectory scheme by demonstrating the properties of the maximum torque-to-voltage ratio curve. S3: Design a new compensation gain scheme based on the demonstrated properties of the maximum torque-to-voltage ratio curve. S4: Design a new magnetic weakening system and a new stator current synthetic vector control trajectory operation scheme for the motor in the deep magnetic weakening domain.

2. The control method according to claim 1, characterized in that: In step S1, the mathematical model of the permanent magnet synchronous motor under magnetic field weakening control is: In the formula, u d 、u q is the d and q axis stator voltage, i d 、i q is the d and q axis stator current, L d , L q is the stator inductance of d and q axes, ω e is the rotor electrical angular velocity, R s is the stator winding resistance, ψ d , q is the stator flux of d and q axes, where the flux equation components are: In the formula, ψ f is the permanent magnet flux, and the electromagnetic torque equation is: T e =1.5n p (ψ d I q -ψ q I d ) Where, T e is the electromagnetic torque, n p is the number of pole pairs of the motor. Since the stator resistance of the permanent magnet motor is small, when the motor is running in the weak magnetic field area, the voltage drop caused by the resistance part can be ignored due to the high motor speed. And when the motor is in steady state, the differential term within the unit cycle of the current loop can be regarded as zero, so the steady-state voltage equation of the IPMSM can be expressed as: When the IPMSM is running in self-control mode, its operating state will be limited by the motor's own characteristics and the inverter output capacity. The limitation includes current and voltage, where the maximum current limit is set to I max , the voltage amplitude limit is set to U max These two limiting parameters are related to the motor parameters and the DC bus voltage and can be expressed by the following equations: In the formula, i s is the stator current synthesis vector, u s is the voltage synthesis vector. From the above formula, on the dq axis plane, the circle surrounded by the stator current synthesis vector with the origin as the center is the current limit circle, and the circle surrounded by the voltage synthesis vector is The ellipse with as the center is the voltage limit ellipse. When the motor is below the base speed, in order to ensure minimum copper loss and achieve optimal efficiency, the maximum torque per ampere (MTPA) control strategy is adopted. According to Lagrange's extreme value theorem, the motor parameter formula is introduced to obtain the MTPA curve equation below the base speed: In the stator current synthesis vector i s When the current limit circle is reached along the MTPA curve, the speed cannot be increased along the MTPA curve. At this time, the voltage closed-loop feedback control can be used to make i s At the same time, the current limit circle and voltage limit ellipse run, so that the motor continues to increase speed. If the characteristic current point If it is within the current limiting circle, there is a deep magnetic weakening domain, and the stator current synthesis vector i s The motor can continue to increase its speed by running along the new trajectory. If the motor has a deep magnetic weakening domain, the maximum torque can be obtained by limiting along the maximum torque per voltage (MTPV) trajectory. According to Lagrange's extreme value theorem, the motor parameter formula is introduced to obtain the MTPV curve equation: However, the MTPV curve equation is more complicated when controlling the current trajectory, and the MTPV curve can be simplified.

3. The control method according to claim 1, characterized in that: In step S2, by analyzing the properties of the maximum torque-to-voltage ratio curve, a new stator current synthetic vector control trajectory scheme is designed: Defining the convex machine ratio Characteristic current The MTPV curve equation can be changed to: Further order We can get: Where m and n are constants composed of motor parameters, then analyzing the above formula, we can get the hyperbolic properties of the MTPV curve. In the dq axis coordinate system, the center point of the hyperbola is (m, 0), and the slopes of the two asymptotes are The asymptote L of the MTPV curve passing through the second quadrant is: The asymptote intersects the current limit circle at a point in the second quadrant, which is defined as Q. The horizontal coordinate of point Q is i dQ for: The vertical coordinate of point Q is i qQ for: The intersection of the MTPV curve and the current limit circle is defined as P, then the horizontal coordinate of point P is i dP for: On the dq axis voltage plane, if u d and u q If both are saturated, both current regulators fail, which is called total loss of control. In the second quadrant, the slope of the line connecting the saturation loss of control point and the origin is -1. Converted to the current plane, the saturation loss of control line is a straight line with a slope of -1 passing through the characteristic current point. The intersection of this straight line and the current limit circle is defined as R, and the straight line is defined as L. RT The motor turns at point P and does not reach the saturation out-of-control line. Ideally, the motor should continue to increase speed along the MTPV curve. In order to avoid complex calculations, the intersection of the line passing through the characteristic current point and perpendicular to the d axis and the current limit circle is defined as N, L NT It is the common stator current synthetic vector running trajectory. From the coordinates of point Q and the coordinates of characteristic current point T We can get a straight line L QT , the slope of the straight line k n for: Characteristic current point horizontal coordinate On the dq axis current plane, we can get: i dP <i dQ <T According to the geometric properties of hyperbolas and straight lines, the coordinates of point Q are located to the upper right of point P. If point P does not reach the saturation out-of-control line, then point Q is still within the limit out-of-control range. It is easy to obtain: k n <-1 It is easy to find that the straight line intersects the MTPV curve at the characteristic current point and a point in the second quadrant, and the straight line is close to the MTPV curve. The straight line equation can be substituted into the motor formula for advance calculation to obtain the slope constant, thus avoiding real-time calculation of the MTPV curve formula. QT The equation of the line is: i q =k n (i d -T) L QT This is the straight line fitted by the MTPV curve (LMTPV), but the advance calculation of the slope is still somewhat complicated, which will be further simplified below. The torque formula of the motor can be expressed as: T e =1.5n p I q ((L d -L q )i d +ψ f ) On the dq axis current plane, the equal torque curve is an inverse proportional function. The built-in permanent magnet synchronous motor L d <L q , where L d -L q The absolute value is small, ψ f The value is large, and the torque is mainly affected by parameter i q The influence of the stator current synthesis vector i s The running trend is i q Reduce, d Negative increase, i q The reduction of torque T e The size of the d The reverse increase of torque T e The size of the gain is smaller. It is easy to infer that, when the magnetic field is weakened along the current limiting circle and the straight line perpendicular to the characteristic current point is passed, under the condition of not losing control, i d The greater the negative increase, the more conducive it is to output high torque, and the straight line L passing through the characteristic current point is defined. MT The slope is k l , if the straight line is between L QT With L NT The closer to L QT The greater the maximum output torque that can be achieved on this trajectory, the better the performance in the high-speed section. l The range is: k l <k n <-1 Since the slope k l is a negative value, take k l The smaller it is, until -∞, the straight line L MT The closer to the L selected by the traditional method NT , take k l The larger it is, the closer it is to -1, and the closer it is to the saturation out-of-control line L RT .

4. The control method according to claim 1, characterized in that: In step S3, a new compensation gain scheme is designed by analyzing the properties of the maximum torque-to-voltage ratio curve demonstrated: In the traditional method of negative q-axis current increment compensation, in i dT The limit is performed at i d Δi generated by the negative increasing trend under weak magnetic control d The gain k m Compensation produces Δi q , making i q Lower to continue increasing speed. k m for: k m The essence is that the MTPV curve is i q is the slope of the variable curve. The purpose of this method is to use the change law of MTPV as the compensation gain to dynamically reduce i q To compensate for the limited i d , and a current trajectory is obtained that is consistent with the contraction trend of the voltage ellipse. From the above analysis, we can see that MTPV is a hyperbola, so At negative infinity on the d axis, the slope approaches the asymptote. The slope of the straight line at k approaches the slope of the straight line perpendicular to the d-axis. m Absolute value k' m , then we can get in the process of weak magnetic control As the speed increases, i q The gain is getting bigger and bigger, and i q In the deep weak magnetic field area, the current is prone to fluctuate, which can easily cause the current loop to saturate and lose control. In fact, the control range of deep magnetic weakening is always limited by the voltage closed-loop feedback within the voltage limit ellipse. The new compensation gain k is defined as n , similarly, take k n Absolute value k' n . Smaller k' n It can relieve the pressure of current loop integration and reduce the possibility of current loop out of control, but too small k' n This will limit the speed increase range. With k' n Pick For example, at this time, the voltage closed-loop feedback compensation amount is limited, Δi d Maximum is I max , then Δi q Maximum is I max / ρ, if the Q point defined above is taken as the turning point, the Q point is geometrically close to the intersection point P of the MTPV curve and the current limit circle, and the turning point i qQ Δi q The gain can approach or cross the d axis, and the speed increase range is sufficient. Due to the performance limitations of the motor itself, the motor with an infinite speed increase system cannot actually increase speed infinitely. If the demand for speed increase is not large, a smaller constant k' can be selected. n , further reducing the possibility of current loop out of control, that is, reducing the difficulty of PI adjustment of current loop and voltage closed-loop feedback.

5. The control method according to claim 1, characterized in that: In step S4, a new weak magnetic field system is designed, and a new stator current synthetic vector control trajectory operation scheme of the motor in the deep weak magnetic field is designed: Based on the traditional voltage closed-loop feedback negative q-axis current incremental compensation method, the leading angle weakening method is adopted. The control framework is basically the same, which simplifies the calculation burden and speeds up the response speed. QT For example, select i dQ is the d-axis current limit point, and the compensation gain is k n , add MAX comparison module, after compensation Fitting straight line L to MTPV QT The corresponding point id* will be limited by the comparison module, and the current trajectory will be limited to the MTPV fitting straight line and its right side, so that the stator current synthesis vector i s The limit running trajectory runs along the current limit circle to point Q, and then switches to the straight line L QT The operation process is always limited by the voltage closed-loop feedback and the current limit circle. The new method has a larger stator current speed regulation range, a larger maximum output torque at the same voltage in the deep weak magnetic field, a stronger load capacity in the high-speed section, and uses LMTPV to replace the traditional MTPV curve, which simplifies the calculation. Then a fitting MTPV deep magnetic weakening control scheme based on the lead angle combined with negative q-axis current compensation can be constructed.

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