Motor control device and electrically driven vehicle

Abstract

A motor control device that outputs a voltage applied to an AC motor on the basis of an output voltage command value includes: a control period determination unit that determines a control period on the basis of a fundamental frequency of the AC motor and a cutoff frequency of current control; a control interruption unit that determines a control interruption on the basis of the control cycle determined by the control cycle determination unit; a current control unit that controls the output voltage command value on the basis of the control interruption determined by the control interruption unit; and a PWM control unit that controls the output of the voltage by PWM corresponding to the output voltage command value controlled by the current control unit. Consequently, when control is performed in which the control period and the inverter frequency are not synchronized, vibration of the current caused by the 6n-order harmonic current component can be more easily suppressed.

Description

Technical Field

[0001] The present invention relates to a motor control device and an electric drive vehicle. Background Art

[0002] In a control system of an AC motor mounted on an electric drive vehicle or the like, in order to improve the voltage utilization rate of an inverter, there are known techniques for operating in an overmodulation mode (so-called overmodulation region) where the output voltage command of the inverter exceeds the maximum output level that can be output as a sine wave, and synchronous PWM that outputs PWM pulses in synchronization with the inverter output frequency.

[0003] On the other hand, in the overmodulation region or synchronous PWM, due to the switching of the inverter, the current in the dq-axis coordinate system used in vector control includes high-order harmonic currents of 6n times (where n is a natural number), and there is a problem that current vibration is generated due to the high-order harmonic currents.

[0004] As a technique for suppressing such high-order harmonic currents, for example, the techniques described in Patent Documents 1 and 2 are known.

[0005] Patent Document 1 discloses a current detection device for a synchronous PWM power converter, which includes: a power conversion main circuit that applies a voltage for driving an AC power load; a voltage command value generation unit that outputs command values of a modulation rate and a phase angle of a voltage respectively; a PWM control signal generation unit that generates a PWM control signal based on the command values of the modulation rate and the phase angle and outputs it to the power conversion main circuit; a current detection unit that detects the current supplied to the AC power load and outputs a current signal corresponding to the current; and a sampling unit that samples and holds the current detection value based on the current signal and outputs a first sampled and held value, wherein the sampling unit includes a voltage phase angle setting unit that sets a prescribed phase angle as the sampling time, and samples and holds the current detection value at the moment when the command value of the phase angle and the set value of the prescribed phase angle become the same value.

[0006] In addition, in Non-Patent Document 1, a filter design method based on the method of using a filter is disclosed for the purpose of preventing deterioration of control performance by reducing the sensitivity of a current control system to interference components overlapping in the overmodulation region. Prior Art Documents Patent Documents

[0007] Patent Document 1: Japanese Patent Laid-Open No. 2004-15949 Non-Patent Documents

[0008] Non-Patent Document 1: Yosuke Nakayama, Shinji Michiki, "Design of Band-Rejection Filter for PMSM Vector Control in Inverter Overmodulation Region", Transactions of the Institute of Electrical Engineers of Japan, D, Vol. 138, No. 11, pp. 884-893 (2018) Summary of the Invention Problems to be Solved by the Invention

[0009] In Patent Document 1, the carrier frequency is synchronized with the inverter output frequency, and the current is detected at the moment when the harmonic current is zero, thereby achieving suppression of the harmonic current. However, for example, when applied to an application such as an automobile that requires rapid acceleration and deceleration, since the control period changes frequently according to the frequency, it is difficult to synchronize the control period with the inverter frequency.

[0010] In addition, in Non-Patent Document 1, the sixth component of the current is removed in the overmodulation region, and the control period can be made fixed. However, the band-rejection filter used to remove the harmonic needs to be changed according to the inverter output frequency, and in order to remove the target frequency, a high-speed operation period needs to be supported.

[0011] The present invention has been completed in view of the above problems, and an object thereof is to provide a motor control device and an electric drive vehicle that can more easily suppress current vibration caused by 6n-th harmonic current components when performing control in which the control period is not synchronized with the inverter frequency. Technical Means for Solving the Problem

[0012] The present invention includes a plurality of means for solving the above problems. If one example is cited, it is a motor control device that outputs a voltage applied to an AC motor according to an output voltage command value. The motor control device includes: a control period determination unit that determines a control period according to the fundamental frequency of the AC motor and the cut-off frequency of current control; a control interruption unit that determines a control interruption based on the control period determined by the control period determination unit; a current control unit that controls the output voltage command value based on the control interruption determined by the control interruption unit; and a PWM control unit that controls the output of the voltage by PWM, and the PWM corresponds to the output voltage command value controlled by the current control unit. Effects of the Invention

[0013] According to the present invention, when performing control in which the control period is not synchronized with the inverter frequency, it is possible to more easily suppress current vibration caused by 6n-th harmonic current components. Description of the Drawings

[0014] Figure 1 It is a functional block diagram showing the motor control device of the first embodiment together with related components. Figure 2 It is a diagram showing the relationship among the three-phase fundamental current, the dq-axis 6th harmonic current, and the reading time. Figure 3 It is a diagram showing an example of the look-up table of the first embodiment. Figure 4 It is a diagram showing the first modified example of the look-up table. Figure 5 It is a diagram showing the second modified example of the look-up table. Figure 6 It is a diagram showing the third modified example of the look-up table. Figure 7 It is a functional block diagram showing the motor control device of the second embodiment together with the associated components. Figure 8 It is a diagram showing the relationship among the three-phase fundamental current, the dq-axis 6th harmonic current, and the reading time. Figure 9 It is a diagram showing an example of the look-up table of the second embodiment. Figure 10 It is a diagram showing the relationship among the three-phase fundamental current, the dq-axis 6th harmonic current, and the reading time. Figure 11 It is a functional block diagram showing the motor control device of the third embodiment together with the associated components. Figure 12 It is a functional block diagram showing the processing content of the control cycle calculation unit of the third embodiment. Figure 13 It is a diagram showing an example of the look-up table of the third embodiment. Figure 14 It is a functional block diagram showing the motor control device of the fourth embodiment together with the associated components. Figure 15 It is a functional block diagram showing the processing content of the control cycle calculation unit of the fourth embodiment. Figure 16 It is a functional block diagram showing the motor control device of the fifth embodiment together with the associated components. Figure 17 It is a functional block diagram showing the processing content of the control cycle calculation unit of the fifth embodiment. Figure 18 It is a functional block diagram showing the motor control device of the sixth embodiment together with the associated components. Figure 19 It is a diagram showing the relationship among the three-phase fundamental current, the dq-axis 6th harmonic current, and the reading time. Figure 20 It is a diagram schematically showing the extraction of the drive system and the control system of an electric drive vehicle equipped with an AC motor controlled by a motor control device. Detailed implementation mode

[0015] Hereinafter, the implementation mode of the present invention will be described with reference to the drawings. In the following description, a permanent magnet synchronous motor (PMSM: Permanent Magnet Synchronous Motor) of an AC motor is exemplified as the control object of the motor control device, but it is not limited thereto. For example, as long as it is an AC machine such as a synchronous reluctance motor, a permanent magnet synchronous generator, a wound synchronous machine, an induction motor, or an induction generator, the present invention can be applied to obtain the same effect. In addition, IGBT is exemplified as the semiconductor switching element of the inverter device, but it is not limited thereto. For example, even when using MOSFET or other power semiconductor elements, the present invention can be applied to obtain the same effect.

[0016] <First Embodiment> Refer to Figures 1 - 6 and Figure 20 describe the first embodiment of the present invention.

[0017] Figure 1 is a functional block diagram showing the motor control device of the present invention together with related components. In addition, Figure 20 is a diagram schematically showing the drive system and the control system of an electric drive vehicle equipped with an AC motor controlled by the motor control device taken out together.

[0018] As Figure 20 shown, the motor control device 100 of the present embodiment controls the power supplied from the DC voltage source 9 (such as a battery) to the PMSM 1 via the power converter 2 (inverter).

[0019] The PMSM 1 is connected to the drive shaft 105 via the transmission 101 and the differential gear 103, and supplies power to the wheels 107.

[0020] In addition, in the present embodiment, the case where the power from the PMSM 1 is transmitted via the transmission 101 is taken as an example for description, but it is not limited thereto. For example, in a configuration where the PMSM is directly connected to the differential gear, or in a configuration where the front wheels and the rear wheels respectively have a PMSM and an inverter, the present invention can also be applied.

[0021] As Figure 1As shown, the motor control device 100 controls the power supplied from the DC voltage source 9 to the PMSM 1 via the power converter 2, and generally includes: a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, coordinate transformation units 7, 11, a current reading unit 8, a PWM controller 12, a control cycle calculation unit 15, a control interrupt generation unit 17, and a current control unit 19. In addition, the coordinate transformation units 7, 11 and the current control unit 19 constitute the control operation unit 13.

[0022] The power converter 2 converts the DC power from the DC voltage source 9 (such as a battery) into AC power according to the gate signal from the PWM controller 12 and outputs it to drive the PMSM 1.

[0023] The phase current detector 3 is composed of, for example, a Hall CT (Current Transformer), etc., and detects the current waveforms Iud, Ivd, Iwd of the U-phase, V-phase, and W-phase of the three-phase current flowing from the power converter 2 to the PMSM 1.

[0024] The magnetic pole position detector 4 is composed of, for example, a resolver, etc., and detects the magnetic pole position of the PMSM 1 and outputs it as magnetic pole position information θ * Output.

[0025] The frequency calculation unit 5 calculates the angular velocity, for example, by differential operation, as the speed information ω1 according to the magnetic pole position information θ detected by the magnetic pole position detector 4 * Output. * Output.

[0026] The current reading unit 8 reads the current waveforms Iud, Ivd, Iwd detected by the phase current detector 3 at the moment when the current reading signal cl from the control interrupt generation unit 17 becomes ON, and outputs them as the three-phase current Iuc, Ivc, Iwc to the coordinate transformation unit 7 of the control operation unit 13.

[0027] The coordinate transformation unit 7 performs coordinate transformation on the three-phase current Iuc, Ivc, Iwc read by the current reading unit 8 according to the magnetic pole position information θ detected by the magnetic pole position detector 4 * Thereby calculating and outputting the dq-axis current detection values Idc, Iqc.

[0028] The current control unit 19 calculates and outputs the dq-axis voltage command values Vd * , Iq * In such a way that they are consistent with the dq-axis current detection values Idc, Iqc from the coordinate transformation unit 7 * , Vq * .

[0029] The coordinate transformation unit 11 performs coordinate transformation on the dq-axis voltage command values Vd * and Vq * from the current control unit 19 according to the pole position information θ detected by the pole position detector 4 * , and thereby calculates and outputs the three-phase voltage command values Vu * , Vv * , and Vw * .

[0030] The control cycle calculation unit 15 calculates and outputs the control cycle tc according to the speed information ω1 calculated by the frequency calculation unit 5 * .

[0031] The control interrupt generation unit 17 generates a control interrupt signal int and a current read signal cl according to the control cycle f from the control cycle calculation unit 15 c . Additionally, in the present embodiment, the control interrupt signal int and the current read signal cl are synchronized and are exactly the same signal.

[0032] The coordinate transformation unit 7, the coordinate transformation unit 11, and the current control unit 19 that constitute the control operation unit 13 operate according to the call of the control interrupt signal int from the control interrupt generation unit 17

[0033] The DC voltage detection device 6 detects the voltage of the DC voltage source 9 and outputs the detection result as DC voltage information Vdc

[0034] The PWM controller 12 calculates a duty ratio signal according to the three-phase voltage command values Vu * , Vv * , and Vw * from the coordinate transformation unit 11 and the DC voltage information Vdc from the DC voltage detection device 6, and generates and outputs a gate signal by comparing the calculated duty ratio signal and a carrier wave

[0035] Here, the basic principle of the present invention will be described

[0036] First, the generation principle of current vibration in the overmodulation region of the AC motor and in the control of synchronous PWM will be described

[0037] Figure 2 is a diagram showing the relationship between the three-phase fundamental current, the dq-axis 6th-order harmonic current, and the read time

[0038] In AC motor control, it is known that in the overmodulation region, a 6nth-order component is generated in the dq-axis current (for example, refer to Non-Patent Document 1), and when the control cycle f c is near 6n times the inverter output frequency (n is a natural number), current vibration occurs

[0039] Figure 2 It represents the influence of the three-phase fundamental current 61, the dq-axis 6th harmonic current 63, and the read-in time, and the 6th component of the 6nth component where n = 1 on the control. In this embodiment, the current read-in unit 8 reads in the current synchronously with the control operation unit 13 at the control period f c reads in the current. In Figure 2 it represents the case where the control period f c deviates from 6 times the inverter output frequency f0 by Δf (i.e., the case where f c = 6f0 + Δf). At this time, the 6th harmonic component i ’ h6 detected at each point is obtained by the following (Equation 1) to (Equation 4).

[0040] [Formula 1] i′ h6 (n)=I h cos θ…(Equation 1)

[0041] [Formula 2]

[0042] [Formula 3]

[0043] [Formula 4]

[0044] According to the above (Equation 1) to (Equation 4), the current is detected at the frequency (6f0 + Δf). Therefore, if observed in the dq coordinate system, it seems that the current vibrates at the frequency (-Δf). The amplitude at this time is the amplitude I h of the harmonic current.

[0045] Here, if the frequency Δf is within the response band of the current control, the current that should actually vibrate at the 6th harmonic is regarded as a current vibrating at the frequency Δf, and the current control acts to cancel the vibration at the frequency Δf. Since the vibration at this frequency Δf is not originally generated, when the current control acts, a vibration at Δf is generated due to the current control. This is the mechanism for generating the current vibration.

[0046] Therefore, in this embodiment, the generation of current vibration is suppressed by the control of the control period operation unit 15. Hereinafter, the basic principle and details will be described.

[0047] The control period operation unit 15 is composed of a look-up table that takes the speed information ω1 * as an input and outputs the control period f c as an output.

[0048] Figure 3This is a diagram showing an example of a lookup table used in the control cycle operation unit. The horizontal axis represents the speed information ω1 * , and the vertical axis represents the control cycle f c .

[0049] As Figure 3 shown, the relationship between the speed information ω1 * and the control cycle f c (line 51) is selected in such a way as to avoid the selection prohibited frequency band defined by the upper limit 53 and the lower limit 55. Here, an example is illustrated where the control cycle f * <angular velocity a, or angular velocity b ≤ speed information ω1 * is set to f1 in the range, and the control cycle f c = f2 in the range where angular velocity a ≤ speed information ω1 * <angular velocity b. Among them, angular velocity a is the angular velocity when f c = f1 intersects with the upper limit 53 of the selection prohibited frequency band, and angular velocity b is the angular velocity when f c = f1 intersects with the lower limit 55 of the selection prohibited area. c

[0050] The upper limit 53 and the lower limit 55 of the selection prohibited frequency band are determined by the following (Equation 5).

[0051] [Equation 5] 6f0 - m×f acr <f c <6f0 + m×f acr …(Equation 5)

[0052] Here, in the above (Equation 5), f0 represents the inverter output frequency, and f acr represents the cut-off frequency of the current control unit 19. In addition, the variable m is a positive value, and its setting method will be described below.

[0053] Since the current apparently generated in the dq-axis current being within the response frequency band of the current control is the cause of current vibration, in this embodiment, control is performed in such a way that the frequency Δf of the apparent dq-axis current high-order harmonics is outside the response frequency band of the current control. In addition, generally, with the cut-off frequency f of the current control acrA frequency more than 10 times that becomes out of the response band controlled by current. Therefore, for example, when the number of current detections is 1 with respect to the control period, if the variable m is set to 10, the frequency Δf can be set outside the response band. Additionally, if a slight current vibration is tolerable, it can be said that the variable m is set to around 5. In this way, by setting the control period in a manner that avoids the selection prohibition band shown in the above (Equation 5), even if a 6th harmonic component is generated in the dq-axis current in the overmodulation region, low-frequency vibration of the current can be avoided.

[0054] In addition, in this embodiment, the case of the 6th harmonic component (i.e., when n = 1) is taken as an example, but in the 6nth harmonic components such as the 12th, 18th, and 24th (i.e., when n > 1), the above principle is fully applicable.

[0055] In this way, when detecting the current used in current control at a moment consistent with the control interruption int, by avoiding integer multiples of 6 as the prohibition band for not generating vibration due to the 6th harmonic current, vibration can be avoided.

[0056] In addition, in this embodiment, the case where the control interruption int and the current read-in signal cl are exactly the same is taken as an example, but if the periods of these signals are the same, a slight deviation in the allowable moment is tolerated, and the same effect can be obtained.

[0057] In addition, in this embodiment, the case of changing the control period according to the angular velocity (speed information ω1 * ) is exemplified, but when changing the frequency according to the inverter output frequency (which is inconsistent with the speed in the case of an induction machine) instead of the angular velocity, the same effect can be obtained by applying the present invention. Additionally, even when changing the frequency according to information other than the angular velocity (such as torque), as long as the control period is set in a manner that avoids the prohibition band determined by the angular velocity, the same effect can be obtained.

[0058] In addition, in an automobile, since it is an application with strict requirements for rapid acceleration and deceleration performance and vibration, it can be said that the effect of the present invention is more significantly manifested than in other applications. Similarly, in a railway, like in an automobile, it is an application with strict requirements for rapid acceleration and deceleration performance and vibration, and it is easy to exhibit the effect of the present invention. That is, in an automobile and a railway, by applying the present invention in the "control of making the control period out of sync with the inverter frequency" which is suitable for improving rapid acceleration and deceleration performance, vibration caused by 6f can be suppressed, and the riding comfort of the driver or passengers can be improved.

[0059] In addition, in this embodiment, as Figure 3As shown, the example illustrates the case of using a lookup table to determine the control period in a rectangular manner and avoiding the selection prohibited frequency band to the minimum, but is not limited to this. For example, as shown in the following 1st to 3rd modified examples, a lookup table that avoids the selection prohibited frequency band for the control period can be used to obtain the same effect as the first embodiment.

[0060] Figure 4 FIG. 4 is a diagram showing a lookup table according to a first variant of the present embodiment. Figure 4 In the equation, speed information ω1 * and control period f c The relationship (line 51A) is set to a rectangular shape with a margin to avoid the selection prohibition band to determine the control cycle. Here, the speed information ω1 is exemplified. * <angular velocity a, or angular velocity b≤ velocity information ω1 * The control period is set to f in the range of c =f1, when angular velocity a≤velocity information ω1 * <The control period f is set to be within the range of angular velocity b c =f2. However, the angular velocities a, b and the control periods f1, f2 are set so that the control periods f1, f2 do not take values above the upper limit 53 and the lower limit 55 of the selection prohibited band, that is, they are set to values having a margin with the selection prohibited region.

[0061] Figure 5 FIG. 4 is a diagram showing a lookup table of a second variant of the present embodiment. Figure 5 In the equation, speed information ω1 * and control period f c The relationship (line 51B) is set in a step-like manner and the control cycle is determined by avoiding the selection prohibition band to the minimum extent. * <The control period f is set within the range of angular velocity a c =f1, when angular velocity a≤velocity information ω1 * The control period is set to f within the range c =f2. Here, the angular velocity a is f c = the angular velocity when f1 intersects the upper limit 53 of the selection prohibited frequency band.

[0062] Figure 6 FIG. 4 is a diagram showing a lookup table according to a third variant of the present embodiment. Figure 6 In the equation, speed information ω1 * and control period f c The relationship (line 51C) is selected so as to avoid the selection prohibition band defined by the upper limit 53 and the lower limit 55. Here, the speed information ω1 is exemplified. * <angular velocity a, or angular velocity b≤ velocity information ω1 *is set to the control period f within the range of c = f1, and when the angular velocity a ≤ the velocity information ω1 * within the range of <angular velocity b, the control period f c is set to the case along the value of the lower limit 55. Among them, the angular velocity a is when f c = f1 intersects with the upper limit 53 of the selection prohibited band, and the angular velocity b is when f c = f1 intersects with the lower limit 55 of the selection prohibited region.

[0063] <Second Embodiment> Refer to Figures 7 - 10 to describe the second embodiment of the present invention.

[0064] This embodiment shows a control method in which the current read-in signal becomes ON at a period twice that of the interrupt signal, and control is performed based on the deviation of the secondary three-phase current. The secondary three-phase current is read in according to the current read-in signal.

[0065] In this embodiment, only the differences from the first embodiment will be described. In the drawings used in this embodiment, the same components as those in the first embodiment are labeled with the same reference numerals, and the description will be appropriately omitted.

[0066] Figure 7 is a functional block diagram showing the motor control device of this embodiment together with the associated configuration.

[0067] As Figure 7 shown, the motor control device 100A controls the power supplied from the DC voltage source 9 to the PMSM1 via the power converter 2, and is generally composed of the following: a phase current detector 3, a magnetic pole position detector 4, a frequency operation unit 5, a DC voltage detection device 6, coordinate transformation units 7A, 11, a current read-in unit 8A, a PWM controller 12, a control period operation unit 15A, a control interrupt generation unit 17A, and a current control unit 19. In addition, the coordinate transformation units 7A, 11 and the current control unit 19 constitute a control operation unit 13A.

[0068] The control interrupt generation unit 17A generates and outputs a current read-in signal cl in such a manner that the current read-in signal cl becomes ON at a period twice that of the interrupt signal int.

[0069] The current read-in unit 8A reads the current waveforms Iud, Ivd, Iwd of the three-phase current when the current read-in signal cl becomes ON. At this time, the current at the read-in moment is output as the three-phase current Iuc1, Ivc1, Iwc1, and the current read in last time is output as the three-phase current Iuc2, Ivc2, Iwc2 to the coordinate transformation unit 7A of the control operation unit 13A.

[0070] The coordinate transformation unit 7A corrects the deviation amount at the time of reading the three-phase currents Iuc1, Ivc1, Iwc1 and the three-phase currents Iuc2, Ivc2, Iwc2 into a rotation angle, and transforms its coordinates into dq coordinates. Then, the dq-axis currents after coordinate transformation are averaged by addition and output to the current control unit 19 as the dq-axis current detection values Idc and Iqc.

[0071] Here, the operation and the operation and effect of the present embodiment configured as described above will be described.

[0072] Figure 8 It is a diagram showing the relationship between the three-phase fundamental current, the dq-axis 6th harmonic current, and the reading time. Figure 8 It shows the three-phase fundamental current 61, the dq-axis 6th harmonic current 63, and the reading time.

[0073] In Figure 8 it shows that the current reading unit 8A reads the current at a double period fcl = 2f synchronously with the control arithmetic unit 13A c and the control period f c is deviated by Δf from 6 times the output frequency f0 of the inverter (that is, the case where fcl = 2f c = 12f0 + 2Δf). Thus, the 6th harmonic component i ’ h6 detected at each point is obtained by the following (Equation 6) to (Equation 9).

[0074] [Formula 6] i′ h6 (n) = I h cos θ…(Equation 6)

[0075] [Formula 7]

[0076] [Formula 8]

[0077] [Formula 9]

[0078] In addition, since the average value of the twice-detected current is used in the current control, the 6th harmonic component i ’ h6_c contained in the current used in the current control is obtained by the following (Equation 10) and (Equation 11).

[0079] [Formula 10]

[0080] [Formula 11]

[0081] According to the above (Equation 10) and (Equation 11), the current is detected at the frequency (12f0 + 2Δf). Therefore, when observed from the dq coordinates, the current appears to oscillate at the frequency (-2Δf). The amplitude at this time is the amplitude of the harmonic current (I h ×2π×0.5Δf÷(12f0 + 2Δf)). As the amplitude, it is smaller compared to the case where the current is detected once in one cycle of the control period (refer to the first embodiment), but if the frequency Δf becomes larger, the amplitude also increases, and thus vibration is generated in the same way (however, the frequency and amplitude of the vibration are different).

[0082] Therefore, in the present embodiment, by changing the control period f from the control period calculation unit 15A according to the speed ω1 * c ,the generation of current vibration is suppressed.

[0083] The control period calculation unit 15A is composed of a look-up table that takes the speed information ω1 * as an input and the control period f c as an output.

[0084] Figure 9 is a diagram showing an example of the look-up table used in the control period calculation unit. The horizontal axis represents the speed information ω1 * and the vertical axis represents the control period f c .

[0085] As Figure 9 shown, the relationship between the speed information ω1 * and the control period f c (line 51D) is selected in a manner that avoids the selection prohibited frequency band defined by the upper limit 53A and the lower limit 55A, as well as the selection prohibited region defined by the upper limit 53B and the lower limit 55B. Here, an example is illustrated where the control period f * <angular velocity a, angular velocity b ≤ speed information ω1 * <angular velocity c, or angular velocity d ≤ speed information ω1 * is set to f1 within the range, the control period f c = f2 within the range of angular velocity a ≤ speed information ω1 * <angular velocity b, and the control period f c = f3 within the range of angular velocity c ≤ speed information ω1 * <angular velocity d. Among them, the angular velocity a is the angular velocity at which f c = f1 intersects with the upper limit 53A of the selection prohibited frequency band, and the angular velocity b is the angular velocity at which f c = f1 intersects with the upper limit 53A of the selection prohibited frequency band, and the angular velocity b is the angular velocity at which f c ​= The angular velocity when f1 intersects with the lower limit 55A of the selection prohibited region. The angular velocity c is f c = The angular velocity when f1 intersects with the upper limit 53B of the selection prohibited region. The angular velocity d is f c = The angular velocity when f1 intersects with the lower limit 55B of the selection prohibited region.

[0086] The upper limit 53A and the lower limit 55A of the selection prohibited frequency band are determined by the following (Equation 12).

[0087] [Formula 12] 6f0 - m1×f acr <f c <6f0 + m1×f acr …(Equation 12)

[0088] Here, in the above (Equation 12), f0 represents the inverter output frequency, and f acr represents the cut-off frequency of the current control unit 19 (current control cut-off frequency). In addition, the variable m1 is a positive value, and its setting method will be described below.

[0089] When the number of current detections is 2 times relative to the control period, it becomes 2 times the case of the frequency being 1 time. Therefore, if the variable m1 = 5 is set, the frequency becomes outside the response frequency band of the current control at 10 times the current control cut-off frequency f acr In addition, considering that a slight current vibration can be tolerated, it can also be said that the variable m1 is set to around 2.5.

[0090] In addition, when the number of current detections is 2 times, since the same vibration also occurs near 3 times the inverter output frequency in the control period, a setting prohibited region defined by the upper limit 53B and the lower limit 55B determined by the following (Equation 13) is added.

[0091] [Formula 13] 3f0 - k×f acr <f c <3f0 + k×f acr …(Equation 13)

[0092] Here, in the above (Equation 13), k is a positive value, and basically k = m1 / 2 is set.

[0093] Next, the reason why the same vibration also occurs near 3 times the inverter output frequency in the control period will be described below.

[0094] Figure 10 is a diagram showing the relationship between the three-phase fundamental current, the dq-axis 6th harmonic current, and the reading time. Figure 10 Represents the three-phase fundamental current 61, the dq-axis 6th harmonic current 63, and the reading time.

[0095] In Figure 10 it, the current reading unit 8A reads the current synchronously with the control arithmetic unit 13A at a double cycle fcl = 2f c and the control cycle f c is deviated by Δf from three times the inverter output frequency f0 (i.e., the case where fcl = 2f c = 6f0 + 2Δf). At this time, the sixth harmonic component i″ h6 detected at each point is obtained by the following (Equation 14) to (Equation 17).

[0096] [Formula 14] i″ h6 (n) = I h cosθ …(Equation 14)

[0097] [Formula 15]

[0098] [Formula 16]

[0099] [Formula 17]

[0100] In addition, since the average value of the twice-detected current is used in the current control, the sixth harmonic component i″ h6_c included in the current used in the current control is obtained by the following (Equation 18) and (Equation 19).

[0101] [Formula 18]

[0102] [Formula 19]

[0103] According to the above (Equation 18) and (Equation 19), since the current is detected at the frequency (6f0 + 2Δf), when observed from the dq coordinate, the current seems to vibrate at the frequency 4Δf. The amplitude at this time is the amplitude I h of the harmonic current. As the amplitude, it is equal to the case where the current is detected once in one cycle of the control period (refer to the first embodiment), and the vibration is generated in the same way. However, since the frequency is twice, the forbidden band for suppressing the vibration only needs to be half.

[0104] Other configurations are the same as those in the first embodiment.

[0105] In the present embodiment configured as above, the same effects as those in the first embodiment can also be obtained.

[0106] In addition, when detecting the current used in current control multiple times at equal intervals within a control period and taking the additive average thereof, it is necessary to avoid integer multiples of (6 ÷ number of current detections) as the prohibited band for preventing vibration caused by the six currents. An example of two times is shown in this embodiment, but even if the number of times increases, vibration can be avoided if the setting is made in the same way. In addition, if the periods of the control interrupt int and the current read signal cl are twice as long, the same effect can be obtained even with a phase shift. In addition, if the control period f is set in a manner that avoids the prohibited band c , and a lookup table is set, the same effect can be obtained even if it is different from the lookup table exemplified in Figure 9 this embodiment.

[0107] <Third Embodiment> Refer to Figures 11 - 13 to describe the third embodiment of the present invention.

[0108] This embodiment shows the case of controlling the control period according to the modulation rate, and the modulation rate is obtained by operating based on the dq-axis voltage command value and the DC voltage.

[0109] In this embodiment, only the differences from the first embodiment will be described. In the drawings used in this embodiment, the same reference numerals are given to the same components as those in the first embodiment, and the description will be appropriately omitted.

[0110] Figure 11 is a functional block diagram showing the motor control device of this embodiment together with the related configuration.

[0111] As Figure 11 shown, the motor control device 100B controls the power supplied from the DC voltage source 9 to the PMSM 1 via the power converter 2, and is generally composed of: a phase current detector 3, a magnetic pole position detector 4, a frequency operation unit 5, a DC voltage detection device 6, a coordinate transformation unit 7, 11, a current reading unit 8, a PWM controller 12, a control period operation unit 15B, a control interrupt generation unit 17, a modulation rate operation unit 18, and a current control unit 19. In addition, the coordinate transformation units 7, 11, the modulation rate operation unit 18, and the current control unit 19 constitute a control operation unit 13B.

[0112] The modulation rate operation unit 18 operates the modulation rate Mf based on the dq-axis voltage command values Vd * , Vq * from the current control unit 19 and the DC voltage Vdc from the DC voltage detection device 6, and outputs it to the control period operation unit 15B.

[0113] Figure 12It is a functional block diagram showing the processing content of the control cycle operation unit.

[0114] As Figure 12 shown, the control cycle operation unit 15B calculates the control cycle f based on the speed information ω1 from the frequency operation unit 5 * and the modulation rate Mf from the modulation rate operation unit 18, and outputs it to the control interrupt generation unit 17. It is generally composed of the following: control cycle selection unit (normal time) 21, control cycle selection unit 23 (overmodulation time), modulation mode determination unit 25, and switching unit 22. c

[0115] The modulation mode determination unit 25 determines whether the control of the PMSM1 is performed in the overmodulation region based on the modulation rate Mf.

[0116] The control cycle selection unit (overmodulation time) 23 selects the control cycle f to be used when performing control in the overmodulation region according to the look-up table c . In the control cycle selection unit (overmodulation time) 23, for example, the look-up table shown in the first embodiment is used Figure 2 . Thus, the control cycle f that avoids the current vibration that may occur during control in the overmodulation region is selected c .

[0117] The control cycle selection unit (normal time) 21 selects the control cycle f to be used when the control is not in the overmodulation region according to the Figure 13 look-up table shown c . Figure 13 It is a diagram showing an example of the look-up table used in the control cycle operation unit. The horizontal axis represents the speed information ω1 * , and the vertical axis represents the control cycle f c . When performing control within the range of the non-overmodulation region, since there is no need to consider the forbidden band, as Figure 13 shown, the relationship between the speed information ω1 * and the control cycle f c is used, and the control cycle f * becomes a look-up table of f = f1 regardless of the speed information ω1 c . At this time, the control cycle f1 does not need to consider the suppression of current vibration, so it can be set higher.

[0118] When the modulation mode determination unit 25 determines that the control is being performed in the overmodulation region, the switching unit 22 switches the output to the control interrupt generation unit 17 to the control cycle f selected by the control cycle selection unit (overmodulation time) 23 c ​Further, when the switching unit 22 determines in the modulation mode determination unit 25 that the control is not in the overmodulation region, the output to the control interruption generation unit 17 is switched to the control period f selected by the control period selection unit (normal time) 21 c (e.g., f c = f1).

[0119] The other configurations are the same as those of the first embodiment.

[0120] In the present embodiment configured as described above, the same effects as those of the first embodiment can also be obtained.

[0121] Further, in the present embodiment, considering that no 6f vibration is generated in the dq-axis current in the normal region (when it is not in the overmodulation region), the control period f that avoids the prohibited band is selected only in the overmodulation region where vibration occurs c , so vibration can be suppressed during overmodulation, and the control period f during normal times c can be set high, improving control performance.

[0122] <Fourth Embodiment> Refer to Figure 14 and Figure 15 to describe the fourth embodiment of the present invention.

[0123] This embodiment shows a case where the operation method for changing the control period and the PWM control method are based on whether the control is synchronous PWM or asynchronous PWM.

[0124] In the present embodiment, only the differences from the first embodiment will be described. In the drawings used in the present embodiment, the same reference numerals are assigned to the same components as those of the first embodiment, and the description will be appropriately omitted.

[0125] Figure 14 is a functional block diagram showing the motor control device of the present embodiment together with associated configurations.

[0126] As Figure 14 shown, the motor control device 100C controls the power supplied from the DC voltage source 9 to the PMSM 1 via the power converter 2, and is generally composed of: a phase current detector 3, a magnetic pole position detector 4, a frequency operation unit 5, a DC voltage detection device 6, coordinate transformation units 7, 11, a current reading unit 8, a PWM controller 12A, a control period operation unit 15C, a synchronous PWM signal generation unit 16, a control interruption generation unit 17, and a current control unit 19. In addition, the coordinate transformation units 7, 11, the synchronous PWM signal generation unit 16, and the current control unit 19 constitute a control operation unit 13C.

[0127] The synchronous PWM signal generation unit 16 calculates the ignition angles αu * 、αv * 、αw * used in synchronous PWM based on the dq-axis voltage command values Vd * 、Vq * from the current control unit 19 and the DC voltage Vdc from the DC voltage detection device 6, and outputs them to the PWM controller 12A.

[0128] In addition, the synchronous PWM signal generation unit 16 calculates the modulation rate based on the dq-axis voltage command values Vd * 、Vq * from the current control unit 19 and the DC voltage Vdc from the DC voltage detection device 6, determines whether it is synchronous PWM control (hereinafter referred to as synchronous PWM mode) or asynchronous PWM control (hereinafter referred to as asynchronous PWM mode) according to the calculated modulation rate, and outputs a PWM mode signal indicating which mode to the control cycle calculation unit 15C and the PWM controller 12A.

[0129] When the PWM mode signal from the synchronous PWM signal generation unit 16 indicates the synchronous PWM mode, the PWM controller 12A compares the ignition angles αu * 、αv * 、αw * from the synchronous PWM signal generation unit 16 with the magnetic pole position θ * from the magnetic pole position detector 4 to generate a gate signal. On the other hand, when the PWM mode signal from the synchronous PWM signal generation unit 16 indicates the asynchronous PWM mode, the PWM controller 12A calculates a duty ratio signal based on the three-phase voltage command values Vu * 、Vv * 、Vw * from the coordinate transformation unit 11 and the DC voltage information Vdc from the DC voltage detection device 6, compares the calculated duty ratio signal with a carrier wave, thereby generates a gate signal, and outputs it to the power converter 2.

[0130] Figure 15 is a functional block diagram showing the processing content of the control cycle calculation unit.

[0131] As Figure 15 shown, the control cycle calculation unit 15C calculates the control cycle f * based on the speed information ω1 c from the frequency calculation unit 5 and the modulation rate Mf from the modulation rate calculation unit 18, and outputs it to the control interrupt generation unit 17. It is roughly composed of: a control cycle selection unit (normal time) 21, a control cycle selection unit 24 (synchronous PWM time), and a switching unit 22.

[0132] The control period selection unit (during synchronous PWM) 24 selects the control period f used in the case of the synchronous PWM mode according to the look-up table c . In the control period selection unit (during synchronous PWM) 24, for example, the look-up table shown in the first embodiment is used Figure 2 . Thus, the control period f that avoids the current vibration that may occur during the control of synchronous PWM is selected c .

[0133] The control period selection unit (during normal operation) 21 selects the control period f used in the case of the asynchronous PWM mode according to the look-up table c . In the control period selection unit (during normal operation), for example, the look-up table shown in the third embodiment is used Figure 13 . Thus, in the asynchronous PWM mode where there is no need to consider current vibration suppression and no need to consider the forbidden band, the control period f c can be set higher

[0134] When the PWM mode signal from the synchronous PWM signal generation unit 16 indicates the synchronous PWM mode, the switching unit 22 switches the output to the control period f selected by the control period selection unit (during synchronous PWM) 24 c . In addition, when the PWM mode signal from the synchronous PWM signal generation unit 16 indicates the asynchronous PWM mode, the switching unit 22 switches the output to the control period f selected by the control period selection unit (during normal operation) 21 c .

[0135] Other configurations are the same as those in the first embodiment

[0136] In the present embodiment configured as described above, the same effects as those in the first embodiment can also be obtained

[0137] In addition, in the present embodiment, considering that no 6f vibration occurs in the dq-axis current in the normal region, the control period f that avoids the forbidden band is selected only in the overmodulation region where vibration occurs c , so vibration can be suppressed during synchronous PWM, and the control period f during normal operation c can be set higher, and the control performance can be improved

[0138] <Fifth Embodiment> Refer to Figure 16 and Figure 17 to describe the fifth embodiment of the present invention

[0139] This embodiment shows a case where the gain of current control is changed according to the cut-off frequency calculated based on speed information.

[0140] In this embodiment, only the differences from the third embodiment will be described. In the drawings used in this embodiment, the same reference numerals are given to the same components as in the third embodiment, and the description will be appropriately omitted.

[0141] Figure 16 It is a functional block diagram showing the motor control device of this embodiment together with the associated configuration.

[0142] As Figure 16 shown, the motor control device 100D controls the power supplied from the DC voltage source 9 to the PMSM 1 via the power converter 2, and is generally composed of: a phase current detector 3, a magnetic pole position detector 4, a frequency calculation unit 5, a DC voltage detection device 6, a coordinate transformation unit 7, 11, a current reading unit 8, a PWM controller 12, a current control cut-off frequency calculation unit 14, a control period calculation unit 15D, a control interruption generation unit 17, a modulation rate calculation unit 18, and a current control unit 19A. In addition, the coordinate transformation unit 7, 11, the modulation rate calculation unit 18, and the current control unit 19A constitute a control calculation unit 13D.

[0143] The current control cut-off frequency calculation unit 14 calculates and changes the current control cut-off frequency f * based on the speed information ω1 from the frequency calculation unit 5 acr , and outputs it to the current control unit 19.

[0144] The current control unit 19A changes the gain of current control according to the current control cut-off frequency f acr from the current control cut-off frequency calculation unit 14.

[0145] Figure 17 It is a functional block diagram showing the processing content of the control period calculation unit.

[0146] As Figure 17 shown, the control period calculation unit 15D calculates the control period f * based on the speed information ω1 from the frequency calculation unit 5, the modulation rate Mf from the modulation rate calculation unit 18, and the current control cut-off frequency f acr from the current control cut-off frequency calculation unit 14 c , and outputs it to the control interruption generation unit 17. It is generally composed of a modulation mode determination unit 25 and a control period selection unit 31, 33.

[0147] The modulation mode determination unit 25 determines whether the control of the PMSM 1 is performed in the overmodulation region according to the modulation rate Mf.

[0148] The control period selection unit 31 selects the control period f according to, for example, Figure 13 the look-up table shown c . Figure 13 The look-up table shown sets the control period f when the stop band does not need to be considered c .

[0149] When it is determined in the modulation mode determination unit 25 that control is being performed in the overmodulation region, the control period selection unit 33 changes the control period (before restriction) from the control period selection unit 31 to not enter the stop band (refer to (Equation 5) of the first embodiment, acr etc.), and outputs it as the control period f Figure 13 to the control interruption generation unit 17. c

[0150] Other configurations are the same as those of the third embodiment.

[0151] In the present embodiment configured as described above, the same effects as those of the third embodiment can also be obtained.

[0152] In addition, in the present embodiment, it is also possible to cope with the case where the current control cut-off frequency is variable. Even if the current control cut-off frequency f acr varies according to the speed information ω1 * , current vibration can be suppressed.

[0153] In addition, in the present embodiment, a configuration in which the current control cut-off frequency is changed according to the speed information ω1 * is illustrated, but it is not limited thereto. For example, even if the current control cut-off frequency is changed according to other information such as the torque command value, the same effects as those of the present embodiment can be obtained.

[0154] <Sixth Embodiment> Refer to Figure 18 and Figure 19 to describe the sixth embodiment of the present invention.

[0155] This embodiment shows a case where the current read signal is controlled to be ON at a cycle twice that of the interrupt signal, and the gain of the current control is changed according to the cut-off frequency calculated based on the speed information, and the two-phase three-phase current is read according to the current read signal.

[0156] In the present embodiment, only the differences from the second embodiment will be described. In the drawings used in the present embodiment, the same reference numerals are assigned to the same components as those of the second embodiment, and the description will be appropriately omitted.

[0157] ​Figure 18 This is a functional block diagram showing the motor control device of the present embodiment together with associated components.

[0158] As Figure 18 shown, the motor control device 100E controls the power supplied from the DC voltage source 9 to the PMSM 1 via the power converter 2, and generally consists of the following components: a phase current detector 3, a magnetic pole position detector 4, a frequency operation unit 5, a DC voltage detection device 6, coordinate transformation units 7B and 11, a current reading unit 8A, a PWM controller 12, a current control cut-off frequency operation unit 14, a control cycle operation unit 15E, a control interruption generation unit 17A, and a current control unit 19A. In addition, the coordinate transformation units 7B and 11 and the current control unit 19A constitute the control operation unit 13E.

[0159] The control interruption generation unit 17A generates and outputs a current reading signal cl, and makes the current reading signal cl become ON at a period twice that of the interruption signal int.

[0160] The current control cut-off frequency operation unit 14 calculates and changes the current control cut-off frequency f * based on the speed information ω1 from the frequency operation unit 5 acr , and outputs it to the current control unit 19.

[0161] The current control unit 19A changes the gain of the current control according to the current control cut-off frequency f acr from the current control cut-off frequency operation unit 14.

[0162] In the control cycle operation unit 15E, for example, the control cycle f Figure 13 is selected according to the look-up table shown c . Figure 13 The look-up table shown c sets the control cycle f when the forbidden band does not need to be considered.

[0163] The coordinate transformation unit 7B determines whether the control cycle f * enters the forbidden band based on the speed information ω1 from the frequency operation unit 5 c , the control cycle f from the control cycle operation unit 15E acr , and the current control cut-off frequency f from the current control cut-off frequency operation unit 14 c (refer to (Equation 12), (Equation 13) of the second embodiment, Figure 9 etc.).

[0164] When it is determined that the control cycle f cWhen the prohibited frequency band is not entered, the coordinate transformation unit 7B corrects the deviation at the moment of reading the three-phase currents Iuc1, Ivc1, Iwc1 and the three-phase currents Iuc2, Ivc2, Iwc2 into a rotation angle, and transforms their coordinates into dq coordinates, adds and averages the dq axis currents after the coordinate transformation, and outputs them as dq axis current detection values Idc and Iqc to the current control unit 19A.

[0165] In addition, when it is determined that the control cycle f c When entering the prohibited band, the coordinate conversion unit 7B switches to a method of alternately calculating the dq-axis current detection values Idc and Iqc, and outputs the calculated dq-axis current detection values Idc and Iqc to the current control unit 19A.

[0166] Figure 19 This is a diagram showing the relationship between the three-phase fundamental current, the dq-axis sixth-order harmonic current, and the reading time. Figure 19 The three-phase fundamental current 61C, the dq-axis sixth-order harmonic current 63C, and the reading time are shown.

[0167] like Figure 19 As shown, in the method of alternately calculating the dq axis current detection values Idc and Iqc, the value of (n) is used in interrupt 1, the value of (n+3) is used in interrupt 2, and the value of (n+4) is used in interrupt 3. As a result, since the sixth-order component appears as a high-order harmonic component on the dq axis, it can be controlled outside the response band by current control. That is, the control period can be fixed, and the control response can be improved.

[0168] The other configurations are the same as those of the second embodiment.

[0169] In the present embodiment configured as described above, the same effects as those of the second embodiment can be obtained.

[0170] <Notes> In addition, the present invention is not limited to the above-mentioned embodiment, and includes various modifications and combinations within the scope of the gist thereof. In addition, the present invention is not limited to having all the configurations described in the above-mentioned embodiment, and includes a configuration in which a part of the configuration is deleted.

[0171] For example, the configurations of the third embodiment and the fourth embodiment may be combined to avoid the prohibited frequency band only in the synchronous PWM or overmodulation region. The synchronous PWM and overmodulation regions share a common feature that the switching cycle is not fixed. Therefore, if both the synchronous PWM and overmodulation regions are considered, it can be said that the prohibited frequency band can be avoided only in the region where the switching cycle is not fixed. Explanation of symbols

[0172] 2... Power converter, 3... Phase current detector, 4... Pole position detector, 5... Frequency calculation unit, 6... DC voltage detection device, 7, 7A, 7B... Coordinate transformation unit, 8, 8A... Current reading unit, 9... DC voltage source, 11... Coordinate transformation unit, 12, 12A... PWM controller, 13, 13A, 13B, 13C, 13D, 13E... Control operation unit, 14... Current control cut-off frequency calculation unit, 15, 15A, 15B, 15C, 15D, 15E... Control cycle calculation unit, 16... Synchronous PWM signal generation unit, 17, 17A... Control interruption generation unit, 18... Modulation rate calculation unit, 19, 19A... Current control unit, 21, 23, 24, 33... Control cycle selection unit, 22... Switching unit, 25... Modulation mode determination unit, 31... Control cycle selection unit, 61, 61C... Three-phase fundamental current, 63, 63C... dq-axis 6th harmonic current, 100, 100A, 100B, 100C, 100D, 100E... Motor control device, 101... Transmission, 103... Differential gear, 105... Drive shaft, 107... Wheel.

Claims

1. A motor control device that outputs a voltage applied to an AC motor according to an output voltage command value, The motor control device is characterized by comprising: A control period calculation unit that determines a control period according to the fundamental wave frequency of the AC motor and the cut-off frequency of current control; A control interruption generation unit that determines a control interruption based on the control period determined by the control period calculation unit; A current control unit that controls the output voltage command value based on the control interruption determined by the control interruption generation unit; and A PWM controller that controls the output of the voltage by PWM, and the PWM corresponds to the output voltage command value controlled by the current control unit.

2. The motor control device according to claim 1, wherein The control period calculation unit determines the control period to be in a range greater than 6n times the fundamental wave frequency ± 10 times the cut-off frequency of current control.

3. The motor control device according to claim 2, wherein The control period calculation unit determines the control period to be in a range greater than 6n times the fundamental wave frequency ± 10 times the cut-off frequency of current control only when the switching period of the PWM controller is not fixed.

4. The motor control device according to claim 1, wherein The control period calculation unit determines the control period according to the fundamental wave frequency of the AC motor, the cut-off frequency of current control, and the modulation rate.

5. The motor control device according to claim 1, wherein The current control unit changes the gain of current control based on the control interruption determined by the control interruption generation unit and the cut-off frequency.

6. An electric drive vehicle equipped with an AC motor, characterized in that, Comprising: The motor control device according to any one of claims 1 to 5 that outputs a voltage applied to the AC motor.

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