Multistage filtering and amplifying suppressor

Abstract

The invention discloses a multi-stage filtering and amplifying suppressor which comprises a signal acquisition module, a space attenuation and calculation module, a multi-stage filtering and amplifying module and a simulation verification module. The signal acquisition module calculates the oscillator distribution spacing according to the frequency of the received signal by acquiring the frequency of the received signal and acquiring an interference signal, and the space attenuation and calculation module calculates the space attenuation amount according to the spacing between the transmitting antenna and the receiving antenna; the multi-stage filtering and amplifying module is used for performing multi-stage filtering and amplifying on a received signal according to an interference signal and a space attenuation amount, and processing insertion loss generated by multi-stage filtering and amplifying at the same time; through structural isolation and a multi-stage filtering amplification circuit, a transmitting signal A1 is gradually suppressed from 40dBm to-75dBm, a receiving signal B1 is increased from-100dBm to-56dBm, and finally, the signal A1 is lower than the signal B1, so that the problem of interference of the transmitting signal of the short message system on a receiving channel is solved.

Description

Technical Field

[0001] The present invention relates to the technical field of navigation communication devices, and particularly relates to a multi-stage filtering amplifier suppressor. Background Art

[0002] The airborne short message transceiver system belongs to the navigation communication system, and is composed of components such as a transceiver antenna, a short message transceiver, a short message control handle, a locking cross-linking box, a mounting bracket, a handle bracket, and a radio frequency cable. It is mainly used to receive, send, and display short message information, obtain the current position through the on-board navigation system, and realize point-to-point communication.

[0003] Due to requirements such as size limitations and installation methods, it is usually necessary to integrally design the short message system and the receiving antenna. Among them, the frequency point of the signal received by the short message system in B1 of BD3 is 1575.42 MHz ± 16.368 MHz, and the frequency point range of the signal received by the transmitting antenna is from 1615 MHz to 1627 MHz. Since the two frequency points are relatively close, there will be a phenomenon that the transmitting signal of the short message system directly interferes with the receiving signal channel of the receiving antenna, resulting in saturation distortion of the receiving antenna signal channel and causing the receiver to be unable to perform positioning and calculation.

[0004] Traditional interference suppression methods mainly rely on filtering technology, and filter out or attenuate interference signals by designing filters with specific frequency responses. However, this method has obvious limitations. On the one hand, while suppressing interference, the filter often also attenuates the target signal to a certain extent, especially when the spectra of the interference signal and the target signal overlap, the filtering effect will be greatly reduced. On the other hand, for a complex and changing interference environment, a single filter is difficult to achieve all-round interference suppression, and it is necessary to design a more complex filtering network or adopt a multi-stage filtering architecture, which undoubtedly increases the complexity and cost of the system. Summary of the Invention

[0005] The present application provides a multi-stage filtering amplifier suppressor, which gradually suppresses the transmitted signal A1 from 40 dBm to -75 dBm and simultaneously boosts the received signal B1 from -100 dBm to -56 dBm through structural isolation and a multi-stage filtering amplifier circuit, and finally makes the A1 signal lower than the B1 signal, solving the problem of interference of the transmitted signal of the short message system to the receiving channel.

[0006] The present application provides a multi-stage filtering amplifier suppressor, comprising: a signal acquisition module, a space attenuation and calculation module, a multi-stage filtering amplifier module, and a simulation verification module; the signal acquisition module calculates the oscillator distribution pitch according to the frequency of the received signal by acquiring the frequency of the received signal and collecting interference signals, converts the oscillator distribution pitch into the distance between the transmitting and receiving antennas, and the space attenuation and calculation module calculates the space attenuation amount through the distance between the transmitting and receiving antennas; the multi-stage filtering amplifier module performs multi-stage filtering and amplification on the received signal according to the interference signal and the space attenuation amount, and simultaneously processes the insertion loss generated by the multi-stage filtering and amplification; the simulation verification module constructs a simulation model according to the results of the multi-stage filtering and amplification to verify the results of the multi-stage filtering and amplification.

[0007] Preferably, the oscillator distribution pitch is calculated according to the acquired frequency of the received signal, and the formula is: , where L is the oscillator distribution pitch; f is the frequency of the received signal; C is the speed of light, with a value of 3×10 8 m / s.

[0008] Preferably, the unit of the oscillator distribution pitch is converted to kilometers and then used as the distance between the transmitting and receiving antennas, and the space attenuation amount is calculated through the distance between the transmitting and receiving antennas. The formula is: , where is the space attenuation amount; 32.45 is a constant term, representing the parameter of the automatic space path loss model, that is, the power attenuation of the signal from the transmitting end to the receiving end due to the propagation distance and frequency; F is the frequency; D is the distance between the transmitting and receiving antennas.

[0009] Preferably, the interference signal is subjected to multi-stage filtering and amplification through the space attenuation amount. First, a filter is used to perform primary filtering on the interference signal, and at the same time, the signal at the transmitting end is attenuated; then, the interference signal after primary filtering is subjected to primary amplification, and the primary amplification is performed using a low-noise amplifier; subsequently, the interference signal after primary amplification is subjected to secondary filtering; then, the interference signal passing through the secondary filtering is subjected to secondary amplification, and the secondary amplification is also performed using a low-noise amplifier; finally, the interference signal after secondary amplification is subjected to tertiary filtering.

[0010] Preferably, the multi-stage filtering amplifier module is used for processing the insertion loss, specifically: S201, using a power splitter to equally divide the received signal into multiple paths to form multiple receiving channels, calculating the noise figure in a single channel through the insertion loss of the filter, and at the same time, calculating the time delay difference; S202, the multiple receiving channels share the same local oscillator; S203, using a signal combining algorithm to calculate the insertion loss of multiple channels, and calculating the noise figure of multiple channels according to the noise figure in a single channel.

[0011] Preferably, the noise figure of a single channel in multiple channels is calculated, and the formula is: , where is the noise figure of a single channel; is the noise figure of the power splitter; is the noise figure of the filter; is the gain of the power splitter; is the gain of the filter; is the noise figure of the primary low-noise amplifier; is the gain of the primary low-noise amplifier; is the noise figure of the secondary low-noise amplifier. The formula for calculating the noise figure of multiple channels is: , where is the noise figure of multiple channels, is the noise figure of a single channel, N is the number of channels, and the noise figure of the single channel is the noise figure of the first receiving channel or the second receiving channel.

[0012] Preferably, the calculation formula for the insertion loss of multiple channels is: , where is the equivalent insertion loss after combining multiple channels, is the insertion loss of a single channel, is the signal-to-noise ratio gain brought by MRC combining, and N is the number of channels.

[0013] Preferably, the steps for the space attenuation and calculation module to calculate the time delay difference are as follows: S301: Divide the signal at the transmitting end into real-time data and calibration data, and divide the interference levels according to the power and signal-to-noise ratio of the received signal at the receiving end; S302: Perform time delay matching on multiple channels, and obtain the time delay difference of multiple channels according to the time delay matching; S303: Compensate multiple channels according to the obtained time delay difference.

[0014] Preferably, the real-time data is communication data. The transmitting end is equipped with a pseudo-random sequence generation module, and the pseudo-random sequence generation module generates a pseudo-random sequence according to preset algorithms and parameters, and the pseudo-random sequence is used as calibration data.

[0015] One or more technical solutions provided in this application have at least the following technical effects or advantages: Through structural isolation (10 cm spacing) and a multi-stage filtering and amplification circuit (three-stage filtering + two-stage amplification), the transmitted signal A1 is gradually suppressed from 40 dBm to -75 dBm, while the received signal B1 is increased from -100 dBm to -56 dBm. Finally, the A1 signal is lower than the B1 signal, solving the interference problem of the transmitted signal of the short message system on the receiving channel. The simulation and measured data verify the effectiveness of the design, achieving efficient suppression of strong interference signals and precise amplification of weak received signals; Through the dual-channel diversity reception architecture and the maximum ratio combining technology, the system can significantly improve the reception performance of the B1 signal: By reducing signal correlation through space / polarization diversity and enhancing the anti-multipath fading ability, the dual-channel system optimizes the insertion loss of a single channel from 3.5 dB to 0.5 dB through MRC combining. After dual-channel MRC combining, the noise figure is reduced from 3.92 dB of a single channel to 2.42 dB; By dynamically adjusting the insertion density of calibration data, accurate dual-channel time delay estimation and compensation are achieved under different interference scenarios, thereby ensuring the correct reception and processing of signals, improving the performance and stability of the communication system, classifying data, dynamically scheduling calibration data, calculating the time delay difference, and calibrating the channel signals according to the time delay difference to improve the time delay compensation accuracy and reduce the pseudorange error; By actively injecting reverse interference signals, the system can achieve precise suppression and effective control of specific interference signals, improve signal quality, optimize spectral characteristics, reduce the impact of high-frequency leakage and low-frequency baseband noise on the communication system, thereby significantly improving the performance and reliability of the communication system. The generated reverse interference signals are divided into four sub-bands: the core suppression band, the edge suppression band, the high-frequency leakage suppression band, and the low-frequency leakage suppression band according to the signal frequency domain characteristics. Each sub-band is designed for different interference characteristics, achieving precise control of interference signals. Description of the Drawings

[0016] Figure 1 It is the schematic diagram of a multi-stage filtering and amplification suppressor of the present invention; Figure 2 It is the simulation result diagram of multi-stage filtering, amplification and suppression of the present invention; Figure 3 It is the actual test result diagram of multi-stage filtering, amplification and suppression of the present invention; Figure 4 It is the block diagram of a multi-stage filtering and amplification suppressor of the present invention; Figure 5 It is the schematic flow diagram of reducing insertion loss by using multiple receiving channels in the embodiment of the present invention; Figure 6 It is the schematic flow diagram of calculating the time delay difference in the embodiment of the present invention; Figure 7 This is a schematic flow chart of controlling the original interference by injecting reverse interference signals in the embodiments of the present invention. Specific embodiments

[0017] To facilitate the understanding of the present invention, the present application will be described more comprehensively with reference to the relevant drawings. The preferred embodiments of the present invention are shown in the drawings. However, the present invention can be implemented in many different forms and is not limited to the embodiments described herein. On the contrary, these embodiments are provided to make the disclosure of the present invention more thorough and comprehensive.

[0018] It should be noted that the terms "vertical", "horizontal", "upper", "lower", "left", "right" and similar expressions used herein are for illustrative purposes only and do not represent the only embodiments.

[0019] Unless otherwise defined, all technical and scientific terms used herein have the same meaning as commonly understood by those skilled in the technical field to which the present invention belongs. The terms used in the specification of the present invention are only for the purpose of describing specific embodiments and are not intended to limit the present invention. The term "and / or" used herein includes any and all combinations of one or more of the related listed items.

[0020] Embodiment 1: Figure 4 This is a schematic flow chart of a multi-stage filtering amplifier suppressor in the embodiments of the present invention, including: a signal acquisition module, a space attenuation and calculation module, a multi-stage filtering amplifier module, and a simulation verification module. The signal acquisition module is electrically connected to the space attenuation and calculation module, the space attenuation and calculation module is electrically connected to the multi-stage filtering amplifier module, and the multi-stage filtering amplifier module is electrically connected to the simulation verification module; The signal acquisition module calculates the oscillator distribution spacing according to the frequency of the received signal by obtaining the frequency of the received signal and the interference signal, converts the oscillator distribution spacing into the spacing between the transmitting and receiving antennas, and the space attenuation and calculation module calculates the space attenuation amount through the spacing between the transmitting and receiving antennas; The frequency of the received signal is obtained by using a spectrum analyzer. The signal of the standard signal generator and the received signal are input into the spectrum analyzer at the same time, and the standard signal frequency is adjusted to make its spectral line coincide with the spectral line of the received signal. At this time, the frequency of the standard signal is the frequency of the received signal. The oscillator distribution spacing is calculated according to the obtained frequency of the received signal. The formula is: , where L is the oscillator distribution spacing; f is the frequency of the received signal; C is the speed of light, with a value of 3×10 8 m / s, convert the oscillator distribution spacing into km as the spacing between the transmitting and receiving antennas, and calculate the space attenuation amount through the spacing between the transmitting and receiving antennas. The formula is: , where is the spatial attenuation; 32.45 is a constant term representing the parameters of the automatic spatial path loss model, that is, the power attenuation of the signal from the transmitter to the receiver due to the propagation distance and frequency; F is the frequency (the signal transmission frequency is the same as the reception frequency); D is the distance between the transmitting and receiving antennas.

[0021] A specific example is as follows: An airborne short message system adopts an integrated design. The transmitted signal is A1, and the signal received by the receiving antenna is B1, with both frequencies being 1575.42 MHz. To enhance the anti-interference ability, the system uses a multi-element antenna, and the spacing between the oscillators needs to be optimized according to the signal frequency and spatial attenuation. C is 3×10 8 m / s, f is 1575.42 MHz. Substitute them into the formula: , D = 0.0001 km, F is 1575.42 MHz. Substitute F and D into the formula: = 16.53 dB. In this embodiment, the spatial attenuation is simplified to 15 dB.

[0022] The multi-stage filtering and amplification module performs multi-stage filtering and amplification on the received signal according to the interference signal and the spatial attenuation; connect to the antenna under test through a spectrum analyzer, set the scanning parameters covering the target frequency band, combine a directional antenna to locate the interference direction, and use a high-resolution bandwidth and detection mode to capture transient or interference characteristics. Obtain the interference signal through time-frequency analysis software. Since the obtained interference signal will cause distortion of the received signal, filter out the interference signal.

[0023] Perform multi-stage filtering and amplification on the interference signal through the spatial attenuation, such as Figure 1As shown, first, a high-power-tolerant filter is used to perform primary filtering on the interference signal. This filter filters out the interference signals outside the target frequency band through its high-selectivity suppression characteristics, and at the same time implements controllable attenuation on the signals in the transmitting channel to balance the signal strength and system security; then, the interference signal after primary filtering is subjected to primary amplification. The primary amplification is carried out using a low-noise amplifier. The low-noise amplifier controls the noise contribution while providing appropriate gain, avoiding non-linear distortion of the amplifier caused by signal overload. The amplifier precisely matches the input signal strength and, on the premise of ensuring that the transmitted signal will not cause saturation, performs basic enhancement on the weak received signal; subsequently, the interference signal after primary amplification is subjected to secondary filtering. The secondary filtering uses a high-power-tolerant surface acoustic wave filter. This filter uses surface acoustic wave technology to achieve higher-precision spectrum shaping. Its out-of-band rejection ability can further eliminate the remaining out-of-band interference components, and at the same time maintain continuous attenuation of the transmitted signal. At this stage, through a steeper roll-off characteristic, a clearer spectrum isolation is formed between the transmitted signal and the received signal; then, the interference signal passing through secondary filtering is subjected to secondary amplification. The secondary amplification is also carried out using a low-noise amplifier. This amplifier provides higher gain while maintaining low-noise characteristics, and gradually increases the received signal strength through the cascading effect of multiple amplification stages. At this stage, while continuously attenuating the transmitted signal strength, a stepped enhancement is implemented on the received signal to ensure that a controllable level difference is always maintained between the two in the processing path; finally, the interference signal after secondary amplification is subjected to tertiary filtering. The filter is a cavity filter. This cavity filter uses the high-Q value characteristic of the metal resonant cavity to achieve ultra-wideband out-of-band rejection, and implements ultimate attenuation on the remaining transmitted signals to ensure the purity of the receiving channel. While eliminating all out-of-band interference, through a fine impedance matching design, signal distortion is minimized to the greatest extent; the high-power-tolerant filter, low-noise amplifier, high-power-tolerant surface acoustic wave filter, and cavity filter are connected in series to form a circuit.

[0024] A specific example is as follows: A satellite communication system needs to suppress the interference of the transmitted signal (marked as A1 signal) on the receiving channel (marked as B1 channel) to ensure that the B1 satellite signal (-100 dBm initial intensity) is not overwhelmed at the receiving end. The system adopts a series architecture of three-stage filtering and two-stage amplification, and the specific implementation is as follows: In the primary filtering stage, a high-power-resistant ceramic dielectric filter is selected. The out-of-band rejection of the ceramic dielectric filter is ≥ 40 dB. The initial intensity of the A1 signal is 40 dBm, and the antenna coupling path attenuation is 15 dB. The intensity of the A1 signal after coupling = initial A1 signal intensity - antenna coupling path attenuation, P 耦 = 40 dBm - 15 dB = 25 dBm. The intensity of the A1 signal after primary filtering = intensity of the A1 signal after coupling - out-of-band rejection of the filter, P 处滤= 25 dBm - 40 dB = -15 dBm. The intensity of signal A1 drops to -15 dBm. Since signal B1 does not enter the filter frequency band, its intensity remains at -100 dBm. In the primary amplification stage, a gallium arsenide low-noise amplifier (LNA) is selected. The gain of the gallium arsenide low-noise amplifier is 20 dB, and the noise figure is 0.6 dB. After the first-stage amplification, the intensity of signal A1 = the intensity of signal A1 after filtering + the amplifier gain, PA 一放 = -15 dBm + 20 dB = 5 dBm. After the first-stage amplification, the intensity of signal B1 = the initial intensity of signal B1 + the amplifier gain, PB 一放 = -100 dBm + 20 dB = -80 dBm. The intensity of signal A1 is 5 dBm, and the intensity of signal B1 increases to -80 dBm. In the second-stage filtering stage, a high-power-resistant surface acoustic wave filter is selected. The out-of-band rejection of the high-power-resistant surface acoustic wave filter is ≥ 40 dB, and the filter insertion loss (in-band) is 3 dB. After the second-stage filtering, the intensity of signal A1 = the intensity of signal A1 after the first-stage amplification - the out-of-band rejection of the filter, PA 二滤 = 5 dBm - 40 dB = -35 dBm. After the second-stage filtering, the intensity of signal B1 = the intensity of signal B1 after the first-stage amplification - the filter insertion loss, PB 二滤 = -80 dBm - 3 dB = -83 dBm. The intensity of signal A1 drops to -35 dBm, and the intensity of signal B1 drops to -83 dBm. In the second-stage amplification stage, a gallium nitride low-noise amplifier is selected. The gain of the gallium nitride low-noise amplifier is 30 dB, and the noise figure is 1 dB. After the second-stage amplification, the intensity of signal A1 = the intensity of signal A1 after the second-stage filtering + the amplifier gain, PA 二放 = -35 dBm + 30 dB = -5 dBm. The intensity of the amplified signal B1 = the intensity of signal B1 after the second-stage filtering + the amplifier gain, PB 二放 = -83 dBm + 30 dB = -53 dBm. The intensity of signal A1 is -5 dBm, and the intensity of signal B1 increases to -53 dBm. In the third-stage filtering stage, a metal cavity filter is selected. The out-of-band rejection of the metal cavity filter is ≥ 70 dB, and the insertion loss (in-band) is 3 dB. After the third-stage filtering, the intensity of signal A1 = the intensity of signal L after the second-stage amplification - the out-of-band rejection of the filter, PA 三滤 = -5 dBm - 70 dB = -75 dBm. After the third-stage filtering, the intensity of signal B1 = the intensity of signal B1 after the second-stage amplification - the filter insertion loss, PB 三滤 = -53 dBm - 3 dB = -56 dBm. The intensity of signal A1 drops to -75 dBm, and the intensity of signal B1 drops to -56 dBm.

[0025] The simulation and verification module constructs a simulation model based on the results of multi-stage filtering and amplification, verifies the results of multi-stage filtering and amplification, constructs a simulation model using ADS, uses the constructed simulation model to simulate the parameters of each stage of the filter and the gain / phase curve of the amplifier respectively to ensure that the single-stage performance meets the standard, connects the filter and the amplifier according to the actual circuit topology, analyzes the frequency response characteristics after cascading, inputs signal A1 and signal B1, observes the output signal strength of channel B1 and the interference strength of signal A1, calculates the interference rejection ratio, and generates a simulation result graph such as Figure 2 According to the above steps S101 to S102, conduct actual tests on it. Use a finished antenna with multi-stage filtering, amplification and suppression design. The frequency and power acquisition results are as shown in Figure 3 Compare the measured data with the simulation results. If the error is large, check whether the actual parameters of the filter are consistent with the simulation model.

[0026] The technical solutions in the embodiments of the present application at least have the following technical effects or advantages: Through structural isolation (10 cm spacing) and a multi-stage filtering and amplification circuit (three-stage filtering + two-stage amplification), the transmitted signal A1 is gradually suppressed from 40 dBm to -75 dBm, and at the same time, the received signal B1 is increased from -100 dBm to -56 dBm. Finally, signal A1 is lower than signal B1, solving the interference problem of the transmitted signal of the short message system to the receiving channel. The simulation and measured data verify the effectiveness of the design, and achieve efficient suppression of strong interference signals and precise amplification of weak received signals.

[0027] Embodiment 2: Based on the insertion loss of the filter in Embodiment 1, each stage of the filter will also cause a certain insertion loss to the target signal (signal B1) while suppressing the interference signal. For example, there is a 3 dB loss of signal B1 after two-stage filtering. If the cumulative loss of multi-stage filtering is too large, resulting in too low signal strength of signal B1, it will affect the receiving sensitivity and reduce the overall performance of the system. Especially when receiving weak signals, it cannot meet the signal-to-noise ratio required by the system. In this embodiment, by using multiple receiving channels (hierarchical reception), each channel is filtered separately and then the signals are combined. The influence of the insertion loss of a single channel is offset by statistical gain, as shown in Figure 5 shown.

[0028] S201, use a power divider to evenly divide the received signal at the receiving end into multiple paths to form multiple receiving channels, and calculate the noise figure in a single channel through the insertion loss of the filter; Further, the received signal is input into the input port of a Wilkinson power divider through an antenna. The Wilkinson power divider is a passive power distribution device used to evenly divide the input signal into two outputs while maintaining the isolation between the output ports. The two output ports of the Wilkinson power divider are respectively connected to two independent radio frequency receiving channels, namely the first receiving channel and the second receiving channel. Both the first receiving channel and the second receiving channel include the same filter, low-noise amplifier, and ADC.

[0029] Calculate the insertion loss of the first receiving channel. In the first receiving channel, the insertion loss mainly comes from the power divider loss and the filter insertion loss. Calculate the noise figure of the first receiving channel according to the noise figures of the power divider, filter, and low-noise amplifier. The formula is: , where is the noise figure of the first receiving channel; is the noise figure of the power divider; is the noise figure of the filter; is the gain (linear value) of the power divider; is the gain of the filter; is the noise figure of the primary low-noise amplifier; is the gain of the primary low-noise amplifier; is the noise figure of the secondary low-noise amplifier.

[0030] S202, the first receiving channel and the second receiving channel share the same local oscillator; Specifically, the first receiving channel and the second receiving channel share the same local oscillator (LO) signal, making the phases of the signals after frequency conversion in the first receiving channel and the second receiving channel consistent. The local oscillator signal provides a unified carrier frequency and phase reference for the first receiving channel and the second receiving channel. Pilot signals with known frequencies and phases are superimposed in the first receiving channel and the second receiving channel. The Fourier transform is performed on the pilot signals of the two channels to extract their phase information, and the phase difference of the second receiving channel relative to the first receiving channel is calculated, and phase rotation compensation is performed on the second receiving channel.

[0031] S203, use a signal combining algorithm to calculate the insertion loss of multiple channels, and calculate the noise figures of multiple channels according to the noise figures within a single channel; Specifically, in this embodiment, the multiple channels are two channels. In a two-channel receiving system, the insertion loss of the two channels is calculated by using a combining algorithm. This combining algorithm is the maximum ratio combining (MRC) technique. MRC is a linear combining technique. By performing weighted summation on the signals of each channel, the signal-to-noise ratio of the combined signal is maximized. If the signal-to-noise ratios of the two channels are the same, the weights are both 0.5. For two independent channels, the signal-to-noise ratio after MRC combination is the sum of the signal-to-noise ratios of each channel: SNR合并 = SNR1 + SNR2, where SNR 合并 is the combined signal-to-noise ratio, SNR1 is the signal-to-noise ratio of the first receiving channel, and SNR2 is the signal-to-noise ratio of the second receiving channel. Since SNR1 = SNR2 = SNR, then SNR 合并 = 2SNR. The formula for the improvement of the multi-channel equivalent insertion loss is: , where is the equivalent insertion loss after multi-channel combination, referring to two channels here, is the insertion loss of a single channel, is the signal-to-noise ratio gain brought by MRC combination, and N = 2.

[0032] After the two channels are combined by MRC, the formula for calculating the combined noise figure is: , where is the combined two-channel noise figure, is the noise figure of the first receiving channel, and N is the number of channels.

[0033] A specific example is: The single-channel insertion loss mainly comes from the power splitter and the filter. The loss of the power splitter is 0.5 dB, and the loss of the filter is 3 dB, = 0.5 dB + 3 dB = 3.5 dB, = 3.5 dB - 3.0 dB = 0.5 dB. For the calculation of the single-channel noise figure: , where = 1, (power splitter), = 10 3 / 10 ≈ 2.0, (filter), = 10 0.6 / 10 ≈ 1.15, = 10 20 / 10 = 100 (primary LNA), = 10 1 / 10 ≈ 1.26, = 10 30 / 10 = 1000 (secondary LNA). Substituting it in, we get: = 1 + 1.122 + 0.336 + 0.0058 = 2.4638. Converting it to dB representation: NF 单 = 10 lg(2.4638) ≈ 3.92 dB, ≈ 1.232. Converting it to dB representation: NF MRC = 10 lg(1.232) ≈ 0.9 dB.

[0034] The technical solutions in the embodiments of the present application at least have the following technical effects or advantages: Through the dual-channel diversity reception architecture and the maximum ratio combining technology, the system can significantly improve the reception performance of the B1 signal: By reducing the signal correlation through space / polarization diversity and enhancing the anti-multipath fading ability, the dual-channel system optimizes the insertion loss of a single channel from 3.5 dB to 0.5 dB through MRC combining. After the dual-channel MRC combining, the noise figure is reduced from 3.92 dB of a single channel to 0.9 dB.

[0035] Embodiment 3: Based on the inter-symbol interference caused by the time delay mismatch in the technical solutions of Embodiment 1 and Embodiment 2 above, this embodiment dynamically calibrates the time delay difference to achieve signal phase alignment and improve efficiency, as Figure 6 shown.

[0036] S301: Divide the signal at the transmitting end into real-time data and calibration data, and divide the interference level according to the power and signal-to-noise ratio of the received signal at the receiving end; Furthermore, the real-time data is the data for normal communication, and the real-time data contains the information that needs to be transmitted in the actual service. The transmitting end is equipped with a dedicated pseudo-random sequence generation module, which generates a pseudo-random sequence (PRN code) according to the preset algorithm and parameters. The pseudo-random sequence is used as calibration data. The pseudo-random sequence has good auto-correlation and cross-correlation characteristics and can meet the requirements of time delay calibration.

[0037] Measure the power of the received signal using the acquisition module at the receiving end, separate the signal component and the noise component from the acquired signal, obtain the signal-to-noise ratio using a signal processing algorithm, and comprehensively determine the power threshold based on factors such as the system's performance requirements, signal transmission characteristics, and actual application scenarios. Compare the power of the received signal measured in real time with the set power threshold. If the power of the received signal is less than the set power threshold, it is determined that the current communication environment is a low-interference scenario. In a low-interference scenario, the system inserts calibration data once every 10 ms and only transmits the basic time stamp to reduce power consumption while still meeting the basic time-delay calibration requirements. If the power of the received signal is greater than or equal to the set power threshold, it is determined that the current communication environment is a high-interference scenario. At this time, the system inserts calibration data once every 2 ms and can dynamically adjust the insertion density according to the interference intensity to more timely track and compensate for the time-delay change and ensure signal quality. The system pre-sets a signal-to-noise ratio threshold according to the actual system performance and communication environment requirements. The signal-to-noise ratio threshold is obtained through a large number of experiments and simulation analyses and is used to measure the interference intensity and guide the adjustment of the calibration data insertion frequency. When the monitored signal-to-noise ratio is lower than the signal-to-noise ratio threshold, it indicates that the interference intensity has increased. The calibration data insertion control module will adjust the calibration data insertion frequency from once every 2 ms to once every 1.5 ms, which can more timely track the time-delay change and improve the accuracy of time-delay calibration.

[0038] S302. Perform time-delay matching on the first receiving channel and the second receiving channel, and obtain the time-delay difference between the two channels according to the time-delay matching. Specifically, time-delay matching is a mathematical operation used to measure the similarity between two signals. In time-delay estimation, by performing time-delay matching on the signals of the first receiving channel and the second receiving channel, the best matching position between the two signals can be found, thereby determining the time-delay difference. The calculation formula for performing time-delay matching is: , where is the time-delay difference; is the maximum parameter estimation, that is, among all possible time-delay candidate values τ, the value that makes the cross-correlation function maximum is selected as the optimal time-delay estimation; N is the sequence length of time-delay matching, which determines the accuracy and computational complexity of time-delay matching. The longer the sequence length, the more accurate the result of time-delay matching, but the greater the computational amount; represents the signal sequence of the first receiving channel, that is, the received PRN code sequence; τ is the time-delay variable, representing the time-delay offset of the second receiving channel signal relative to the first receiving channel signal.

[0039] S303. Compensate the second receiving channel according to the obtained time-delay difference. Furthermore, according to the calculated time-delay difference , the time-delay difference is a continuous physical quantity, while digital signal processing is carried out in the discrete time domain. Therefore, it is necessary to convert Δτ into the number of discrete sampling points. The sampling period Ts is an important parameter in digital signal processing, which represents the time interval between two adjacent sampling points. By dividing the time delay difference Δτ by the sampling period Ts, the number of sampling points to be compensated can be obtained. The formula for compensating the second receiving channel is: , where is for compensating the second receiving channel, and Ts is the sampling period; is the time delay difference; represents the floor operation. The second receiving channel is aligned with the signal of the first receiving channel through digital delay line compensation.

[0040] The technical solutions in the embodiments of the present application at least have the following technical effects or advantages: By dynamically adjusting the insertion density of calibration data, accurate two-channel time delay estimation and compensation are achieved in different interference scenarios, thereby ensuring the correct reception and processing of signals, improving the performance and stability of the communication system, classifying data, dynamically scheduling calibration data, calculating the time delay difference, calibrating the channel signal according to the time delay difference, improving the time delay compensation accuracy, and reducing the pseudorange error.

[0041] Embodiment 4: According to Embodiment 1 above, passive suppression of interference is achieved through multi-stage filtering, but the target signal is attenuated. In this embodiment, by actively injecting a reverse interference signal, the limitation of sacrificing the target signal strength by actively suppressing the original interference with the reverse interference signal to reduce interference is shown as Figure 7 shown.

[0042] S401, generate a reverse interference signal using a frequency conversion module according to the interference signal; Furthermore, the system uses an NVMe SSD as the storage module, in which the spectral characteristic parameters of the A1 signal are pre-stored. The center frequency, modulation mode, and bandwidth of the A1 signal are read from the NVMe SSD, and the DPUCVD accelerator on the Xilinx RFSoC ZCU111 frequency conversion module is used to generate baseband I / Q data. The baseband generation algorithm set in the DPUCVD accelerator is used to generate I / Q data, and the conjugate of the generated baseband I / Q data is taken to generate a reverse interference signal.

[0043] S402, divide the reverse interference signal into multiple sub-bands according to the signal frequency domain characteristics; Specifically, the generated reverse interference signal is divided into four sub-bands according to the signal frequency domain characteristics, namely the core suppression band, the edge suppression band, the high-frequency leakage suppression band, and the low-frequency leakage suppression band. For the core suppression band, the frequency band range is 1571.34 MHz - 1579.50 MHz, and the bandwidth is 8.16 MHz. The core suppression band covers the center frequency (1575.42 MHz) of the original interference and the range of ±4 MHz, suppressing the main energy. For the edge suppression band, the frequency band range is 1563.25 MHz - 1571.34 MHz and 1579.50 MHz - 1587.59 MHz, and the bandwidth is 16.32 MHz. The edge suppression band suppresses the transition frequency band where the main energy spreads to both sides, preventing interference tails. For the high-frequency leakage suppression band, the frequency band range is 1587.59 MHz - 1595.68 MHz, and the bandwidth is 8.09 MHz. The high-frequency leakage suppression band targets the high-order harmonics and out-of-band spurs of the A1 signal. For the low-frequency leakage suppression band, the frequency band range is 1559.16 MHz - 1563.25 MHz, and the bandwidth is 4.09 MHz. The low-frequency leakage suppression band suppresses the low-frequency baseband noise, avoiding the uplift of the interference lower sideband.

[0044] S403. Obtain the powers of the core suppression band, the edge suppression band, the high-frequency leakage suppression band, and the low-frequency leakage suppression band, and calculate the reverse interference amplitudes of each sub-band according to the powers of each sub-band. Further, for the core suppression band, determine its center frequency point f c = 1575.42 MHz according to the frequency range of the core suppression band. Use a spectrum analysis device to measure the power at the center frequency point to obtain the core frequency point power, and calculate the reverse interference amplitude of the core suppression band according to the core frequency point power. The formula is: , where is the reverse interference amplitude of the core suppression band, is the core frequency point power; for the edge suppression band, according to the frequency range of the edge suppression band, this frequency range is adjacent to the low-frequency and high-frequency boundaries of the core suppression band respectively. By monitoring the average power of these two edge frequency bands, where represents the average power of the low-frequency and high-frequency bands of the edge suppression band, calculate the reverse interference amplitude of the edge suppression band according to the average power of the edge band. The formula is: , where is the reverse interference amplitude of the edge suppression band, is the average power of the edge band; for the high-frequency leakage suppression band, determine its center frequency point as the high-frequency leakage peak power according to the frequency range of the high-frequency leakage suppression band, and calculate the reverse interference amplitude of the high-frequency leakage suppression band according to the high-frequency leakage peak power: , where is the reverse interference amplitude of the high-frequency leakage suppression band, is the high-frequency leakage peak power; for the low-frequency leakage suppression band, determine its center frequency point as the low-frequency base noise power according to the frequency range of the low-frequency leakage suppression band, and calculate the reverse interference amplitude of the low-frequency leakage suppression band according to the low-frequency base noise power: , where is the reverse interference amplitude of the low-frequency leakage suppression band, is the low-frequency base noise power, and in the above formula , , are all the maximum allowable amplitudes.

[0045] S404, control the original interference according to the obtained reverse interference amplitudes of each sub-band; Specifically, for the core suppression band, the generated high-amplitude core suppression band reverse interference signal is inversely superimposed on the main lobe of the original interference signal. Inverse superposition means that the phase of the reverse interference signal is 180 degrees different from the phase of the main lobe of the original interference signal. When these two signals meet in space, due to their opposite phases, destructive interference occurs. The result of destructive interference is to form a "spectrum pit" on the spectrum of the core suppression band. This "spectrum pit" significantly reduces the power of the main lobe of the original interference signal. By precisely controlling the amplitude and phase of the reverse interference signal, the main lobe power can be forced to be reduced by 20 dB, thereby effectively suppressing the interference energy in the core suppression band and ensuring that the main lobe energy of the A1 signal is significantly suppressed. For the edge suppression band, the generated reverse interference signal is applied to the transition region of the edge suppression band. Different from the high-amplitude reverse interference in the core suppression band, the reverse interference amplitude in the edge suppression band is relatively low. The purpose is to gently suppress the interference energy in the transition region. This gentle suppression can prevent the interference energy from spreading excessively to both sides of the core suppression band, thereby maintaining the relative stability of the signal spectrum. By gently suppressing the transition region, the power attenuation of the transition band in the edge suppression band is ≥15 dB. This attenuation effect effectively reduces the distribution of interference energy in the transition region and avoids the occurrence of the interference "shoulder" lifting phenomenon, further optimizing the spectral characteristics of the signal. For the high-frequency leakage suppression band, the generated reverse interference signal is a narrowband signal targeting high-frequency harmonics. This narrowband reverse signal is superimposed on the high-frequency harmonics in the original interference signal. Since the frequency and phase of the reverse interference signal match those of the high-frequency harmonics, destructive interference occurs between them. Through the destructive interference between the narrowband reverse signal and the high-frequency harmonics, the high-frequency harmonics can be effectively eliminated, making the spurious suppression ratio ≥30 dBc. This means that the power of the spurious signals in the high-frequency leakage suppression band is reduced by at least 30 dB relative to the power of the carrier signal, thereby significantly improving the quality and purity of the signal. For the low-frequency leakage suppression band, the generated low-amplitude reverse interference signal is applied to the low-frequency leakage suppression band. Due to the low amplitude of the reverse interference signal, it can interact with the baseband noise in the original interference signal in a gentle manner. Through the interaction between the low-amplitude reverse interference and the baseband noise, the baseband noise power is reduced by ≥10 dB. This reduction in baseband noise directly improves the sensitivity of the receiver, enabling the receiver to detect weaker signals. For example, the receiver sensitivity can be improved to -130 dBm, which means that the receiver can operate normally at a lower signal power level, improving the performance and reliability of the communication system.

[0046] The technical solutions in the embodiments of the present application described above have at least the following technical effects or advantages: By actively injecting reverse interference signals, the system can achieve precise suppression and effective control of specific interference signals, improve signal quality, optimize spectral characteristics, reduce the influence of high-frequency leakage and low-frequency baseband noise on the communication system, thereby significantly enhancing the performance and reliability of the communication system. The generated reverse interference signals are divided into four sub-bands according to the signal frequency domain characteristics: the core suppression band, the edge suppression band, the high-frequency leakage suppression band, and the low-frequency leakage suppression band. Each sub-band is designed for different interference characteristics, achieving precise control of the interference signals.

[0047] The above are only the preferred embodiments of the present invention and are not intended to limit the present invention. For those skilled in the art, the present invention can have various changes and modifications. Any modifications, equivalent replacements, improvements, etc. made within the spirit and principle of the present invention shall be included within the protection scope of the present invention.

Claims

1. A multi-stage filtering amplifier suppressor, characterized in that, Including: a signal acquisition module, a spatial attenuation and calculation module, a multi-stage filtering and amplification module, and a simulation verification module; The signal acquisition module calculates the oscillator distribution pitch according to the frequency of the received signal by acquiring the frequency of the received signal and collecting interference signals, converts the oscillator distribution pitch into the pitch between the transmitting and receiving antennas, and the spatial attenuation and calculation module calculates the spatial attenuation amount through the pitch between the transmitting and receiving antennas; the multi-stage filtering and amplification module performs multi-stage filtering and amplification on the received signal according to the interference signal and the spatial attenuation amount, and simultaneously processes the insertion loss generated by the multi-stage filtering and amplification; the simulation verification module constructs a simulation model according to the results of the multi-stage filtering and amplification to verify the results of the multi-stage filtering and amplification.

2. The multi-stage filtering amplifier suppressor according to claim 1, wherein Calculate the oscillator distribution spacing according to the frequency of the received signal obtained. The formula is: , where L is the oscillator distribution spacing; f is the frequency of the received signal; C is the speed of light, with a value of 3×10 8 m / s.

3. The multi-stage filtering amplifier suppressor according to claim 2, characterized in that, Convert the unit of the oscillator distribution spacing to kilometers and use it as the spacing between the transmitting and receiving antennas. Calculate the spatial attenuation based on the spacing between the transmitting and receiving antennas. The formula is as follows: , where is the spatial attenuation; 32.45 is a constant term representing the parameter of the automatic space path loss model, that is, the power attenuation of the signal from the transmitting end to the receiving end due to the propagation distance and frequency; F is the frequency; D is the distance between the transmitting and receiving antennas.

4. A multi-stage filtering amplifier suppressor according to claim 1, characterized in that, The interference signal is subjected to multi-stage filtering and amplification through the spatial attenuation amount. First, a filter is used to perform primary filtering on the interference signal, and at the same time, the signal at the transmitting end is attenuated; then the interference signal after primary filtering is subjected to primary amplification, and the primary amplification is performed by a low-noise amplifier; subsequently, the interference signal after primary amplification is subjected to secondary filtering; then, the interference signal passing through the secondary filtering is subjected to secondary amplification, and the secondary amplification is also performed by a low-noise amplifier; finally, the interference signal after secondary amplification is subjected to tertiary filtering.

5. A multi-stage filtering amplifier suppressor according to claim 1, characterized in that, The multi-stage filtering and amplification module is used for the processing of insertion loss, specifically: S201, using a power splitter to evenly divide the received signal into multiple paths to form multiple receiving channels, calculating the noise figure within a single channel through the insertion loss of the filter, and at the same time, calculating the time delay difference; S202, the multiple receiving channels share the same local oscillator; S203, using a signal combining algorithm to calculate the insertion loss of multiple channels, and calculating the noise figure of multiple channels according to the noise figure within a single channel.

6. The multi-stage filtering amplifier suppressor according to claim 5, characterized in that, Calculate the noise figure of a single channel in multiple channels. The formula is: , where is the noise figure of the single channel; is the noise figure of the power splitter; is the noise figure of the filter; is the gain of the power splitter; is the gain of the filter; is the noise figure of the primary low-noise amplifier; is the gain of the primary low-noise amplifier; is the noise figure of the secondary low-noise amplifier. The formula for calculating the noise figure of multiple channels is: , where is the noise figure of the multiple channels, is the noise figure of the single channel, N is the number of channels, and the noise figure of the single channel is the noise figure of the first receiving channel or the second receiving channel.

7. The multistage filtering amplifier suppressor according to claim 5, wherein, The calculation formula for the insertion loss of multiple channels is as follows: , where is the equivalent insertion loss after multi-channel combination, is the insertion loss of a single channel, is the signal-to-noise ratio gain brought by MRC combination, and N is the number of channels.

8. A multi-stage filtering amplifier suppressor according to claim 1, characterized in that, The steps for the spatial attenuation and calculation module to calculate the time delay difference are: S301, dividing the signal at the transmitting end into real-time data and calibration data, and dividing the interference level according to the power and signal-to-noise ratio of the received signal at the receiving end; S302, performing time delay matching on multiple channels, and obtaining the time delay difference of multiple channels according to the time delay matching; S303, compensating multiple channels according to the obtained time delay difference.

9. The multistage filtering amplifier suppressor according to claim 8, characterized in that, The real-time data is communication data, and the transmitting end is equipped with a pseudo-random sequence generation module. The pseudo-random sequence generation module generates a pseudo-random sequence according to preset algorithms and parameters, and the pseudo-random sequence is used as calibration data.

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