Light load efficiency optimization modulation method and system based on three-phase LLC resonant converter

Abstract

The invention discloses a light load efficiency optimization modulation method and system based on a three-phase LLC resonant converter. The method comprises the steps that the voltage gain M and the output power po of the three-phase LLC resonant converter are calculated; substituting the voltage gain M and the output power po into the optimal modulation relational expression, and calculating the optimal duty ratios dp, op of the primary side and the optimal duty ratios ds, op of the secondary side; calculating the deviation between the output voltage acquisition value vo and the output voltage given values vo and ref of the three-phase LLC resonant converter, performing PI control, and superposing the control output quantity on the optimal duty ratios dp and op of the primary side so as to compensate the dead time and the error generated by the theoretical model; and calculating the phase shift angle d phi of the primary side and the secondary side so as to generate driving signals of all switching tubes. According to the method, the voltage regulation range of the three-phase LLC resonant converter can be widened under the light-load working condition, the efficiency of the converter is improved, and the method has a wide prospect in the field of high-power battery charging equipment.

Description

Technical Field

[0001] The present invention belongs to the field of power electronic resonant converter control, and relates to a light-load efficiency optimization modulation method and system based on a three-phase LLC resonant converter. Background Art

[0002] With the booming development of the new energy industry, power electronic converters are increasingly widely used. As an important type of application, battery charging equipment has become an indispensable part of products such as electric vehicles, energy storage systems, and uninterruptible power supplies. Due to advantages such as electrical isolation, a wide soft-switching range, and high efficiency, the LLC resonant converter stands out from numerous topologies and is applied to battery charging equipment. Compared with the traditional single-phase LLC resonant converter, the three-phase LLC resonant converter has smaller current ripple and filter volume, and higher efficiency at high power, so it is more suitable for high-power charging scenarios.

[0003] The light-load condition of battery charging equipment accounts for a relatively high proportion, and at the same time, a wide voltage regulation range is required. Therefore, it is crucial to improve the light-load performance of the converter. The three-phase LLC resonant converter usually uses pulse frequency modulation (PFM) to control the output voltage, but PFM has two problems under light-load conditions: 1) In the over-resonant region, affected by parasitic capacitance, the voltage gain and switching frequency are not monotonic, which limits the output voltage regulation range; 2) The high switching frequency increases the switching loss and core loss of the converter, seriously affecting the efficiency. Therefore, under light-load conditions, PFM not only affects the normal operation of the three-phase LLC resonant converter but also increases power loss.

[0004] To improve the light-load operating performance of the three-phase LLC resonant converter, its modulation strategy needs to be improved. Currently, there are mainly two types of solutions. The first type is the phase-cut control method. Utilizing the characteristics of the interleaved parallel connection of the three-phase LLC resonant converter, one or two phases are blocked under light-load conditions to reduce the switching loss and improve the efficiency. This type of solution requires mode switching, increasing the control complexity, and some methods also add additional hardware. At the same time, the phase-cut method does not solve the problem of non-monotonic voltage gain and switching frequency under light-load conditions. The second type of solution is the phase-shift control based on the primary and secondary sides. Under a wide voltage regulation range, the efficiency of this type of solution drops severely and it is difficult to be applied to battery charging equipment. Therefore, a new modulation strategy is needed to ensure a wide voltage regulation range and improve the operating efficiency under light-load conditions. Summary of the Invention

[0005] To overcome the above-mentioned disadvantages of the prior art, the present invention provides a light-load efficiency optimization modulation method and system based on a three-phase LLC resonant converter, enabling the converter to have a wide voltage regulation range and high efficiency under light-load conditions to improve the light-load performance of the converter.

[0006] To achieve the above object, the present invention provides the following technical solutions: In a first aspect, the present invention provides a light-load efficiency optimization modulation method based on a three-phase LLC resonant converter, characterized by including: S1: Calculate the voltage gain M and the output power p of the three-phase LLC resonant converter o ; S2: Input the voltage gain M and the output power p o into the optimal modulation relational expression to calculate the primary optimal duty cycle d p,op and the secondary optimal duty cycle d s,op ; S3: Calculate the deviation between the output voltage acquisition value v o and the output voltage given value v o,ref of the three-phase LLC resonant converter, and perform PI control. Superimpose the control output quantity on the primary optimal duty cycle d p,op to compensate for the dead time and the error generated by the theoretical model; S4: Calculate the phase shift angle d φ between the primary and secondary sides of the three-phase LLC resonant converter, and then generate the drive signal of the switching tube.

[0007] As a further improvement of the present invention, the three-phase LLC resonant converter includes a primary three-phase bridge H1 and a secondary three-phase bridge H2. H1 includes switching tubes S1, S2, S3, S4, S5, and S6. Among them, S1, S3, and S5 are the upper tubes of phases A, B, and C respectively, and S2, S4, and S6 are the lower tubes of phases A, B, and C respectively. The midpoints of the three-phase bridge arms of H1 are respectively connected to one end of the primary side of the three-phase resonant cavity, and the other ends of the primary side of the three-phase resonant cavity are commonly connected as a common point O1; H2 includes switching tubes Q1, Q2, Q3, Q4, Q5, and Q6. Among them, Q1, Q3, and Q5 are the upper tubes of phases A, B, and C respectively, and Q2, Q4, and Q6 are the lower tubes of phases A, B, and C respectively. The midpoints of the three-phase bridge arms of H2 are respectively connected to one end of the secondary side of the three-phase resonant cavity, and the other ends of the secondary side of the three-phase resonant cavity are commonly connected as a common point O2.

[0008] As a further improvement of the present invention, in the primary three-phase bridge H1, the drive signals of the upper and lower tubes of phases A, B, and C are complementary, the switching frequency f s is equal to the resonant frequency f r , and the duty cycle of the drive signal of the upper tube is d p , 0 < d p ≤0.5, and the phase differences of the drive signals between the three-phase bridge arms are 2π / 3 from each other; In the secondary three-phase bridge H2, the drive signals of the upper and lower tubes of phases A, B, and C are complementary, the switching frequency f s is equal to the resonant frequency fr and the duty cycle of the driving signal of the upper tube is d s , 0 < d s ≤ 0.5, and the phases of the driving signals between the three-phase bridge arms are mutually different by 2π / 3; the driving signals of the switching tubes S n and Q n at the same position on the primary and secondary sides are centrosymmetric. For n = 1 to 6, define the phase difference φ ps between the rising edges of the driving signals of the upper tubes on the primary and secondary sides as the phase shift angle between the primary and secondary sides, and its duty cycle form is d φ = φ ps / 2π. Then, the primary duty cycle d p , the secondary duty cycle d s and the phase shift angle d φ between the primary and secondary sides satisfy the following constraint relationship:

[0009] The resonant frequency f r is:

[0010] where L r is the resonant inductor of the resonant cavity, and C r is the resonant capacitor of the resonant cavity.

[0011] As a further improvement of the present invention, the theoretical voltage gain is calculated according to the duty cycle:

[0012] where d p is the primary duty cycle, and d s is the secondary duty cycle. As a further improvement of the present invention, according to the changes in the primary and secondary duty cycles, the three-phase LLC resonant converter has four types of operating modes, and the constraint conditions for each type of operating mode are as follows: Mode I:

[0013] Mode II:

[0014] Mode III:

[0015] Mode IV:

[0016] where the primary duty cycle is d p , the secondary duty cycle is d s and the phase shift angle between the primary and secondary sides is d φ .

[0017] As a further improvement of the present invention, the voltage gain M and the output power p o are calculated by collecting the voltage and current of the converter, specifically as follows:

[0018] wherein, v o is the collected value of the output voltage, i o is the collected value of the output current, v in is the collected value of the input voltage, and n is the turns ratio of the transformer.

[0019] As a further improvement of the present invention, the optimal modulation relation is obtained by means of off-line calculation, and the specific method includes: S21: Determine the optimization objective and constraint conditions of the optimal modulation; S22: According to the optimization objective of the optimal modulation, solve the optimal duty cycle under different voltage gains M and output powers p o and study the relationship and law between the optimal duty cycle and the voltage gain M and the output power p o ; S23: According to the obtained relationship law, provide the analytical expressions of the optimal duty cycle and the voltage gain M and the output power p o by means of fitting or look-up table.

[0020] As a further improvement of the present invention, the optimization objective of the optimal modulation is that the effective value of the resonant current of the converter is minimized under the given voltage gain M and output power p o ; the constraint condition of the optimal modulation is that all the switching tubes of the converter achieve zero-voltage turn-on; the judgment condition for the zero-voltage turn-on is that the junction capacitance of the switching tube is fully charged and discharged within the dead time, that is, the turn-on current of the switching tube satisfies the minimum current required for ZVS. The expressions of the optimization objective and the constraint conditions are as follows:

[0021] wherein i rms is the effective value of the primary side current of the resonant cavity, f rms represents its relationship with the secondary side duty cycle d s , the voltage gain M and the output power p o , i 0S1 is the turn-on current of S1, i 0S2 is the turn-on current of S2, i 0Q1 is the turn-on current of Q1, i 0Q2 is the turn-on current of Q2, C oss is the junction capacitance of the switching tube, and t d is the dead time.

[0022] As a further improvement of the present invention, as the voltage gain changes, the operating modes that appear within the full power range are based on the relationship between d s,op and p o There are a total of four paradigms. Paradigm 1 includes operating modes II and III. Paradigm 2 includes operating modes I, II, and III. Paradigm 3 includes operating modes I and II. Paradigm 4 includes operating mode I. As the voltage gain increases, the relationship between d s,op and p o gradually transitions from Paradigm 1 to Paradigm 4. The boundary gains between different paradigms are solved by the following constraint conditions:

[0023] where P o,r is the rated power, P o,l is the minimum power at light load, M b1 is the boundary gain between Paradigm 1 and 2, M b2 is the boundary gain between Paradigm 2 and 3, M b3 is the boundary gain between Paradigm 3 and 4; The optimal modulation expressions for the four paradigms are obtained separately by the method of piecewise linear fitting. The total expression is as follows:

[0024] where g1, g2, g3, and g4 are the fitting functions corresponding to Paradigm 1 to 4 respectively, and the four paradigms are fitted as follows: Paradigm 1 includes operating modes II and III. Among them, mode II is linearly fitted with one segment, and mode III is non-linear and requires two segments of linear fitting at the beginning and end. The fitting expressions are as follows:

[0025] where d s0 , d s,l , k II , k III1 , k III2 , d s,t1 and p o,t1 are coefficients to be solved. d s0 is the optimal duty cycle at the rated power, d s,l is the optimal duty cycle at the minimum power, k II is the slope of the linear fitting of mode II, k III1 and k III2 are the slopes of the two segments of linear fitting of mode III respectively. All coefficients to be solved are only related to the voltage gain M; Paradigm 2 includes operating modes I, II, and III. Modes II and III are linearly fitted with one segment respectively, and mode I is non-linear and requires two segments of linear fitting at the beginning and end. The fitting expressions are as follows:

[0026] Among them, d s0 , k I1 , k I2 , k II , k III , d s,t1 , p o,t1 , d s,t2 and p o,t2 are coefficients to be solved, and k I1 and k I2 are the slopes of the two-segment linear fitting in Mode I respectively. All coefficients to be solved are only related to the voltage gain M; Normal form 3 includes Modes I and II. Mode II is linearly fitted with one segment, and Mode I is non-linear and needs to be linearly fitted with two segments at the beginning and the end. The fitting expressions are as follows:

[0027] Among them, d s0 , k I1 , k I2 , k II , d s,t2 and p o,t2 are coefficients to be solved, and they are only related to the voltage gain M; Normal form 4 includes Mode I. Mode I is non-linear and needs to be linearly fitted with two segments at the beginning and the end. The fitting expressions are as follows:

[0028] Among them, d s0 , d s,l , k I1 and k I2 are coefficients to be solved, and they are only related to the voltage gain M; All coefficients to be solved within the above four normal forms are fitted using a cubic polynomial:

[0029] Among them, x represents the coefficient to be solved, and c0, c1, c2, and c3 represent the coefficients of the fitting polynomial.

[0030] In a second aspect, the present invention provides a light-load efficiency optimization modulation system based on a three-phase LLC resonant converter, including: A parameter calculation module that calculates the voltage gain M and output power p of the converter o ; An optimal modulation calculation module that obtains the relationship between the optimal duty cycle and the voltage gain M and output power p through offline calculation, and uses the voltage gain M and output power p calculated by the parameter calculation module o of the above.o Substitute into the optimal modulation relation formula to calculate the optimal duty cycle d p,op and d s,op ; Error compensation module, calculate the sampled value v of the output voltage of the converter o and the given value v of the output voltage o,ref between the deviations, perform PI control on the deviation value, and superimpose the control quantity on the original side optimal duty cycle d p,op to compensate for the errors caused by the dead time and the theoretical model; Drive signal generation module, calculate the phase shift angles d of the primary and secondary sides according to the duty cycles of the primary and secondary sides φ and generate drive signals for the switching tubes S1 - S6 and Q1 - Q6.

[0031] Compared with the prior art, the present invention has the following advantages: The present invention proposes a light - load optimization modulation method for a three - phase LLC resonant converter. The duty cycles of both the primary and secondary sides of this modulation method are freely adjustable, broadening the voltage regulation range under light - load conditions; further, the modulation strategy is optimized and designed. On the basis of ensuring a wide voltage regulation range, the light - load efficiency of the converter is improved by optimizing the effective value of the current. The three - phase LLC resonant converter based on this light - load optimization modulation strategy can be applied to battery charging equipment. The improvement of efficiency can reduce the power loss of the equipment, lower the usage cost, and improve the economic benefits. This method can broaden the voltage regulation range of the three - phase LLC resonant converter under light - load conditions and improve the efficiency of the converter, and has broad prospects in the field of high - power battery charging equipment. Brief Description of the Drawings

[0032] In order to more clearly illustrate the technical solutions in the embodiments of the present invention or the prior art, the following will briefly introduce the drawings required for use in the description of the embodiments or the prior art. Obviously, the drawings in the following description are some embodiments of the present invention. For those of ordinary skill in the art, without creative efforts, other drawings can also be obtained based on these drawings.

[0033] Figure 1 is a schematic diagram of the light - load efficiency optimization modulation method based on a three - phase LLC resonant converter of the present invention; Figure 2 is a topology diagram of the three - phase LLC resonant converter of the present invention; Figure 3 is a basic working principle diagram of the modulation strategy of the present invention; Figure 4 is a schematic diagram of four working modes of the modulation strategy of the present invention; Figure 5 is a relationship diagram of the effective value of the resonant current and the duty cycle under a fixed voltage gain of the present invention; Figure 6 are four relational paradigm diagrams of the optimal duty cycle and output power of the present invention; Figure 7 are four relational paradigm fitting diagrams of the optimal duty cycle and output power of the present invention; Figure 8 is the soft-switching waveform diagram of the embodiment of the present invention; Figure 9 is the load-shedding waveform diagram of the embodiment of the present invention; Figure 10 is the comparison diagram of efficiency improvement of the present invention. Detailed implementation manners

[0034] In order to enable those skilled in the art to better understand the solution of the present invention, the technical solution of the present invention will be clearly and completely described below in conjunction with the accompanying drawings and specific embodiments. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without making creative efforts belong to the scope of protection of the present invention.

[0035] Refer to Figure 1 , the present invention provides a light-load efficiency optimization modulation method based on a three-phase LLC resonant converter, including the following steps: calculating the voltage gain M and output power p of the three-phase LLC resonant converter o ; substituting the voltage gain M and output power p o into the optimal modulation relational formula to calculate the primary optimal duty cycle d p,op and the secondary optimal duty cycle d s,op ; calculating the deviation between the output voltage acquisition value v o of the three-phase LLC resonant converter and the output voltage given value v o,ref , and performing PI control, and superimposing the control output quantity on the primary optimal duty cycle d p,op to compensate for the error caused by the dead time and the theoretical model; calculating the primary-secondary phase-shift angle d φ , and then generating the drive signals of all switching tubes.

[0036] Its basic working principle is described as follows: Such as Figure 2As shown in the figure, the three-phase LLC resonant converter includes a primary three-phase bridge H1 and a secondary three-phase bridge H2. The primary three-phase bridge H1 includes switching tubes S1, S2, S3, S4, S5, and S6. Among them, S1, S3, and S5 are the upper tubes of phases A, B, and C respectively, and S2, S4, and S6 are the lower tubes of phases A, B, and C respectively. The midpoints of the three-phase bridge arms of the primary three-phase bridge H1 are respectively connected to one end of the primary side of the three-phase resonant cavity, and the other ends of the primary side of the three-phase resonant cavity are commonly connected to a common point O1. The secondary three-phase bridge H2 includes switching tubes Q1, Q2, Q3, Q4, Q5, and Q6. Among them, Q1, Q3, and Q5 are the upper tubes of phases A, B, and C respectively, and Q2, Q4, and Q6 are the lower tubes of phases A, B, and C respectively. The midpoints of the three-phase bridge arms of H2 are respectively connected to one end of the secondary side of the three-phase resonant cavity, and the other ends of the secondary side of the three-phase resonant cavity are commonly connected to a common point O2.

[0037] Among them, the parameters of the three-phase resonant cavity are the same, including a resonant inductor L r , a resonant capacitor C r and a transformer with a turns ratio of n:1. The magnetizing inductor of the transformer is L m . v 1x and v 2x (x = a, b, c) are the primary side voltage and secondary side voltage of the resonant cavity respectively, i 1x and i 2x (x = a, b, c) are the primary side current and secondary side current of the resonant cavity respectively, v in and v o respectively represent the input voltage and the output voltage, and i o represents the output current.

[0038] Figure 3 The basic working waveforms of this modulation strategy are provided. The driving signals of the upper and lower tubes of phases A, B, and C of the primary three-phase bridge H1 are complementary. The switching frequency f s is equal to the resonant frequency f r , and the duty cycle of the driving signal of the upper tube is d p , 0 < d p ≤0.5. The phase difference between the driving signals of the three-phase bridge arms is T s / 3, and T s is the switching period. The driving signals of the upper and lower tubes of phases A, B, and C of the secondary three-phase bridge H2 are complementary. The switching frequency f s is equal to the resonant frequency f r , and the duty cycle of the driving signal of the upper tube is d s , 0 < d s ≤0.5. The phase difference between the driving signals of the three-phase bridge arms is T s / 3. The switching tubes S n and Q nThe driving signals of (n = 1 to 6) are centrosymmetric, and the phase difference φ between the rising edges of the driving signals of the primary and secondary upper switches is defined ps as the phase shift angle between the primary and secondary sides, and its duty cycle form is d φ = φ ps / 2π, then the primary duty cycle d p , the secondary duty cycle d s and the phase shift angle d φ between the primary and secondary sides satisfy the following constraint relations:

[0039] The resonant frequency f r is:

[0040] According to the changes in the primary and secondary duty cycles, the operating modes of the three-phase LLC resonant converter can be divided into four categories, and the constraint conditions corresponding to the four operating modes are shown in Table 1: Table 1 Operating Modes and Constraint Conditions

[0041] Figure 4 The theoretical operating waveforms of the four operating modes are provided, where S 1x (x = a, b, c) is the switching function of the primary three-phase bridge arm, 1 indicates that the upper switch (S1 / S3 / S5) is turned on, and 0 indicates that the lower switch (S2 / S4 / S6) is turned on; S 2x (x = a, b, c) is the switching function of the secondary three-phase bridge arm, 1 indicates that the upper switch (Q1 / Q3 / Q5) is turned on, and 0 indicates that the lower switch (Q2 / Q4 / Q6) is turned on; the main difference between the four operating modes lies in the different resonant cavity voltages of the primary and secondary sides, corresponding to Figure 4 (a), (b), (c), (d) in, respectively.

[0042] Furthermore, the theoretical voltage gain can be calculated according to the duty cycle:

[0043] As Figure 1 shown, the light load efficiency optimization modulation method based on a three-phase LLC resonant converter provided by the present invention includes the following steps: Step 1: Calculate the voltage gain M and output power p of the three-phase LLC resonant converter by collecting the voltage and current of the converter o :

[0044] Step 2: Substitute the voltage gain M and output power p o into the optimal modulation relation formula to calculate the optimal primary duty cycle dp,op and the optimal duty cycle d of the secondary side s,op , the optimal modulation relationship is obtained by off-line calculation, and the specific steps are as follows: 21) Determine the optimization objective and constraint conditions of the optimal modulation. The optimization objective of the optimal modulation is that under the given voltage gain M and output power p o , the effective value of the resonant current of the converter is minimized; the constraint condition of the optimal modulation is that all the switching tubes of the converter can achieve zero-voltage switching (ZVS). The judgment condition of ZVS is that the junction capacitance of the switching tube can be fully charged and discharged within the dead time, that is, the on-current of the switching tube meets the minimum current required for ZVS. The expressions of the optimization objective and constraint conditions are as follows:

[0045] where i rms is the effective value of the primary-side current of the resonant cavity, and f rms represents its relationship with the secondary-side duty cycle d s , voltage gain M and output power p o , i 0S1 is the on-current of S1, i 0S2 is the on-current of S2, i 0Q1 is the on-current of Q1, i 0Q2 is the on-current of Q2, C oss is the junction capacitance of the switching tube, and t d is the dead time.

[0046] Figure 5 provides the relationship between the effective value of the primary-side current of the resonant cavity and the secondary-side duty cycle when the voltage gain M = 0.8. On the premise of realizing ZVS, the closer the duty cycle is to the ZVS boundary, the smaller the effective value of the current. Therefore, it can be concluded that the ZVS boundary of the converter is the optimal solution under the constraint conditions:

[0047] Since it is the most difficult to realize ZVS for the upper tube on the primary side, the ZVS boundary of S1 is used as the ZVS boundary of the converter. The on-current i 0S1 of S1 can be calculated according to d s , M, p o . The minimum current required to realize ZVS can be calculated from the dead time t d and the junction capacitance C oss of the switching tube. k b is the ZVS margin coefficient.

[0048] Furthermore, the optimal duty cycle d o under the given voltage gain M and output power p p,op and ds,op :

[0049] Among them, g represents d s,op as a function of the voltage gain M and the output power p o which has only numerical solutions and no analytical expressions.

[0050] 22) Study the relationship between the optimal duty cycle and the voltage gain M and the output power p o of Figure 6 provides the relationship between the optimal duty cycle d s,op and the output power p o at different voltage gains. At any fixed voltage gain, d s,op decreases as p o decreases and goes through different operating modes in turn. The relationship between d s,op and p o is different in different operating modes. In particular, the relationship between the optimal duty cycle and the output power in Mode III and Mode IV is similar and can be combined into Mode III; the boundary points between the operating modes are called turning points, and their classification and constraint conditions are shown in Table 2: Table 2 Turning points and their constraint conditions

[0051] In Table 2, d p,t1 and d s,t1 are the primary and secondary optimal duty cycles at turning point 1 respectively, p o,t1 is the output power at turning point 1 and can be obtained by solving according to the voltage gain M; d p,t2 and d s,t2 are the primary and secondary optimal duty cycles at turning point 2 respectively, p o,t2 is the output power at turning point 2 and can be obtained by solving according to the voltage gain M.

[0052] Refer to Figure 6 , as the voltage gain changes, the operating modes that appear in the full power range are different, and the relationship between d s,op and p o altogether has four paradigms, corresponding to Figure 6 (a), (b), (c), (d) in s,op respectively. Paradigm 1 includes operating modes II and III, paradigm 2 includes operating modes I, II, and III, paradigm 3 includes operating modes I and II, and paradigm 4 includes operating mode I. As the voltage gain increases, the relationship between d o and p

[0053] Among them, P o,r is the rated power, and P o,l is the minimum power under light load, M b1 is the boundary gain between paradigms 1 and 2, and M b2 is the boundary gain between paradigms 2 and 3, and M b3 is the boundary gain between paradigms 3 and 4.

[0054] 23) Refer to Figure 7 , and the optimal modulation expressions for the four paradigms can be obtained respectively by using the piecewise linear fitting method. The total expression is as follows:

[0055] Among them, g1, g2, g3, and g4 are the fitting functions corresponding to paradigms 1 to 4 respectively, and the fitting for the four paradigms is as follows: Refer to Figure 7 in (a). Paradigm 1 includes working modes II and III. Among them, mode II can be linearly fitted with one segment, and mode III is non-linear and requires two linear fittings at the beginning and end. The fitting expressions are as follows:

[0056] Among them, d s0 , d s,l , k II , k III1 , k III2 , d s,t1 and p o,t1 are coefficients to be solved. d s0 is the optimal duty cycle at the rated power, d s,l is the optimal duty cycle at the minimum power, k II is the slope of the linear fitting of mode II, and k III1 and k III2 are the slopes of the two linear fittings of mode III respectively. All coefficients to be solved are only related to the voltage gain M.

[0057] Refer to Figure 7 in (b). Paradigm 2 includes working modes I, II, and III. Modes II and III can be linearly fitted with one segment respectively, and mode I is non-linear and requires two linear fittings at the beginning and end. The fitting expressions are as follows:

[0058] Among them, d s0 , k I1 , k I2 , k II , k III , d s,t1 , po,t1 and d s,t2 and p o,t2 are coefficients to be solved, and k I1 and k I2 are the slopes of the two - segment linear fitting in Mode I respectively. All coefficients to be solved are only related to the voltage gain M.

[0059] Refer to Figure 7 in (c). Normal form 3 includes operating modes I and II. Mode II can be linearly fitted with one segment, and Mode I is non - linear and needs to be linearly fitted with two segments at the beginning and the end. The fitting expressions are as follows:

[0060] Among them, d s0 , k I1 , k I2 , k II , d s,t2 and p o,t2 are coefficients to be solved and are only related to the voltage gain M.

[0061] Refer to Figure 7 in (d). Normal form 4 includes operating mode I. Mode I is non - linear and needs to be linearly fitted with two segments at the beginning and the end. The fitting expressions are as follows:

[0062] Among them, d s0 , d s,l , k I1 and k I2 are coefficients to be solved and are only related to the voltage gain M.

[0063] All coefficients to be solved within the above four normal forms can be fitted using a cubic polynomial:

[0064] where x represents the coefficient to be solved, and c0, c1, c2, and c3 represent the coefficients of the fitting polynomial.

[0065] Step 3: Calculate the deviation between the measured value v o of the output voltage of the three - phase LLC resonant converter and the given value v o,ref of the output voltage, and perform PI control on the deviation value. Superimpose the control output on the original - side optimal duty cycle d p,op to compensate for the dead - time and the error generated by the theoretical model; Step 4: Calculate the phase - shift angle d φ between the primary and secondary sides according to the constraint conditions of the modulation strategy, and generate the switching - tube drive signals v S1 -v S6 and v Q1 -vQ6 。

[0066] The light-load efficiency optimization modulation system based on a three-phase LLC resonant converter disclosed by the present invention further includes: A parameter calculation module that calculates the voltage gain M and the output power p by collecting the voltage and current of the converter o ; An optimal modulation calculation module that obtains the relationship between the optimal duty cycle and the voltage gain M and the output power p through off-line calculation, substitutes the voltage gain M and the output power p calculated by the parameter calculation module into the optimal modulation relationship, and calculates the optimal duty cycle d o and d o ; p,op and d s,op ; An error compensation module that calculates the deviation between the collected value v of the output voltage of the converter and the given value v of the output voltage, performs PI control on the deviation value, and superimposes the control quantity on the primary optimal duty cycle d o to compensate for the errors caused by the dead time and the theoretical model; o,ref A drive signal generation module that calculates the phase shift angles d p,op of the primary and secondary sides according to the duty cycles of the primary and secondary sides, and generates the drive signals v -v φ and v S1 -v S6 and v Q1 -v Q6 ;

[0067] In order to make the objectives and technical solutions of the present invention clearer and easier to understand, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. The specific embodiments described herein are only used to explain the present invention and are not used to limit the present invention.

[0068] As Figure 1 shown in the light-load efficiency optimization modulation method and system based on a three-phase LLC resonant converter, in combination with the embodiments, the specific parameters of the converter are given in Table 3. The input voltage of the converter is fixed at 400V, the output voltage is 267V - 400V, and the variation range is 1.5 times. The transformer turns ratio is set to 1, so when the output voltage is 267V, it corresponds to the maximum gain adjustment range. The rated power of the converter is 4.5kW, and the minimum power is 500W, which is 11% of the rated power. The switching tube is selected as a silicon carbide MOSFET, and the model is CI60N120SM.

[0069] Table 3 System parameters

[0070] Based on the above parameters, the optimal duty cycle d s,op and the output power po Boundary gain of the four normal forms of the relationship:

[0071] The fitting results of all coefficients to be solved within the four normal forms are shown in Table 4: Table 4 Fitting results of the coefficients to be solved in the fitting expression

[0072] Based on all the above parameters, PLECS is used for simulation to verify the feasibility of this modulation strategy. The simulation results are as shown in Figure 8 、 Figure 9 and Figure 10 .

[0073] Figure 8 shows the ZVS situation of the three-phase LLC resonant converter under this modulation strategy. From top to bottom are the switching signals of the primary switches S1 and S2 under steady state, the primary resonant current i 1a , the switching signals of the secondary switches Q1 and Q2, and the secondary resonant current i 2a . Figure 8 In (a), the converter operates at the lowest output voltage and the lowest power, Figure 8 in (b), the converter operates at the lowest output voltage and the rated power. By observation, the current at the opening moment of all switches meets the ZVS requirements. Therefore, all switches can achieve ZVS, proving that the constraint conditions of the optimal modulation strategy are satisfied and the converter can achieve ZVS.

[0074] Figure 9 shows the load-shedding dynamic process of the three-phase LLC resonant converter. From top to bottom in the figure are the output voltage v o , the output current i o and the waveforms of the primary resonant current i 1a . Before 2 ms, the converter operates under the half-load condition, and the output voltage is 271 V. At t = 2.5 ms, the load is switched from half-load to full-load, and the converter operates under the full-load condition after 2.5 ms, with the output voltage being 271 V. The entire dynamic response is smooth and rapid, and the working state of the converter is always stable, proving the effectiveness of the proposed control strategy.

[0075] Figure 10 Compares the converter operating efficiency of this modulation strategy with the single-degree-of-freedom strategy and PFM. Figure 10 In (a), the efficiency of different strategies is provided when the converter output voltage is 267 V, Figure 10 in (b), the efficiency of different strategies is provided when the converter output voltage is 320 V. PFM loses the voltage regulation ability under light load, while the proposed modulation strategy can ensure the wide voltage range regulation ability under light load.

[0076] Furthermore, this modulation strategy has the highest efficiency across the entire load range, and the efficiency at the lowest load is greater than 92%.

[0077] The above results demonstrate that the proposed modulation strategy has a wide voltage regulation range under light load, and can improve the light-load efficiency while ensuring the operating efficiency across the entire range.

[0078] The above embodiments are only used to illustrate the technical solutions of the present invention and are not intended to limit it. Although the present invention has been described in detail with reference to the above embodiments, those of ordinary skill in the art can still modify or equivalently replace the specific implementation manners of the present invention. Any modification or equivalent replacement that does not depart from the spirit and scope of the present invention shall fall within the protection scope of the claims of the present invention.

Claims

1. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter, characterized in that, including: S1: Calculate the voltage gain M and output power p of the three-phase LLC resonant converter o ; S2: Input the voltage gain M and the output power p o into the optimal modulation relationship formula to calculate the primary optimal duty cycle d p,op and the secondary optimal duty cycle d s,op ; S3: Calculate the sampled value v of the output voltage of the three-phase LLC resonant converter o and the given value v o,ref of the output voltage, perform PI control, and superimpose the control output on the optimal duty cycle d p,op on the primary side to compensate for the dead time and the error generated by the theoretical model; S4: Calculate the primary-secondary phase shift angle d of the three-phase LLC resonant converter φ , and then generate the driving signals of the switching tubes.

2. The light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 1, wherein The three-phase LLC resonant converter includes a primary three-phase bridge H1 and a secondary three-phase bridge H2. H1 includes switching tubes S1, S2, S3, S4, S5, and S6. Among them, S1, S3, and S5 are the upper tubes of phases A, B, and C respectively, and S2, S4, and S6 are the lower tubes of phases A, B, and C respectively. The midpoints of the three-phase bridge arms of H1 are respectively connected to one end of the primary side of the three-phase resonant cavity, and the other ends of the primary side of the three-phase resonant cavity are commonly connected to a common point O1; H2 includes switching tubes Q1, Q2, Q3, Q4, Q5, and Q6. Among them, Q1, Q3, and Q5 are the upper tubes of phases A, B, and C respectively, and Q2, Q4, and Q6 are the lower tubes of phases A, B, and C respectively. The midpoints of the three-phase bridge arms of H2 are respectively connected to one end of the secondary side of the three-phase resonant cavity, and the other ends of the secondary side of the three-phase resonant cavity are commonly connected to a common point O2.

3. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 2, characterized in that, In the original three-phase bridge H1, the driving signals of the upper and lower switches of phases A, B, and C are complementary, and the switching frequency f s is equal to the resonance frequency f r , and the duty cycle of the driving signal of the upper switch is d p , 0 < d p ≤ 0.5, and the phases of the driving signals between the three-phase bridge arms are mutually different by 2π / 3; In the secondary three-phase bridge H2, the driving signals of the upper and lower switches of phases A, B, and C are complementary, and the switching frequency f s is equal to the resonance frequency f r , and the duty cycle of the driving signal of the upper switch is d s , 0 < d s ≤ 0.5, and the phases of the driving signals between the three-phase bridge arms are mutually different by 2π / 3; the driving signals of the switching tubes S n and Q n at the same position on the primary and secondary sides are centrosymmetric, n = 1 to 6, and the phase difference φ ps between the rising edges of the driving signals of the upper switches on the primary and secondary sides is defined as the phase shift angle between the primary and secondary sides, and its duty cycle form is d φ = φ ps / 2π, then the primary duty cycle d p , the secondary duty cycle d s and the phase shift angle d φ between the primary and secondary sides satisfy the following constraint relationship: The resonant frequency f r is as follows: Among them, L r is the resonant inductor of the resonant cavity, and C r is the resonant capacitor of the resonant cavity.

4. A light-load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 3, characterized in that The theoretical voltage gain can be calculated according to the duty cycle: Among them, d p is the duty cycle of the primary side, and d s is the duty cycle of the secondary side.

5. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 3, characterized in that According to the changes in the primary and secondary duty cycles, the three-phase LLC resonant converter has four types of operating modes, and the constraint conditions for each type of operating mode are as follows: Mode I: Mode II: Mode III: Mode IV: Among them, the duty cycle of the primary side is d p , the duty cycle of the secondary side is d s and the phase shift angle between the primary and secondary sides is d φ .

6. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 1, characterized in that The voltage gain M and the output power p o are calculated by collecting the voltage and current of the converter, specifically as follows: Among them, v o is the acquired value of the output voltage, i o is the acquired value of the output current, v in is the acquired value of the input voltage, and n is the turns ratio of the transformer.

7. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 1, characterized in that The optimal modulation relation is obtained by an off-line calculation method. The specific method includes: S21: Determine the optimization objective and constraint conditions of the optimal modulation; S22: Solve for the optimal duty cycle under different voltage gains M and output powers p according to the optimization objective of the optimal modulation, and study the relationship and pattern between the optimal duty cycle, voltage gain M, and output power p; o o ​ S23: According to the obtained relationship law, use the method of fitting or lookup table to provide the analytical expressions of the optimal duty cycle, voltage gain M, and output power p. o ​ 8. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 7, characterized in that The optimization objective of the optimal modulation is to minimize the effective value of the resonant current of the converter under the given voltage gain M and output power p o The constraint condition of the optimal modulation is that all the switching tubes of the converter achieve zero-voltage turn-on; The judgment condition for zero-voltage turn-on is that the switching tube junction capacitance is fully charged and discharged within the dead time, that is, the switching tube turn-on current meets the minimum current required for ZVS. The expressions of the optimization objective and constraint conditions are as follows: where i rms is the effective value of the primary side current of the resonant cavity, and f rms represents its relationship with the duty cycle d s , voltage gain M, and output power p o . i 0S1 is the turn-on current of S1, i 0S2 is the turn-on current of S2, i 0Q1 is the turn-on current of Q1, i 0Q2 is the turn-on current of Q2, and C oss is the junction capacitance of the switching tube, and t d is the dead time.

9. A light load efficiency optimization modulation method based on a three-phase LLC resonant converter according to claim 7, characterized in that As the voltage gain changes, the operating modes that occur within the full power range are based on the relationship between d s,op and p o There are a total of four paradigms. Paradigm 1 includes operating modes II and III. Paradigm 2 includes operating modes I, II, and III. Paradigm 3 includes operating modes I and II. Paradigm 4 includes operating mode I. As the voltage gain increases, the relationship between d s,op and p o gradually transitions from Paradigm 1 to Paradigm 4. The boundary gains between different paradigms are solved by the following constraint conditions: Among them, P o,r is the rated power, P o,l is the minimum light load power, M b1 is the boundary gain of paradigms 1 and 2, M b2 is the boundary gain of paradigms 2 and 3, M b3 is the boundary gain of paradigms 3 and 4; The optimal modulation expressions of the four paradigms are respectively obtained by the piecewise linear fitting method, and the total expression is as follows: Among them, g1, g2, g3, and g4 are the fitting functions corresponding to paradigms 1 to 4 respectively, and the four paradigms are respectively fitted as follows: Paradigm 1 includes operating modes II and III. Among them, mode II is linearly fitted with one segment, and mode III is non-linear and requires two segments of linear fitting at the head and tail. The fitting expression is as follows: where d s0 , d s,l , k II , k III1 , k III2 , d s,t1 and p o,t1 are coefficients to be solved, d s0 is the optimal duty cycle at the rated power, d s,l is the optimal duty cycle at the minimum power, k II is the slope of the linear fitting in Mode II, k III1 and k III2 are the slopes of the two - segment linear fittings in Mode III respectively. All coefficients to be solved are only related to the voltage gain M; Paradigm 2 includes operating modes I, II, and III. Modes II and III are linearly fitted with one segment respectively, and mode I is non-linear and requires two segments of linear fitting at the head and tail. The fitting expression is as follows: Among them, d s0 , k I1 , k I2 , k II , k III , d s,t1 , p o,t1 , d s,t2 and p o,t2 are coefficients to be solved. k I1 and k I2 are the slopes of the two linear fittings in Mode I respectively. All coefficients to be solved are only related to the voltage gain M; Paradigm 3 includes operating modes I and II. Mode II is linearly fitted with one segment, and mode I is non-linear and requires two segments of linear fitting at the head and tail. The fitting expression is as follows: where d s0 , k I1 , k I2 , k II , d s,t2 and p o,t2 are coefficients to be solved, and are all only related to the voltage gain M; Paradigm 4 includes operating mode I. Mode I is non-linear and requires two segments of linear fitting at the head and tail. The fitting expression is as follows: Among them, d s0 , d s,l , k I1 and k I2 are coefficients to be solved, and all are only related to the voltage gain M; All the coefficients to be solved within the above four paradigms are fitted using a cubic polynomial: Where x represents the coefficient to be solved, and c0, c1, c2, and c3 represent the coefficients of the fitting polynomial.

10. A light-load efficiency optimization modulation system based on a three-phase LLC resonant converter, characterized in that, including: Parameter calculation module, calculating the voltage gain M and output power p of the converter o ; The optimal modulation calculation module obtains the relationship between the optimal duty cycle, voltage gain M, and output power p through offline calculation, and substitutes the voltage gain M and output power p calculated by the parameter calculation module into the optimal modulation relationship to calculate the optimal duty cycle d o and calculates the optimal duty cycle d o by substituting the voltage gain M and output power p calculated by the parameter calculation module into the optimal modulation relationship p,op and d s,op ; The error compensation module calculates the sampled value v of the output voltage of the converter o and the given value v of the output voltage o,ref to obtain the deviation therebetween, performs PI control on the deviation value, and superimposes the control quantity on the optimal duty ratio d of the primary side p,op to compensate for the errors caused by the dead time and the theoretical model; The driving signal generation module calculates the primary and secondary phase-shift angles d according to the primary and secondary duty ratios φ and generates the driving signals for the switching tubes S1 - S6 and Q1 - Q6.

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