Directional coupling processing device and radio frequency power synthesis device

Abstract

The invention discloses a directional coupling processing device and a radio frequency power synthesis device. The device comprises a first bi-directional coupler, a second bi-directional coupler, a first vector adder and a second vector adder, wherein the input end of the first bi-directional coupler and the input end of the second bi-directional coupler are respectively used for receiving one of two paths of same-amplitude orthogonal radio frequency signals; the straight-through ends of the first bi-directional coupler and the second bi-directional coupler respectively output respective radio-frequency signals and respectively receive reflected signals of the radio-frequency signals, and the two paths of reflected signals are orthogonal signals with the same amplitude; output of coupling ends of the first bi-directional coupler and the second bi-directional coupler is added by a first vector adder and then is output; the output of the isolation end of the first bi-directional coupler and the output of the isolation end of the second bi-directional coupler are added through the second vector adder and then output. The device is used for radio frequency signal power and standing wave detection, the requirement for the power capacity of a single coupler can be effectively lowered, and the stability and reliability of a transmitter can be improved.

Description

Technical Field

[0001] The present invention relates to the field of communication technology, and in particular to a directional coupling processing device and a radio frequency power synthesis device comprising the directional coupling processing device. Background Art

[0002] In RF broadband high-power transmitter systems, it is necessary to detect output power and standing waves to effectively control system output power and accurately evaluate system performance.

[0003] Currently, the main equipment for standing wave detection is the directional coupler. Its working principle is to sample the signal amplitude on the RF channel and then output it. Generally, a dual directional coupler is used to couple and sample both the forward output power and the reflected power. In theory, the coupling end only samples the forward signal, and the signal amplitude output by this port represents the forward output power; the isolation end only samples the reverse signal, and the signal amplitude output by this port represents the reverse (reflected) power. Substituting both into the standing wave ratio formula can obtain the standing wave ratio. Because the coupler is affected by factors such as the actual material uniformity, processing accuracy, circuit form, spatial radiation, and grounding effect, the actual directionality cannot be infinite. At this time, the coupled signal (including forward coupling and reverse coupling) contains information on both forward and reverse power, of which the reflection coefficient Γ and phase difference are the most affected. Their influence will vary with the standing wave load and operating frequency, resulting in uncertainty in the detection of the coupled signal. Because these sampled coupled signals contain unexpected values ​​(interference information), they ultimately lead to errors in power control and standing wave detection.

[0004] For example, when a 20dB directional coupler is used to measure a component with a 15dB return loss, the measurement uncertainty range is -4 to +7dB, meaning the possible measured value is between 11.1 and 22dB, depending on the phase of the forward and reflected signals in the profile at the detection point. If the coupler's directivity is low, or if it results in significant ripple within the coupling band when operating in a system with large standing waves, this can cause significant differences in the detection voltage at different frequencies. This can significantly affect the performance and reliability of the amplifier itself, resulting in significant differences in the power output of the amplifier at different frequencies within the operating band. Summary of the Invention

[0005] In view of this, embodiments of the present invention provide a directional coupling processing device and a radio frequency power synthesis device including the directional coupling processing device, aiming to improve the accuracy of radio frequency coupling detection and solve the problems existing in the background technology.

[0006] To achieve the above object, according to a first aspect of an embodiment of the present invention, a directional coupling processing device is provided, comprising a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein,

[0007] The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude;

[0008] The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal;

[0009] The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the first vector adder;

[0010] The isolation end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by the second vector adder and then output.

[0011] To achieve the above object, according to a second aspect of an embodiment of the present invention, there is provided a radio frequency power combining device, comprising a directional coupling processing device and a two-way orthogonal power combiner;

[0012] The directional coupling processing device includes a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein,

[0013] The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude;

[0014] The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal;

[0015] The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by a first vector adder;

[0016] The isolation end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by a second vector adder and then output;

[0017] The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler are connected to the input end of the two-way orthogonal power combiner.

[0018] According to a third aspect of the embodiments of the present invention, the present invention further provides a radio frequency power combining device, comprising a directional coupling processing device and a two-way co-directional power combiner;

[0019] The directional coupling processing device includes an orthogonal phase shift network, a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein,

[0020] The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude;

[0021] The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal;

[0022] The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the first vector adder;

[0023] The isolation end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the second vector adder;

[0024] The through ends of the first dual directional coupler and the second dual directional coupler are connected to the orthogonal phase shift network;

[0025] The orthogonal phase shift network performs orthogonal phase shift processing on the two RF signals at the through end of the first dual directional coupler and the through end of the second dual directional coupler, and inputs the obtained two RF signals with the same amplitude and phase into the two-way in-phase power combiner.

[0026] According to a fourth aspect of the embodiments of the present invention, the present invention further provides a radio frequency power combining device, comprising a directional coupling processing device and a two-way co-directional power combiner;

[0027] The directional coupling processing device includes a first orthogonal phase shift network, a second orthogonal phase shift network, a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein,

[0028] The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude;

[0029] The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal;

[0030] The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the first vector adder;

[0031] The isolation end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the second vector adder;

[0032] The input ends of the first dual directional coupler and the second dual directional coupler are connected to a first quadrature phase shift network, and the through ends of the first dual directional coupler and the second dual directional coupler are connected to a second quadrature phase shift network;

[0033] The first orthogonal phase shift network processes the two same-amplitude and same-phase radio frequency signals into the two same-amplitude orthogonal radio frequency signals;

[0034] The second orthogonal phase shift network processes the two RF signals from the through-end of the first dual directional coupler and the through-end of the second dual directional coupler into two RF signals with the same amplitude and phase, which are input into the two-way in-phase power combiner.

[0035] Compared with the prior art, the embodiments of the present invention can achieve at least the following beneficial effects:

[0036] When the coupler's directivity cannot be improved, this embodiment of the present invention processes the coupled signal based on two dual-directional couplers and two vector adders, taking into account the phases of the direct and reflected signals. This reduces the interference components of the coupled signal, thereby improving the accuracy of coupling detection. This embodiment of the present invention is applied to scenarios where coupling detection is performed on the power and standing waves of orthogonal RF signals of the same amplitude or same amplitude and phase. This can effectively reduce the power capacity requirements of a single coupler, thereby improving the stability and reliability of the RF transmitter and system.

[0037] In the present invention, the above-mentioned technical solutions can be combined with each other to achieve more preferred combinations. Other features and advantages of the present invention will be described in the following description, and some advantages will become apparent from the description or be learned through practice of the present invention. The objectives and other advantages of the present invention can be realized and obtained through the contents particularly pointed out in the description and drawings. BRIEF DESCRIPTION OF THE DRAWINGS

[0038] The accompanying drawings are only for the purpose of illustrating particular embodiments and are not to be considered limiting of the present invention. Like reference symbols denote like parts throughout the drawings.

[0039] Figure 1 FIG. 1 is a schematic diagram of the composition of a directional coupling processing device according to an embodiment of the present invention.

[0040] Figure 2 It is a schematic diagram of the composition of a directional coupling processing device according to another embodiment of the present invention.

[0041] Figure 3 is based on Figure 2The schematic diagram of the composition of the radio frequency power synthesis device of the directional coupling processing device shown.

[0042] Figure 4 Schematic diagram of the principle of type A and type B phase shift networks according to an embodiment of the present invention.

[0043] Figure 5 is the orthogonal network phase difference α provided by the embodiment of the present invention k and χ o and χ m relationship diagram.

[0044] Figure 6 Schematic diagram of the principle of improving coupling detection accuracy according to an embodiment of the present invention.

[0045] Figure 7 FIG. 1 is a schematic diagram of the composition of a radio frequency power synthesis device according to an embodiment of the present invention.

[0046] Figure 8 FIG. 1 is a schematic diagram of the composition of a radio frequency power synthesis device according to another embodiment of the present invention.

[0047] Figure 9 FIG. 1 is a schematic diagram of the composition of a radio frequency power synthesis device according to another embodiment of the present invention. DETAILED DESCRIPTION

[0048] The preferred embodiments of the present invention will be described in detail below in conjunction with the accompanying drawings, wherein the accompanying drawings constitute a part of this application and are used together with the embodiments of the present invention to illustrate the principles of the present invention, and are not used to limit the scope of the present invention.

[0049] To improve the accuracy of RF coupling detection, embodiments of the present invention provide a directional coupling processing device and an RF power combiner incorporating the directional coupling processing device for detecting RF output power and standing waves. Under the condition of a constant coupler directionality, embodiments of the present invention process the coupled signal accordingly to reduce undesired interference components in the coupled signal, thereby improving coupling detection accuracy.

[0050] Figure 1 FIG. 1 is a schematic diagram of a directional coupling processing device according to an embodiment of the present invention. Figure 1 As shown, the directional coupling processing device in this embodiment can improve the coupling detection accuracy, and includes a first directional coupler (dual directional coupler 1), a second directional coupler (dual directional coupler 2), a first vector adder (vector adder 1) and a second vector adder (vector adder 2).

[0051] The input end of the dual-directional coupler 1 and the input end of the dual-directional coupler 2 are respectively used to receive one of the two orthogonal RF signals with the same amplitude; the through-end of the dual-directional coupler 1 and the through-end of the dual-directional coupler 2 respectively output their respective RF signals and respectively receive reflected signals of the RF signals; the coupled-end outputs of the dual-directional coupler 1 and the dual-directional coupler 2 are vector-added by the vector adder 1 and then output; the isolated-end outputs of the dual-directional coupler 1 and the dual-directional coupler 2 are vector-added by the vector adder 2 and then output.

[0052] Specifically, such as Figure 1 As shown, the phases of the two same-amplitude orthogonal RF signals are θ and θ-90°, respectively, with a phase difference of 90°. The phase shifts of the first dual-directional coupler and the second dual-directional coupler are the same, both ψ. The signals at the through ends of the first dual-directional coupler and the second dual-directional coupler are still orthogonal signals, with phases of θ+ψ and θ-90°+ψ, respectively. The reflected signals of the two output signals are same-amplitude orthogonal signals, and are inversely quadrature signals with respect to the two same-amplitude orthogonal RF signals, that is, the phase difference of the two reflected signals is opposite in direction (opposite in sign) to the phase difference of the two same-amplitude orthogonal RF signals. For example, as Figure 1 As shown, the phase of one reflected signal (reflected signal 1) after the back-end device is α, and the phase of the other reflected signal (reflected signal 2) is α+90°. The corresponding phase difference between the two reflected signals is -90°.

[0053] Furthermore, in this embodiment and some embodiments of the present invention, the input ends of the dual directional coupler 1 and the dual directional coupler 2 are connected to a first quadrature phase-shift network, and / or the through ends of the dual directional coupler 1 and the dual directional coupler 2 are connected to a second quadrature phase-shift network; wherein the first quadrature phase-shift network processes two RF signals with the same amplitude and phase into the two orthogonal RF signals with the same amplitude and phase; and the second quadrature phase-shift network processes the two RF signals at the through ends of the dual directional coupler 1 and the through ends of the dual directional coupler 2 into two RF signals with the same amplitude and phase.

[0054] Furthermore, in this embodiment and some embodiments of the present invention, the first orthogonal phase-shifting network includes a first phase-shifting network and a second phase-shifting network, and the second orthogonal phase-shifting network includes a third phase-shifting network and a fourth phase-shifting network; the input end of the first phase-shifting network is connected to the first RF signal of the two RF signals with the same amplitude and phase, the output end of the first phase-shifting network is connected to the input end of the dual-directional coupler 1, and the through end of the dual-directional coupler 1 is connected to the input end of the third phase-shifting network; the input end of the second phase-shifting network is connected to the second RF signal of the two RF signals with the same amplitude and phase, the output end of the second phase-shifting network is connected to the input end of the dual-directional coupler 2, and the through end of the dual-directional coupler 2 is connected to the input end of the fourth phase-shifting network.

[0055] Furthermore, in this embodiment and some embodiments of the present invention, the phase shift value of the first phase shift network differs from the phase shift value of the second phase shift network by 90 degrees or -90 degrees; correspondingly, the phase shift value of the third phase shift network differs from the phase shift value of the fourth phase shift network by -90 degrees or 90 degrees.

[0056] Furthermore, in this embodiment and some embodiments of the present invention, the first phase-shifting network and the fourth phase-shifting network have the same circuit structure, and the second phase-shifting network and the third phase-shifting network have the same circuit structure.

[0057] Furthermore, in this embodiment and some embodiments of the present invention, the first phase-shifting network, the second phase-shifting network, the third phase-shifting network, and the fourth phase-shifting network include two capacitors, two inductors, and a coaxial cable segment. Specifically, the first end of the first capacitor is connected to the input end of the phase-shifting network, the second end of the first capacitor is connected to the first end of the inner conductor of the coaxial cable segment, the second end of the first capacitor is connected to the first end of the first inductor, and the second end of the first inductor is grounded; the first end of the second inductor is connected to the input end of the phase-shifting network, the second end of the second inductor is connected to the first end of the second capacitor, the second end of the second inductor is connected to the first end of the outer conductor of the coaxial cable segment, and the second end of the second capacitor is grounded; the second end of the outer conductor of the coaxial cable segment is grounded, and the second end of the inner conductor of the coaxial cable segment is connected to the output end of the phase-shifting network.

[0058] Furthermore, in this embodiment and some embodiments of the present invention, in the same circuit structure of the first phase-shifting network and the fourth phase-shifting network, the component parameters of the capacitor, the inductor, and the coaxial cable segment at corresponding positions are the same; in the same circuit structure of the second phase-shifting network and the third phase-shifting network, the component parameters of the capacitor, the inductor, and the coaxial cable segment at corresponding positions are the same.

[0059] The embodiment of the present invention reasonably sets the positions of two dual-directional couplers in a two-way synthesized RF circuit and utilizes the phase of the signal and the orthogonal phase transformation of the phase shift network to achieve the two-way orthogonal signals with the same amplitude arriving at the input end of the two-way in-phase synthesizer in the same phase, or the two-way orthogonal signals with the same amplitude arriving at the input end of the two-way orthogonal synthesizer in the orthogonal phase, thereby ultimately achieving power synthesis output. At the same time, in the coupled signals output from the coupling end and the isolation end of the dual-directional coupler, the useful signals are added in phase, and the interference signals are canceled out in opposite phases, thereby achieving accurate detection of RF power and realizing high directivity of the coupled detection.

[0060] For details, see Figure 1 The signals output from the coupling end of dual directional coupler 1 and the signals output from the coupling end of dual directional coupler 2 are added together by vector adder 1, and the output is a forward coupling signal. The forward coupling signal has no phase component and is only related to the RF input power P0, the coupling degree C of the coupler, the directivity D, and the reflection coefficient Γ, greatly improving the accuracy of the directional coupling processing device for RF detection. The signals output from the isolation end of coupler 1 and the signals output from the isolation end of coupler 2 are added together by vector adder 2, and the output is a reverse coupling signal. The reverse coupling signal has no phase component and is only related to the input power P0, the coupling degree C of the coupler, the directivity D, and the reflection coefficient Γ, also greatly improving the accuracy of the directional coupling processing device for RF detection.

[0061] Figure 2 FIG. 1 is a schematic diagram of a directional coupling processing device according to an embodiment of the present invention. Figure 2 As shown, the directional coupling processing device in this embodiment can improve coupling detection accuracy and includes a quadrature phase shift network 1, a dual directional coupler 1, a dual directional coupler 2, a quadrature phase shift network 2, a vector adder 1, and a vector adder 2. The quadrature phase shift network 1 includes a first phase shift network and a second phase shift network. The quadrature phase shift network 2 includes a third phase shift network and a fourth phase shift network.

[0062] The input end of the first phase-shifting network in the orthogonal phase-shifting network 1 is connected to the input end 1, the output end of the first phase-shifting network in the orthogonal phase-shifting network 1 is connected to the input end of the dual-directional coupler 1, the coupling end of the dual-directional coupler 1 is connected to the input end 1 of the vector adder 1, the isolation end of the dual-directional coupler 1 is connected to the input end 2 of the vector adder 2, the output end of the dual-directional coupler 1 is connected to the input end of the third phase-shifting network in the orthogonal phase-shifting network 2, and the output end of the third phase-shifting network in the orthogonal phase-shifting network 2 is the first output end (output end 1) of the directional coupling processing device.

[0063] The input end of the second phase-shifting network in the orthogonal phase-shifting network 1 is connected to the input end 2, the output end of the second phase-shifting network in the orthogonal phase-shifting network 1 is connected to the input end of the dual-directional coupler 2, the coupling end of the dual-directional coupler 2 is connected to the input end 2 of the vector adder 1, the isolation end of the dual-directional coupler 2 is connected to the input end 1 of the vector adder 2, the output end of the dual-directional coupler 2 is connected to the input end of the fourth phase-shifting network in the orthogonal phase-shifting network 2, and the output end of the fourth phase-shifting network in the orthogonal phase-shifting network 2 is the second output end (output end 2) of the directional coupling processing device.

[0064] Input terminal 1 and input terminal 2 are used to input a first RF signal and a second RF signal of two RF signals with the same amplitude and phase, respectively. The output terminal of vector adder 1 is connected to the forward coupling output terminal, and the output terminal of vector adder 2 is connected to the reverse coupling output terminal.

[0065] Specifically, in this embodiment and some embodiments of the present invention, the first and second phase-shifting networks in quadrature phase-shifting network 1 have a phase difference of approximately ±90°, and the fourth and third phase-shifting networks in quadrature phase-shifting network 2 have a phase difference of approximately ±90°. The first and fourth phase-shifting networks have the same circuit form and component parameters, and the second and third phase-shifting networks have the same circuit form and component parameters. The positions of quadrature phase-shifting network 1 and quadrature phase-shifting network 2 are interchangeable.

[0066] See also Figure 2 Specifically, the first phase-shifting network includes a first capacitor C1, a second capacitor C2, a first inductor L1, a second inductor L2, and a coaxial cable segment 1. One end of the first capacitor C1 is connected to the input terminal 1, and the other end is connected to one end of the first inductor L1. One end of the first inductor L1 is connected to one end of the inner conductor of the coaxial cable segment 1, and the other end of the first inductor L1 is grounded. One end of the second inductor L2 is connected to the input terminal 1, and the other end of the second inductor L2 is connected to one end of the outer conductor of the coaxial cable segment 1. One end of the second capacitor C2 is connected to one end of the outer conductor of the coaxial cable segment 1, and the other end of the second capacitor C2 is grounded. The other end of the outer conductor of the coaxial cable segment 1 is grounded, and the other end of the inner conductor of the coaxial cable segment 1 is connected to the output terminal of the first phase-shifting network.

[0067] See also Figure 2Specifically, the second phase-shifting network includes a third capacitor C3, a fourth capacitor C4, a third inductor L3, a fourth inductor L4, and a coaxial cable segment 2. One end of the third capacitor C3 is connected to the input terminal 2, and the other end is connected to one end of the third inductor L3. One end of the third inductor L3 is connected to one end of the inner conductor of the coaxial cable segment 2, and the other end of the third inductor L3 is grounded. One end of the fourth inductor L4 is connected to the input terminal 2, and the other end of the fourth inductor L4 is connected to one end of the outer conductor of the coaxial cable segment 2. One end of the fourth capacitor C4 is connected to one end of the outer conductor of the coaxial cable segment 2, and the other end of the fourth capacitor C4 is grounded. The other end of the outer conductor of the coaxial cable segment 2 is grounded, and the other end of the inner conductor of the coaxial cable segment 2 is connected to the output terminal of the second phase-shifting network.

[0068] See also Figure 2 Specifically, the third phase-shifting network includes a fifth capacitor C5, a sixth capacitor C6, a fifth inductor L5, a sixth inductor L6, and a coaxial cable segment 3. One end of the fifth capacitor C5 is connected to the output end of the dual directional coupler 1, and the other end is connected to one end of the fifth inductor L5. The other end of the fifth inductor L5 is grounded. One end of the fifth inductor L5 is connected to one end of the inner conductor of the coaxial cable segment 3. One end of the sixth inductor L6 is connected to the output end of the dual directional coupler 1, and the other end of the sixth inductor L6 is connected to one end of the outer conductor of the coaxial cable segment 3. One end of the sixth capacitor C6 is connected to one end of the outer conductor of the coaxial cable segment 3, and the other end of the sixth capacitor C6 is grounded. The other end of the outer conductor of the coaxial cable segment 3 is grounded. The other end of the inner conductor of the coaxial cable segment 3 serves as the output end 1 of the directional coupling processing device.

[0069] See also Figure 2 Specifically, the fourth phase-shifting network includes a seventh capacitor C7, an eighth capacitor C8, a seventh inductor L7, an eighth inductor L8, and a coaxial cable segment 4. One end of the seventh capacitor C7 is connected to the output end of the dual-directional coupler 2, and the other end is connected to one end of the seventh inductor L7. The other end of the seventh inductor L7 is grounded. One end of the seventh inductor L7 is connected to one end of the inner conductor of the coaxial cable segment 4. One end of the eighth inductor L8 is connected to the output end of the dual-directional coupler 2, and the other end of the eighth inductor L8 is connected to one end of the outer conductor of the coaxial cable segment 4. One end of the eighth capacitor C8 is connected to one end of the outer conductor of the coaxial cable segment 4, and the other end of the eighth capacitor C8 is grounded. The other end of the outer conductor of the coaxial cable segment 4 is grounded. The other end of the inner conductor of the coaxial cable segment 4 serves as the output end 2 of the directional coupling processing device.

[0070] Figure 3 is based on Figure 2 The schematic diagram of the composition of the radio frequency power synthesis device of the directional coupling processing device is shown. Figure 3 As shown, the RF power synthesis device in this embodiment can improve the coupling detection accuracy. Figure 2The radio frequency power synthesis device of the directional coupling processing device includes an orthogonal phase shift network 1, a dual directional coupler 1, a dual directional coupler 2, an orthogonal phase shift network 2, a vector adder 1, a vector adder 2, and a two-way co-directional power synthesizer. The orthogonal phase shift network 1 includes a first phase shift network and a second phase shift network. The orthogonal phase shift network 2 includes a third phase shift network and a fourth phase shift network. The output end 1 of the directional coupling processing device is connected to the distribution end 1 of the two-way co-directional power divider. The output end 2 of the directional coupling processing device is connected to the distribution end 2 of the two-way co-directional power divider. The synthesis end and the output end of the two-way co-directional power combiner are connected.

[0071] The phase and amplitude of the distribution end 1 and the distribution end 2 of the two-way unidirectional power combiner are consistent, and the input characteristic impedance of the three ports is Z. The lower limit of the operating frequency range of the two-way unidirectional power combiner is f L , upper frequency limit f H .set up In this embodiment, χ m Its value is generally between 2 and 3.

[0072] The circuit form and component parameters of the dual directional coupler 1 and the dual directional coupler 2 are consistent, with forward coupling (coupling end output) and reverse coupling (isolation end output) functions, and the lower limit of the operating frequency is ≤f L , upper frequency limit ≥ f H .

[0073] The circuit form and component parameter selection of the first phase-shifting network in the orthogonal phase-shifting network 1 and the fourth phase-shifting network in the orthogonal phase-shifting network 2 are consistent, and the circuit form and component parameter selection of the second phase-shifting network in the orthogonal phase-shifting network 1 and the third phase-shifting network in the orthogonal phase-shifting network 2 are consistent.

[0074] It can be understood that after the same-amplitude and same-phase signals at input terminals 1 and 2 pass through the first phase-shifting network and the second phase-shifting network in the orthogonal phase-shifting network 1, the phase difference between the two phase-shifting networks is 90° (or -90°). After passing through the third phase-shifting network and the fourth phase-shifting network in the orthogonal phase-shifting network 2, respectively, the phase difference between the two phase-shifting networks is -90° (or 90°). After the cross-conversion of the phase difference, the signals become the same-amplitude and same-phase signals at the input terminals of the two-way co-directional combiner, thereby realizing power synthesis output.

[0075] Figure 4 Schematic diagram of the principle of type A and type B phase shift networks according to an embodiment of the present invention.

[0076] See also Figure 4In the A-type phase-shifting network, the first end of the first capacitor C1 is connected to the input end of the phase-shifting network, the second end of the first capacitor C1 is connected to the first end of the inner conductor of the coaxial cable segment, the second end of the first capacitor C1 is connected to the first end of the first inductor L1, and the second end of the first inductor L1 is grounded; the first end of the second inductor L2 is connected to the input end of the phase-shifting network, the second end of the second inductor L2 is connected to the first end of the second capacitor C2, the second end of the second inductor L2 is connected to the first end of the outer conductor of the coaxial cable segment, and the second end of the second capacitor C2 is grounded; the second end of the outer conductor of the coaxial cable segment is grounded, and the second end of the inner conductor of the coaxial cable segment is connected to the output end of the phase-shifting network.

[0077] See also Figure 4 In the B-type phase-shifting network, the first end of the first inductor L1 is connected to the input end of the phase-shifting network, the second end of the first inductor L1 is connected to the first end of the inner conductor of the coaxial cable segment, the second end of the first inductor L1 is connected to the first end of the first capacitor C1, and the second end of the first capacitor C1 is grounded; the first end of the second capacitor C2 is connected to the input end of the phase-shifting network, the second end of the second capacitor C2 is connected to the first end of the second inductor L2, the second end of the second capacitor C2 is connected to the first end of the outer conductor of the coaxial cable segment, and the second end of the second inductor L2 is grounded; the second end of the outer conductor of the coaxial cable segment is grounded, and the second end of the inner conductor of the coaxial cable segment is connected to the output end of the phase-shifting network.

[0078] In some other embodiments of the present invention, the first phase shift network and the second phase shift network of the orthogonal phase shift network 1, and the third phase shift network and the fourth phase shift network of the orthogonal phase shift network 2 may also adopt BB type combination, AB type combination and BA type combination.

[0079] In order to facilitate actual use, appropriate methods should be adopted as needed. Figure 4 As shown, the first and second phase-shifting networks can be either Type A (with DC blocking) or Type B. The first and second phase-shifting networks in the quadrature phase-shifting network 1 can be freely selected as either Type A or Type B, i.e., AA, AB, BA, or BB combinations. The component parameters required for each phase-shifting network combination are identical, except that the phase difference between the first and second phase-shifting networks is either 90° or -90°. The phase differences for each combination are shown in Table 1.

[0080] Table 1

[0081]

[0082] Understandably, see Figure 4 , Figure 2 and Figure 3In the illustrated embodiment, the first and second phase shift networks in the quadrature phase shift network 1 adopt an AA type combination, and the third and fourth phase shift networks in the quadrature phase shift network 2 adopt an AA type combination.

[0083] The port impedance values ​​of the input and output ends of the first phase shift network in the orthogonal phase shift network 1 are equal, which is Z a The port impedance values ​​of the input and output ends of the second phase-shift network in the orthogonal phase-shift network 1 are equal, which is Z b .

[0084] Figure 2 and Figure 3 In the first phase-shifting network shown, the input signal of the coaxial cable segment 1 is drawn from the connection point of the two LCs, which is a two-end signal; the output signal of the coaxial cable segment 1 is a single-end signal with one end grounded. The coaxial cable segment 1 plays the role of a 1:1 balanced-unbalanced impedance converter 1 in the circuit, with the end connected to the inductor and capacitor as the balanced end and the end with the outer conductor grounded as the unbalanced end; the length of the coaxial cable segment 1 is l1, l1≤0.5*λ, where λ is the wavelength of the highest operating frequency in the coaxial cable segment 1 medium; the characteristic impedance of the coaxial cable segment 1 is Z1, Z1=Z=Za=Zin=Zout, that is, its characteristic impedance is consistent with the port impedance of the phase-shifting network. Requirements are met: L1=L2, C1=C2, Request f a <f L , f L It is the lower limit of the operating frequency range of the two-way co-directional power combiner.

[0085] Figure 2 and Figure 3 In the second phase-shifting network shown, the input signal of the coaxial cable segment 2 is drawn out from the connection point of the two LCs, which is a two-end signal; the output signal of the coaxial cable segment 2 is a single-ended signal with one end grounded. The coaxial cable segment 2 plays the role of a 1:1 balanced-unbalanced impedance converter 2 in the circuit, with the end connected to the inductor and capacitor as the balanced end, and the end where the outer conductor is grounded as the unbalanced end; the line length of the coaxial cable segment 2 is l2, l2≤0.5*λ, where λ is the wavelength of the highest operating frequency in the medium of the coaxial cable segment 2. The parameters of the coaxial cable segment 1 and the coaxial cable segment 2 are the same, that is, l1=v2, and both are ≤0.5*λ. In addition, the characteristic impedance Z2 of the coaxial cable segment 2, Z2=Z=Zb=Zin=Zout, that is, its characteristic impedance is consistent with the port impedance of the phase-shifting network. Requirements are met: L3=L4, C3=C4, Request f H <f b , f HIt is the upper frequency limit of the operating frequency range of the two-way unidirectional power combiner.

[0086] The first phase shift network and the second phase shift network in the orthogonal phase shift network 1 can be respectively adopted as follows Figure 4 The Type A or Type B phase-shifting network shown in the figure both meet the following requirements: Z = Za = Zb = Zin = Zout. Although the absolute phase shift at the output of the Type A or Type B phase-shifting network at a given frequency depends on the length of the phase-shifting network and the coaxial cable segment, the relative phase shift between the two remains relatively constant, approximately 90° (or -90°), achieving quadrature phase shifting.

[0087] Figure 2 and Figure 3 In the embodiment shown, the first phase shift network and the second phase shift network in the orthogonal phase shift network 1 are set Its value is generally between 2 and 3. a <f b and The angular frequency of the first phase shift network is ω 0a =2πf a , phase shift value α a =-2arctan(ω / ω 0a ), the angular frequency of the second phase-shifting network is ω 0b =2πf b , the phase shift is α b =-2arctan(ω / ω 0b ), approximately equal to 90° or -90°.

[0088] Figure 5 is the orthogonal network phase difference α provided by the embodiment of the present invention k and χ o and χ m In order to facilitate the observation of the difference in phase shift and χ o And the relationship between the working frequency, set

[0089] set up

[0090] χ0=ω ob / ω m =ω m / ω oa ,

[0091] χ m =ω / ω m ,

[0092] Then α a =-2arctan(χ o *χ m ),

[0093] α b =-2arctan(χ m / χ o ),

[0094] at this time

[0095] According to this formula, we can get Figure 5 , which shows that the phase shift difference α between the two phase shift networks is k and χ o and χ m Therefore, it is necessary to select and determine the parameter χ according to the given phase shift offset and operating frequency bandwidth. o and χ m ,from Figure 5 It can be seen from the figure that under the conditions of the specified differential phase shift offset and working bandwidth, first determine χ o The value range of χ o After determining, ω can be calculated 0a and ω 0b , then according to the formula Determine the circuit parameter values.

[0096] The phase shift network used in the embodiment of the present invention is an orthogonal phase shift network, and the phase shift difference α k =α b -α a The phase shift is approximately 90° or -90°. Figure 5 It can be seen that χ o When the value is selected between 2.5 and 2.8, the bandwidth (χ m ) several times, the phase shift difference remains close to 90 degrees, which can meet most orthogonal usage scenarios.

[0097] The operating frequency of the embodiment of the present invention is 100MHz~500Mhz, and ω can be calculated according to the above steps. m ≈1404MHz, χ m =2.236, check Figure 5 α k -χ m Follow χ o For a graph of changes, select χ o = 2.7 ensures the differential phase shift αk is within the range of 81.9° to 98.7°. L1 is 96.2nH, C1 is 38.3pF, L2 is 13.5nH, and C2 is 5.3pF. A 1:1 balun impedance transformer uses a coaxial cable segment with a length of l ≤ 0.5*λ, where λ is the wavelength of the highest operating frequency in the cable dielectric. If the dielectric constant is 2.1, then ι ≤ 172.4mm.

[0098] It is understandable that the positions of the orthogonal phase-shift network 1 and the orthogonal phase-shift network 2 can be interchanged. That is, it does not matter which of the first and second phase-shift networks in the orthogonal phase-shift network 1 has a phase that leads by 90° or lags by 90°. As long as the phase difference between the third and fourth phase-shift networks in the orthogonal phase-shift network 2 is consistent in amplitude but opposite in phase compared to the phase difference between the first and second phase-shift networks, the final phase difference of the RF signal after passing through the above two orthogonal phase-shift networks is 0. This ensures that the two RF signals with consistent amplitudes and phases at input ends 1 and 2 have consistent amplitudes and phases at the distribution ends 1 and 2 of the two-way co-directional power combiner, completing co-directional power combination.

[0099] The above content is a detailed description of how to implement orthogonal phase shifting of RF signals and ultimately achieve equal amplitude and phase at distribution end 1 and distribution end 2 of a two-way co-directional power combiner.

[0100] The following combination Figures 2 to 6 , describes in detail the working process of improving coupling accuracy of the directional coupling processing device in the embodiment of the present invention, including the radio frequency power synthesis device of the directional coupling processing device.

[0101] Figure 6 This is a schematic diagram of the principle of improving the coupling detection accuracy of an embodiment of the present invention. Assuming the coupling degree of the coupler is C, since the directivity D of the coupler cannot be infinite, there are two signals in the coupled signal at the coupled end of the coupler, namely the forward coupled signal and the reverse (reflected) coupled signal. Let's start with the dual directional coupler 1. Figure 6 As shown, assuming that the power of the two input ends is P0 and the phase is 0°, assuming that the phase after the first phase shift network is θ, where θ is a function related to the operating frequency, and the subsequent β, φ and ψ are all frequency functions, then the phase after the second phase shift network is θ+90°. The forward coupling signal at the coupling end of the dual directional coupler 1 can be expressed by formula (1): ff1 =A*cos(ωt+θ)<Equation (1)>, where A is the amplitude of the forward coupled signal, which can be expressed by equation (2): A 2 / R=P0*C<Equation (2)>, R is the port impedance, generally 50 ohms, ω=2πf is the operating angular frequency, and the initial phase of the dual directional coupler 1 is θ; the reverse coupling signal at the coupling end of the dual directional coupler 1 can be expressed by equation (3): fr1=B*cos(ωt+θ+β+φ+ψ+90°)<Equation (3)>, where B is the amplitude of the reverse coupling signal, β is the initial phase of the reflected signal at the combined end of the two-way co-directional power combiner, φ is the phase shift of the two-way co-directional power combiner, ψ is the phase shift of the dual-directional coupler 1, and (β+φ+ψ+90°) is the phase difference between the reverse signal and the forward signal at the forward coupling end; therefore, the coupled signal at the forward coupling end of coupler 1 can be expressed by equation (4): f1 =y ff1 +y fr1 =A*cos(ωt+θ)+B*cos(ωt+θ+β+φ+ψ+90°)<Equation (4)>. The coupled signal can be detected by a power detector, and its DC detection voltage amplitude can be expressed by equation (5): V f1 2 =A 2 +B 2 +2A*B*cos(β+φ+ψ+90°)<Equation (5)>. Since the coupler directivity is D and the reflection coefficient is Γ, the DC detection voltage at the coupling end can be expressed by equation (6): V f1 2 =A 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ+ψ+90°) / D<Equation (6)>. From equation (6), we can see that assuming the directivity of the coupler is =log 10 D 2 =20dB, that is, D=10. At this time, the second term in equation (6) is small and can be ignored, while the third term is large, which has a greater impact on the detection accuracy. In addition, it is affected by the reflection coefficient Γ and the phase difference (β+φ+ψ+90°). Because θ, β, φ and ψ are all frequency functions, their influence will change with the change of standing wave load and operating frequency, resulting in uncertainty in the detection of coupled signals.

[0102] The forward coupled signal at the isolation end of the dual directional coupler 1 can be expressed by equation (7): rf1 =A / D*cos(ωt+ψ+θ)<Equation (7)>; The reverse coupled signal at the isolation end of the dual directional coupler 1 can be expressed by equation (8): rr1 =A / D*Γ*cos(ωt+θ+β+φ+90°)<Equation (8)>, (β+φ+90°-ψ) is the phase difference between the reverse signal and the forward signal at the isolation end; therefore, the coupled signal at the isolation end of coupler 1 can be expressed by equation (9): r1 =y rf1 +y rr1=A / D*cos(ωt+ψ+θ)+A / D*Γ*cos(ωt+θ+β+φ+90°)<Equation (9)>. The coupled signal can be detected by a power detector, and its DC detection voltage amplitude can be expressed by equation (10): V r1 2 =A 2 / D 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ-ψ+90°) / D<Equation (10)>. As shown in Equation (10), assuming the directivity of the coupler is 20 dB, that is, D = 10, the second term in Equation (10) is small and can be ignored. The third term is large and has a greater impact on the detection accuracy. It is also affected by the reflection coefficient Γ and the phase difference (β+φ+90°-ψ). Because θ, β, φ, and ψ are all frequency functions, their influence will change with changes in the standing wave load and operating frequency, resulting in uncertainty in the detection of the coupled signal.

[0103] The design parameters and performance indicators of dual directional coupler 2 are consistent with those of dual directional coupler 1, with a coupling degree of C and a directivity of D. Similarly, we can calculate the signal at the coupled end of dual directional coupler 2 according to the above calculation.

[0104] Let’s analyze the dual directional coupler 2 again. The RF signal flow direction is shown in Figure 6 As shown, assuming that the power of the two input ends is P0 and the phase is 0°, the phase after passing through the first phase shift network is θ, and the phase after passing through the second phase shift network is θ+90°, the forward coupling signal of the coupling end of the dual directional coupler 2 can be expressed by formula (11): ff2 =A*cos(ωt+θ+90°)<Equation (11)>, where A is the amplitude of the forward coupled signal, which can be expressed by equation (12): A 2 / R=P0*C<Equation (12)>, R is the port impedance, generally 50 ohms, ω=2πf is the operating angular frequency, θ+90° is the initial phase of the dual directional coupler 2; the reverse coupling signal at the coupled end of the dual directional coupler 2 can be expressed by equation (13): fr2 =B*cos(ωt+θ+β+φ+ψ)<Equation (13)>, where B is the amplitude of the reverse coupling signal, β is the initial phase of the reflected signal at the combined end of the two-way co-directional combiner, φ is the phase shift of the two-way co-directional combiner, ψ is the phase shift of the dual directional coupler 2, and (β+φ+ψ-90°) is the phase difference between the reverse signal and the forward signal at the forward coupling end; therefore, the coupled signal at the forward coupling end of coupler 2 can be expressed by equation (14): f2 =y ff2 +y fr2=A*cos(ωt+θ+90°)+B*cos(ωt+θ+β+φ+ψ)<Equation (14)>. The coupled signal can be detected by a power detector, and its DC detection voltage amplitude can be expressed by equation (15): V f2 2 =A 2 +B 2 +2A*B*cos(β+φ+ψ-90°)<Equation (15)>. Since the coupler directivity is D and the reflection coefficient is Γ, the DC detection voltage at the coupling end can be expressed by equation (16): V f2 2 =A 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ+ψ-90°) / D<Equation (16)>. As shown in Equation (16), assuming the directivity of the coupler is 20 dB, that is, D = 10, the second term in Equation (16) is small and can be ignored. The third term is large and has a greater impact on the detection accuracy. It is also affected by the reflection coefficient Γ and the phase difference (β+φ+ψ-90°). Because θ, β, φ, and ψ are all frequency functions, their influence will change with changes in the standing wave load and operating frequency, resulting in uncertainty in the detection of the coupled signal.

[0105] The forward coupled signal at the isolation port of the dual directional coupler 2 can be expressed by equation (17):

[0106] y rf2 =A / D 0.5 *cos(ωt+ψ+θ+90°)<Formula (17)>;

[0107] The reverse coupled signal at the isolation end of the dual directional coupler 2 can be expressed by equation (18):

[0108] y rr2 =A / D*Γ*cos(ωt+θ+β+φ)<Formula (18)>,

[0109] (β+φ-ψ-90°) is the phase difference between the reverse signal and the forward signal at the isolation end; therefore, the coupled signal at the isolation end of coupler 2 can be expressed by formula (19):

[0110] y r2 =y rf2 +y rr2 =A / D*cos(ωt+ψ+θ+90°)+A / D*Γ*cos(ωt+θ+β+φ) <Formula (19)>.

[0111] The coupled signal can be detected by a power detector, and its DC detection voltage amplitude can be expressed by formula (20): Vr2 2 =A 2 / D 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ-ψ-90°) / D<Equation (20)>. As shown in Equation (20), assuming the directivity of the coupler is 20 dB, that is, D = 10, the second term in Equation (20) is small and can be ignored. The third term is large and has a greater impact on the detection accuracy. It is also affected by the reflection coefficient Γ and the phase difference (β+φ-ψ-90°). Because θ, β, φ, and ψ are all frequency functions, their influence will change with changes in the standing wave load and operating frequency, resulting in uncertainty in the detection of the coupled signal.

[0112] The coupled end signal V of the dual directional coupler 1 f1 2 =A 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ+ψ+90°) / D and the coupled end signal V of the dual directional coupler 2 f2 2 =A 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ+ψ-90°) / D are connected to input terminal 1 and input terminal 2 of vector adder 1 respectively, then the forward coupling signal output from the output terminal of vector adder 1 can be expressed as formula (21):

[0113]

[0114] From formula (21), it can be seen that the forward coupling signal output from the output end of the vector adder 1 has no phase component at this time and is only related to the input power P0, the coupling degree C of the coupler, the directivity D, and the reflection coefficient Γ, which greatly improves the detection accuracy of the coupler.

[0115] The isolated end signal V of the dual directional coupler 1 r1 2 =A 2 / D 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ-ψ+90°) / D and the isolated end signal V of the dual directional coupler 2 r22 =A 2 / D 2 +A 2 *Γ 2 / D 2 +2A 2 *Γ*cos(β+φ-ψ-90°) / D are connected to input terminal 2 and input terminal 1 of vector adder 2 respectively, then the reverse coupling signal output from the output terminal of vector adder 2 can be expressed as formula (22):

[0116]

[0117] From formula (22), it can be seen that the reverse coupling signal output from the output end of vector adder 2 has no phase component at this time and is only related to the input power P0, the coupling degree C of the coupler, the directivity D, and the reflection coefficient Γ, which greatly improves the detection accuracy of the coupler.

[0118] In summary, based on Figure 2 and Figure 3 The directional coupling processing device and the radio frequency power synthesis device including the directional coupling processing device in the embodiment of the present invention shown utilize the characteristics of the previous and next stage signals, or use the orthogonal characteristics of the orthogonal phase shift network in conjunction with the phase characteristics (in-phase characteristics and orthogonal characteristics) of the previous and next stage signals to achieve a phase-free output signal of the forward coupling and reverse coupling of the coupler, thereby improving the detection accuracy of the coupler.

[0119] Furthermore, in the embodiment of the present invention, Figure 2 The phase shift function circuits of the first and second phase shift networks of quadrature phase shift network 1 are moved up to the preceding circuit, for example, to a two-way power divider preceding two amplifiers 1 and 2 with identical performance indicators. The two-way divider then becomes a two-way quadrature power divider. Similarly, the phase shift function circuits of quadrature phase shift network 2 are moved back to the next-stage two-way unidirectional power combiner. The two-way unidirectional combiner then becomes a two-way quadrature power combiner.

[0120] Specifically, the radio frequency power synthesis device in some embodiments of the present invention includes a directional coupling processing device and a two-way orthogonal power synthesizer. The directional coupling processing device includes a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein the input end of the first dual directional coupler and the input end of the second dual directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude; the through-end of the first dual directional coupler and the through-end of the second dual directional coupler respectively output their respective one of the radio frequency signals and respectively receive the reflected signals of the one of the radio frequency signals; the coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after vector addition processing by the first vector adder; the isolated end outputs of the first dual directional coupler and the second dual directional coupler are output after vector addition processing by the second vector adder; the through-end of the first dual directional coupler and the through-end of the second dual directional coupler are connected to the input end of the two-way orthogonal power synthesizer.

[0121] Specifically, the radio frequency power combining device in some embodiments of the present invention includes a directional coupling processing device and a two-way co-directional power combiner. The directional coupling processing device includes an orthogonal phase shift network, a first dual directional coupler, a second dual directional coupler, a first vector adder, and a second vector adder; wherein the input end of the first dual directional coupler and the input end of the second dual directional coupler are respectively used to receive one of the two orthogonal RF signals with the same amplitude; the through-end of the first dual directional coupler and the through-end of the second dual directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal; the coupled end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by the first vector adder and then output; the isolated end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by the second vector adder and then output; the through-ends of the first dual directional coupler and the second dual directional coupler are connected to the orthogonal phase shift network; the orthogonal phase shift network performs orthogonal phase shift processing on the two RF signals of the through-end of the first dual directional coupler and the through-end of the second dual directional coupler, and inputs the obtained two RF signals with the same amplitude and phase into the two-way in-phase power combiner.

[0122] Therefore, in addition to Figure 3 The RF power synthesis device shown can be based on Figure 1 The directional coupling processing device in the embodiment of the present invention is used to construct other radio frequency power synthesis devices in the embodiment of the present invention, such as Figures 7 to 9 shown.

[0123] Figures 7 to 9In the specific embodiment of the present invention shown, the two-way orthogonal power divider includes the orthogonal phase shift network 1 in the embodiment of the present invention, and the two-way orthogonal power combiner includes the orthogonal phase shift network 2 in the embodiment of the present invention.

[0124] Specifically, Figure 9 The process of improving the detection accuracy of the coupler in these embodiments is described in detail using the example of FIG. After the input signal passes through the two-way orthogonal distributor, it is input to two power amplifiers 1 and 2 with consistent performance indicators for amplification. Then, after passing through two dual-directional couplers with consistent parameters and performance, it reaches the two-way orthogonal power combiner, and finally realizes the orthogonal synthesis output. At this time, the forward coupling signal and the reverse coupling signal output after the vector adder have no phase component, and are only affected by the input power size P0, the coupling degree C of the coupler, the directivity D, and the reflection coefficient Γ (the reflection power size P r ), which improves the coupler detection accuracy. Assume that the phase of the signal input at the synthesis end of the two-way orthogonal power divider is 0°, and the output signal amplitudes of the distribution ends 1 and 2 are the same. At this time, the output signal phase of the distribution end 1 is θ. Since the input impedance of amplifier 1 is non-ideal Z, there must be a mismatch phenomenon, that is, there is a standing wave. The reflected signal is phase-inverted at the reflection interface. At this time, the phase of the reflected signal is θ+180°; at this time, the output signal phase of the distribution end 2 is θ-90°. Similarly, since the input impedance of amplifier 2 is non-ideal Z, there must be a mismatch phenomenon, that is, there is a standing wave. The reflected signal is phase-inverted at the reflection interface. At this time, the phase of the reflected signal is θ+90°. Because amplifiers 1 and 2 have the same performance indicators, the amplitudes of the reflected signals are consistent. After being fed back through the two distribution ends of the two-way orthogonal divider, the phase difference of the two signals inside the two-way orthogonal power divider is θ+180°-(θ+90°-90°)=180°, and the phases are opposite. Therefore, they will be consumed on the internal balancing resistor of the two-way orthogonal power divider, and finally manifested as Figure 9 There is no reflected signal output at the input end of amplifier 1 and amplifier 2, that is, the reflected signal caused by the standing wave condition at the ports of amplifier 1 and amplifier 2 cannot reach the upper-stage device, and has little impact on the working status of the upper-stage device. It can be considered that the circuit has a certain front-end and back-end isolation capability, which helps to stabilize and reliably operate the devices when the system is cascaded.

[0125] It can be seen from the detailed description of the above embodiments of the present invention that, compared with the prior art, the embodiments of the present invention can achieve at least one of the following beneficial effects:

[0126] The directional coupling processing device of the embodiment of the present invention and the radio frequency power synthesis device including the directional coupling processing device avoid detection inaccuracy caused by antenna standing wave differences, can complete the detection of output power and standing waves, achieve effective control of system output power, and can also achieve accurate evaluation of system working efficiency through accurate detection of power size.

[0127] The directional coupling processing device of the embodiment of the present invention and the radio frequency power synthesis device including the directional coupling processing device are applied to the radio frequency broadband high-power transmitter system. For the scenario where single-channel amplification cannot achieve broadband high-power output (for example, the amplified output of the target power level cannot be achieved through a single amplifier circuit), a solution of amplifying two or more channels of power and then synthesizing them can be selected as needed to achieve low-loss synthesis of multiple power signals to output a signal of a higher power level.

[0128] In the prior art, adding a coupler increases circuit complexity, and RF broadband, high-power transmitter systems generally employ only a single coupler. The directional coupling processing device and RF power combiner incorporating the same in the embodiments of the present invention break with this prior art convention. While the use of two dual-directional couplers increases circuitry, the power capacity of a single coupler is reduced. Two couplers can be implemented on the same printed circuit board using striplines, providing excellent consistency and minimal area and space requirements. This not only improves the directionality of coupling detection (effectively reducing the impact of standing wave differences on airborne antennas), but also allows the quadrature phase-shifting circuit to achieve isolation between the front and rear stages, contributing to the stable operation of the amplification system.

[0129] The positions of the first dual-directional coupler and the second dual-directional coupler in the circuit of the embodiment of the present invention are symmetrical, and the stripline coupler or microstrip line coupler can be designed using the same printed circuit board, which helps to ensure the consistency of performance indicators of the two couplers.

[0130] The embodiment of the present invention uses two dual directional couplers to respectively perform coupling detection on the magnitudes of two input powers, which can effectively reduce the power capacity requirement of a single dual directional coupler.

[0131] The embodiment of the present invention implements quadrature phase shifting through two sets of quadrature phase shift networks. The reverse coupling signal has no phase component and is only related to the input power P0, the coupling degree C of the coupler, the directivity D, and the reflection coefficient Γ. This greatly improves the coupling detection accuracy, especially the reverse coupling signal detection accuracy, which helps to improve the stability and reliability of equipment and systems.

[0132] The two sets of orthogonal phase-shift networks in the embodiment of the present invention have a certain ability to isolate the front and back stages. When standing waves exist at the input or output ends of the two-way unidirectional power combiner, they have little impact on the working state of the upper-level equipment, which helps to ensure stable and reliable working state of each device when the system is cascaded.

[0133] The quadrature phase-shift network circuit of the present invention has a simple circuit topology and is a lossless LC network. In embodiments employing two quadrature phase-shift networks, the circuit is interchangeable, minimizing the increase in overall insertion loss caused by the introduction of the quadrature phase-shift network. Furthermore, the use of a lumped parameter design helps reduce size and volume at lower operating frequencies, facilitating miniaturization.

[0134] Compared to 3dB quadrature bridge phase-shifting networks, the quadrature phase-shifting network of this invention achieves a wider operating bandwidth and lower in-band ripple, making it more suitable for broadband, high-power operation and maintaining a near-90° phase shift across a wider operating frequency band. The use of lumped-parameter LC components allows operation at lower frequencies, facilitating the miniaturization of phase-shifting networks in these lower-frequency bands.

[0135] When the AA-type phase-shifting network is used in the embodiment of the present invention, the signal passing through the inner conductor of the coaxial cable segment is DC-isolated by the capacitor C of the phase-shifting network, and the signal passing through the inductor L of the phase-shifting network is grounded via the outer conductor of the coaxial cable segment, achieving the DC-isolating effect of the front and rear stages.

[0136] The quadrature phase-shifting networks employed in the embodiments of the present invention offer a variety of combinations, allowing for the appropriate configuration to be used in practical applications. The two phase-shifting networks can be either Type A (with DC blocking) or Type B. The two phase-shifting networks in the quadrature phase-shifting network can be freely selected as either Type A or Type B, i.e., AA, AB, BA, or BB combinations. The component parameters required for each phase-shifting network combination are identical, with the only difference being that the phase difference between the two phase-shifting networks is either 90° or -90°.

[0137] Those skilled in the art will appreciate that all or part of the process steps of the above-described embodiments can be implemented by instructing related hardware through a computer program, and the program can be stored in a computer-readable storage medium, such as a magnetic disk, an optical disk, a read-only memory, or a random access memory.

[0138] The above description is only a preferred specific embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any changes or substitutions that can be easily thought of by any technician familiar with this technical field within the technical scope disclosed by the present invention should be covered by the scope of protection of the present invention.

Claims

1. A directional coupling processing device, characterized in that: It includes a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein, The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude; The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal; The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the first vector adder; The isolation end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by the second vector adder and then output.

2. The directional coupling processing device according to claim 1, wherein: The input ends of the first dual directional coupler and the second dual directional coupler are connected to a first quadrature phase shift network, and / or the through ends of the first dual directional coupler and the second dual directional coupler are connected to a second quadrature phase shift network; in, The first orthogonal phase shift network processes the two same-amplitude and same-phase radio frequency signals into the two same-amplitude orthogonal radio frequency signals; The second orthogonal phase-shift network processes two radio frequency signals from the through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler into two radio frequency signals with the same amplitude and phase.

3. The directional coupling processing device according to claim 2, wherein: The first orthogonal phase-shifting network includes a first phase-shifting network and a second phase-shifting network, and the second orthogonal phase-shifting network includes a third phase-shifting network and a fourth phase-shifting network; The input end of the first phase-shifting network is connected to the first RF signal of the two RF signals with the same amplitude and phase, the output end of the first phase-shifting network is connected to the input end of the first dual-directional coupler, and the through end of the first dual-directional coupler is connected to the input end of the third phase-shifting network; The input end of the second phase-shifting network is connected to the second RF signal of the two RF signals with the same amplitude and phase, the output end of the second phase-shifting network is connected to the input end of the second dual-directional coupler, and the through end of the second dual-directional coupler is connected to the input end of the fourth phase-shifting network.

4. The directional coupling processing device according to claim 3, wherein: The phase shift value of the first phase shift network and the phase shift value of the second phase shift network differ by 90 degrees or -90 degrees; The phase shift value of the third phase shift network and the phase shift value of the fourth phase shift network differ by -90 degrees or 90 degrees.

5. The directional coupling processing device according to claim 4, wherein: The first phase-shifting network and the fourth phase-shifting network have the same circuit structure, and the second phase-shifting network and the third phase-shifting network have the same circuit structure.

6. The directional coupling processing device according to claim 5, wherein: The first phase-shifting network, the second phase-shifting network, the third phase-shifting network and the fourth phase-shifting network each include two capacitors, two inductors and a coaxial cable segment; wherein, A first end of the first capacitor is connected to the input end of the phase shift network, a second end of the first capacitor is connected to the first end of the inner conductor of the coaxial cable segment, a second end of the first capacitor is connected to the first end of the first inductor, and a second end of the first inductor is grounded; A first end of the second inductor is connected to the input end of the phase shift network, a second end of the second inductor is connected to the first end of the second capacitor, a second end of the second inductor is connected to the first end of the outer conductor of the coaxial cable segment, and a second end of the second capacitor is grounded; The second end of the outer conductor of the coaxial cable segment is grounded, and the second end of the inner conductor of the coaxial cable segment is connected to the output end of the phase shift network.

7. The directional coupling processing device according to claim 6, wherein: In the same circuit structure of the first phase-shifting network and the fourth phase-shifting network, the component parameters of the capacitor, the inductor, and the coaxial cable segment in corresponding positions are the same; in the same circuit structure of the second phase-shifting network and the third phase-shifting network, the component parameters of the capacitor, the inductor, and the coaxial cable segment in corresponding positions are the same.

8. A radio frequency power synthesis device, characterized in that: It includes a directional coupling processing device and a two-way orthogonal power combiner; The directional coupling processing device includes a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein, The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude; The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal; The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by a first vector adder; The isolation end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by a second vector adder and then output; The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler are connected to the input end of the two-way orthogonal power combiner.

9. A radio frequency power synthesis device, characterized in that: It includes a directional coupling processing device and a two-way co-directional power combiner; The directional coupling processing device includes an orthogonal phase shift network, a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein, The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude; The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal; The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the first vector adder; The isolation end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by the second vector adder and then output; The through ends of the first dual directional coupler and the second dual directional coupler are connected to the orthogonal phase shift network; The orthogonal phase shift network performs orthogonal phase shift processing on the two RF signals at the through end of the first dual directional coupler and the through end of the second dual directional coupler, and inputs the obtained two RF signals with the same amplitude and phase into the two-way in-phase power combiner.

10. A radio frequency power synthesis device, characterized in that: It includes a directional coupling processing device and a two-way co-directional power combiner; The directional coupling processing device includes a first orthogonal phase shift network, a second orthogonal phase shift network, a first dual directional coupler and a second dual directional coupler, a first vector adder and a second vector adder; wherein, The input end of the first dual-directional coupler and the input end of the second dual-directional coupler are respectively used to receive one of the two orthogonal radio frequency signals with the same amplitude; The through-end of the first dual-directional coupler and the through-end of the second dual-directional coupler respectively output the respective one RF signal and respectively receive the reflected signal of the one RF signal; The coupled end outputs of the first dual directional coupler and the second dual directional coupler are output after being subjected to vector addition processing by the first vector adder; The isolation end outputs of the first dual directional coupler and the second dual directional coupler are vector-added by the second vector adder and then output; The input ends of the first dual directional coupler and the second dual directional coupler are connected to a first quadrature phase shift network, and the through ends of the first dual directional coupler and the second dual directional coupler are connected to a second quadrature phase shift network; The first orthogonal phase shift network processes the two same-amplitude and same-phase radio frequency signals into the two same-amplitude orthogonal radio frequency signals; The second orthogonal phase shift network processes the two RF signals from the through-end of the first dual directional coupler and the through-end of the second dual directional coupler into two RF signals with the same amplitude and phase, which are input into the two-way in-phase power combiner.

Images