Broadband operation control method for electrically excited synchronous motor of M3C matrix frequency converter

Abstract

The invention relates to the technical field of multi-level power electronic converters, in particular to a broadband operation control method for an electrically excited synchronous motor of an M3C matrix frequency converter, which comprises the following steps of: establishing a mathematical model of the M3C frequency converter under a double alpha-beta coordinate system for completing alternating current-alternating current bidirectional power conversion of the M3C frequency converter; modeling is carried out on the bridge arm capacitor voltage and the bridge arm power of the M3C frequency converter, and decoupling of energy flow is realized through double alpha-beta conversion; decoupling control is carried out on the M3C frequency converter, and independent control over the input side and the output side of the M3C frequency converter and internal circulation and common-mode voltage of the M3C frequency converter is completed. The method has the advantages that through accurate modeling and decoupling control, the harmonic content is reduced, and the input current harmonic distortion rate can be effectively reduced; output side decoupling control enables the voltage of the motor end to be more stable and sinusoidal, reduces the torque ripple of the motor, and improves the operation stability of the motor; the input side active and reactive decoupling control reduces reactive power transmission, reduces power grid burden and electric energy loss, and enhances electric energy transmission efficiency.

Description

Technical Field

[0001] The present invention relates to the technical field of multi-level power electronic converters, and in particular to a method for controlling wide-band operation of an electrically excited synchronous motor of an M3C matrix inverter. Background Art

[0002] Electrically excited synchronous motors (EESMs), a crucial power plant in the industrial sector, have traditionally been controlled using two-level or three-level inverters. However, these traditional control methods have numerous limitations. Firstly, the topology of two-level and three-level inverters restricts their performance in wide-band operation, resulting in severe harmonic distortion and impacting motor smoothness and efficiency. Secondly, during wide-band operation, traditional control methods place significant stress on switching devices, which not only reduces their lifespan but can also cause system instability.

[0003] The modular multilevel matrix converter (M3C), with its unique high-order modulation capabilities and flexible modular structure, offers a new solution for wide-band operation of electrically excited synchronous motors. M3C's ability to achieve stable operation across a wider frequency range, reduce harmonic distortion, and improve overall system efficiency demonstrates significant potential in the field of electrically excited synchronous motor control, potentially overcoming challenges faced by traditional control methods.

[0004] However, despite the many advantages of M3C, existing control methods for M3C still face some challenges that need to be addressed in wideband operation mode:

[0005] First, voltage balance control is a key issue. During wide-band operation, unit capacitor voltages fluctuate significantly, directly impacting the stability of the power unit. Especially under co-frequency conditions, the DC side voltage is prone to instability, threatening the safe and stable operation of the entire system. For this reason, the maximum operating frequency range of conventional M3C inverters and M3C low-frequency transmission is limited to 0-40Hz, making it difficult to fully utilize the performance advantages of M3C over a wider frequency range.

[0006] Secondly, the problem of circulating current suppression also plagues the application of M3C in wide-band operation mode. The existing control strategy often leads to large circulating current in the same-frequency mode, which not only increases the system loss but also may affect the overall efficiency and stability of the system.

[0007] In summary, both traditional electrically excited synchronous motor control methods and existing M3C control strategies have significant shortcomings in wideband operation, making them unable to meet the demands of modern industrial production for efficient, stable, and wideband operation of motor systems. Therefore, there is an urgent need to further optimize M3C control strategies to improve their performance in wideband operation, fully realize the technical potential of M3C, and expand the application range and operating conditions of electrically excited synchronous motors. Summary of the Invention

[0008] The purpose of the present invention is to provide a wide-band operation control method for an electrically excited synchronous motor of an M3C matrix inverter, thereby reducing the DC voltage fluctuation of the power unit during co-frequency operation and improving system stability; achieving minimum circulating current injection and reducing additional losses; and improving the dynamic and steady-state performance of the frequency conversion system through dual-frequency dq feedforward decoupling control.

[0009] To achieve the above object, the present invention is implemented through the following technical solutions:

[0010] A method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter, specifically comprising:

[0011] S1. Based on the voltages on the three-phase input and three-phase output sides of the M3C inverter, as well as the currents and output voltages of each bridge arm, a mathematical model of the M3C inverter in the dual αβ coordinate system is established using dual αβ coordinate transformation. This model is used to complete AC-AC bidirectional power conversion of the M3C inverter.

[0012] S2. Modeling the bridge arm capacitor voltage and power of the M3C inverter is used to establish a quantitative relationship between the bridge arm capacitor voltage and power, and decoupling the energy flow through double αβ transformation;

[0013] S3. Decoupling control is performed on the M3C inverter to achieve independent control of the input side, output side, internal circulating current, and common mode voltage of the M3C inverter.

[0014] In S1, the corresponding Kirchhoff equations are listed for each bridge arm of the M3C inverter, and their matrix expressions are as follows:

[0015]

[0016] In formula (1), v su , v sv , v sw Indicates the voltage on the input side of the M3C inverter, v ma , v mb , v mc Indicates the output side voltage of the M3C inverter. L indicates that the M3C inverter is connected to the reactor. Represents the current of each bridge arm, Represents the total output voltage of each bridge arm cascade sub-module, v N Indicates the common mode voltage between the neutral points of the three-phase input / three-phase output terminals of the M3C inverter;

[0017] When the M3C inverter completes AC-AC bidirectional power conversion, according to Kirchhoff's current law (KCL);

[0018] The relationship between the current of each bridge arm of the M3C inverter and the phase current of the three-phase input / three-phase output terminal of the M3C inverter is as follows:

[0019]

[0020] In formula (2), i u 、i v 、i w Respectively represent the phase current at the input side of the M3C inverter; i a 、i b 、i c Respectively represent the output side current of the M3C inverter; by regulating the internal circulating current component of the M3C inverter, the capacitance of each bridge arm and the common mode voltage v of the M3C inverter are optimized N and each bridge arm current Collaboratively participate in the internal energy conversion of the M3C inverter;

[0021] The mathematical model of the M3C inverter in the dual αβ coordinate system is established as follows:

[0022] Formula (1) uses double αβ coordinate transformation, and the expression is as follows:

[0023] C dual-αβ0 (G 3*3 )=C αβ0 .G 3*3 .C αβ0 T

[0024] Among them G 3*3 is the matrix to be transformed, C αβ0 . is the Clark transformation matrix, C dual-αβ0 (G 3*3 ) is the result after double αβ transformation;

[0025] The αβ transformation matrix expression is as follows:

[0026]

[0027] Perform double αβ transformation on the M3C frequency conversion system to obtain the M3C frequency conversion model after double αβ transformation, which is expressed as follows:

[0028]

[0029] In formula (4), v sα and v sβ Indicates the voltage v on the input side of the M3C inverter su 、v sv 、v sw The component in the αβ coordinate system, v mα and v mβ Indicates the output voltage v of the M3C inverter ma 、v mb 、v mc The component in the αβ coordinate system, i sα and i sβ Indicates the phase current i on the input side of the M3C inverter u 、i v 、i w The component in the αβ coordinate system, i mα and i mβ Indicates the output current i of the M3C inverter a 、i b 、i c Components in the αβ coordinate system;

[0030] The M3C inverter decoupling model is expressed as four independent equations:

[0031] The input side system equation is as follows:

[0032]

[0033] The output side system equation is as follows:

[0034]

[0035] The circulation system equation is as follows:

[0036]

[0037] The common-mode voltage equation is as follows:

[0038]

[0039] After double conversion of formula (2), the relationship between the phase currents on the input and output sides of the M3C variable frequency system is expressed as follows:

[0040]

[0041] Among them, v sα and v sβ is the input port voltage v su ,v sv ,v swThe component in the αβ coordinate system, v mα and v mβ is the output port voltage v ma , v mb ,v mc The component in the αβ coordinate system, i sα and i sβ is the input port phase current i u ,i v ,i w The component in the αβ coordinate system, i mα and i mβ is the output port current i a ,i b ,i c The components in the αβ coordinate system, Proportional to the input current i sα ,i sβ , Proportional to the output current i mα ,i mβ , It is the circulating voltage and circulating current components in the dual αβ coordinate system, which are decoupled from the input and output side voltages and currents and exist independently;

[0042] The input and output side equations of the M3C variable frequency system are organized into two decoupled models. The relationship between each phase voltage and phase current is shown below:

[0043]

[0044] The combined formulas (7), (8), (11) and (12) are used to obtain the decoupling equivalent model of the input side, output side, four circulating current components and common mode voltage after double αβ transformation.

[0045] In S2, the bridge arm capacitor voltage and bridge arm power of the M3C inverter are modeled as follows:

[0046] The relationship between the bridge arm capacitor voltage and the bridge arm power of the M3C inverter is as follows:

[0047]

[0048] in is the capacitor voltage of each bridge arm, Indicates the capacitor voltage command value, is the power of each bridge arm;

[0049] Expanding formula (13) to the matrix form of the bridge arm capacitor voltage and bridge arm power of the M3C inverter is as follows:

[0050]

[0051] In formula (14), Represents the capacitor voltage command value, C represents the equivalent unit capacitance value, and double αβ transformation is applied to the matrix form of the capacitor voltage-power model to decouple the energy flow between the bridge arms of the M3C inverter. The capacitor voltage-power model containing the capacitor voltage ripple component and DC bias is obtained as shown below:

[0052]

[0053] In formula (15), Indicates the active power flowing into or out of the M3C inverter. and It represents the energy exchange between the three parallel subconverters (subconverter a, subconverter b, subconverter c) from the perspective of the input three-phase AC system (three-phase AC power). and It represents the energy exchange between the three parallel sub-inverters from the perspective of the output three-phase system (output three-phase AC power). and Indicates the power flow inside the corresponding subconverter (subconverter a, subconverter b, subconverter c);

[0054] The power expression of each bridge arm in the dual αβ coordinate system is shown in formula (16-23):

[0055]

[0056]

[0057] Substitute the uncontrolled components of the power expressions of each bridge arm in formula (16-23) into formula (15), and calculate the capacitor voltage ripple of the M3C inverter in the dual αβ coordinate system through trigonometric function:

[0058]

[0059]

[0060] In the formula Indicates the voltage imbalance between the three parallel subconverters (subconverter a, subconverter b, subconverter c) from the perspective of the input three-phase AC system. Indicates the voltage imbalance between the three parallel sub-inverters from the output three-phase AC system. Indicates that the voltage inside the sub-inverter is unbalanced;

[0061] In the process of double αβ transformation, the four frequency fluctuation components in the original abc stationary coordinate system are partially decoupled, including ω s -ω m and ω s +ω m The frequency component of the output side will diverge when the frequency of the input side is equal to the frequency of the output side. and Contains 2ω m The frequency component will diverge when the output frequency is zero; and Contains 2ω s The frequency component will diverge when the input frequency is zero.

[0062] In S3, the decoupling control of the M3C inverter includes the input side control of the M3C inverter. The input side control of the M3C inverter is to convert the input side voltage and current AC quantities of the M3C inverter into the corresponding active and reactive components in the dq coordinate system for control; the control component i of the input side current sα and i sβ The phase-locked loop angle on the input side is converted into a DC value in the dq rotating coordinate system. and And adopt PI controller to perform zero static error control;

[0063] The Park transformation matrix is ​​shown in formula (32):

[0064]

[0065] Please give the formula (32), C dq is the component of the voltage or current vector in the dq coordinate system, and wt is the angle of the dq rotation coordinate system;

[0066] The system voltage on the input side of the M3C inverter after Park transformation is as follows:

[0067]

[0068] In formula (33), and Indicates the bridge arm voltage after double αβ conversion on the input side and Components in the dq coordinate system; and Represents the components of the input side grid voltage in the dq coordinate system; and Represents the component of the input side bridge arm current in the dq coordinate system; active power control is used to maintain the average value of the total capacitor voltage of the 9 bridge arms in the M3C inverter stable, and reactive power control is set to 0;

[0069] In order to achieve decoupling control of the input current of the M3C inverter, the bridge arm voltage in the dq coordinate system is given as follows:

[0070]

[0071] In formula (34), PIs represents the PI controller transfer function of the current loop on the input side of the M3C inverter; and Represents the command value of the input side current in the dq coordinate system; the output of the current loop PI controller is transformed to the αβ coordinate system through the inverse Park transformation to obtain the voltage reference component required for input system control and

[0072] The output system control model after Park transformation is as follows:

[0073]

[0074] In formula (35), and Indicates the bridge arm voltage after double αβ conversion and Expression in dq coordinate system; and Represents the components of the output side voltage in the dq coordinate system; and Represents the components of the output side current in the dq coordinate system;

[0075] In order to complete the decoupling control of the output side current of the M3C inverter, the expression of the given component of the bridge arm voltage in the dq coordinate system is as follows:

[0076]

[0077] In formula (36), PI m Represents the PI controller transfer function matrix of the output current loop control;

[0078] Similarly, the output side motor control introduces the feedforward component of the motor voltage. The output component of the current loop PI controller is converted to the αβ coordinate system through the Park inverse transformation to obtain the voltage reference component required for the motor control output. and

[0079] The decoupling control of the M3C inverter also includes capacitor voltage balance control in the M3C low-frequency-high-frequency mode. The content of capacitor voltage balance control in the M3C low-frequency-high-frequency mode is as follows:

[0080] 1) Input sub-converter imbalance control

[0081] When the common-mode voltage injected into the input side of the M3C inverter is 0, the power ripple expression between the input sub-converters is as follows:

[0082]

[0083] right The components are injected into the following circulation: d1 , I d2 The amplitude of the circulating current injected to balance the input sub-converter voltage;

[0084]

[0085] By simplifying the power-capacitor voltage model, the following control model can be obtained:

[0086]

[0087] 2) Output sub-converter imbalance control

[0088] The power ripple between output sub-converters is given by the following formula:

[0089]

[0090] By injecting a circulating current with the same frequency as the output side of the M3C inverter, and Internally generates DC power;

[0091] right Inject the following circulation: where I d3 , I d4 The amplitude of the circulating current injected to balance the output sub-converter voltage is:

[0092]

[0093] The output side control model is obtained by simplifying the power-capacitor voltage model:

[0094]

[0095] 3) Internal imbalance control of sub-converters

[0096] The power ripple inside each sub-converter in the M3C inverter is as follows:

[0097]

[0098] Injecting a circulating current with the same frequency as the input system generates a DC power flow. When the injected circulating current frequency is the same as the input side frequency of the M3C inverter, the expression of the circulating current system transformed by trigonometric functions is: d5, I d6 The amplitude of the circulating current injected to balance the internal voltage of the balancing sub-converter is:

[0099]

[0100] The power-capacitor voltage model is used to obtain the unbalanced control model inside each sub-converter, which is expressed as follows:

[0101]

[0102]

[0103] The decoupling control of the M3C inverter also includes M3C circulating current control. The circulating current control consists of two or more frequency components. The proportional controller is selected to track and control the circulating current setting. The circulating current control reference signal instruction includes the following three balance control components:

[0104] 1) Sub-converter internal voltage balance control command signal and

[0105] 2) Voltage balance control command signal between sub-converters and

[0106] 3) Differential frequency balance control command signal in the same frequency control mode and

[0107] The output of the circulating current proportional controller is the reference value of the circulating current control voltage of each bridge arm in the M3C inverter in the dual αβ coordinate system. and

[0108] The decoupling control of the M3C inverter also includes the unit voltage suppression control under the M3C same-frequency working condition. The unit voltage suppression control under the M3C same-frequency working condition is as follows:

[0109] Introducing diagonal transformation, its transformation matrix is ​​as follows: Where, C D is the diagonal transformation matrix:

[0110]

[0111] After double αβ transformation and diagonal transformation, the power flow inside the sub-converter is represented by and Convert to p cir1_α 、p cir1_β 、p cir2_α and p cir2_β , further divided into two categories:

[0112] 1) Energy balance in the forward diagonal direction, i.e. p after diagonal transformation cir1_α and p cir1_β ;

[0113] 2) Energy balance in the reverse diagonal direction, that is, p after diagonal transformation cir2_α and p cir2_β ;

[0114] The expression is as follows: cir2_α ,i cir2_β ,i cir1_α ,i cir1_β Circulation After diagonal transformation, the αβ components are:

[0115]

[0116] Substituting the specific expressions of voltage and current into this power component, equations (59) and (60) can be simplified to:

[0117]

[0118] When the M3C inverter meets V S I M =V M I S , and the input and output phases of the M3C inverter meet When the diagonal transformation is decoupled by Park transformation, p cir1_α and p cir1_β Transformed into a DC quantity in the dq coordinate system for control, the angular velocity of the rotating coordinate system is ω s -ω m , combined with the capacitor voltage-power model, the power component expression in the dq coordinate system is as follows:

[0119]

[0120]

[0121] The control of equal frequency condition adopts full control component, i.e. and in and The circulating currents output by the DC voltage loop d-axis and q-axis control loop are given respectively, and the power component p cir1_ α and p cir1_β Convert to dq coordinate system for control, at this time the common mode voltage v N The circulating current component in the dq coordinate system is in phase with the circulating current component in the dq coordinate system, generating adjustable DC power; the circulating current of the injected dq axis is and is a high-frequency signal, and the injected circulating current in the dq coordinate system is and common mode voltage v N The form is as follows:

[0122]

[0123] in is the zero-sequence circulating current component

[0124] Substituting the circulating current and common-mode voltage into equations (69) and (70), we obtain the following expressions:

[0125]

[0126] In the closed-loop control in the dq coordinate system, the voltage control in the dq axis introduces feedforward, and the dq axis feedforward components are shown in Equations (74) and (75):

[0127]

[0128] In formula (74) and formula (75), and They represent the reference values ​​of the low-frequency fluctuation suppression voltage loop output in the dq coordinate system, and Represents the feedforward component of the voltage loop;

[0129] Substituting the feedforward component into the equivalent closed-loop control model, the expression is as follows:

[0130]

[0131] When the voltage frequency on the output side of the M3C inverter is equal to the voltage frequency on the input side, the control is performed using formulas (76) and (77). The PI controller output is proportional to the zero-sequence component f n (t) multiplied, and finally the inverse Park transform was performed to convert i cir1 The circulation is transformed into the dual αβ coordinate system for control.

[0132] The M3C inverter has a 3×3 modular multilevel matrix converter M3C topology, specifically including 9 bridge arms. Each bridge arm includes N sub-module units and a bridge arm inductor. Each sub-module unit includes an H-bridge and a capacitor. Decoupling control is performed on the M3C inverter to achieve independent control of the input side, output side, and internal circulating current and common-mode voltage of the M3C inverter.

[0133] Compared with the prior art, the present invention has the following beneficial effects:

[0134] 1. Through precise modeling and decoupling control, the voltage and current waveforms on the input and output sides are made closer to the ideal state, reducing the harmonic content. For example, on the input side, active and reactive power decoupling control and feedforward control can effectively reduce the input current harmonic distortion rate; the output side dq decoupling control makes the motor terminal voltage more stable and sinusoidal, reduces motor torque pulsation, and improves motor operation stability; the input side active and reactive power decoupling control can stabilize the input power factor at an ideal state close to 1, reduce reactive power transmission, reduce the burden on the grid and power loss, and enhance the efficiency and economy of power transmission;

[0135] 2. Decomposing the complex coupled system of the M3C inverter into independent subsystems such as input, output, circulating current, and common-mode voltage, each subsystem can be designed with a targeted controller, greatly simplifying the control logic and improving control accuracy. For example, the input-side dq decoupling control can achieve independent and precise regulation of the active and reactive components of the input current; the output-side dq decoupling control can accurately control the motor stator voltage, current frequency, and amplitude.

[0136] 3. Circulating current control includes multiple balancing control components, which can accurately suppress circulating current fluctuations caused by factors such as system imbalance and nonlinear load, ensure stable operation of the inverter, and improve the control accuracy and dynamic performance of the electrically excited synchronous motor;

[0137] 4. Bridge arm capacitor voltage-power modeling and a series of control measures, such as capacitor voltage balance control in low-frequency-high-frequency mode, can ensure stable and balanced bridge arm capacitor voltages. This prevents M3C inverter control failures caused by capacitor voltage fluctuations or imbalances, improves inverter reliability and stability, and extends its service life.

[0138] 5. The control method fully considers the characteristics of the M3C inverter under different frequencies, loads and working conditions, and has strong adaptability and robustness to disturbances such as grid voltage fluctuations and load mutations. For example, the unit voltage suppression control under the same frequency working condition can effectively suppress voltage fluctuations under the same frequency working condition, ensuring the stable operation of the inverter under complex working conditions.

[0139] 6. This control method can give full play to the advantages of the M3C matrix inverter, realize the wide-band operation of the electrically excited synchronous motor, expand the motor speed regulation range, meet the requirements of motor speed regulation performance in different industrial scenarios, and improve production efficiency and equipment flexibility;

[0140] 7. The synergistic effect of decoupling control and circulating current control enables the M3C inverter to quickly respond to changes in motor load and torque demand fluctuations, shorten system response time, improve dynamic response speed and control accuracy, and enhance the operating stability and reliability of the motor system under complex working conditions. BRIEF DESCRIPTION OF THE DRAWINGS

[0141] Figure 1This is a structural diagram of the M3C modular multi-level matrix converter.

[0142] Figure 2 It is an equivalent decoupling model in the double αβ coordinate system.

[0143] Figure 3 This is the overall control strategy block diagram for M3C wide frequency stable operation.

[0144] Figure 4 This is the block diagram of the M3C input side control strategy.

[0145] Figure 5 This is the block diagram of the M3C output side control strategy.

[0146] Figure 6 This is the block diagram of the M3C low-frequency / high-frequency operating control strategy.

[0147] Figure 7 This is the unit voltage suppression control block diagram under M3C same-frequency working conditions.

[0148] Figure 8 This is a schematic diagram of the M3C motor test platform.

[0149] Figure 9 It is the voltage and current waveform at the input side of the M3C converter.

[0150] Figure 10 It is the voltage and current waveform of the output side of the M3C converter (wideband 0-50Hz).

[0151] Figure 11 It is the current waveform of the 9th bridge arm of the M3C converter (0-50Hz).

[0152] Figure 12 Motor speed and motor torque waveforms (wideband 0-50Hz).

[0153] Figure 13 This is the waveform (0-50Hz) of an M3C 10kV inverter driving an electrically excited synchronous motor with load and wide frequency operation.

[0154] Figure 14 It is the voltage and current waveform at the input side of the M3C converter.

[0155] Figure 15 It is the voltage and current waveform of the output side of the M3C converter (120Hz).

[0156] Figure 16 It is the current waveform of the bridge arm of the M3C converter 9 (output frequency 120Hz).

[0157] Figure 17 It is the DC side voltage waveform of the M3C converter (output frequency 120Hz). DETAILED DESCRIPTION

[0158] The present invention will be described in detail below with reference to the accompanying drawings, but it should be noted that the implementation of the present invention is not limited to the following embodiments.

[0159] The following examples are implemented under the premise of the technical solution of the present invention, and provide detailed implementation methods and specific operating processes, but the scope of protection of the present invention is not limited to the following examples. The methods used in the following examples are conventional methods unless otherwise specified.

[0160] Example 1

[0161] See Figure 1 The M3C inverter adopts an H-bridge cascaded modular multi-level structure. The three-phase input side and the three-phase output side of the M3C inverter are connected by 9 bridge arms in a 3×3 pattern, realizing AC-AC bidirectional power conversion. Compared with the traditional AC-DC-AC conversion method, it reduces the intermediate DC link and improves energy transmission efficiency. During operation, the M3C inverter can ensure:

[0162] 1) The current on the power supply side (the three-phase input side of the M3C inverter) and the motor side (the three-phase output side of the M3C inverter) both maintain a sinusoidal waveform;

[0163] 2) It has the ability to adjust the power factor to suit different application requirements;

[0164] 3) The circuit parameters of each bridge arm branch are the same, ensuring balanced power distribution among modules and improving system reliability.

[0165] A method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter, specifically comprising:

[0166] S1. Based on the voltages on the three-phase input and three-phase output sides of the M3C inverter, as well as the currents and output voltages of each bridge arm, a mathematical model of the M3C inverter in the dual αβ coordinate system is established using dual αβ coordinate transformation. This model is used to complete AC-AC bidirectional power conversion of the M3C inverter.

[0167] The matrix expressions of Kirchhoff equations corresponding to each bridge arm of the M3C inverter are as follows:

[0168]

[0169] In formula (1), v su , v sv , v sw Indicates the voltage on the input side of the M3C inverter, v ma , v mb , v mc Indicates the output side voltage of the M3C inverter. L indicates that the M3C inverter is connected to the reactor. Represents the current of each bridge arm, Represents the total output voltage of each bridge arm cascade sub-module, v N Indicates the common mode voltage between the neutral points of the three-phase input / three-phase output terminals of the M3C inverter;

[0170] The M3C inverter completes AC-AC bidirectional power conversion. According to Kirchhoff's current law (KCL), the relationship between the current of each bridge arm of the M3C inverter and the phase current of the three-phase input / three-phase output terminal of the M3C inverter is as follows:

[0171]

[0172] In formula (2), i u 、i v 、i w Respectively represent the phase current at the input side of the M3C inverter; i a 、i b 、i c Respectively represent the output side current of the M3C inverter; by controlling the internal circulating current component of the M3C inverter, the capacitance and voltage control of each bridge arm of the M3C inverter are optimized, and the common mode voltage v N and each bridge arm current Used for internal energy conversion of M3C inverter;

[0173] From the number of variables in the equation group of formula (2), it can be seen that even if the input and output side currents of the M3C inverter are determined, the current state of the 9 bridge arms cannot be determined. Therefore, there is a circulating current between the M3C sub-inverter and the sub-inverter. The circulating current does not affect the input and output side phase currents, but can realize the internal energy conversion of the M3C inverter. Therefore, the control of the capacitor voltage of each bridge arm of the system can be optimized by controlling the internal circulating current component of the M3C inverter. In addition, the common mode voltage v N and bridge arm current The function can also realize the internal energy conversion of the inverter. Therefore, this patent provides a wide-band control scheme based on the internal circulating current and common-mode voltage as the control freedom, as follows:

[0174] Formula (1) uses double αβ coordinate transformation: C dual-αβ0 (G 3*3 )=C αβ0 .G 3*3 .C αβ0 T

[0175] Among them G 3*3 is the matrix to be transformed, C αβ0 . is the Clark transformation matrix, C dual-αβ0 (G 3*3) is the result after double αβ transformation;

[0176] Among them, the αβ transformation matrix expression is as follows:

[0177]

[0178] Perform double αβ transformation on the M3C frequency conversion system to obtain the M3C frequency conversion model after double αβ transformation, which is expressed as follows:

[0179]

[0180] In formula (4), v sα and v sβ Indicates the voltage v on the input side of the M3C inverter su 、v sv 、v sw The component in the αβ coordinate system, v mα and v mβ Indicates the output voltage v of the M3C inverter ma 、v mb 、v mc The component in the αβ coordinate system, i sα and i sβ Indicates the phase current i on the input side of the M3C inverter u 、i v 、i w The component in the αβ coordinate system, i mα and i mβ Indicates the output current i of the M3C inverter a 、i b 、i c Components in the αβ coordinate system; the double αβ transformation realizes the decoupling of the two three-phase AC systems on the input and output sides of the M3C inverter;

[0181] The M3C inverter decoupling model is expressed as four independent equations:

[0182] The input side system equation is as follows:

[0183]

[0184] The output side system equation is as follows:

[0185]

[0186] The circulation system equation is as follows:

[0187]

[0188] The common-mode voltage equation is as follows:

[0189]

[0190] Performing double αβ transformation on formula (2) yields the relationship between the input and output phase currents of the M3C variable frequency system in the double αβ coordinate system. The expression is as follows:

[0191]

[0192] Among them, v sα and v sβ is the input port voltage v su ,v sv ,v sw The component in the αβ coordinate system, v mα and v mβ is the output port voltage v ma , v mb ,v mc The component in the αβ coordinate system, i sα and i sβ is the input port phase current i u ,i v ,i w The component in the αβ coordinate system, i mα and i mβ is the output port current i a ,i b ,i c The components in the αβ coordinate system, Proportional to the input current i sα ,i sβ , Proportional to the output current i mα ,i mβ , It is the circulating voltage and circulating current components in the dual αβ coordinate system, which are decoupled from the input and output side voltages and currents and exist independently;

[0193] The input and output side equations of the M3C variable frequency system are organized into two decoupled models. The relationship between each phase voltage and phase current is shown as follows:

[0194]

[0195] The combined formulas (7), (8), (11) and (12) are used to obtain the decoupling equivalent model of the input side, output side, four circulating current components and common mode voltage after double αβ transformation.

[0196] S2. Modeling the bridge arm capacitor voltage and power of the M3C inverter is used to establish a quantitative relationship between the bridge arm capacitor voltage and power, and decoupling the energy flow through double αβ transformation;

[0197] The relationship between the bridge arm suspension capacitor voltage and the bridge arm power of the M3C inverter is as follows:

[0198]

[0199] in is the capacitor voltage of each bridge arm, Indicates the capacitor voltage command value, is the power of each bridge arm;

[0200] Expanding formula (13) to the matrix form of the capacitor voltage and bridge arm power of the M3C inverter is as follows:

[0201]

[0202] In formula (14), Represents the capacitor voltage command value, C represents the equivalent unit capacitance value, and double αβ transformation is applied to the matrix form of the capacitor voltage-power model to decouple the energy flow between the bridge arms of the M3C inverter. The capacitor voltage-power model containing the capacitor voltage ripple component and DC bias is obtained as shown below:

[0203]

[0204] In formula (15), Indicates the active power flowing into or out of the M3C inverter. and It represents the energy exchange between the three parallel subconverters (subconverter a, subconverter b, subconverter c) from the perspective of the input three-phase AC system (three-phase AC power). and It represents the energy exchange between the three parallel sub-inverters from the perspective of the output three-phase system (output three-phase AC power). and Indicates the power flow inside the corresponding subconverter (subconverter a, subconverter b, subconverter c);

[0205] The power components of the M3C bridge arm can be divided into three categories:

[0206] 1) Uncontrollable Component: This component is entirely caused by the three-phase AC voltage and current on the input and output sides. It does not change with changes in the circulating current and common-mode voltage of the M3C inverter's internal control variables. It reflects the power flow and balance between the two three-phase AC systems. The uncontrollable component also indicates that the exchange of active power in the M3C inverter can cause energy pulsation within the bridge arm, which in turn causes fluctuations in the unit capacitor voltage.

[0207] 2) Semi-controlled component: Semi-controlled components are generated by the controllable circulating current or common-mode voltage of the M3C inverter and the voltage and current of the three-phase system on the input and output sides of the M3C converter. When the input and output systems are determined, only the internal circulating current or common-mode voltage can be controlled, so this component belongs to the semi-controlled component.

[0208] 3) Fully controlled component: The fully controlled component is entirely composed of the internal control quantity circulating current and common mode voltage of the M3C inverter. Both can be freely configured. It is a fully controllable component and its control is completely independent of the input and output systems.

[0209] The power expression of each bridge arm in the dual αβ coordinate system is shown in formula (16-23):

[0210]

[0211] Substitute the uncontrolled components of the power expressions of each bridge arm in formula (16-23) into formula (15), and calculate the capacitor voltage ripple of the M3C inverter in the dual αβ coordinate system through trigonometric function:

[0212] Substitute the uncontrolled components of the power expressions of each bridge arm in formula (16-23) into formula (15), and calculate the capacitor voltage ripple of the M3C inverter in the dual αβ coordinate system through trigonometric function:

[0213]

[0214]

[0215] In the formula Indicates the voltage imbalance between the three parallel subconverters (subconverter a, subconverter b, subconverter c) from the perspective of the input three-phase AC system. Indicates the voltage imbalance between the three parallel sub-inverters from the output three-phase AC system. Indicates that the voltage inside the sub-inverter is unbalanced;

[0216] In the process of double αβ transformation, the four frequency fluctuation components in the original abc stationary coordinate system are partially decoupled, including ω s -ω m and ω s +ω m The frequency component of the output side will diverge when the frequency of the input side is equal to the frequency of the output side. and Contains 2ω m The frequency component will diverge when the output frequency is zero; and Contains 2ω sThe frequency component will diverge when the input frequency is zero.

[0217] Since the input frequency of the power system cannot be zero, and the constant power load and constant torque load on the motor side have a small modulation degree in the low frequency band, there is no 2ω m and 2ω s To avoid the risk of voltage instability in the low-frequency band system unit, the present invention converts the difference frequency component (output system frequency and input system frequency) in the motor control system into s -ω m The working condition of |<5Hz is the key research content, which is called same-frequency control in this invention.

[0218] S3. Decoupling control is performed on the M3C inverter to complete independent control of the input side, output side and internal circulating current and common-mode voltage of the M3C inverter; further decoupling of the circulating current model is achieved through diagonal transformation. At the same time, analysis of the DC side capacitor voltage fluctuation after diagonal transformation shows that the diagonal transformation decomposes the unbalanced components of the 8 capacitor voltages into four dimensions: horizontal, vertical, diagonal and anti-diagonal. The physical meaning is very clear. Based on the diagonal transformation, a capacitor voltage fluctuation suppression vector model and the corresponding vector control method are given; based on the M3C bridge arm capacitor voltage zero-static difference balancing strategy, the decoupling of the circulating current model is completed through diagonal transformation. At the same time, after the diagonal transformation, the diagonal transformation is completed through analysis of the DC side capacitor voltage fluctuation of the M3C inverter, and the unbalanced components of the 8 capacitor voltages are decomposed into four dimensions: horizontal, vertical, diagonal and anti-diagonal. Based on the diagonal transformation, a capacitor voltage fluctuation suppression vector model and the corresponding vector control are given; the overall control strategy block diagram for M3C wide-frequency stable operation is shown in Figure 3 It is a system hierarchical control based on double αβ transformation and diagonal transformation, including the input and output system control of the M3C inverter, capacitor voltage balance control and power fluctuation suppression under the same frequency working condition; the DC side voltage control loop output of the M3C inverter is used as the reference value of the current loop for circulating current control, and the voltage reference of each bridge arm in the M3C inverter is obtained to perform carrier phase shift modulation and intra-phase unit balance control.

[0219] Through double αβ transformation, M3C can obtain equivalent decoupled models of the input and output systems. The parameters and control variables of the two do not affect each other, so they can be independently designed and controlled according to the application scenarios of the input and output sides. First, the control strategy of the input system is introduced. The control of the M3C input side mainly realizes adjustable power factor grid connection and total capacitor voltage stability, and performs reactive power compensation and voltage support when the input system requires it. The input side control of the M3C inverter includes total active power control and grid reactive power control. Active power control is used to achieve total active power balance, while reactive power control completes the input system power command management while optimizing the circulating current size of the same frequency control.

[0220] The output-side control of the M3C inverter includes flux control and torque decoupling control, which decomposes the stator current into an excitation current component that generates a magnetic field and a torque current component that generates torque. By detecting the rotor position and speed signals of the motor, the d-axis is oriented in the direction of the rotor flux, so that the excitation current component is related to the rotor flux, and the torque current component is related to the electromagnetic torque, thus completing the decoupling control of the magnetic field and torque.

[0221] The nine bridge arm currents of the M3C can be decomposed into input current, circulating current, and output current through double αβ transformation. Therefore, the current loop design can also be divided into input current control, output current control, and circulating current control.

[0222] The main purpose of capacitor voltage balance control is to ensure that the average DC component of the capacitor is stable at a given value and to generate given values ​​of four internal circulating currents according to the unbalanced amount of the capacitor voltage.

[0223] The voltage fluctuation suppression under the same frequency condition is based on double αβ transformation and diagonal transformation, which further decouples the four frequency components of the M3C circulating current. cir1_α and p cir1_β To suppress the violent ω under the equal frequency condition s -ω m Frequency low frequency pulsation;

[0224] The decoupling control of the M3C inverter includes the input side control of the M3C inverter, which is to convert the AC voltage and current of the input side of the M3C inverter into the corresponding active and reactive components in the dq coordinate system for control; the control component i of the input side current sα and i sβ The phase-locked loop angle of the input system is converted into a DC quantity in the dq rotating coordinate system and And adopt PI controller to perform zero static error control;

[0225] The Park transformation matrix is ​​shown in formula (32):

[0226]

[0227] Please give the formula (32), C dq is the component of the voltage or current vector in the dq coordinate system, and wt is the angle of the dq rotation coordinate system;

[0228] The system voltage on the input side of the M3C inverter after Park transformation is as follows:

[0229]

[0230] In formula (33), and Indicates the bridge arm voltage after double αβ conversion and Components in the dq coordinate system; active power control is used to maintain the average value of the total capacitor voltage of the 9 bridge arms in the M3C inverter, and reactive power control is set to 0; because the M3C inverter has the ability to achieve reactive power control at any power factor, the circulating current size of the same frequency control can be optimized through reactive power control; the control block diagram of the M3C inverter input system is shown in Figure 4;

[0231] In order to realize the decoupling control of the input current of the M3C inverter, the bridge arm voltage in the dq coordinate system is given as shown in formula (34):

[0232]

[0233] In formula (34), PIs represents the PI controller transfer function of the current loop on the input side of the M3C inverter; and Represents the command value of the input side current in the dq coordinate system; the output of the current loop PI controller is transformed to the αβ coordinate system through the inverse Park transformation to obtain the voltage reference component required for input system control and

[0234] The feedforward component of the input side grid voltage is introduced into the input side control of the M3C inverter, which can reduce the influence of the input side grid voltage disturbance. The output of the current loop PI controller is transformed to the αβ coordinate system through the inverse Park transformation to obtain the voltage reference component required for input system control. and

[0235] The output system control model after Park transformation is shown in formula (35):

[0236]

[0237] In formula (35), and Indicates the bridge arm voltage after double αβ conversion and Expression in dq coordinate system;

[0238] See Figure 5 The motor control strategy block diagram in the dq coordinate system of the output side (the output side refers to the output side of the M3C inverter) is given. The control of the output current loop is similar to that of the input current loop. The d-axis and q-axis given components of the current loop output are obtained by the motor control algorithm. In the dq coordinate system, the output current d-axis component i md Adjust the motor rotor flux ψ r , through the output current q-axis component i mq Adjust the motor torque T m .

[0239] In order to complete the decoupling control of the output side current of the M3C inverter, the expression of the given component of the bridge arm voltage in the dq coordinate system is as follows:

[0240]

[0241] In formula (36), PI m Represents the PI controller transfer function matrix of the output current loop control;

[0242] Similarly, the output side motor control introduces the feedforward component of the motor voltage. The output component of the current loop PI controller is converted to the αβ coordinate system through the Park inverse transformation, which is the voltage given component required for the motor control output. and

[0243] M3C inverter decoupling control also includes capacitor voltage balance control in M3C low frequency-high frequency mode.

[0244] Under low-frequency or high-frequency output conditions, the power distribution between the bridge arms of M3C is uniform, so the voltage fluctuation is small. Due to the inherent integral characteristics of the capacitor element, any slight DC offset in the bridge arm current will cause the capacitor voltage of each unit in the bridge arm to diverge. The DC bias of the capacitor voltage of the bridge arm of the M3C inverter is controlled to stabilize the capacitor voltage during low-frequency or high-frequency operation. The control freedom of M3C includes two types: internal circulating current and common-mode voltage. The voltage balance control method based on the injection of bridge arm circulating current is discussed. First, the decoupled capacitor voltage is low-pass filtered, and the DC bias instruction to be suppressed is obtained through the DC side voltage closed-loop control. In order to offset these DC quantities, a specific form of circulating current can be injected to generate a DC power component in the opposite direction for suppression. The block diagram of the capacitor balance control strategy in the M3C low-frequency-high-frequency mode is shown in Fig. Figure 6 ; The capacitor voltage balance control in M3C low-frequency-high-frequency mode specifically includes:

[0245] 1) Input sub-converter imbalance control

[0246] When the common-mode voltage injected into the input side of the M3C inverter is 0, the power ripple expression between the input sub-converters is as follows:

[0247]

[0248] right The components are injected into the following circulation: d1 , I d2 The amplitude of the circulating current injected to balance the input sub-converter voltage;

[0249]

[0250] By simplifying the power-capacitor voltage model, the following control model can be obtained:

[0251]

[0252] 2) Output sub-converter imbalance control

[0253] The power ripple between output sub-converters is given by the following formula:

[0254]

[0255] By injecting a circulating current with the same frequency as the output side of the M3C inverter, and Internally generates DC power;

[0256] right Inject the following circulation: where I d3 , I d4 The amplitude of the circulating current injected to balance the output sub-converter voltage is:

[0257]

[0258] The output side control model is obtained by simplifying the power-capacitor voltage model:

[0259]

[0260] 3) Internal imbalance control of sub-converters

[0261] The power ripple inside each sub-converter in the M3C inverter is as follows:

[0262]

[0263] Injecting a circulating current with the same frequency as the input system generates a DC power flow. When the injected circulating current frequency is the same as the input side frequency of the M3C inverter, the expression of the circulating current system transformed by trigonometric functions is: d5 , I d6 The amplitude of the circulating current injected to balance the internal voltage of the balancing sub-converter is:

[0264]

[0265] The power-capacitor voltage model is used to obtain the unbalanced control model inside each sub-converter, which is expressed as follows:

[0266]

[0267] M3C inverter decoupling control also includes M3C circulating current control, which is composed of two or more frequency components. A proportional controller is selected to track and control the circulating current setting.

[0268] The circulating current control reference signal command includes the following three balance control components:

[0269] 1) Sub-converter internal voltage balance control command signal and

[0270] 2) Voltage balance control command signal between sub-converters and

[0271] 3) Differential frequency balance control command signal in the same frequency control mode and

[0272] The output of the circulating current proportional controller is the reference value of the circulating current control voltage of each bridge arm in the M3C inverter in the dual αβ coordinate system. and

[0273] The decoupling control of the M3C inverter also includes the unit voltage suppression control under the M3C same frequency working condition, because after the double αβ conversion and The component is still coupled to ω s -ω m and ω s +ω m The energy balance mechanism of the two frequency components cannot be directly analyzed, so a diagonal transformation is introduced, and the transformation matrix is ​​shown as follows:

[0274]

[0275] After double αβ transformation and diagonal transformation, the power flow inside the sub-converter is represented by and Convert to p cir1_α 、p cir1_β 、p cir2_α and p cir2_β , further divided into two categories:

[0276] 1) Energy balance in the forward diagonal direction: p after diagonal transformation cir1_α and p cir1_β ;

[0277] 2) Energy balance in the reverse diagonal direction: p after diagonal transformation cir2_α and p cir2_β ;

[0278] The expression is as follows:

[0279]

[0280]

[0281] When running under equal frequency conditions, ω s -ω m The direct reason for the instability of the frequency power component is that the uncontrollable component in the power component has become a low-frequency or even DC pulsating component. Substituting the specific expressions of the voltage and current of the M3C input and output system into this part of the power component, formulas (59) and (60) can be simplified to:

[0282]

[0283] When the M3C inverter meets V S I M =V M I S , and the input and output phases of the M3C inverter meet When there is a special working state, which makes p cir1_α and p cir1_β The uncontrollable component in the frequency converter is zero. At this time, even if the input and output system frequencies are equal, the M3C inverter can operate stably without additional control strategies. This patent uses this as an optimization condition to reduce the circulating current. On this basis, a general low-frequency pulsation suppression method for equal-frequency working conditions is proposed. The p after diagonal transformation decoupling is converted into cir1_α and p cir1_β Transformed into a DC quantity in the dq coordinate system for control, the angular velocity of the rotating coordinate system is ω s -ω m , combined with the capacitor voltage-power model, the power component expression in the dq coordinate system is as follows:

[0284]

[0285] When full control component is used for control, the circulating current i cir1 and common-mode voltage. Since both the circulating current and the common-mode voltage are controllable degrees of freedom, they can be set arbitrarily. It is only necessary to ensure that the product of the two in the dq coordinate system contains a DC component. The circulating current can be injected in the form of high frequency to reduce the impact on other power components of the system. The control of the equal-frequency working condition adopts the full control component, i.e. and in and The circulating currents output by the DC voltage loop d-axis and q-axis control loop are given respectively, and the power component p cir1_ α and p cir1_β Convert to dq coordinate system for control, then the common mode voltage v NThe circulating current component in the dq coordinate system is in phase with the circulating current component in the dq coordinate system, generating adjustable DC power; the circulating current of the injected dq axis is and It is a high frequency signal, so the additional semi-controllable component introduced has a small fluctuation, and the circulating current injected in the dq coordinate system and common mode voltage v N The form is as follows:

[0286]

[0287] f n (t) and h(t) are both relatively high-frequency waveforms No matter what algebraic operation is performed on the fully controlled component and the input and output frequency components of the M3C inverter, no low-frequency power fluctuation will be generated. Substituting the circulating current and common-mode voltage into equations (69) and (70), the following expressions are obtained:

[0288]

[0289] In the closed-loop control in the dq coordinate system, feedforward is introduced into the voltage control in the dq axis to improve the rapidity of the system; the dq axis feedforward components are shown in Equations (74) and (75):

[0290]

[0291] In formula (74) and formula (75), and They represent the reference values ​​of the low-frequency fluctuation suppression voltage loop output in the dq coordinate system, and Represents the feedforward component of the voltage loop;

[0292] Substituting the feedforward component into the equivalent closed-loop control model, the expression is as follows:

[0293]

[0294] The low-frequency power fluctuation suppression control block diagram under the same frequency working condition is shown in Figure 7 When the voltage frequency on the output side of the M3C inverter is equal to the voltage frequency on the input side, equations (76) and (77) are used for control. The influence of disturbances in the control loop is reduced by introducing uncontrollable component feedforward. The output of the PI controller is related to the zero-sequence component f n (t) multiplied, and finally the inverse Park transform was performed to convert i cir1 The circulation is transformed into the dual αβ coordinate system for control.

[0295] Example 2

[0296] In this embodiment, a wide-band operation control method for an electrically excited synchronous motor using an M3C matrix inverter is the same as that in embodiment 1, with an M3C-based electrically excited synchronous motor control application added thereto.

[0297] In order to conduct the M3C full-power inverter driving electric excitation high-voltage synchronous motor test, the 10kV4MWM3C full-power inverter (hereinafter referred to as the M3C inverter) provides a test driving platform. The platform adopts a 4MW salient pole electric excitation synchronous motor + 1MW asynchronous motor driving scheme, which can realize high-voltage load testing of CHB inverters, M3C inverters, SFC inverters and other equipment. It fully verifies the M3C motor (M3C motor refers to 4MW salient pole electric excitation synchronous motor + 1MW asynchronous motor) vector control and M3C wide-band operation control related technologies. The main wiring diagram of the control platform is as follows Figure 9-12 :

[0298] The rated input voltage of the M3C inverter is 10kV, the input frequency is 50Hz, the output rated voltage is 10kV, and the output frequency is 0-50Hz. The motor (the motor here refers to the M3C motor) constant torque load starting and steady-state operation test is carried out. The rated power is 4MW. Figure 13 The waveforms of the M3C inverter's input and output voltages and currents, the M3C inverter's 9-arm current, the M3C inverter's DC-side voltage, and the M3C motor's speed and torque are presented. These waveforms show that the half-peak fluctuation of the DC-side voltage is less than 20V during high-frequency operation. Wide-band load operation verifies the effectiveness of the M3C inverter's circulating current injection (low frequency f < 40Hz) and coordinated zero-sequence voltage and circulating current injection control strategies (same frequency 40Hz ≤ f < 55Hz) under wide-band operating conditions.

[0299] The rated input voltage of the M3C inverter is 10kV, the input frequency is 50Hz, the output rated voltage is 10kV, and the output frequency is 120Hz; Figure 14-17 The voltage and current waveforms of the input side, output side, arm 9, and DC side voltage of the M3C inverter are given. It can be seen from the waveforms that the half-peak fluctuation of the DC side voltage is less than 20V when the M3C inverter is running, which verifies the effectiveness of the control strategy under the high-frequency output condition of the M3C inverter.

[0300] Conclusion: The present invention reduces the DC voltage fluctuation of the power unit when the M3C inverter is operating at the same frequency through a closed-loop comprehensive control strategy of tertiary zero-sequence voltage and tertiary circulating current; minimizes the injected circulating current during the same frequency operation by collaboratively controlling the power factor of the M3C inverter input side to be equal to the power factor of the electrically excited synchronous motor, thereby reducing additional losses; and realizes independent regulation of active power and reactive power through dq coordinate transformation of different rotation angular frequencies on the input side and output side (i.e., the motor side) of the M3C inverter, while decoupling the independent control of magnetic flux and torque, which not only ensures the steady-state difference of the closed-loop control under wide-band operation of the M3C inverter, but also makes it easier to achieve high dynamic performance of the frequency conversion system.

[0301] Through precise modeling and decoupling control, the present invention makes the voltage and current waveforms on the input and output sides closer to the ideal state and reduces the harmonic content. For example, on the input side, active and reactive decoupling control and feedforward control can effectively reduce the input current harmonic distortion rate; the output side dq decoupling control makes the motor terminal voltage more stable and sinusoidal, reduces the motor torque pulsation, and improves the motor operation stability; the input side active and reactive decoupling control can stabilize the input power factor at an ideal state close to 1, reduce reactive power transmission, reduce the grid burden and power loss, and enhance the power transmission efficiency and economy; the complex coupling system of the M3C inverter is decomposed into independent subsystems such as input, output, circulating current and common mode voltage. Each subsystem can be designed with a targeted controller, which greatly simplifies the control logic and improves the control accuracy. For example, input-side dq decoupling control enables independent and precise regulation of the active and reactive components of the input current. Output-side dq decoupling control precisely controls the motor stator voltage, current frequency, and amplitude. Circulating current control, which includes multiple balancing control components, accurately suppresses circulating current fluctuations caused by system imbalances, nonlinear loads, and other factors, ensuring stable inverter operation and improving the control accuracy and dynamic performance of electrically excited synchronous motors. Bridge arm capacitor voltage-power modeling and a series of control measures, such as capacitor voltage balancing control in low-frequency-high-frequency mode, ensure stable and balanced bridge arm capacitor voltages. This prevents M3C inverter control failures caused by capacitor voltage fluctuations or imbalances, improves inverter reliability and stability, and extends its service life. The control method fully considers the characteristics of the M3C inverter under different frequencies, loads, and operating conditions, and exhibits strong adaptability and robustness to disturbances such as grid voltage fluctuations and sudden load changes. For example, unit voltage suppression control under equal-frequency conditions can effectively suppress voltage fluctuations under equal-frequency conditions, ensuring stable operation of the inverter under complex working conditions. This control method can give full play to the advantages of the M3C matrix inverter, realize wide-band operation of the electrically excited synchronous motor, expand the motor speed regulation range, meet the requirements of motor speed regulation performance in different industrial scenarios, and improve production efficiency and equipment flexibility. The synergistic effect of decoupling control and circulating current control enables the M3C inverter to quickly respond to changes in motor load and torque demand fluctuations, shorten the system response time, improve dynamic response speed and control accuracy, and enhance the operating stability and reliability of the motor system under complex working conditions.

Claims

1. A method for controlling the wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter, characterized in that: Specifically include: S1. Based on the voltages on the three-phase input and three-phase output sides of the M3C inverter, as well as the currents and output voltages of each bridge arm, a mathematical model of the M3C inverter in the dual αβ coordinate system is established using dual αβ coordinate transformation. This model is used to complete AC-AC bidirectional power conversion of the M3C inverter. S2. Modeling the bridge arm capacitor voltage and power of the M3C inverter is used to establish a quantitative relationship between the bridge arm capacitor voltage and power, and decoupling the energy flow through double αβ transformation; S3. Decoupling control is performed on the M3C inverter to achieve independent control of the input side, output side, internal circulating current, and common mode voltage of the M3C inverter.

2. The method for controlling the wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 1, wherein: In S1, the corresponding Kirchhoff equations are listed for each bridge arm of the M3C inverter, and their matrix expressions are as follows: In formula (1), v su , v sv , v sw Indicates the voltage on the input side of the M3C inverter, v ma , v mb , v mc Indicates the output side voltage of the M3C inverter. L indicates that the M3C inverter is connected to the reactor. Represents the current of each bridge arm, Represents the total output voltage of each bridge arm cascade sub-module, v N Indicates the common mode voltage between the neutral points of the three-phase input / three-phase output terminals of the M3C inverter; When the M3C inverter completes AC-AC bidirectional power conversion, according to Kirchhoff's current law (KCL); The relationship between the current of each bridge arm of the M3C inverter and the phase current of the three-phase input / three-phase output terminal of the M3C inverter is as follows: In formula (2), i u 、i v 、i w Respectively represent the phase current at the input side of the M3C inverter; i a 、i b 、i c Respectively represent the output side current of the M3C inverter; by regulating the internal circulating current component of the M3C inverter, the capacitance of each bridge arm and the common mode voltage v of the M3C inverter are optimized N and each bridge arm current Collaboratively participate in the internal energy conversion of the M3C inverter; The mathematical model of the M3C inverter in the dual αβ coordinate system is established as follows: Formula (1) uses double αβ coordinate transformation, and the expression is as follows: C dual-αβ0 (G 3*3 )=C αβ0 .G 3*3 .C αβ0 T Among them G 3*3 is the matrix to be transformed, C αβ0 . is the Clark transformation matrix, C dual-αβ0 (G 3*3 ) is the result after double αβ transformation; The αβ transformation matrix expression is as follows: Perform double αβ transformation on the M3C frequency conversion system to obtain the M3C frequency conversion model after double αβ transformation, which is expressed as follows: In formula (4), v sα and v sβ Indicates the voltage v on the input side of the M3C inverter su 、v sv 、v sw The component in the αβ coordinate system, v mα and v mβ Indicates the output voltage v of the M3C inverter ma 、v mb 、v mc The component in the αβ coordinate system, i sα and i sβ Indicates the phase current i on the input side of the M3C inverter u 、i v 、i w The component in the αβ coordinate system, i mα and i mβ Indicates the output current i of the M3C inverter a 、i b 、i c Components in the αβ coordinate system; The M3C inverter decoupling model is expressed as four independent equations: The input side system equation is as follows: The output side system equation is as follows: The circulation system equation is as follows: The common-mode voltage equation is as follows: After double conversion of formula (2), the relationship between the phase currents on the input and output sides of the M3C variable frequency system is expressed as follows: Among them, v sα and v sβ is the input port voltage v su ,v sv ,v sw The component in the αβ coordinate system, v mα and v mβ is the output port voltage v ma , v mb ,v mc The component in the αβ coordinate system, i sα and i sβ is the input port phase current i u ,i v ,i w The component in the αβ coordinate system, i mα and i mβ is the output port current i a ,i b ,i c The components in the αβ coordinate system, Proportional to the input current i sα ,i sβ , Proportional to the output current i mα ,i mβ , It is the circulating voltage and circulating current components in the dual αβ coordinate system, which are decoupled from the input and output side voltages and currents and exist independently; The input and output side equations of the M3C variable frequency system are organized into two decoupled models. The relationship between each phase voltage and phase current is shown below: The combined formulas (7), (8), (11) and (12) are used to obtain the decoupling equivalent model of the input side, output side, four circulating current components and common mode voltage after double αβ transformation.

3. The method for controlling the wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 1, wherein: In S2, the bridge arm capacitor voltage and bridge arm power of the M3C inverter are modeled as follows: The relationship between the bridge arm capacitor voltage and the bridge arm power of the M3C inverter is as follows: in is the capacitor voltage of each bridge arm, Indicates the capacitor voltage command value, is the power of each bridge arm; Expanding formula (13) to the matrix form of the bridge arm capacitor voltage and bridge arm power of the M3C inverter is as follows: In formula (14), Represents the capacitor voltage command value, C represents the equivalent unit capacitance value, and double αβ transformation is applied to the matrix form of the capacitor voltage-power model to decouple the energy flow between the bridge arms of the M3C inverter. The capacitor voltage-power model containing the capacitor voltage ripple component and DC bias is obtained as shown below: In formula (15), Indicates the active power flowing into or out of the M3C inverter. and It represents the energy exchange between the three parallel sub-converters (subconverter a, subconverter b, subconverter c) from the perspective of the input three-phase AC system (three-phase AC power). and It represents the energy exchange between the three parallel sub-inverters from the perspective of the output three-phase system (output three-phase AC power). and Indicates the power flow inside the corresponding subconverter (subconverter a, subconverter b, subconverter c); The power expression of each bridge arm in the dual αβ coordinate system is shown in formula (16-23): Substitute the uncontrolled components of the power expressions of each bridge arm in formula (16-23) into formula (15), and calculate the capacitor voltage ripple of the M3C inverter in the dual αβ coordinate system through trigonometric function: In the formula Indicates the voltage imbalance between the three parallel subconverters (subconverter a, subconverter b, subconverter c) from the perspective of the input three-phase AC system. Indicates the voltage imbalance between the three parallel sub-inverters from the output three-phase AC system. Indicates that the voltage inside the sub-inverter is unbalanced; In the process of double αβ transformation, the four frequency fluctuation components in the original abc stationary coordinate system are partially decoupled, including ω s -ω m and ω s +ω m The frequency component of the output side will diverge when the frequency of the input side is equal to the frequency of the output side. and Contains 2ω m The frequency component will diverge when the output frequency is zero; and Contains 2ω s The frequency component will diverge when the input frequency is zero.

4. The method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 1, wherein: In S3, the decoupling control of the M3C inverter includes the input side control of the M3C inverter, which is to convert the input side voltage and current AC quantities of the M3C inverter into the corresponding active and reactive components in the dq coordinate system for control; the control component i of the input side current sα and i sβ The phase-locked loop angle on the input side is converted into a DC value in the dq rotating coordinate system. and And adopt PI controller to perform zero static error control; The Park transformation matrix is ​​shown in formula (32): Please give the formula (32), C dq is the component of the voltage or current vector in the dq coordinate system, and wt is the angle of the dq rotation coordinate system; The system voltage on the input side of the M3C inverter after Park transformation is as follows: In formula (33), and Indicates the bridge arm voltage after double αβ conversion on the input side and Components in the dq coordinate system; and Represents the components of the input side grid voltage in the dq coordinate system; and Represents the component of the input side bridge arm current in the dq coordinate system; active power control is used to maintain the average value of the total capacitor voltage of the 9 bridge arms in the M3C inverter stable, and reactive power control is set to 0; In order to achieve decoupling control of the input current of the M3C inverter, the bridge arm voltage in the dq coordinate system is given as follows: In formula (34), PIs represents the PI controller transfer function of the current loop on the input side of the M3C inverter; and Represents the command value of the input side current in the dq coordinate system; the output of the current loop PI controller is transformed to the αβ coordinate system through the inverse Park transformation to obtain the voltage reference component required for input system control and The output system control model after Park transformation is as follows: In formula (35), and Indicates the bridge arm voltage after double αβ conversion and Expression in dq coordinate system; and Represents the components of the output side voltage in the dq coordinate system; and Represents the components of the output side current in the dq coordinate system; In order to complete the decoupling control of the output side current of the M3C inverter, the expression of the given component of the bridge arm voltage in the dq coordinate system is as follows: In formula (36), PI m Represents the PI controller transfer function matrix of the output current loop control; Similarly, the output side motor control introduces the feedforward component of the motor voltage. The output component of the current loop PI controller is converted to the αβ coordinate system through the Park inverse transformation to obtain the voltage reference component required for the motor control output. and 5. The method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 4, wherein: The decoupling control of the M3C inverter also includes capacitor voltage balance control in the M3C low-frequency-high-frequency mode. The content of the capacitor voltage balance control in the M3C low-frequency-high-frequency mode is as follows: 1) Input sub-converter imbalance control When the common-mode voltage injected into the input side of the M3C inverter is 0, the power ripple expression between the input sub-converters is as follows: right The components are injected into the following circulation: d1 , I d2 The amplitude of the circulating current injected to balance the input sub-converter voltage; By simplifying the power-capacitor voltage model, the following control model can be obtained: 2) Output sub-converter imbalance control The power ripple between output sub-converters is given by the following formula: By injecting a circulating current with the same frequency as the output side of the M3C inverter, and Internally generates DC power; right Inject the following circulation: where I d3 , I d4 The amplitude of the circulating current injected to balance the output sub-converter voltage is: The output side control model is obtained by simplifying the power-capacitor voltage model: 3) Internal imbalance control of sub-converters The power ripple inside each sub-converter in the M3C inverter is as follows: Injecting a circulating current with the same frequency as the input system generates a DC power flow. When the injected circulating current frequency is the same as the input side frequency of the M3C inverter, the expression of the circulating current system transformed by trigonometric functions is: d5 , I d6 The amplitude of the circulating current injected to balance the internal voltage of the balancing sub-converter is: The power-capacitor voltage model is used to obtain the unbalanced control model inside each sub-converter, which is expressed as follows:

6. The method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 4, wherein: The decoupling control of the M3C inverter also includes M3C circulating current control. The circulating current control is composed of two or more frequency components. A proportional controller is selected to track and control the circulating current setting. The circulating current control reference signal instruction includes the following three balancing control components: 1) Sub-converter internal voltage balance control command signal and 2) Voltage balance control command signal between sub-converters and 3) Differential frequency balance control command signal in the same frequency control mode and The output of the circulating current proportional controller is the reference value of the circulating current control voltage of each bridge arm in the M3C inverter in the dual αβ coordinate system. and 7. The method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 4, wherein: The decoupling control of the M3C inverter also includes unit voltage suppression control under the M3C same-frequency working condition. The unit voltage suppression control under the M3C same-frequency working condition is as follows: Introducing diagonal transformation, its transformation matrix is ​​as follows: Where, C D is the diagonal transformation matrix: After double αβ transformation and diagonal transformation, the power flow inside the sub-converter is represented by and Convert to p cir1_α 、p cir1_β 、p cir2_α and p cir2_β , further divided into two categories: 1) Energy balance in the forward diagonal direction, i.e. p after diagonal transformation cir1_α and p cir1_β ; 2) Energy balance in the reverse diagonal direction, that is, p after diagonal transformation cir2_α and p cir2_β ; The expression is as follows: cir2_α ,i cir2_β ,i cir1_α ,i cir1_β Circulation After diagonal transformation, the αβ components are: Substituting the specific expressions of voltage and current into this power component, equations (59) and (60) can be simplified to: When the M3C inverter meets V S I M =V M I S , and the input and output phases of the M3C inverter meet When the diagonal transformation is decoupled by Park transformation, p cir1_α and p cir1_β Transformed into a DC quantity in the dq coordinate system for control, the angular velocity of the rotating coordinate system is ω s -ω m , combined with the capacitor voltage-power model, the power component expression in the dq coordinate system is as follows: The control of equal frequency condition adopts full control component, i.e. and in and The circulating currents output by the DC voltage loop d-axis and q-axis control loop are given respectively, and the power component p cir1_ α and p cir1_ β is converted to the dq coordinate system for control, at this time the common mode voltage v N The circulating current component in the dq coordinate system is in phase with the circulating current component in the dq coordinate system, generating adjustable DC power; the circulating current of the injected dq axis is and is a high-frequency signal, the injected circulating current in the dq coordinate system and common mode voltage v N The form is as follows: in is the zero-sequence circulating current component Substituting the circulating current and common-mode voltage into equations (69) and (70), we obtain the following expressions: In the closed-loop control in the dq coordinate system, the voltage control in the dq axis introduces feedforward, and the dq axis feedforward components are shown in Equations (74) and (75): In formula (74) and formula (75), and They represent the reference values ​​of the low-frequency fluctuation suppression voltage loop output in the dq coordinate system, and Represents the feedforward component of the voltage loop; Substituting the feedforward component into the equivalent closed-loop control model, the expression is as follows: When the voltage frequency on the output side of the M3C inverter is equal to the voltage frequency on the input side, the control is performed using formulas (76) and (77). The PI controller output is proportional to the zero-sequence component f n (t) multiplied, and finally the inverse Park transform was performed to convert i cir1 The circulation is transformed into the dual αβ coordinate system for control.

8. The method for controlling wide-band operation of an electrically excited synchronous motor using an M3C matrix inverter according to claim 1, wherein: The M3C inverter has a 3×3 modular multilevel matrix converter M3C topology, specifically including 9 bridge arms, each of which includes N submodule units and a bridge arm inductor. Each submodule unit includes an H-bridge and a capacitor. Decoupling control is performed on the M3C inverter to achieve independent control of the input side, output side, and internal circulating current and common-mode voltage of the M3C inverter.

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