Current harmonic suppression method based on generalized active disturbance rejection controller

Abstract

The invention belongs to the technical field of motor control, and particularly relates to a current harmonic suppression method based on a generalized active disturbance rejection controller. Comprising the following steps: S1, establishing a permanent magnet synchronous motor current loop mathematical model containing disturbance, and constructing a d-axis current equation and a q-axis current equation containing disturbance by taking respective current components of a d-axis and a q-axis as state variables; s2, defining the lumped disturbance of the d axis and the q axis as an extended state variable, and constructing an extended state observer to estimate the lumped disturbance; s3, constructing an improved resonance controller; and S4, establishing an active-disturbance-rejection control law, feeding back the lumped disturbance estimated in the step S2 to the active-disturbance-rejection control law for aperiodic disturbance compensation, and expanding the improved resonance controller in the step 3 and embedding the expanded improved resonance controller into the active-disturbance-rejection control law for periodic harmonic compensation. According to the invention, harmonic interference generated in the operation process of permanent magnet synchronization can be suppressed, so that the robustness of a motor control system is improved.

Description

Technical Field

[0001] This invention belongs to the field of motor control technology, and particularly relates to a current harmonic suppression method based on a generalized active disturbance rejection controller. Background Technology

[0002] Permanent magnet synchronous motors (PMSMs) occupy an important position in modern industrial drive systems due to their excellent power density, energy conversion efficiency, and torque output characteristics. Precise current control is crucial for improving the system's steady-state performance. However, during actual operation of PMSMs, effects such as cogging torque, magnetic circuit saturation, and inverter nonlinearity can cause current distortion. These harmonic disturbances not only lead to additional losses and torque fluctuations but also affect the stability and control accuracy of the control system. In recent years, active disturbance rejection control (ADRC) has been widely used in PMSM control systems due to its excellent disturbance suppression effect. Its key module—the extended state observer—can estimate the internal state of the system and external disturbances online, thereby enhancing the system's anti-interference capability. However, traditional extended state observers are limited by their bandwidth characteristics, and their estimation ability is limited when facing high-frequency harmonic interference, affecting the control accuracy of the current loop. Therefore, it is urgent to introduce more targeted harmonic suppression methods to further improve the current control quality of the system under complex operating conditions. Summary of the Invention

[0003] In view of this, the present invention aims to provide a current harmonic suppression method based on a generalized active disturbance rejection controller, in order to solve the problem that the existing traditional extended state observer is limited by its own bandwidth characteristics, and its estimation ability is limited when facing high-frequency harmonic interference, which affects the control accuracy of the current loop. The present invention can suppress harmonic interference generated by permanent magnet synchronous motor during operation, thereby improving the robustness of the motor control system.

[0004] To achieve the above objectives, the technical solution created by this invention is implemented as follows:

[0005] A current harmonic suppression method based on a generalized active disturbance rejection controller is used to suppress current harmonics in a permanent magnet synchronous motor, specifically including the following steps:

[0006] S1: Establish a mathematical model of the current loop of the permanent magnet synchronous motor containing disturbances, and construct the d and q axis current equations containing disturbances using the current components of the d-axis and q-axis as state variables.

[0007] S2: Define the lumped disturbances of the d-axis and q-axis as extended state variables, and construct an extended state observer to estimate the lumped disturbances;

[0008] S3: Constructing an improved resonant controller:

[0009]

[0010] γ d =k pd ;

[0011] γ q =k pq ;

[0012] Where, γ d and γ q γ is an intermediate parameter with no physical meaning. d =k pd γ q =k pq K r For the resonant gain, ω c ω is the resonant bandwidth. h k is the resonant frequency. pd and k pq These are the current loop controller coefficients for the d-axis and q-axis, respectively. Let be the transfer function of the d-axis improved resonant controller. Let be the transfer function of the q-axis improved resonant controller, and s be a Laplace complex variable;

[0013] S4: Establish an active disturbance rejection control law, feed back the lumped disturbance estimated in step S2 to the active disturbance rejection control law for compensation of aperiodic disturbances, and extend the improved resonant controller in step 3 and embed it into the active disturbance rejection control law for compensation of periodic harmonics, thereby obtaining a generalized active disturbance rejection controller based on the improved resonant controller, and realizing the suppression of current harmonics of permanent magnet synchronous motor.

[0014] Furthermore, lumped disturbances include periodic disturbances and non-periodic disturbances.

[0015] Furthermore, in step S1, the mathematical model of the permanent magnet synchronous motor current loop containing the disturbance is as follows:

[0016]

[0017] Where, ΔL s =L s -L s0 ΔR s =R s -R s0 , Δψ f =ψ f -ψ f0 L s0 R s0 and ψ f0 ψ represents the nominal values ​​of inductance, resistance, and magnetic flux, respectively. dh and ψ qh The magnetic flux harmonics along the d-axis and q-axis are respectively, ψ dh(6k)and ψ qh(6k) These represent the amplitudes of the 6kth harmonics along the d-axis and q-axis, respectively, where k is a positive number, and Δu sdh and Δu sqh Voltage harmonics on the d-axis and q-axis, respectively, T dead and T s These represent the dead time and sampling period, U dc For DC voltage, δ sd and δ sq These represent the unmodeled perturbations along the d-axis and q-axis, respectively.

[0018] Furthermore, step S1 includes the equations for the perturbation d-axis and q-axis currents:

[0019]

[0020]

[0021] Among them, f dap and f qap f represents the non-periodic disturbances along the d-axis and q-axis, respectively. dp and f qp The periodic perturbations along the d-axis and q-axis are respectively, f d and f q These represent the total disturbance along the d-axis and the total disturbance along the q-axis, respectively.

[0022] Furthermore, in step S2, the extended state observer is:

[0023]

[0024] Where γ0=1 / L s0 γ0 is a constant, z 1d For the estimated value z d The first element, z d z is an estimate of the state matrix containing the motor output current. 2d For the estimated value z d The second element, z 3d For the estimated value z d The third element, z 1q For the estimated value z q The first element, z q z is an estimate of the state variable matrix containing internal and external disturbances. 2q For the estimated value z q The second element, z 3q For the estimated value z q The third element is n, which is the order of the extended state observer, and i is the index value.

[0025] Furthermore, the improved resonant controller is extended to the following equation to suppress the 6th and 12th harmonics caused by flux harmonics and inverter nonlinearity:

[0026]

[0027] in, For the extended d-axis improved resonant controller, For the extended q-axis improved resonant controller, K r1 For the resonant gain corresponding to the 6th harmonic, ω c1 The resonant bandwidth corresponding to the 6th harmonic is given by s, where s is the Laplace complex frequency variable and K is the resonant bandwidth corresponding to the 6th harmonic. r2 For the resonant gain corresponding to the 12th harmonic, ω c2 ω is the resonant bandwidth corresponding to the 12th harmonic. h1 ω is the resonant frequency corresponding to the 6th harmonic. h2 This is the resonant frequency corresponding to the 12th harmonic.

[0028] Furthermore, the active disturbance rejection control law includes the control law of the d-axis active disturbance rejection controller and the control law of the q-axis active disturbance rejection controller, wherein the control law of the d-axis active disturbance rejection controller is u sd for:

[0029]

[0030] in, k is the rate of change of the d-axis output current. pd i is the d-axis current loop controller coefficient. sd This refers to the d-axis stator current. Z represents the rate of change of the d-axis reference current. 2d For the estimated value z d The second element, For the extended d-axis improved resonant controller, This is the current reference value for the d-axis;

[0031] The control law of the q-axis active disturbance rejection controller is:

[0032]

[0033] in, k is the rate of change of the q-axis output current. pq i is the q-axis current loop controller coefficient. sq This is the q-axis stator current. The rate of change of the q-axis reference current. For the extended q-axis improved resonant controller, This is the current reference value for the q-axis.

[0034] Furthermore, the lumped disturbance estimated in step S2 is z. 2d and z 2q .

[0035] Compared with the prior art, the present invention can achieve the following beneficial effects:

[0036] (1) The present invention provides a current harmonic suppression method based on a generalized active disturbance rejection controller. This method achieves precise control of the current, reduces the impact of harmonic interference on the stability and control accuracy of the control system, makes the current response of the motor control system more stable, and improves the steady-state performance of the system.

[0037] (2) The current harmonic suppression method based on the generalized active disturbance rejection controller described in this invention combines the advantages of generalized active disturbance rejection control and improved resonant control. It can not only suppress low-frequency non-periodic disturbances, but also effectively suppress medium- and high-frequency periodic harmonic disturbances, thus achieving effective suppression of wideband disturbances and improving the anti-interference performance of the system under complex working conditions. Attached Figure Description

[0038] The accompanying drawings, which form part of this invention, are used to provide a further understanding of the invention. The illustrative embodiments and descriptions of the invention are used to explain the invention and do not constitute an undue limitation of the invention. In the drawings:

[0039] Figure 1 The structural block diagram of the permanent magnet synchronous motor control system described in the embodiment of the present invention;

[0040] Figure 2 A flowchart of the current harmonic suppression method based on a generalized active disturbance rejection controller as described in the embodiments of the present invention;

[0041] Figure 3 Different ω as described in the embodiments of the present invention o Bode plot of the generalized extended state observer;

[0042] Figure 4 Different Ks described in the embodiments of the present invention r and ω c Bode plot of the improved resonant controller;

[0043] Figure 5 The structural block diagram of the generalized active disturbance rejection controller based on the improved resonant controller described in the embodiments of the present invention;

[0044] Figure 6 G as described in the embodiments of the present invention N (s) and G R Bode plot of (s);

[0045] Figure 7 Bode plots of four different active disturbance rejection controllers described in the embodiments of the present invention;

[0046] Figure 8 Simulation waveform diagram of conventional active disturbance rejection control as described in the embodiments of the present invention;

[0047] Figure 9 Simulation waveform diagram of the generalized active disturbance rejection control described in the embodiments of the present invention;

[0048] Figure 10 Simulation waveform diagram of the generalized active disturbance rejection control based on a quasi-resonant controller as described in the embodiments of the present invention;

[0049] Figure 11 The simulation waveform diagram of the generalized active disturbance rejection controller based on the improved resonant controller described in the embodiment of the present invention. Detailed Implementation

[0050] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and specific embodiments. It should be understood that the specific embodiments described herein are merely illustrative of the invention and do not constitute a limitation thereof.

[0051] It should be noted that, unless otherwise specified, the embodiments and features described in the present invention can be combined with each other.

[0052] In the description of this invention, it should be understood that the terms "center," "longitudinal," "lateral," "upper," "lower," "front," "rear," "left," "right," "vertical," "horizontal," "top," "bottom," "inner," and "outer," etc., indicating orientations or positional relationships based on the orientations or positional relationships shown in the accompanying drawings, are only for the convenience of describing this invention and simplifying the description, and do not indicate or imply that the device or element referred to must have a specific orientation, or be constructed and operated in a specific orientation, and therefore should not be construed as a limitation on this invention. Furthermore, the terms "first," "second," etc., are used for descriptive purposes only and should not be construed as indicating or implying relative importance or implicitly specifying the number of indicated technical features. Thus, features defined with "first," "second," etc., may explicitly or implicitly include one or more of that feature. In the description of this invention, unless otherwise stated, "a plurality of" means two or more.

[0053] In the description of this invention, it should be noted that, unless otherwise explicitly specified and limited, the terms "installation," "connection," and "linking" should be interpreted broadly. For example, they can refer to a fixed connection, a detachable connection, or an integral connection; they can refer to a mechanical connection or an electrical connection; they can refer to a direct connection or an indirect connection through an intermediate medium; and they can refer to the internal connection of two components. Those skilled in the art will understand the specific meaning of the above terms in this invention based on the specific circumstances.

[0054] The invention will now be described in detail with reference to the accompanying drawings and embodiments.

[0055] like Figure 1 As shown, in one embodiment of the present invention, PI represents a proportional-integral regulator, SVPWM is space vector pulse width modulation, Inverter is a three-phase inverter, and U... dc This is the DC power supply for the three-phase inverter. In this invention, due to the adoption of a field-oriented vector control strategy, the d-axis current command is 0. The speed loop is controlled by a proportional-integral controller, whose input is the difference between the speed command and the actual speed, and whose output is the q-axis current command. The current loop is controlled by a generalized active disturbance rejection controller based on an improved resonant controller, which can be divided into a d-axis current controller (input is the difference between the d-axis current command and the d-axis feedback current, output is the d-axis voltage command) and a q-axis current controller (input is the difference between the d-axis current command and the d-axis feedback current, output is the q-axis voltage command). Subsequently, the d-axis voltage command and the q-axis voltage command are transformed into α voltage command and β voltage command through coordinate transformation. The α voltage command and the β voltage command are then applied to the three-phase inverter after space vector pulse width modulation, thereby outputting three-phase voltages to control the motor.

[0056] The following is combined Figure 2 The present invention will be described in detail below:

[0057] Step 1: Construct a mathematical model of the permanent magnet synchronous motor current loop considering disturbances: First, based on the mathematical model of the permanent magnet synchronous motor current loop under the synchronous rotating coordinate system dq axis, the lumped disturbances existing in the current loop are modeled. The d-axis current component and the q-axis current component are selected as key state variables, and the d-axis and q-axis current equations containing disturbances are established, laying a mathematical foundation for the design of subsequent advanced control algorithms.

[0058] The stator current equation of a permanent magnet synchronous motor can be expressed as:

[0059]

[0060] Among them, u sd and u sq The stator voltages along the d-axis and q-axis are respectively, i sd and i sqThe stator currents R are the d-axis and q-axis currents, respectively. s For the stator resistance, ω e Let ψ be the electric angular velocity of the permanent magnet synchronous motor. f For permanent magnet flux linkage, L sd and L sq These are the d-axis and q-axis inductances, respectively. Since the d- and q-axis magnetic circuits of a surface-mount permanent magnet synchronous motor are symmetrical, therefore, L... sd =L sq =L s0 L s0 This is the nominal value of the inductance.

[0061] According to equation (1), the mathematical model of the current loop of the permanent magnet synchronous motor considering periodic disturbances (flux harmonics and inverter nonlinearity) and non-periodic disturbances (parameter perturbations and internal and external uncertainties) can be expressed as:

[0062]

[0063]

[0064] Where, ΔL s =L s -L s0 ΔR s =R s -R s0 , Δψ f =ψ f -ψ f0 L s0 R s0 and ψ f0 ψ represents the nominal values ​​of inductance, resistance, and magnetic flux, respectively. dh and ψ qh The magnetic flux harmonics along the d-axis and q-axis are respectively, ψ dh(6k) and ψ qh(6k) Δu represents the amplitude of the 6kth harmonic along the d-axis and q-axis, respectively, where k is a positive integer. sdh and Δu sqh Voltage harmonics on the d-axis and q-axis, respectively, T dead and T s These represent the dead time and sampling period, U dc For DC voltage, δ sd and δ sq These represent the unmodeled perturbations along the d-axis and q-axis, respectively.

[0065] According to equations (2)-(4), define f d and f q Let the lumped disturbances be the d-axis and q-axis respectively. Then, the d-axis and q-axis current equations of the permanent magnet synchronous motor including the lumped disturbances can be expressed as:

[0066]

[0067] Among them, f dap and f qap f represents the non-periodic disturbances along the d-axis and q-axis, respectively. dp and f qp These are the periodic harmonic disturbances along the d-axis and q-axis, respectively.

[0068] Based on equations (5) and (6), consider an extended-order system with n additional states:

[0069]

[0070] In the formula,

[0071] Where γ0=1 / L s0 x d x q y d and y q All of these are state variables set to construct the extended state observer, where γ0 is a constant and A, B, C, and D are constant matrices.

[0072] Step 2: Construct a generalized extended state observer to obtain a lumped perturbation estimate. Based on the extended-order system equations described above, a generalized extended state observer can be constructed as follows:

[0073]

[0074] In the formula, and They represent x respectively d and x q The estimated value, x d Let x be the state variable matrix containing the d-axis output current and the total d-axis disturbance. q This is a state variable matrix containing the d-axis output current and the total d-axis disturbance; is the gain coefficient of the generalized extended state observer.

[0075] The bandwidth tuning method is used to configure the poles of the generalized extended state observer at -ω. o Therefore, we can obtain:

[0076]

[0077] Where, ω o This is the bandwidth of the generalized extended state observer.

[0078] To achieve a balance between disturbance rejection and noise immunity, this invention selects a second-order generalized extended state observer with two extended states to observe lumped disturbances. Finally, the extended state observer is constructed as follows:

[0079]

[0080] Among them, z 1d For the estimated value z d The first element, z 2d For the estimated value z d The second element, z 3d For the estimated value z d The third element, z 1q For the estimated value z q The first element, z 2q For the estimated value z q The second element, z 3q For the estimated value z q The third element.

[0081] Combining (7) and (12), based on the extended state observer, the unknown disturbance estimation transfer function G can be obtained. z2d (s) and the unknown disturbance estimation error transfer function G e2d (s) is:

[0082]

[0083] Figure 3 Different ω were displayed o Download G e2d (s) and G z2d Bode plot of (s). By Figure 3 It is known that the generalized extended state observer is essentially a low-pass filter, capable of effectively estimating low-frequency aperiodic disturbances. Theoretically, increasing the observer bandwidth can expand the frequency range for disturbance estimation, but this method also amplifies high-frequency measurement noise. Especially for harmonic disturbances with pronounced periodic characteristics, the generalized extended state observer often struggles to achieve accurate estimation. To address the issue of the generalized extended state observer's poor estimation and compensation performance for periodic disturbances, the system's feedback control loop requires an additional compensation mechanism to suppress residual disturbances. This invention proposes a solution that introduces an improved resonant controller into the feedback control loop, aiming to significantly improve the system's ability to suppress periodic disturbances.

[0084] Step 3: Construct an improved resonant controller for the d-axis and q-axis as follows:

[0085]

[0086] In the formula, γ d =kpd γ q =k pq γ d and γ q K is an intermediate parameter with no physical meaning. r Represents the resonant gain, ω c ω represents the resonant bandwidth. h k represents the resonant frequency. pd and k pq These are the current loop controller coefficients for the d-axis and q-axis, respectively. Let be the transfer function of the d-axis improved resonant controller. This is the transfer function of the q-axis improved resonant controller.

[0087] Figure 4 It showed ω c From 2 rad / s to 30 rad / s, K r Bode plot of the improved resonant controller as the value changes from 10 to 120. Figure 4 As can be seen from the amplitude-frequency response curve, the improved resonant controller exhibits a significant gain peak at the resonant frequency. This characteristic enables it to accurately suppress harmonic disturbances at the target frequency. This can be achieved by adjusting the bandwidth parameter ω. c This expands the effective operating bandwidth of the improved resonant controller, thereby enhancing its robustness against frequency shifts and interference from adjacent frequency bands, and improving system robustness. Simultaneously, increasing the resonant gain K... r It can enhance the regulation effect of the improved resonant controller at the target frequency, further reduce steady-state error, and improve the harmonic suppression effect.

[0088] Step 4: Establish an active disturbance rejection control law based on feedback control. Substitute the lumped disturbance estimated in Step 2 into the control law to compensate for aperiodic disturbances. Embed the improved resonant controller from Step 3 into the control law to compensate for periodic harmonics. Based on this, establish a generalized active disturbance rejection controller based on the improved resonant controller to suppress the current harmonics of the permanent magnet synchronous motor.

[0089] Define the current reference value for the d-axis as follows: The current reference value for the q-axis is Let the current tracking error of the d-axis be The current tracking error on the q-axis is then:

[0090]

[0091] To ensure that the error converges exponentially, the error convergence control law using proportional control is as follows:

[0092]

[0093] in, The rate of change of the d-axis current tracking error. This represents the rate of change of the q-axis current tracking error. The rate of change of the d-axis reference current. The rate of change of the q-axis reference current. The rate of change of the d-axis output current. This represents the rate of change of the q-axis output current.

[0094] To accurately suppress the 6th and 12th harmonics caused by flux harmonics and inverter nonlinearity, the improved resonant controller can be extended to:

[0095]

[0096] in, For the extended d-axis improved resonant controller, For the extended q-axis improved resonant controller, K r1 For the resonant gain corresponding to the 6th harmonic, ω c1 The resonant bandwidth corresponding to the 6th harmonic is given by S, where S is the Laplace complex frequency variable and K is the frequency of the 6th harmonic. r2 For the resonant gain corresponding to the 12th harmonic, ω c2 This is the resonant bandwidth corresponding to the 12th harmonic;

[0097] In addition, the resonant frequency ω corresponding to the 6th harmonic h1 =6ω e The resonant frequency ω corresponding to the 12th harmonic h2 =12ω e .

[0098] Combining equations (17) and (19), the improved resonant controller (19) is implemented in parallel with the proportional control law, and the disturbance estimated by the observer is used to replace the actual disturbance. The control law of the d-axis active disturbance rejection controller is then obtained as follows:

[0099]

[0100] Similarly, the control law of the q-axis active disturbance rejection controller can be designed as follows:

[0101]

[0102] Combining equations (7), (12), and (20), the generalized active disturbance rejection controller based on the improved resonant controller is derived from f d (s) to x 1d Perturbation suppression transfer function of (s) The disturbance rejection transfer function describes the output current of the system under disturbance, that is, the effect of the disturbance on the output current. The smaller this value, the better the system's disturbance rejection capability. Therefore, the purpose of this transfer function is to illustrate the system's disturbance rejection performance in subsequent analysis.

[0103]

[0104] In the formula,

[0105] If we take n = 1 in equation (9) and remove the resonant controller in equation (20), we can obtain the disturbance suppression transfer function of the traditional active disturbance rejection controller by combining the equations.

[0106]

[0107] In the formula,

[0108] To more clearly analyze the system's anti-interference performance, equation (22) is rewritten in the following form:

[0109]

[0110] As can be seen from equation (24), the disturbance suppression transfer function of the generalized active disturbance rejection controller based on the improved resonant controller can be expressed as the disturbance suppression transfer function of the generalized active disturbance rejection controller. With notch filter G N The product relationship of (s).

[0111] like Figure 5 As shown, the generalized active disturbance rejection controller based on the improved resonant controller consists of two parts: a generalized extended state observer and an active disturbance rejection control law. The generalized extended state observer can estimate constant or slowly changing non-periodic disturbances, such as parameter mismatch disturbances, and feed them back to the control law for compensation. The improved resonant controller can compensate for periodic harmonic interferences and acts in parallel with the control law on the permanent magnet synchronous motor, thereby enhancing the robustness of the system. The expressions of the generalized active disturbance rejection controller based on the improved resonant controller are: Equations (20) and (21).

[0112] According to equation (24), when a quasi-resonant controller is used When the disturbance rejection transfer function of the generalized active disturbance rejection controller based on the quasi-resonant controller is expressed as:

[0113]

[0114] In the formula, It can be seen that the disturbance suppression transfer function of the generalized active disturbance rejection controller based on the quasi-resonant controller can be expressed as the disturbance suppression transfer function of the generalized active disturbance rejection controller. With letter GR The product relationship of (s).

[0115] Figure 6 ω is given h =100rad / s, ω c =8 rad / s, k pd =50, K r =100, k r =3K r Time G N (s) and G R Bode plot of (s). By Figure 6 It can be seen that the notch filter G N (s) and transmission G R (s) exhibits significant amplitude attenuation characteristics at specific harmonic frequencies, providing an effective means for harmonic suppression. However, comparative analysis shows that the transfer function G... R (s) Resonant peaks exceeding 0 dB are generated near harmonic frequencies. This undesirable amplification effect can amplify noise and affect system stability. In contrast, the notch filter G N (s) Undesirable peaks were completely eliminated through optimized design, resulting in superior performance.

[0116] Figure 7 Drawn and The Bode plot, where ω h =400rad / s, ω c =8 rad / s, k pd =50, K r =500,k r =3K r It can be observed that the generalized active disturbance rejection controller based on the improved resonant controller of this invention achieves a significant improvement in disturbance suppression performance. Specifically, the generalized active disturbance rejection controller of this invention not only enhances the system's ability to suppress low-frequency aperiodic disturbances, but also significantly improves the suppression effect on mid-to-high frequency periodic harmonic disturbances. It is worth noting that by introducing a notch filter structure, this invention effectively avoids the noise amplification problem common in quasi-resonant control and eliminates the unwanted resonance peak phenomenon, enabling this invention to achieve more effective suppression of broadband disturbances, especially periodic harmonic interference, while inheriting the advantages of generalized active disturbance rejection control.

[0117] Figure 8 , Figure 9 , Figure 10 and Figure 11Simulation waveforms of permanent magnet synchronous motors under four active disturbance rejection (ADNR) control schemes are shown, with the motor speed at 150 r / min and the load torque at 5 N·m. The simulation results, from top to bottom, include q-axis current, three-phase current, and Fast Fourier Transform (FFT) analysis of the q-axis current. Comparative analysis of the simulation data shows that in the traditional ADNR control scheme, the system exhibits significant current fluctuations, with a peak q-axis current ripple of 0.73 A and a total harmonic distortion (THD) of 5.66%. After adopting the scheme proposed in this invention, the system performance is improved to some extent, with the peak q-axis current ripple decreasing to 0.71 A and the THD slightly reduced to 5.56%. Notably, when using the control strategy of this invention, the system control effect is significantly improved, with the peak q-axis current ripple decreasing dramatically to 0.37 A, and the THD significantly improved to only 2.07%. Most notably, this invention demonstrates superior control performance, specifically: the peak value of q-axis current ripple is further reduced to 0.33A, and the THD index is optimized to 1.88%. This series of data fully verifies the significant advantages of this invention in suppressing current harmonics.

[0118] Therefore, compared with traditional active disturbance rejection control methods, the present invention has significant advantages in harmonic suppression performance; at the same time, compared with existing generalized active disturbance rejection control algorithms based on quasi-resonant controllers, the present invention effectively eliminates the problem of unwanted resonance peaks and has a better harmonic suppression effect.

[0119] It should be understood that the various forms of processes shown above can be used to reorder, add, or delete steps. For example, the steps described in this invention disclosure can be executed in parallel, sequentially, or in different orders, as long as the desired result of the technical solution disclosed in this invention can be achieved, and this is not limited herein.

[0120] The specific embodiments described above do not constitute a limitation on the scope of protection of this invention. Those skilled in the art should understand that various modifications, combinations, sub-combinations, and substitutions can be made according to design requirements and other factors. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of this invention should be included within the scope of protection of this invention.

Claims

1. A current harmonic suppression method based on a generalized active disturbance rejection controller, used to suppress current harmonics in a permanent magnet synchronous motor, characterized in that: Specifically, the steps include the following: S1: Establish a mathematical model of the current loop of the permanent magnet synchronous motor containing disturbances, and construct the d and q axis current equations containing disturbances using the current components of the d-axis and q-axis as state variables. S2: Define the lumped disturbances of the d-axis and q-axis as extended state variables, and construct an extended state observer to estimate the lumped disturbances; S3: Constructing an improved resonant controller: c d =k pd ; c q =k pq ; Where, γ d and γ q γ is an intermediate parameter with no physical meaning. d =k pd γ q =k pq K r For the resonant gain, ω c ω is the resonant bandwidth. h k is the resonant frequency. pd and k pq These are the current loop controller coefficients for the d-axis and q-axis, respectively. Let be the transfer function of the d-axis improved resonant controller. Let be the transfer function of the q-axis improved resonant controller, and s be a Laplace complex variable; S4: Establish an active disturbance rejection control law, feed back the lumped disturbance estimated in step S2 to the active disturbance rejection control law for compensation of aperiodic disturbances, and embed the improved resonant controller from step 3 into the active disturbance rejection control law after expansion for compensation of periodic harmonics, thereby obtaining a generalized active disturbance rejection controller based on the improved resonant controller, and realizing the suppression of current harmonics of permanent magnet synchronous motor.

2. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 1, characterized in that: Lumped disturbances include periodic disturbances and non-periodic disturbances.

3. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 1, characterized in that: In step S1, the mathematical model of the permanent magnet synchronous motor current loop containing the disturbance is as follows: Where, ΔL s =L s -L s0 ΔR s =R s -R s0 , Δψ f =ψ f -ψ f0 L s0 R s0 and ψ f0 ψ represents the nominal values ​​of inductance, resistance, and magnetic flux, respectively. dh and ψ qh The magnetic flux harmonics along the d-axis and q-axis are respectively, ψ dh(6k) and ψ qh(6k) Δu represents the amplitude of the 6kth harmonic along the d-axis and q-axis, respectively, where k is a positive number. sdh and Δu sqh Voltage harmonics on the d-axis and q-axis, respectively, T dead and T s These represent the dead time and sampling period, U dc For DC voltage, δ sd and δ sq These are the unmodeled perturbations along the d-axis and q-axis, respectively.

4. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 3, characterized in that: Step S1 includes the equations for the perturbation d-axis and q-axis currents: Among them, f dap and f qap f represents the non-periodic disturbances along the d-axis and q-axis, respectively. dp and f qp The periodic perturbations along the d-axis and q-axis are respectively, f d and f p These represent the total disturbance along the d-axis and the total disturbance along the q-axis, respectively.

5. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 1, characterized in that: In step S2, the extended state observer is: Where γ0=1 / L s0 γ0 is a constant, z 1d For the estimated value z d The first element, z d z is an estimate of the state matrix containing the motor output current. 2d For the estimated value z d The second element, z 3d For the estimated value z d The third element, z 1q For the estimated value z q The first element, z q z is an estimate of the state variable matrix containing internal and external disturbances. 2q For the estimated value z q The second element, z 3q For the estimated value z q The third element is n, which is the order of the extended state observer, and i is the index value.

6. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 1, characterized in that: The improved resonant controller is extended to the following equation to suppress the 6th and 12th harmonics caused by flux harmonics and inverter nonlinearity: in, For the extended d-axis improved resonant controller, For the extended q-axis improved resonant controller, K r1 For the resonant gain corresponding to the 6th harmonic, ω c1 The resonant bandwidth corresponding to the 6th harmonic is given by s, where s is the Laplace complex frequency variable and K is the resonant bandwidth corresponding to the 6th harmonic. r2 For the resonant gain corresponding to the 12th harmonic, ω c2 ω is the resonant bandwidth corresponding to the 12th harmonic. h1 ω is the resonant frequency corresponding to the 6th harmonic. h2 This is the resonant frequency corresponding to the 12th harmonic.

7. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 5, characterized in that: Active disturbance rejection (ADRR) control laws include the control law for the d-axis ADRR and the control law for the q-axis ADRR. The control law for the d-axis ADRR is u... sd for: in, k is the rate of change of the d-axis output current. pd i is the d-axis current loop controller coefficient. sd This refers to the d-axis stator current. Z represents the rate of change of the d-axis reference current. 2d For the estimated value z d The second element, For the extended d-axis improved resonant controller, This is the current reference value for the d-axis; The control law of the q-axis active disturbance rejection controller is: in, k is the rate of change of the q-axis output current. pq i is the q-axis current loop controller coefficient. sq This is the q-axis stator current. The rate of change of the q-axis reference current. For the extended q-axis improved resonant controller, This is the current reference value for the q-axis.

8. The current harmonic suppression method based on a generalized active disturbance rejection controller according to claim 5, characterized in that: The lumped disturbance estimated in step S2 is z 2d and z 2q .

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