Control method and device of permanent magnet synchronous motor, electronic equipment and storage medium

Abstract

The invention provides a permanent magnet synchronous motor control method and device, electronic equipment and a storage medium. The method comprises the following steps: acquiring a sampling rotor electrical angular velocity of the permanent magnet synchronous motor, and determining a first shaft current instruction based on the sampling rotor electrical angular velocity; determining a reference voltage vector according to the first axis current instruction and a preset second axis current instruction; synthesizing a reference voltage vector by using the first preset number of non-zero vectors and the second preset number of zero vectors; determining a candidate voltage vector set by adjusting the number of voltage vectors acting in the control period and the acting time of each voltage vector; constructing a cost function based on the current error, the current harmonic and the switching frequency, and determining an optimal voltage vector combination and action time from the candidate voltage vector set according to the cost function; and determining a driving signal of each switching tube of the inverter based on the optimal voltage vector combination and the action time, and controlling the permanent magnet synchronous motor based on the driving signals.

Description

Technical Field

[0001] This disclosure relates to the field of motor control technology, and in particular to a control method, device, electronic equipment and storage medium for a permanent magnet synchronous motor. Background Technology

[0002] In recent years, model predictive control has been widely studied in the control of permanent magnet synchronous motors due to its simple concept, fast dynamic response, and multivariable control capabilities. However, current model predictive control can only apply a fixed number of voltage vectors in a single control cycle, resulting in relatively large current and torque ripples. Furthermore, the switching frequency is not fixed, leading to poor steady-state performance.

[0003] Therefore, improving the steady-state performance of model predictive control has become an urgent technical problem to be solved. Summary of the Invention

[0004] In view of this, the purpose of this disclosure is to provide a control method, device, electronic equipment and storage medium for a permanent magnet synchronous motor to solve or partially solve the above-mentioned technical problems.

[0005] To achieve the above objectives, the first aspect of this disclosure provides a control method for a permanent magnet synchronous motor, the method comprising:

[0006] Obtain the sampled rotor electric angular velocity of the permanent magnet synchronous motor, and determine the first shaft current command based on the sampled rotor electric angular velocity;

[0007] The reference voltage vector is determined based on the first axis current command and the pre-set second axis current command;

[0008] The reference voltage vector is synthesized using a first preset number of non-zero vectors and a second preset number of zero vectors. Multiple voltage vectors and the duty cycle of each voltage vector are determined based on the volt-second balance algorithm.

[0009] The candidate voltage vector set is determined by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector.

[0010] A cost function is constructed based on current error, current harmonics, and switching frequency. The optimal voltage vector combination and application time are determined from the candidate voltage vector set according to the cost function.

[0011] The drive signal for each switch of the inverter is determined based on the optimal voltage vector combination and the operating time, and the permanent magnet synchronous motor is controlled based on the drive signal.

[0012] Based on the same inventive concept, a second aspect of this disclosure proposes a control device for a permanent magnet synchronous motor, comprising:

[0013] The first axis current command determination module is configured to acquire the sampled rotor electric angular velocity of the permanent magnet synchronous motor and determine the first axis current command based on the sampled rotor electric angular velocity.

[0014] The reference voltage vector determination module is configured to determine the reference voltage vector based on the first axis current command and a preset second axis current command.

[0015] The duty cycle determination module is configured to synthesize the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors, and to determine multiple voltage vectors and the duty cycle of each voltage vector based on a volt-second balance algorithm.

[0016] The candidate voltage vector set determination module is configured to determine the candidate voltage vector set by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector.

[0017] The optimal voltage vector combination determination module is configured to construct a cost function based on current error, current harmonics and switching frequency, and determine the optimal voltage vector combination and application time from the candidate voltage vector set according to the cost function.

[0018] The control module is configured to determine the drive signal for each switch of the inverter based on the optimal voltage vector combination and the operating time, and to control the permanent magnet synchronous motor based on the drive signal.

[0019] Based on the same inventive concept, a third aspect of this disclosure proposes an electronic device including a memory, a processor, and a computer program stored in the memory and executable by the processor, wherein the processor implements the method described above when executing the computer program.

[0020] Based on the same inventive concept, a fourth aspect of this disclosure provides a non-transitory computer-readable storage medium that stores computer instructions for causing a computer to perform the methods described above.

[0021] As can be seen from the above description, the control method, device, electronic equipment, and storage medium for permanent magnet synchronous motors provided in this disclosure are as follows: The sampled rotor electrical angular velocity of the permanent magnet synchronous motor is acquired, and a first axis current command is determined based on the sampled rotor electrical angular velocity. A reference voltage vector is determined according to the first axis current command and a pre-set second axis current command. The reference voltage vector is synthesized using a first preset number of non-zero vectors and a second preset number of zero vectors. Multiple voltage vectors and the duty cycle of each voltage vector are determined based on a volt-second balance algorithm. A candidate voltage vector set is determined by adjusting the number of voltage vectors acting within the control cycle and the acting time of each voltage vector. Thus, the number of voltage vectors in the candidate voltage vector set is not fixed, considering various voltage vector combinations. A cost function is constructed based on current error, current harmonics, and switching frequency. The optimal voltage vector combination and acting time are determined from the candidate voltage vector set according to the cost function, ensuring that the obtained optimal voltage vector combination and acting time represent the optimal voltage vector combination for current error, current harmonics, and switching frequency, further improving steady-state performance. The drive signal for each switch of the inverter is determined based on the optimal voltage vector combination and acting time, and the permanent magnet synchronous motor is controlled based on the drive signal. Attached Figure Description

[0022] To more clearly illustrate the technical solutions in this disclosure or related technologies, the accompanying drawings used in the description of the embodiments or related technologies will be briefly introduced below. Obviously, the accompanying drawings described below are only embodiments of this disclosure. For those skilled in the art, other drawings can be obtained based on these drawings without creative effort.

[0023] Figure 1 This is a flowchart of a control method for a permanent magnet synchronous motor according to an embodiment of the present disclosure;

[0024] Figure 2 This is a hardware structure diagram of a permanent magnet synchronous motor according to an embodiment of this disclosure;

[0025] Figure 3 This is a schematic diagram of the current control structure of a permanent magnet synchronous motor based on a unified multi-vector according to an embodiment of this disclosure;

[0026] Figure 4 This is a schematic diagram of the synthesized reference voltage vector according to an embodiment of this disclosure;

[0027] Figure 5 This is a schematic diagram of the segmentation of harmonic current according to an embodiment of the present disclosure;

[0028] Figure 6 This is a schematic diagram of the motor test results of the 1VB-MFPCC according to an embodiment of this disclosure;

[0029] Figure 7This is a schematic diagram of the motor test results of the SVM-MFPCC according to an embodiment of this disclosure;

[0030] Figure 8 This is a schematic diagram showing the motor test results of the GMV-MFPCC according to an embodiment of this disclosure;

[0031] Figure 9 This is a schematic diagram showing the current harmonic analysis results of a 1VB-MFPCC according to an embodiment of this disclosure;

[0032] Figure 10 This is a schematic diagram of the current harmonic analysis results of the SVM-MFPCC according to an embodiment of this disclosure;

[0033] Figure 11 This is a schematic diagram showing the current harmonic analysis results of the GMV-MFPCC according to an embodiment of this disclosure;

[0034] Figure 12 This is a schematic diagram of the total harmonic distortion of current and switching frequency at different speeds according to an embodiment of the present disclosure.

[0035] Figure 13 This is a schematic diagram of the dynamic response of GMV-MFPCC according to an embodiment of the present disclosure;

[0036] Figure 14 This is a schematic diagram of the experimental results of the GMV-MFPCC of this disclosure when the reference switching frequency jumps from 2kHz to 2.5kHz at rated speed;

[0037] Figure 15 This is a schematic diagram showing the steady-state test results of the GMV-MFPCC of this embodiment under rated load at 2000 r / min;

[0038] Figure 16 This is a schematic diagram of the structure of the control device for a permanent magnet synchronous motor according to an embodiment of the present disclosure;

[0039] Figure 17 This is a schematic diagram of the structure of an electronic device according to an embodiment of the present disclosure. Detailed Implementation

[0040] To make the objectives, technical solutions, and advantages of this disclosure clearer, the following detailed description is provided in conjunction with specific embodiments and the accompanying drawings.

[0041] It should be noted that, unless otherwise defined, the technical or scientific terms used in the embodiments of this disclosure should have the ordinary meaning understood by one of ordinary skill in the art to which this disclosure pertains. The terms "first," "second," and similar terms used in the embodiments of this disclosure do not indicate any order, quantity, or importance, but are merely used to distinguish different components. Terms such as "comprising" or "including" mean that the element or object preceding the word encompasses the elements or objects listed following the word and their equivalents, without excluding other elements or objects. Terms such as "connected" or "linked" are not limited to physical or mechanical connections, but can include electrical connections, whether direct or indirect. Terms such as "upper," "lower," "left," and "right" are used only to indicate relative positional relationships; when the absolute position of the described object changes, the relative positional relationship may also change accordingly.

[0042] Based on the background description, Model Predictive Control (MPC) has been widely studied due to its simple concept, fast dynamic response, and multivariable control capabilities. Compared with field-oriented control, MPC has a simpler structure and faster dynamic response. Compared with direct torque control, MPC achieves more accurate and efficient voltage vector selection by minimizing the cost function. Depending on the controlled object, Model Predictive Control can be divided into Model Predictive Torque Control (MPTC) and Model Predictive Current Control (MPCC). Model Predictive Current Control uses current as the control variable, eliminating the need for torque and flux calculations and weight factor design, making it simpler. In traditional MPC, only one voltage vector is applied in one control cycle, resulting in relatively large current and torque ripple, and the switching frequency is not fixed.

[0043] To address the aforementioned issues, a dual-vector MPC, consisting of a non-zero vector and a zero vector, is employed in the control cycle to reduce current harmonics and improve steady-state performance. However, at high speeds, the optimal vector combination may not be a single non-zero vector and a single zero vector, but rather two non-zero vectors. In some schemes, the restrictions on the second vector are relaxed in the dual-vector combination, further reducing current ripple. In some schemes, three voltage vectors are applied within a control cycle; the additional voltage vectors increase the degrees of freedom for eliminating d-axis and q-axis current errors at sampling time. In some schemes, four voltage vectors are applied within a control cycle; combining MPCC with Space Vector Modulation (SVM) can achieve deadbeat-free control with better stability and a fixed switching frequency. Of course, the complexity of control increases with the number of vector combinations. New methods combining the advantages of single-vector and multi-vector MPC remain attractive. Currently, there is no good method that simultaneously satisfies: 1) pre-establishing candidate vector sets with different numbers of applied voltages; 2) significantly improving steady-state control performance; 3) a fixed switching frequency; and 4) robustness. Therefore, a simple and practical method needs to be developed to achieve better control performance.

[0044] As mentioned above, improving the steady-state performance of model predictive control has become an important research question.

[0045] Based on the above description, such as Figure 1 As shown in this embodiment, the control method for a permanent magnet synchronous motor includes:

[0046] Step 101: Obtain the sampled rotor electric angular velocity of the permanent magnet synchronous motor, and determine the first shaft current command based on the sampled rotor electric angular velocity.

[0047] Step 102: Determine the reference voltage vector based on the first axis current command and the preset second axis current command.

[0048] Step 103: Synthesize the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors, and determine multiple voltage vectors and the duty cycle of each voltage vector based on the volt-second balance algorithm.

[0049] Step 104: Determine the candidate voltage vector set by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector.

[0050] Step 105: Construct a cost function based on current error, current harmonics and switching frequency, and determine the optimal voltage vector combination and application time from the candidate voltage vector set according to the cost function.

[0051] Step 106: Determine the drive signal for each switch of the inverter based on the optimal voltage vector combination and the operating time, and control the permanent magnet synchronous motor based on the drive signal.

[0052] In practice, Figure 2 This is a hardware structure diagram of a permanent magnet synchronous motor according to an embodiment of this disclosure. Figure 2 As shown, the permanent magnet synchronous motor includes: a three-phase voltage source, a synchronous motor, a three-phase diode rectifier bridge, a DC-side capacitor, a voltage and current sampling circuit, a digital signal processor (DSP) controller, and a drive circuit. The voltage and current sampling circuit uses voltage Hall sensors and current Hall sensors to collect the DC-side voltage and the a-phase and b-phase currents of the synchronous motor, respectively. The sampled signals are then processed by a signal conditioning circuit and converted into digital signals by the DSP controller. The control method for the permanent magnet synchronous motor proposed in this embodiment is applied to the DSP controller, which outputs six switching pulses. These pulses then pass through the drive circuit to obtain the final drive signals for the six switching transistors of the inverter.

[0053] Figure 3 This is a schematic diagram of the current control structure of a permanent magnet synchronous motor based on a unified multi-vector method, according to an embodiment of this disclosure. Figure 3 As shown, the entire system adopts a series control structure, with the first axis current command (i.e., the q-axis current command) as the control structure. The speed is obtained through the outer loop PI controller. Ignoring field weakening operation and aiming for the maximum torque / current ratio, a second-axis current command (i.e., the d-axis current command) can be set. According to the q-axis current command and d-axis current command The reference voltage vector can be obtained using a model-free method based on hyperlocal adaptation. Based on the principle of volt-second balance, a reference voltage vector can be synthesized using two non-zero vectors and one zero vector, and the three-segment voltage vector u can be calculated. x u y The duty cycle T of u0 x T y T0. By adjusting the obtained voltage vectors and their application time, a candidate voltage vector set considering different numbers of voltage vectors is obtained. The optimal voltage vector combination is selected by evaluating a cost function with current error, current harmonics, and switching action penalties. The drive signal for each switch of the inverter is constructed based on the optimal voltage vector combination and application time.

[0054] Through the above embodiments, the sampled rotor electrical angular velocity of the permanent magnet synchronous motor is obtained, and the first axis current command is determined based on the sampled rotor electrical angular velocity. A reference voltage vector is determined according to the first axis current command and a pre-set second axis current command. The reference voltage vector is synthesized using a first preset number of non-zero vectors and a second preset number of zero vectors. Multiple voltage vectors and the duty cycle of each voltage vector are determined based on a volt-second balance algorithm. A candidate voltage vector set is determined by adjusting the number of voltage vectors acting within the control cycle and the acting time of each voltage vector. Thus, the number of voltage vectors in the candidate voltage vector set is not fixed, considering various voltage vector combinations. A cost function is constructed based on current error, current harmonics, and switching frequency. The optimal voltage vector combination and acting time are determined from the candidate voltage vector set according to the cost function, ensuring that the obtained optimal voltage vector combination and acting time represent the best voltage vector combination in terms of current error, current harmonics, and switching frequency, further improving steady-state performance. The drive signal for each switch of the inverter is determined based on the optimal voltage vector combination and acting time, and the permanent magnet synchronous motor is controlled based on the drive signal.

[0055] In some embodiments, step 101 includes:

[0056] Step 1011: Obtain the sampled rotor electric angular velocity of the permanent magnet synchronous motor.

[0057] Step 1012: Determine the torque command based on the sampled rotor electrical angular velocity using the proportional-integral regulator of the outer ring speed.

[0058]

[0059] in, For the torque command, k p k is the proportional gain of the proportional-integral controller. i Let ω be the integral gain of the proportional-integral controller, s be the complex variable in the Laplace transform, and ω be the integral gain of the proportional-integral controller. * The reference rotor electric angular velocity is ω, which is the sampled rotor electric angular velocity.

[0060] In practical implementation, the proportional-integral controller is a PI controller. The PI controller uses the given value and the actual output value to form a control deviation. The proportional and integral components of this deviation are then linearly combined to form the control quantity, which controls the controlled object. p k is the proportional gain of the proportional-integral controller, used to adjust the system's response speed. i is the integral gain of the proportional-integral controller, used to eliminate steady-state error. s is a complex variable in the Laplace transform, used for control system analysis. It is the transfer function of the proportional-integral controller, with the proportional term k.p Provides rapid response, while the points item Used to eliminate any steady-state error.

[0061] Step 1013: Determine the first shaft current command based on the torque command.

[0062]

[0063] in, This is the first axis current command. For the torque command, n p ψ is the number of pole pairs of the motor. f It is a permanent magnet flux linkage.

[0064] In practical implementation, in motor control, current is typically decomposed into d-axis (direct axis) and q-axis (quadrature axis) components. The first axis current command is the q-axis current command. The q-axis current command is usually directly related to the motor's torque generation. The second-axis current command is the d-axis current command.

[0065] n p ψ is the number of pole pairs in a motor, a fixed parameter indicating how many pairs of magnetic poles the motor has. f The flux linkage (or magnetic flux density) of a permanent magnet is a measure of the strength of the magnetic field generated by a permanent magnet. It is usually a constant that depends on the material and size of the permanent magnet.

[0066] The above scheme utilizes a proportional-integral regulator of the outer ring speed to accurately determine the torque command based on the sampled rotor electrical angular velocity. This allows for adjustment of the permanent magnet synchronous motor's torque output, ensuring that the sampled rotor electrical angular velocity of the permanent magnet synchronous motor closely approximates the reference rotor electrical angular velocity. Furthermore, the torque command accurately determines the first shaft current command, thereby setting the stator current vector, which helps ensure the permanent magnet synchronous motor operates at both the torque command and the reference rotor electrical angular velocity.

[0067] In some embodiments, step 102 includes:

[0068] Step 1021: Determine the stator current vector according to the first axis current command and the preset second axis current command.

[0069] In practice, the stator current vector is determined by the first axis current command. Second axis current command Composed of, according to the first axis current command Second axis current command Determine the stator current vector.

[0070] Step 1022: Construct a hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system based on the stator current vector.

[0071]

[0072] in, Here is the current equation for the permanent magnet synchronous motor, F is the system disturbance estimation parameter, α is the gain of the input variable, and u s These are the input variables for the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system.

[0073] In practical implementation, the current equation of the permanent magnet synchronous motor is determined based on the stator current vector in the αβ coordinate system.

[0074]

[0075] in, Here is the current equation for a permanent magnet synchronous motor, i s L is the stator current vector, t is time, and L is the current vector. s For synchronous inductors, u s R is the stator voltage vector. s For stator resistance, ψ r It is the rotor flux linkage vector. ψ f Let θ be the flux linkage of a permanent magnet, j be the imaginary part of the complex number, and θ be the flux linkage of the magnet. e For the reference rotor electrical angle, jω is the imaginary part of the angular frequency, ω = dθ e / dt is the reference rotor electric angular velocity.

[0076] Construct a first-order hyperlocal model of a single-input single-output system. The first-order hyperlocal model can be represented as:

[0077]

[0078] Where y is the output variable of the first-order hyperlocal model, F is the disturbance estimation parameter of the system under study, including the known or unknown parts of the system and various disturbances, α is the gain of the input variable, which is a non-physical parameter, and u is the input variable of the first-order hyperlocal model.

[0079] Construct a hyperlocal model of a permanent magnet synchronous motor in a stationary coordinate system.

[0080]

[0081] in, Let F be the current equation for the permanent magnet synchronous motor, F be the disturbance estimation parameter of the system, α be the gain of the input variable, and u be the current equation for the permanent magnet synchronous motor. s These are the input variables for the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system.

[0082] Due to the short control cycle, assuming that the system's disturbance estimation parameter F and the input variable gain α remain constant over the past two control cycles, the stator current relationship is as follows:

[0083]

[0084] Where k-1 represents the first historical moment, Let k-1 represent the first historical stator current vector, and k-2 represent the second historical moment. T is the second historical stator current vector at the second historical moment k-2. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. This represents the second historical stator voltage vector at the second historical time k-2, where k represents the current time. Let k be the current stator current vector at the current time. This is the first historical stator voltage vector at the first historical moment k-1.

[0085] Define the second historical current differential at the second historical time k-2 and the first historical current differential at the first historical time k-1.

[0086]

[0087] in, This is the differential of the second historical current at the second historical moment k-2.

[0088]

[0089] in, This is the differential of the first historical current at the first historical moment k-1.

[0090] Step 1023: Determine the gain of the input variables based on the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system.

[0091]

[0092] Where α is the gain of the input variable, For the first historical moment k-1, the differential of the first historical current. For the second historical time k-2, the differential of the second historical current. This represents the first historical stator voltage vector at the first historical moment k-1. The second historical stator voltage vector is the second historical moment k-2.

[0093] In practice, by subtracting formulas (6) and (7), the expression for the gain α of the input variable can be obtained as follows:

[0094]

[0095] Where α is the gain of the input variable.

[0096] Step 1024: Determine the disturbance estimation parameters of the system based on the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system.

[0097]

[0098] Where F is the interference estimation parameter of the system. Let α be the differential of the first historical current at the first historical time k-1, and let α be the gain of the input variable. This is the first historical stator voltage vector at the first historical moment k-1.

[0099] In practical implementation, by substituting formula (8) into formula (7), the expression for the system's interference estimation parameter F can be calculated.

[0100]

[0101] Where F is the system's disturbance estimation parameter.

[0102] Step 1025: Based on the gain of the input variables and the disturbance estimation parameters of the system, determine the first target stator current vector at the first target time.

[0103]

[0104] in, Let K be the first target stator current at the first target time k+1. Let T be the current stator current vector at time k. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. This is the current stator voltage vector at time k.

[0105] Step 1026: Based on the gain of the input variables and the disturbance estimation parameters of the system, determine the second target stator current vector at the second target time.

[0106]

[0107] in, The second target stator current is at the second target time k+2. Let T be the first target stator current vector at time k+1. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. The first target stator voltage vector is the first target time k+1.

[0108] Step 1027: Determine the second reference stator current vector at the second target time based on the first axis current command and the preset second axis current command.

[0109]

[0110] in, The second reference stator current vector at the second target time k+2. For the pre-set second axis current command, The first axis current command, j is the imaginary part of the complex number, θ e For reference rotor electrical angle, ω r T is the electric angular velocity of the sampled rotor. sv To control the cycle.

[0111] Step 1028: Replace the second target stator current vector with the second reference stator current vector, and determine the reference voltage vector based on the first target stator current vector and the second reference stator current vector.

[0112]

[0113] in, The reference voltage vector, The second reference stator current vector at the second target time k+2. Let T be the first target stator current vector at the first target time k+1. sc The control period is F, the disturbance estimation parameter of the system is F, and the gain of the input variable is α.

[0114] In practical implementation, a one-step delay exists between the actual output voltage and the command voltage in a real digital control system. This one-step delay has a more significant impact when the sampling frequency is low. To improve control accuracy, the current at the (k+1)th sampling moment is considered as the initial value for the next prediction. Based on the deadbeat principle, using... replace An ideal voltage vector can be obtained.

[0115]

[0116] in, For the reference voltage vector, The second reference stator current vector at the second target time k+2. Let T be the first target stator current vector at the first target time k+1. scThe control period is F, the disturbance estimation parameter of the system is F, and the gain of the input variable is α.

[0117] Using the above scheme, the constructed hyperlocal model can accurately predict the first target stator current vector at the first target time and the second target stator current vector at the second target time. The second reference stator current vector at the second target time is determined based on the first axis current command and the pre-set second axis current command. Based on the deadbeat principle, the reference voltage vector can be quickly and accurately determined using the first target stator current vector and the second reference stator current vector.

[0118] In some embodiments, step 103 includes:

[0119] Step 1031: Synthesize the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors; wherein the first preset number of non-zero vectors and the second preset number of zero vectors include: a first voltage vector, a second voltage vector, and a third voltage vector.

[0120] In practice, based on space vector modulation (SVM), the reference voltage vector can be synthesized from two non-zero vectors and one zero vector. Figure 4 This is a schematic diagram of the synthesized reference voltage vector according to an embodiment of this disclosure. Figure 4 As shown, the reference voltage vector It can be composed of a first voltage vector u1 (100) at a first action time T1 and a second voltage vector u2 (110) at a second action time T2. The reference voltage vector... The angle between the reference voltage vector and the first voltage vector u1(100) is θ.

[0121] Step 1032: Determine the first duty cycle of the first voltage vector based on the modulation ratio and the reference voltage vector angle.

[0122]

[0123] Among them, T x θ represents the first duty cycle of the first voltage vector segment, M represents the modulation ratio, and θ represents the reference voltage vector angle.

[0124] Step 1033: Determine the second duty cycle of the second voltage vector based on the modulation ratio and the reference voltage vector angle.

[0125] T y =Msin(θ)

[0126] Among them, T y θ represents the second duty cycle of the second voltage vector segment, M represents the modulation ratio, and θ represents the reference voltage vector angle.

[0127] Step 1034: Based on the volt-second balance algorithm, determine the third duty cycle of the third voltage vector according to the first duty cycle and the second duty cycle.

[0128] T0 = ​​1 - T x -T y

[0129] Where T0 is the third duty cycle of the third segment voltage vector, T x T represents the first duty cycle of the first voltage vector segment. y This is the second duty cycle of the second voltage vector.

[0130] In specific implementation, such as Figure 4 As shown, the three-segment voltage vector u is calculated based on the volt-second balance principle. x u y Duty cycle T of u0 x T y T0,

[0131]

[0132] T y =Msin(θ) (13)

[0133] T0 = ​​1 - T x -T y (13)

[0134] Where M is the modulation ratio. U dc This is the DC bus voltage, and θ is the reference voltage vector angle.

[0135] The first target stator voltage vector at the first target time k+1 can be obtained from formula (11). and the second target stator voltage vector at the second target time k+2

[0136]

[0137] The current error at the second target time k+2 can be represented by the reference voltage vector. and the first target stator voltage vector at the first target time k+1 The difference is used to express the result, that is, subtracting formula (14) from formula (12).

[0138]

[0139] Therefore, minimizing the voltage error can be equivalently achieved by minimizing the current error.

[0140] Using the above scheme, based on the volt-second balance algorithm, the first duty cycle of the first voltage vector, the second duty cycle of the second voltage vector, and the third duty cycle of the third voltage vector can be accurately determined, so as to determine the action time of each voltage vector in the voltage vector combination based on the first duty cycle, the second duty cycle, and the third duty cycle.

[0141] In some embodiments, step 104 includes:

[0142] Step 1041: In response to determining that the number of voltage vectors is one, construct a first voltage vector combination based on the one voltage vector, and determine the first volt-second error corresponding to each first voltage vector combination.

[0143] In practice, when determining the candidate vector set, it is divided into four types according to the number of voltage vectors acting in one control cycle, namely, one, two, three, and four voltage vectors acting in one control cycle. The following uses the first sector as an example to introduce the candidate vector set determined when the reference voltage vector falls in the first sector.

[0144] When a control cycle applies a voltage vector, taking the first sector as an example, the voltage vectors to be considered are u1, u2, and u0, and their corresponding application times are t1, t2, and t0, respectively, where t1 = T1T. sc t2=T2T sc , t0=T0T sc Following the principle of volt-second balance, we can obtain:

[0145]

[0146] Where u1 is the first non-zero voltage vector, t1 is the first action time, u2 is the second non-zero voltage vector, t2 is the second action time, u0 is the zero voltage vector, and t0 is the third action time.

[0147] T sc =t1+t2+t0 (17)

[0148] The volt-second error (u) caused when the three voltage vectors (u1, u2, u0) act on the entire control cycle is... ε1 u ε2 u ε0 )for:

[0149]

[0150] Among them, u ε0 The first volt-second error caused by the zero voltage vector t0 acting throughout the entire control cycle, t ε1 The first volt-second error caused by the first non-zero voltage vector t1 acting throughout the entire control cycle, tε2 The first volt-second error caused when the second non-zero voltage vector u2 acts throughout the entire control cycle.

[0151] Since |u1|=|u2|, the first volt-second error u ε0 The square value is:

[0152]

[0153] Similarly, the first volt-second error u ε1 u ε2 The square value is:

[0154]

[0155] It's difficult to compare directly |u ε0 | 2 、|u ε1 | 2 、|u ε2 | 2 The values ​​are , but the differences between them can be compared, and these differences are expressed as:

[0156]

[0157] From the above formula, it can be seen that if t0 = max(t0, t1, t2), then |u ε1 | 2 -|u ε0 | 2 >0 and |u ε2 | 2 -|u ε0 | 2 >0. This means |u ε0 |ratio|u ε1 | and |u ε2 | They all need to be small. And u ε0 It is the reference voltage vector The first volt-second error generated when the zero voltage vector u0 acts throughout the entire control cycle is clearly the voltage vector that minimizes the first volt-second error. Similarly, it can be confirmed that when only one voltage vector is applied in a control cycle, the voltage vector with the longest duration is optimal, and the results are summarized in Table 1. To reduce the switching frequency, when the optimal vector is the zero vector, the last voltage vector u0 at the end of the previous cycle is used as the reference. old To select, if the last voltage vector at the end of the previous cycle is u1, u3, u5, or u0, select the zero voltage vector u0; otherwise, select u7.

[0158] Table 1 shows the vector selection and application time when only one voltage vector is applied.

[0159] Time conditions <![CDATA[t0>max(t1,t2)]]> <![CDATA[t1>max(t0,t2)]]> <![CDATA[t2>max(t0,t1)]]> Optimal Vector <![CDATA[u0]]> <![CDATA[u1]]> <![CDATA[u2 <!-- 10 -->]]> Duration of action <![CDATA[T sc ]]> <![CDATA[T sc ]]> <![CDATA[T sc ]]>

[0160] Step 1042: In response to determining that the number of voltage vectors is two, the two voltage vectors are combined to construct a second voltage vector combination, and the second volt-second error corresponding to each second voltage vector combination is determined.

[0161] In practical implementation, when two voltage vectors are applied in one control cycle, the voltage vector combinations to be considered are (u1, u0), (u2, u7), and (u1, u2). Taking (u1, u2) as an example, the second volt-second error generated when the voltage vector combination (u1, u2) applies throughout the entire control cycle is:

[0162] u ε12 = u1t1+u2t2+u0t0-u1t1-u2t2

[0163] =u1t1+u2t2-u1(t1+kt0)-u2(t2+(1-k)t0)

[0164] =-t0(ku1+(1-k)u2) (22)

[0165] Where t′1=t1+kt0 and t′2=t2+(1-k)t0 represent the final action time of u1 and u2, and k is the proportional coefficient that allocates the action time of the unselected voltage vector to the selected voltage vector.

[0166] So, |u ε12 The square of | can be calculated as follows:

[0167] |u ε12 | 2 =|u1| 2 t0 2 (1-k+k 2 ) (twenty three)

[0168] When k = 0.5, we can obtain |u| from the above formula. ε12 | 2 The minimum value. |u ε12 The minimum value of | is:

[0169]

[0170] Similarly, the second volt-second error of the voltage vector combination is minimized when the duration of the unselected voltage vector is evenly distributed among the two voltage vectors in the combination. The minimum second volt-second error generated by the voltage vector combinations (u1, u0) and (u2, u7) is:

[0171]

[0172] It is easy to see from formulas (24) to (26) that the optimal voltage vector combination is the combination with the longest operating time, and the operating time of the unselected voltage vectors is evenly distributed among the two voltage vectors in the voltage vector combination. At the same time, an improved dual voltage vector combination is added. If the two voltage vectors are a non-zero voltage vector and a zero voltage vector, then the pulse sequence is that the voltage vector with fewer switching transitions with the zero voltage vector u0 is placed on both sides. Taking the first sector as an example, that is, (u1,u0,u1), (u2,u7,u2) and (u1,u2,u1).

[0173] The application time of the voltage vectors placed on both sides in the dual-vector system is optimized and improved based on the principle of minimizing current harmonics. Assume that the selected voltage vector sequence at the first target time k+1 is u. x ,u y ,u x The duration of action is λt x ,t y ,(1-λ)t x .

[0174] The formula for calculating the root mean square (RMS) value of harmonic current within one control cycle is as follows:

[0175]

[0176] Among them, i h (t)=i ref (t)-i(t) is the error between the reference value and the actual value of the stator current.

[0177] Since three voltage vectors are applied within the control cycle, the RMS value of the current harmonics can be obtained by adding the values ​​of the three voltage vectors, as expressed by:

[0178]

[0179] Where T(j) is the duration of the applied voltage vector within a control cycle.

[0180] Figure 5 This is a schematic diagram illustrating the segmentation of harmonic current according to an embodiment of this disclosure. For example... Figure 5 As shown, the harmonic current can be divided into three segments with four endpoints: e0, e1, e2, and e3. e0 is the starting point, and its expression is:

[0181]

[0182] in, The reference current vector is the current vector at the first target time k+1.

[0183] When the voltage vector ux Effect λt x Then, the predicted current at this time is:

[0184]

[0185] The predicted reference current at this time is:

[0186]

[0187] The expression for the endpoint e1 of the harmonic current after the application of the first voltage vector is:

[0188]

[0189] in,

[0190] Similarly, the expression for endpoint e2 is:

[0191]

[0192] in,

[0193] Endpoint e3 is the endpoint, and the expression is:

[0194]

[0195] Among them, i k+2 It is the predicted current at the second target time k+2. and

[0196] Let k1 represent Let k2 represent At this point, the expressions for endpoints e1 and e2 are:

[0197] e1=e0+k1λt x (35)

[0198] e2=e1+k2t y (36)

[0199] To avoid complex integral calculations, the squared value of the current harmonics for one control cycle is calculated separately for each of the three current harmonic segments and then summed. The formula for calculating the squared value of the RMS of each current harmonic segment is as follows:

[0200]

[0201] Here, y1 and y2 represent the starting and ending points of the line segment.

[0202] The total square value of current harmonics was calculated. It is a quadratic function of λ. Therefore, by solving... We can obtain λ that minimizes current harmonics. opt The expression is:

[0203]

[0204] To reduce the switching frequency, according to u old To select the order of voltage vectors in the optimal voltage vector combination, the voltage vector combination is related to u old The voltage vector with fewer switching operations is placed first. When the optimal vector combination is (u1, u2), u old When the voltage vector is one of u0, u1, u5, or u6, the traditional vector combination to be considered is (u1, u2), and the improved dual-vector combination is (u1, u2, u1). In summary, the results when two voltage vectors are applied are summarized in Tables 2 and 3.

[0205] Table 2. Traditional dual voltage vector selection and application time when using two voltage vectors.

[0206]

[0207]

[0208] Table 3. Selection and application time of the improved symmetric dual voltage vector when using two voltage vectors.

[0209]

[0210] Step 1043: In response to determining that the number of voltage vectors is three, the three voltage vectors are combined to construct a third voltage vector combination, and the third volt-second error corresponding to each third voltage vector combination is determined.

[0211] In practice, when three voltage vectors are applied in one control cycle, the three voltage vectors can be synthesized into a reference voltage vector. Eliminate current tracking error. Similarly, based on the last voltage vector u at the end of the previous cycle... old The order of vector combinations is selected, and there is at most one switching transition between two control cycles. When the last voltage vector u at the end of the previous cycle... old When u is one of u0, u1, u2, or u7, there is no switch transition between two control cycles, and the switch transitions twice within the current control cycle. old When u is one of u3, u4, u5, or u6, there is one switch transition between two control cycles, and two switch transitions within the current control cycle. When u oldWhen the vector sequence is u3 or u6, there are two possible vector sequences. Taking u3 as an example, u2, u0, and u3 all have only one switching transition, and the current error is 0. However, the current harmonics generated by these two vector sequences may be different. The results are summarized in Table 4.

[0212] Table 4 shows the voltage vector selection and application time when using three voltage vectors.

[0213]

[0214] Step 1044: In response to determining that the voltage vector data is four, the four voltage vectors are combined to construct a fourth voltage vector combination, and the fourth volt-second error corresponding to each fourth voltage vector combination is determined.

[0215] In practical implementation, four voltage vectors are applied within one control cycle to achieve deadbeat control with a fixed switching frequency, essentially equivalent to deadbeat control based on the SVM principle. However, the performance of the traditional 0127 sequence degrades at higher speeds and lower carrier ratios, necessitating consideration of bus clamp SVM technology. Considering a maximum of three switching operations within one control cycle, five vector sequences are considered: 1) Sequence 0127 / 7210; 2) Sequence 0121 / 1210; 3) Sequence 1012 / 2101; 4) Sequence 2721 / 1272; 5) Sequence 7212 / 2127. The current harmonic content of sequences 0127 and 7210, 0121 and 1210, 1012 and 2101, 2721 and 1272, and 7212 and 2127 is equal within one control cycle. To reduce the switching frequency, the order of vector combinations is selected based on the last voltage vector at the end of the previous cycle, ensuring at most one switching transition between two control cycles. The results are summarized in Table 5.

[0216] Table 5 shows the voltage vector selection and application time when using four voltage vectors.

[0217]

[0218] Step 1045: Select multiple combinations of the first voltage vectors, multiple combinations of the second voltage vectors, multiple combinations of the third voltage vectors, and multiple combinations of the fourth voltage vectors as a candidate voltage vector set.

[0219] In specific implementation, the candidate voltage vector set includes voltage vector combinations obtained by combining different numbers of voltage vectors. Specifically, the first voltage vector combination consists of one voltage vector, the second voltage vector combination consists of two voltage vectors, the third voltage vector combination consists of three voltage vectors, and the fourth voltage vector combination consists of four voltage vectors.

[0220] The above scheme uses multiple combinations of first voltage vectors, multiple combinations of second voltage vectors, multiple combinations of third voltage vectors, and multiple combinations of fourth voltage vectors as candidate voltage vector sets. Thus, the number of voltage vectors in the candidate voltage vector sets is not fixed, considering various voltage vector combinations.

[0221] In some embodiments, step 105 includes:

[0222] Step 1051: Construct a cost function based on current error, current harmonics, and switching frequency.

[0223]

[0224] Where J is the cost function, The second reference stator current vector at the second target time k+2. Ih is the second target stator current vector at the second target time k+2. rms For the current harmonics, g sw The weighting coefficients for the switching frequency are given, k+1 is the first target time, and f is the weighting coefficient for the switching frequency. sw The switching frequency is [value].

[0225] In practical implementation, when one or two voltage vectors are applied in a control cycle, a current tracking error exists at the end of the control cycle; when three or four voltage vectors are applied, there is no current tracking error at the end of the control cycle. However, different sequences generate different current harmonics. Furthermore, in order to control the switching frequency, a cost function J, which includes current error, current harmonics, and switching frequency, is established.

[0226]

[0227] Among them, g sw ih is the weighting coefficient for the switching frequency. rms For current harmonics, f sw This represents the switching frequency.

[0228] The current error can be converted into the voltage error using formula (15), and the cost function J can also be written as:

[0229]

[0230] To calculate current harmonics, we first need to calculate the current i(n) after applying each voltage vector in the candidate voltage vector combination:

[0231]

[0232] Where m is the number of voltage vectors applied in the (k+1)th control cycle, i(n) is the current after applying the nth voltage vector, and u s(j) is the j-th voltage vector, T sc (j) represents the duration of the j-th voltage vector. The i corresponding to i(n) ref The formula for calculating (n) is:

[0233]

[0234] Then the harmonic current i after applying the nth voltage vector h The formula for calculating (t) is:

[0235] i h (t)=i ref (n)-i(n) (n=1,2,…,m) (43)

[0236] RMS value of harmonic current The calculation formula is:

[0237]

[0238] Switching frequency f within one control cycle sw The defining formula is:

[0239]

[0240] Where N represents the total number of switching actions within one control cycle, l is the number of phases, and z is the number of switching devices per phase. For a three-phase two-level inverter, the switching frequency f sw It can be represented as:

[0241]

[0242] g sw The size of g affects the priority of the switching action penalty. sw The larger the value of g, the more the controller tends to select voltage vector combinations with fewer switching actions, while relaxing control over current tracking and steady-state performance. To ensure a constant switching frequency in the control method, g... sw The value f is calculated based on the switching frequency. sw With switching frequency reference value The size relationship between them can be adjusted online.

[0243]

[0244] Where Δg > 0 represents the adjustment range of the weighting coefficient. Generally, a larger Δg will result in a faster dynamic response. However, the accuracy of the switching frequency control will decrease. In order to accurately control the switching frequency while ensuring a fast dynamic response, after multiple comparative experiments, the embodiments of this disclosure set Δg as follows:

[0245]

[0246] The above scheme constructs a cost function based on current error, current harmonics, and switching frequency. In this way, the cost function can balance the relationship between current error, current harmonics, and switching frequency, thereby improving steady-state performance.

[0247] In some embodiments, step 105 includes:

[0248] Step 1052: Substitute the voltage vector combinations in the candidate voltage vector set into the cost function in sequence to obtain the cost value corresponding to each voltage vector combination.

[0249] Step 1053: Determine the voltage vector combination with the minimum substitution value from the candidate voltage vector set, take the voltage vector combination with the minimum substitution value as the optimal voltage vector combination, and determine the action time corresponding to the optimal voltage vector combination.

[0250] In practice, the candidate vector voltage set is substituted into the cost function, and the solution that minimizes the cost function is the optimal solution (U). opt ,T opt The cost function effectively balances the relationship between current tracking error, current harmonics, and switching frequency, thereby improving steady-state performance.

[0251] The above scheme involves sequentially substituting the voltage vector combinations from the candidate voltage vector set into the cost function to obtain the cost value for each combination. The voltage vector combination with the minimum cost value is then determined from the candidate voltage vector set and selected as the optimal voltage vector combination. The corresponding application time for this optimal combination is then determined. In this way, the cost function can balance the relationship between current error, current harmonics, and switching frequency, thereby improving steady-state performance.

[0252] Through the above embodiments, the sampled rotor electrical angular velocity of the permanent magnet synchronous motor is obtained, and the first axis current command is determined based on the sampled rotor electrical angular velocity. A reference voltage vector is determined according to the first axis current command and a pre-set second axis current command. The reference voltage vector is synthesized using a first preset number of non-zero vectors and a second preset number of zero vectors. Multiple voltage vectors and the duty cycle of each voltage vector are determined based on a volt-second balance algorithm. A candidate voltage vector set is determined by adjusting the number of voltage vectors acting within the control cycle and the acting time of each voltage vector. Thus, the number of voltage vectors in the candidate voltage vector set is not fixed, considering various voltage vector combinations. A cost function is constructed based on current error, current harmonics, and switching frequency. The optimal voltage vector combination and acting time are determined from the candidate voltage vector set according to the cost function, ensuring that the obtained optimal voltage vector combination and acting time represent the best voltage vector combination in terms of current error, current harmonics, and switching frequency, further improving steady-state performance. The drive signal for each switch of the inverter is determined based on the optimal voltage vector combination and acting time, and the permanent magnet synchronous motor is controlled based on the drive signal.

[0253] The effectiveness of the method proposed in this disclosure can be verified by examining the implementation of the method in this embodiment. Figures 6 to 11 The experimental results shown are as follows. Figure 6 This is a schematic diagram illustrating the motor test results of the 1VB-MFPCC according to an embodiment of this disclosure. Figure 6 The figure shows the experimental results of a motor operating at 2000 r / min with rated load using 1VB Model-Free Predictive Current Control (1VB-MFPCC). Figure 7 This is a schematic diagram illustrating the motor test results of the SVM-MFPCC according to an embodiment of this disclosure. Figure 7 The figure shows the experimental results of a motor operating at 2000 r / min with rated load using Space Vector Modulation Model-Free Predictive Current Control (SVM-MFPCC). Figure 8 This is a schematic diagram illustrating the motor test results of the GMV-MFPCC according to an embodiment of this disclosure. Figure 8 The figure shows the experimental results of a model-free predictive current control (GMV-MFPCC) motor operating at 2000 r / min with rated load, based on the optimization objective. The sampling frequencies for the three methods were set to 22 kHz, 2.5 kHz, and 9 kHz, respectively, achieving the same switching frequency of 2.5 kHz. Figures 6 to 8In the middle, the waveforms from top to bottom represent the rotational speed ω. r q-axis current and i q d-axis current and i d Switching frequency f sw and the stator terminal a-phase current i a .

[0254] Figure 9 This is a schematic diagram of the current harmonic analysis results of the 1VB-MFPCC according to an embodiment of this disclosure. Figure 10 This is a schematic diagram of the current harmonic analysis results of the SVM-MFPCC according to an embodiment of this disclosure. Figure 11 This is a schematic diagram showing the current harmonic analysis results of the GMV-MFPCC according to an embodiment of this disclosure. Figures 9 to 11 There are three methods in Figures 6 to 8 Current harmonic analysis results under operating conditions. At this time, the switching frequencies of the 1VB-MFPCC, SVM-MFPCC, and GMV-MFPCC methods are 2486Hz, 2500Hz, and 2479Hz, respectively. Compared with SVM-MFPCC and 1VB-MFPCC, the GMV-MFPCC proposed in this embodiment reduces the total harmonic distortion (THD) by 47.29% and 1.2%, respectively.

[0255] Figure 12 This diagram illustrates the total harmonic distortion (THD) of the current and the switching frequency at different speeds according to embodiments of this disclosure. Figure 12 As shown, the current THD and switching frequency of the three methods under different speeds and rated loads are compared. Compared with 1VB-MFPCC and SVM-MFPCC, the proposed GMV-MFPCC method exhibits better steady-state performance at different speeds and lower current THD at the same switching frequency. Figure 12 As can be seen, the current THD of 1VB-MFPCC and GMV-MFPCC is basically equal at rated speed, but under other operating conditions, GMV-MFPCC has better steady-state performance and lower current THD. SVM-MFPCC has a current THD that is basically equal to GMV-MFPCC at speeds of 400-1000 r / min; however, at higher speeds and with lower carrier ratios, the performance of SVM-MFPCC degrades. Compared with SVM-MFPCC and 1VB-MFPCC, the method proposed in this disclosure reduces current THD by a maximum of 47.29% and 46.85%, respectively. Simultaneously, the GMV-MFPCC method achieves frequency following of the reference value, enabling a constant switching frequency across the entire operating range.

[0256] Figure 13 This is a schematic diagram illustrating the dynamic response of GMV-MFPCC according to an embodiment of this disclosure. Figure 13 As shown, the dynamic response of the GMV-MFPCC from +2000 r / min to -2000 r / min is presented. The forward and reverse switching is smooth, and it can quickly and accurately track the q-axis current reference value. A step change. Figure 14 This is a schematic diagram illustrating the experimental results of the GMV-MFPCC according to an embodiment of this disclosure when the reference switching frequency increases from 2kHz to 2.5kHz at rated speed. Figure 14 The results shown are experimental findings of the GMV-MFPCC at rated speed and rated load, stepping from 2kHz to 2.5kHz. It can be clearly seen that, due to online adjustment, it quickly tracks the reference value within approximately 0.07s.

[0257] Figure 15 This is a schematic diagram illustrating the steady-state test results of the GMV-MFPCC according to an embodiment of this disclosure at 2000 r / min with rated load. Figure 15 The figure shows the steady-state experimental results of the GMV-MFPCC method for the hyperlocal model under rated load at 2000 r / min when parameters are mismatched. From top to bottom, the waveforms represent the rotor speed ω. r q-axis current and i q d-axis current and i d The ratio of the actual inductance value to the nominal value L r / L and A phase stator current i a The machine parameters used in the control method are initially accurate, such as... Figure 15 As shown in the left half of the diagram. When the inductor in the controller becomes 180% of its nominal value, as... Figure 15 As shown in the right half, since no parameters are used in the hyperlocal model, the performance of the GMV-MFPCC proposed in this disclosure is not affected by parameter mismatch. Experimental results clearly demonstrate that the proposed GMV-MFPCC has strong robustness to changes in machine parameters.

[0258] It should be noted that the method of this disclosure embodiment can be executed by a single device, such as a computer or server. The method of this embodiment can also be applied to a distributed scenario, where multiple devices cooperate to complete the task. In such a distributed scenario, one of these devices may execute only one or more steps of the method of this disclosure embodiment, and the multiple devices will interact with each other to complete the method described.

[0259] It should be noted that the above description describes some embodiments of this disclosure. Other embodiments are within the scope of the appended claims. In some cases, the actions or steps recorded in the claims can be performed in a different order than that shown in the above embodiments and still achieve the desired result. Furthermore, the processes depicted in the drawings do not necessarily require a specific or sequential order to achieve the desired result. In some embodiments, multitasking and parallel processing are also possible or may be advantageous.

[0260] Based on the same inventive concept, corresponding to any of the above embodiments, this disclosure also provides a control device for a permanent magnet synchronous motor.

[0261] refer to Figure 16 The control device for the permanent magnet synchronous motor includes:

[0262] The first axis current command determination module 301 is configured to acquire the sampled rotor electric angular velocity of the permanent magnet synchronous motor and determine the first axis current command based on the sampled rotor electric angular velocity.

[0263] The reference voltage vector determination module 302 is configured to determine the reference voltage vector based on the first axis current command and a preset second axis current command.

[0264] The duty cycle determination module 303 is configured to synthesize the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors, and to determine multiple voltage vectors and the duty cycle of each voltage vector based on a volt-second balance algorithm.

[0265] The candidate voltage vector set determination module 304 is configured to determine the candidate voltage vector set by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector.

[0266] The optimal voltage vector combination determination module 305 is configured to construct a cost function based on current error, current harmonics and switching frequency, and determine the optimal voltage vector combination and application time from the candidate voltage vector set according to the cost function.

[0267] The control module 306 is configured to determine the drive signal for each switch of the inverter based on the optimal voltage vector combination and the operating time, and to control the permanent magnet synchronous motor based on the drive signal.

[0268] In some embodiments, the first shaft current command determination module 301 includes:

[0269] The sampling rotor electric angular velocity acquisition unit is configured to acquire the sampling rotor electric angular velocity of the permanent magnet synchronous motor;

[0270] The torque command determination unit is configured to determine the torque command based on the sampled rotor electrical angular velocity using a proportional-integral regulator of the outer loop speed.

[0271]

[0272] in, For the torque command, k p k is the proportional gain of the proportional-integral controller. i Let ω be the integral gain of the proportional-integral controller, s be the complex variable in the Laplace transform, and ω be the integral gain of the proportional-integral controller. * The reference rotor electric angular velocity is ω, which is the sampled rotor electric angular velocity.

[0273] The first axis current command determination unit is configured to determine the first axis current command based on the torque command.

[0274]

[0275] in, This is the first axis current command. For the torque command, n p ψ is the number of pole pairs of the motor. f It is a permanent magnet flux linkage.

[0276] In some embodiments, the reference voltage vector determination module 302 includes:

[0277] The stator current vector determination unit is configured to determine the stator current vector based on the first axis current command and a preset second axis current command;

[0278] The hyperlocal model construction unit is configured to construct a hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system based on the stator current vector.

[0279]

[0280] in, Here is the current equation for the permanent magnet synchronous motor, F is the system disturbance estimation parameter, α is the gain of the input variable, and u s These are the input variables for the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system;

[0281] The gain determination unit is configured to determine the gain of the input variables based on the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system.

[0282]

[0283] Where α is the gain of the input variable, For the first historical moment k-1, the differential of the first historical current. For the second historical time k-2, the differential of the second historical current. This represents the first historical stator voltage vector at the first historical moment k-1. The second historical stator voltage vector at the second historical moment k-2;

[0284] The disturbance estimation parameter determination unit is configured to determine the disturbance estimation parameters of the system based on the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system.

[0285]

[0286] Where F is the interference estimation parameter of the system. Let α be the differential of the first historical current at the first historical time k-1, and let α be the gain of the input variable. The first historical stator voltage vector at the first historical moment k-1;

[0287] The first target stator current vector determination unit is configured to determine the first target stator current vector at the first target time based on the gain of the input variables and the disturbance estimation parameters of the system.

[0288]

[0289] in, Let K be the first target stator current at the first target time k+1. Let T be the current stator current vector at time k. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. The current stator voltage vector at time k;

[0290] The second target stator current vector determination unit is configured to determine the second target stator current vector at the second target time based on the gain of the input variables and the disturbance estimation parameters of the system.

[0291]

[0292] in, The second target stator current is at the second target time k+2. Let T be the first target stator current vector at time k+1. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. The first target stator voltage vector at the first target time k+1;

[0293] The second reference stator current vector determination unit is configured to determine the second reference stator current vector at the second target time based on the first axis current command and a preset second axis current command.

[0294]

[0295] in, The second reference stator current vector at the second target time k+2. For the pre-set second axis current command, Here, θ is the first axis current command, j is the imaginary part of the complex number, and θ is the current. e For reference rotor electrical angle, ω r T is the electric angular velocity of the sampled rotor. sc To control the cycle;

[0296] The reference voltage vector determination unit is configured to replace the second target stator current vector with the second reference stator current vector, and determine the reference voltage vector based on the first target stator current vector and the second reference stator current vector.

[0297]

[0298] in, The reference voltage vector, The second reference stator current vector at the second target time k+2. Let T be the first target stator current vector at the first target time k+1. sc The control period is F, the disturbance estimation parameter of the system is F, and the gain of the input variable is α.

[0299] In some embodiments, the duty cycle determination module 303 includes:

[0300] The reference voltage vector synthesis unit is configured to synthesize the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors; wherein the first preset number of non-zero vectors and the second preset number of zero vectors include: a first voltage vector, a second voltage vector, and a third voltage vector;

[0301] The first duty cycle determining unit is configured to determine the first duty cycle of the first voltage vector based on the modulation ratio and the reference voltage vector angle.

[0302]

[0303] Among them, T x The first duty cycle of the first voltage vector segment is M, the modulation ratio is θ, and the reference voltage vector angle is θ.

[0304] The second duty cycle determining unit is configured to determine the second duty cycle of the second voltage vector based on the modulation ratio and the reference voltage vector angle.

[0305] T y =M sin(θ)

[0306] Among them, T y The second duty cycle of the second voltage vector is M, where M is the modulation ratio and θ is the reference voltage vector angle.

[0307] The third duty cycle determination unit is configured to determine the third duty cycle of the third voltage vector based on the volt-second balance algorithm, according to the first duty cycle and the second duty cycle.

[0308] T0 = ​​1 - T x -T y

[0309] Where T0 is the third duty cycle of the third segment voltage vector, T x T represents the first duty cycle of the first voltage vector segment. y This is the second duty cycle of the second voltage vector.

[0310] In some embodiments, the candidate voltage vector set determination module 304 includes:

[0311] The first voltage vector combination construction unit is configured to, in response to determining that the number of voltage vectors is one, construct a first voltage vector combination based on a voltage vector, and determine a first volt-second error corresponding to each first voltage vector combination;

[0312] The second voltage vector combination construction unit is configured to combine the two voltage vectors to construct a second voltage vector combination in response to determining that the number of voltage vectors is two, and to determine the second volt-second error corresponding to each second voltage vector combination;

[0313] The third voltage vector combination construction unit is configured to combine the three voltage vectors to construct a third voltage vector combination in response to determining that the number of voltage vectors is three, and to determine the third volt-second error corresponding to each third voltage vector combination;

[0314] The fourth voltage vector combination construction unit is configured to combine the four voltage vectors to construct a fourth voltage vector combination in response to determining that the voltage vector data is four, and to determine the fourth volt-second error corresponding to each fourth voltage vector combination;

[0315] The candidate voltage vector set determination unit is configured to use a plurality of first voltage vector combinations, a plurality of second voltage vector combinations, a plurality of third voltage vector combinations, and a plurality of fourth voltage vector combinations as candidate voltage vector sets.

[0316] In some embodiments, the optimal voltage vector combination determination module 305 includes:

[0317] The cost function construction unit is configured to construct a cost function based on current error, current harmonics, and switching frequency.

[0318]

[0319] Where J is the cost function, The second reference stator current vector at the second target time k+2. Ih is the second target stator current vector at the second target time k+2. rms For the current harmonics, g sw The weighting coefficients for the switching frequency are given, k+1 is the first target time, and f is the weighting coefficient for the switching frequency. sw The switching frequency is [value].

[0320] In some embodiments, the optimal voltage vector combination determination module 305 includes:

[0321] The cost value determination unit is configured to sequentially substitute the voltage vector combinations in the candidate voltage vector set into the cost function to obtain the cost value corresponding to each voltage vector combination.

[0322] The optimal voltage vector combination determination unit is configured to determine the voltage vector combination with the minimum cost value from the candidate voltage vector set, take the voltage vector combination with the minimum cost value as the optimal voltage vector combination, and determine the action time corresponding to the optimal voltage vector combination.

[0323] For ease of description, the above apparatus is described in terms of its functions, divided into various modules. Of course, in implementing this disclosure, the functions of each module can be implemented in one or more software and / or hardware.

[0324] The apparatus of the above embodiments is used to implement the control method of the corresponding permanent magnet synchronous motor in any of the foregoing embodiments, and has the beneficial effects of the corresponding method embodiments, which will not be repeated here.

[0325] Based on the same inventive concept, corresponding to the methods of any of the above embodiments, this disclosure also provides an electronic device, including a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein the processor executes the program to implement the control method of the permanent magnet synchronous motor described in any of the above embodiments.

[0326] Figure 17This embodiment illustrates a more specific hardware structure of an electronic device, which may include a processor 1010, a memory 1020, an input / output interface 1030, a communication interface 1040, and a bus 1050. The processor 1010, memory 1020, input / output interface 1030, and communication interface 1040 are interconnected internally via the bus 1050.

[0327] The processor 1010 can be implemented using a general-purpose CPU (Central Processing Unit), microprocessor, application-specific integrated circuit (ASIC), or one or more integrated circuits, and is used to execute relevant programs to implement the technical solutions provided in the embodiments of this specification.

[0328] The memory 1020 can be implemented in the form of ROM (Read Only Memory), RAM (Random Access Memory), static storage device, dynamic storage device, etc. The memory 1020 can store the operating system and other applications. When the technical solutions provided in the embodiments of this specification are implemented by software or firmware, the relevant program code is stored in the memory 1020 and is called and executed by the processor 1010.

[0329] The input / output interface 1030 is used to connect input / output modules to realize information input and output. Input / output modules can be configured as components within the device (not shown in the figure) or externally connected to the device to provide corresponding functions. Input devices may include keyboards, mice, touchscreens, microphones, various sensors, etc., while output devices may include displays, speakers, vibrators, indicator lights, etc.

[0330] The communication interface 1040 is used to connect a communication module (not shown in the figure) to enable communication between this device and other devices. The communication module can communicate via wired means (such as USB (Universal Serial Bus), network cable, etc.) or wireless means (such as mobile network, WIFI (Wireless Fidelity), Bluetooth, etc.).

[0331] Bus 1050 includes a pathway for transmitting information between various components of the device, such as processor 1010, memory 1020, input / output interface 1030, and communication interface 1040.

[0332] It should be noted that although the above-described device only shows the processor 1010, memory 1020, input / output interface 1030, communication interface 1040, and bus 1050, in specific implementations, the device may also include other components necessary for normal operation. Furthermore, those skilled in the art will understand that the above-described device may only include the components necessary for implementing the embodiments of this specification, and not necessarily all the components shown in the figures.

[0333] The electronic devices described above are used to implement the control methods of the corresponding permanent magnet synchronous motors in any of the foregoing embodiments, and have the beneficial effects of the corresponding method embodiments, which will not be repeated here.

[0334] Based on the same inventive concept, corresponding to the methods of any of the above embodiments, this disclosure also provides a non-transitory computer-readable storage medium that stores computer instructions for causing the computer to execute the control method of the permanent magnet synchronous motor as described in any of the above embodiments.

[0335] The computer-readable medium of this embodiment includes permanent and non-permanent, removable and non-removable media, and information storage can be implemented by any method or technology. Information can be computer-readable instructions, data structures, program modules, or other data. Examples of computer storage media include, but are not limited to, phase-change memory (PRAM), static random access memory (SRAM), dynamic random access memory (DRAM), other types of random access memory (RAM), read-only memory (ROM), electrically erasable programmable read-only memory (EEPROM), flash memory or other memory technologies, CD-ROM, digital versatile optical disc (DVD) or other optical storage, magnetic tape, magnetic disk storage or other magnetic storage devices, or any other non-transfer medium that can be used to store information accessible by a computing device.

[0336] The computer instructions stored in the storage medium of the above embodiments are used to cause the computer to execute the control method of the permanent magnet synchronous motor as described in any of the above embodiments, and have the beneficial effects of the corresponding method embodiments, which will not be repeated here.

[0337] Based on the same inventive concept, corresponding to any of the above embodiments, this application also provides a computer program product, including computer program instructions. When the computer program instructions are run on a computer, the computer executes the control method of the permanent magnet synchronous motor as described in any of the above embodiments, and has the beneficial effects of the corresponding method embodiments, which will not be repeated here.

[0338] It is understood that before using the technical solutions of the various embodiments in this disclosure, users will be informed of the type, scope of use, and usage scenarios of the personal information involved in an appropriate manner, and user authorization will be obtained.

[0339] For example, upon receiving a user's active request, a prompt message is sent to the user to explicitly inform them that the requested operation will require the acquisition and use of the user's personal information. This allows the user to independently choose, based on the prompt message, whether to provide personal information to the software or hardware such as electronic devices, applications, servers, or storage media performing the operations of this disclosed technical solution.

[0340] As an optional but not limited implementation, in response to a user's active request, sending a prompt message to the user can be done via a pop-up window, where the prompt message can be presented in text format. Furthermore, the pop-up window can also include a selection control allowing the user to choose "agree" or "disagree" to provide personal information to the electronic device.

[0341] It is understood that the above notification and user authorization process are merely illustrative and do not constitute a limitation on the implementation of this disclosure. Other methods that comply with relevant laws and regulations may also be applied to the implementation of this disclosure.

[0342] Those skilled in the art should understand that the discussion of any of the above embodiments is merely exemplary and is not intended to imply that the scope of this disclosure is limited to these examples; within the framework of this disclosure, the technical features of the above embodiments or different embodiments can also be combined, the steps can be implemented in any order, and there are many other variations of different aspects of the embodiments of this disclosure as described above, which are not provided in detail for the sake of brevity.

[0343] Additionally, to simplify the description and discussion, and to avoid obscuring the embodiments of this disclosure, the provided drawings may or may not show well-known power / ground connections to integrated circuit (IC) chips and other components. Furthermore, the apparatus may be shown in block diagram form to avoid obscuring the embodiments of this disclosure, and this also takes into account the fact that the details of implementation of these block diagram apparatuses are highly dependent on the platform on which the embodiments of this disclosure will be implemented (i.e., these details should be fully understood by those skilled in the art). While specific details (e.g., circuitry) have been set forth to describe exemplary embodiments of this disclosure, it will be apparent to those skilled in the art that the embodiments of this disclosure may be implemented without these specific details or with variations thereof. Therefore, these descriptions should be considered illustrative rather than restrictive.

[0344] Although this disclosure has been described in conjunction with specific embodiments thereof, many substitutions, modifications, and variations of these embodiments will be apparent to those skilled in the art from the foregoing description. For example, other memory architectures (e.g., dynamic RAM (DRAM)) may be used with the embodiments discussed.

[0345] This disclosure is intended to cover all such substitutions, modifications, and variations that fall within the broad scope of this disclosure. Therefore, any omissions, modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of this disclosure should be included within the protection scope of this disclosure.

Claims

1. A control method for a permanent magnet synchronous motor, characterized in that, The method includes: Obtain the sampled rotor electric angular velocity of the permanent magnet synchronous motor, and determine the first shaft current command based on the sampled rotor electric angular velocity; The reference voltage vector is determined based on the first axis current command and the pre-set second axis current command; The reference voltage vector is synthesized using a first preset number of non-zero vectors and a second preset number of zero vectors. Multiple voltage vectors and the duty cycle of each voltage vector are determined based on the volt-second balance algorithm. The candidate voltage vector set is determined by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector. A cost function is constructed based on current error, current harmonics, and switching frequency. The optimal voltage vector combination and application time are determined from the candidate voltage vector set according to the cost function. The drive signal for each switch of the inverter is determined based on the optimal voltage vector combination and the operating time, and the permanent magnet synchronous motor is controlled based on the drive signal.

2. The method according to claim 1, characterized in that, The process of acquiring the sampled rotor electrical angular velocity of the permanent magnet synchronous motor and determining the first shaft current command based on the sampled rotor electrical angular velocity includes: Obtain the sampled rotor electric angular velocity of the permanent magnet synchronous motor; The torque command is determined based on the sampled rotor electrical angular velocity using a proportional-integral regulator of the outer loop speed. in, For the torque command, k p k is the proportional gain of the proportional-integral controller. i Let ω be the integral gain of the proportional-integral controller, s be the complex variable in the Laplace transform, and ω be the integral gain of the proportional-integral controller. * The reference rotor electric angular velocity is ω, which is the sampled rotor electric angular velocity. The first shaft current command is determined based on the torque command. in, This is the first axis current command. For the torque command, n p ψ is the number of pole pairs of the motor. f It is a permanent magnet flux linkage.

3. The method according to claim 1, characterized in that, The step of determining the reference voltage vector based on the first axis current command and the preset second axis current command includes: The stator current vector is determined according to the first axis current command and the preset second axis current command; A hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system is constructed based on the stator current vector. in, Here is the current equation for the permanent magnet synchronous motor, F is the system disturbance estimation parameter, α is the gain of the input variable, and u s These are the input variables for the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system; The gain of the input variable is determined based on the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system. Where α is the gain of the input variable, For the first historical moment k-1, the differential of the first historical current. For the second historical time k-2, the differential of the second historical current. This represents the first historical stator voltage vector at the first historical moment k-1. The second historical stator voltage vector at the second historical moment k-2; The disturbance estimation parameters of the system are determined based on the hyperlocal model of the permanent magnet synchronous motor in the stationary coordinate system. Where F is the interference estimation parameter of the system. Let α be the differential of the first historical current at the first historical time k-1, and let α be the gain of the input variable. The first historical stator voltage vector at the first historical moment k-1; Based on the gain of the input variables and the disturbance estimation parameters of the system, the first target stator current vector at the first target time is determined. in, Let K be the first target stator current at the first target time k+1. Let T be the current stator current vector at time k. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. The current stator voltage vector at time k; Based on the gain of the input variables and the disturbance estimation parameters of the system, the second target stator current vector at the second target time is determined. in, The second target stator current is at the second target time k+2. Let T be the first target stator current vector at time k+1. sc Let F be the control period, F be the disturbance estimation parameter of the system, and α be the gain of the input variable. The first target stator voltage vector at the first target time k+1; The second reference stator current vector at the second target time is determined based on the first axis current command and the preset second axis current command. in, The second reference stator current vector at the second target time k+2. For the pre-set second axis current command, Here, θ is the first axis current command, j is the imaginary part of the complex number, and θ is the current. e For reference rotor electrical angle, ω r T is the electric angular velocity of the sampled rotor. sc To control the cycle; The second reference stator current vector is used to replace the second target stator current vector, and the reference voltage vector is determined based on the first target stator current vector and the second reference stator current vector. in, The reference voltage vector, The second reference stator current vector at the second target time k+2. Let T be the first target stator current vector at the first target time k+1. sc The control period is F, the disturbance estimation parameter of the system is F, and the gain of the input variable is α.

4. The method according to claim 1, characterized in that, The process of synthesizing the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors, and determining multiple voltage vectors and the duty cycle of each voltage vector based on a volt-second balance algorithm, includes: The reference voltage vector is synthesized using a first preset number of non-zero vectors and a second preset number of zero vectors; wherein the first preset number of non-zero vectors and the second preset number of zero vectors include: a first voltage vector, a second voltage vector, and a third voltage vector; The first duty cycle of the first voltage vector is determined based on the modulation ratio and the reference voltage vector angle. Among them, T x The first duty cycle of the first voltage vector segment is M, the modulation ratio is θ, and the reference voltage vector angle is θ. The second duty cycle of the second voltage vector is determined based on the modulation ratio and the angle of the reference voltage vector. T y =Msin(θ) Among them, T y The second duty cycle of the second voltage vector is M, where M is the modulation ratio and θ is the reference voltage vector angle. Based on the volt-second balance algorithm, the third duty cycle of the third voltage vector is determined according to the first duty cycle and the second duty cycle. T0=1-T x -T y Where T0 is the third duty cycle of the third segment voltage vector, T x T represents the first duty cycle of the first voltage vector segment. y This is the second duty cycle of the second voltage vector.

5. The method according to claim 1, characterized in that, The process of determining the candidate voltage vector set by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector's action includes: In response to determining that the number of voltage vectors is one, a first voltage vector combination is constructed based on the one voltage vector, and a first volt-second error corresponding to each first voltage vector combination is determined; In response to determining that the number of voltage vectors is two, the two voltage vectors are combined to construct a second voltage vector combination, and the second volt-second error corresponding to each second voltage vector combination is determined; In response to determining that the number of voltage vectors is three, the three voltage vectors are combined to construct a third voltage vector combination, and the third volt-second error corresponding to each third voltage vector combination is determined; In response to determining that the voltage vector data is four, the four voltage vectors are combined to construct a fourth voltage vector combination, and the fourth volt-second error corresponding to each fourth voltage vector combination is determined; Multiple combinations of the first voltage vector, multiple combinations of the second voltage vector, multiple combinations of the third voltage vector, and multiple combinations of the fourth voltage vector are used as candidate voltage vector sets.

6. The method according to claim 1, characterized in that, The cost function constructed based on current error, current harmonics, and switching frequency includes: A cost function is constructed based on current error, current harmonics, and switching frequency. Where J is the cost function, The second reference stator current vector at the second target time k+2. Ih is the second target stator current vector at the second target time k+2. rms For the current harmonics, g sw The weighting coefficients for the switching frequency are given, k+1 is the first target time, and f is the weighting coefficient for the switching frequency. sw The switching frequency is [value].

7. The method according to claim 1, characterized in that, The step of determining the optimal voltage vector combination and application time from the candidate voltage vector set according to the cost function includes: Substitute the voltage vector combinations in the candidate voltage vector set into the cost function in sequence to obtain the cost value corresponding to each voltage vector combination; The voltage vector combination with the lowest substitution value is determined from the candidate voltage vector set, and the voltage vector combination with the lowest substitution value is taken as the optimal voltage vector combination. The action time corresponding to the optimal voltage vector combination is then determined.

8. A control device for a permanent magnet synchronous motor, characterized in that, include: The first axis current command determination module is configured to acquire the sampled rotor electric angular velocity of the permanent magnet synchronous motor and determine the first axis current command based on the sampled rotor electric angular velocity. The reference voltage vector determination module is configured to determine the reference voltage vector based on the first axis current command and a preset second axis current command. The duty cycle determination module is configured to synthesize the reference voltage vector using a first preset number of non-zero vectors and a second preset number of zero vectors, and to determine multiple voltage vectors and the duty cycle of each voltage vector based on a volt-second balance algorithm. The candidate voltage vector set determination module is configured to determine the candidate voltage vector set by adjusting the number of voltage vectors acting within the control cycle and the duration of each voltage vector. The optimal voltage vector combination determination module is configured to construct a cost function based on current error, current harmonics and switching frequency, and determine the optimal voltage vector combination and application time from the candidate voltage vector set according to the cost function. The control module is configured to determine the drive signal for each switch of the inverter based on the optimal voltage vector combination and the operating time, and to control the permanent magnet synchronous motor based on the drive signal.

9. An electronic device, characterized in that, It includes a memory, a processor, and a computer program stored in the memory and running on the processor, wherein the processor, when executing the program, implements the method as claimed in any one of claims 1 to 7.

10. A non-transitory computer-readable storage medium, characterized in that, The non-transitory computer-readable storage medium stores computer instructions for causing a computer to perform the method according to any one of claims 1 to 7.

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