Programmable direct current electronic load device and control method thereof

Abstract

The application discloses a kind of program-controlled direct current electronic load device and its control method, device includes control unit and with the digital-analog conversion module of control unit connection, feedback current loop, the feedback current loop includes power dissipation unit and measurement unit;The control unit is by controlling the digital-analog conversion module output voltage, makes the feedback current loop establish load current, periodically by the measurement unit obtains the actual current value, and the actual current value is compared with target current value, to adjust the output voltage of the digital-analog conversion module, so that the actual current value approaches the target current value.Thereby, by feedback current loop, fast current establishment and stability are realized.By hardware negative feedback loop ensures rapidity and stability, again with software fine adjustment realizes precision optimization, to meet the requirement of high response speed and high control precision simultaneously under compact volume and low cost.

Description

Technical Field

[0001] This invention relates to the field of electronic load device technology, and in particular to a programmable DC electronic load device and its control method. Background Technology

[0002] A DC electronic load is an energy-consuming device commonly used in DC power supply development projects to test the load-carrying capacity of the output DC voltage. In addition, DC electronic loads also have significant application needs in battery charging and discharging maintenance.

[0003] Traditional electronic loads typically employ pure hardware feedback to achieve closed-loop control of the current circuit. This involves outputting a fixed reference voltage via a potentiometer to the non-inverting input of an operational amplifier, and then amplifying the voltage across the sampling resistor before feeding it into the inverting input of the operational amplifier. Under this loop control method, factors such as voltage drop when the power supply under test is under load and loop losses can lead to a significant error between the actual load current in the loop and the preset parameters. Generally, the actual current consumed by the loop is slightly lower than the preset current value. The set current value can only be gradually approximated by manually adjusting the reference voltage. Furthermore, the operation is cumbersome and difficult to control accuracy due to factors such as the precision and smoothness of the adjustable potentiometer.

[0004] It is evident that existing programmable DC electronic load devices lack precision in current regulation and suffer from slow response speed, resulting in poor current control performance. Summary of the Invention

[0005] The purpose of this invention is to overcome the shortcomings of the prior art and provide a programmable DC electronic load device and its control method. The DC electronic load device is small in size, low in cost, high in precision, and high in power, and has a good current control effect. It can be equipped with an LCD screen to form an independent device, and can also be networked via a bus to expand the load power range. It can also be easily installed in other devices as a sub-functional module for use. It has strong application prospects in the field of electronic equipment development.

[0006] The objective of this invention is achieved through the following technical solution:

[0007] In a first aspect, this application discloses a programmable DC electronic load device, including a control unit. The control unit is equipped with a digital-to-analog converter module and a feedback current loop. The feedback current loop includes a power dissipation unit and a measurement unit. The power dissipation unit includes at least one dissipation branch, and each dissipation branch includes a high-power MOSFET, a sampling resistor, a current sensing amplifier, and an operational amplifier. The high-power MOSFET is connected to the power supply under test to dissipate power. The sampling resistor is connected in series with the source of the high-power MOSFET. The current sensing amplifier is connected across the sampling resistor. The non-inverting input of the operational amplifier is connected to the output of the digital-to-analog converter module. The inverting input of the operational amplifier is connected to the output of the current sensing amplifier, and the output of the operational amplifier is connected to the gate of the high-power MOSFET. The measuring unit is connected between the power supply under test and the power dissipation unit to measure the actual current value flowing through the power dissipation unit, and the measuring unit is connected to the input of the control unit. The control unit controls the output voltage of the digital-to-analog converter module to establish a load current in the feedback current loop, periodically obtains the actual current value through the measuring unit, compares the actual current value with the target current value, and adjusts the output voltage of the digital-to-analog converter module to make the actual current value approach the target current value.

[0008] The beneficial effects of this invention are as follows: Rapid current build-up and stabilization are achieved through a feedback current loop. The negative feedback loop, composed of operational amplifiers, can respond to current changes in real time. By driving the gate of the MOS transistor in an analog manner, it avoids the sampling delay and algorithm processing delay problems in digital control, thus ensuring a millisecond-level response speed. Simultaneously, the control unit (microcontroller) periodically reads the actual current value through a high-precision measurement unit and compares it with the target value. An incremental fine-tuning strategy is used to adjust the DAC output voltage. This combination of rapid hardware response and fine software correction avoids the drift problems that may exist in purely analog circuits and overcomes the slow speed problem of purely digital control, ultimately achieving high-precision load current control over a wide dynamic range. Therefore, by ensuring speed and stability through a hardware negative feedback loop and then optimizing accuracy through software fine-tuning, the requirements of high response speed and high control accuracy are simultaneously met in a compact size and at low cost.

[0009] Secondly, this application discloses a control method for a programmable DC electronic load device, used to control the programmable DC electronic load device, comprising the following steps: S1: receiving target load parameters, the target load parameters including a target current value in constant current mode, a target voltage value in constant voltage mode, a target resistance value in constant resistance mode, or a target power value in constant power mode; S2: calculating the target current value according to the target load parameters and the voltage of the current power supply under test, using Ohm's law or a power formula; S3: outputting the initial digital-to-analog converter output voltage value to the feedback current loop to quickly establish the load current; S4: periodically acquiring the actual current value through the measurement unit; S5: comparing the actual current value with the target current value; S6: fine-tuning the output voltage value of the digital-to-analog converter module according to the comparison result, so that the actual current value approaches the target current value. Attached Figure Description

[0010] Figure 1 This is a simplified schematic diagram of the core circuit of a programmable DC electronic load device according to some embodiments of this application;

[0011] Figure 2 This is a partial circuit schematic of the feedback current loop according to some embodiments of this application;

[0012] Figure 3 This is a circuit schematic diagram of another part of the feedback current loop according to some embodiments of this application;

[0013] Figure 4 The present invention provides a schematic diagram of a measurement unit according to some embodiments of the present application.

[0014] Figure 5 The present invention provides schematic diagrams of power supply circuits according to some embodiments of the present application.

[0015] Figure 6 The present invention provides a schematic diagram of a temperature sampling circuit according to some embodiments thereof.

[0016] Figure 7 The present invention provides a schematic diagram of a fan drive circuit according to some embodiments of the present application. Detailed Implementation

[0017] The technical solution of the present invention will be clearly and completely described below with reference to the embodiments. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0018] according to Figures 1-7 This embodiment provides a programmable DC electronic load device and its control method.

[0019] The programmable DC electronic load device according to an embodiment of this application includes a control unit, the control unit being equipped with a digital-to-analog conversion module, and a feedback current loop. In subsequent examples, a microcontroller will be used as an example to illustrate the control unit.

[0020] The feedback current loop includes a power dissipation unit and a measurement unit. Specifically, the power dissipation unit may include four dissipation branches, but fewer or more are also possible. Each dissipation branch includes a high-power MOSFET, a sampling resistor, a current sensing amplifier, and an operational amplifier. (First, combine...) Figure 1 The simplified circuit diagram shown illustrates the embodiments of this application. A high-power MOSFET is used to connect to the power supply under test to consume power. A sampling resistor is connected in series with the source of the high-power MOSFET, and a current sensing amplifier is connected across the sampling resistor.

[0021] The non-inverting input of the operational amplifier is connected to the output of the digital-to-analog converter module, the inverting input of the operational amplifier is connected to the output of the current sensing amplifier, and the output of the operational amplifier is connected to the gate of the high-power MOSFET. The measuring unit is connected between the power supply under test and the power dissipation unit to measure the actual current value flowing through the power dissipation unit. The measuring unit is also connected to the input of the control unit to provide the control unit with the measured actual current result.

[0022] You can understand this by referring to a specific example. Figure 2 As shown, the current sensing amplifier is model INA180A2. It is connected across the sampling resistor (such as R8) to amplify the differential voltage across it (for example, by a factor of 50) and then output it as a single-ended signal. The non-inverting input of the operational amplifier (model OP07) is connected to the output of the DAC, and the inverting input is connected to the output of the current sensing amplifier. Its output drives the gate of the MOSFET through a driving resistor (such as R3).

[0023] As mentioned above, the measurement unit is connected to the control unit and is capable of high-precision measurement, used to measure in real time the actual current value flowing through the power dissipation unit, the voltage and power of the power supply under test. In this embodiment, the control unit (i.e., the microcontroller) can output a corresponding DAC voltage to the non-inverting input of the operational amplifier based on the received target current value, so as to quickly establish the load current through a hardware negative feedback loop; then, it periodically reads the actual current value through the measurement unit and compares it with the target current value, and fine-tunes the DAC output voltage to make the actual current value gradually approach the target value, thereby achieving high-precision, fast and stable current control.

[0024] In other words, the microcontroller can combine the current operating mode and load voltage, and use Ohm's law to convert the load parameter settings into DAC channel output voltage values, which are then output to the non-inverting input of the operational amplifier in the load current control loop. Figure 2 and Figure 3 The output of the operational amplifier is connected to the gate (G) of a high-power MOSFET. The drain (D) of the high-power MOSFET is connected to the positive terminal of the power supply under test. The source (S) is connected in series with a sampling resistor and then connected to the negative terminal of the power supply under test. The voltage across the sampling resistor is amplified by a current sensing amplifier and then input to the inverting input of the operational amplifier. Due to the virtual short characteristic of the operational amplifier, when the circuit reaches stability, the DC voltage across the sampling resistor × the current sensing amplifier amplification factor = the DAC output voltage. The current consumption of the power supply under test can be controlled by controlling the DAC output voltage value of the microcontroller, thus realizing constant voltage, constant current, constant resistance, and constant power mode load testing of the power supply under test.

[0025] Understandably, the aforementioned feedback current loop enables rapid current build-up and stabilization. The negative feedback loop, composed of operational amplifiers, responds to current changes in real time, driving the MOSFET gate analogically, thus avoiding the sampling delay and slow algorithm processing time issues of digital control and ensuring millisecond-level response speed. Simultaneously, the control unit (microcontroller) periodically reads the actual current value through a high-precision measurement unit and compares it with the target value, adjusting the DAC output voltage using an incremental fine-tuning strategy. This combination of rapid hardware response and fine-tuning software avoids the drift problems that may exist in purely analog circuits and overcomes the slow speed of purely digital control, ultimately achieving high-precision load current control over a wide dynamic range. Therefore, by ensuring speed and stability through a hardware negative feedback loop and optimizing accuracy through software fine-tuning, the requirements of high response speed and high control precision are simultaneously met in a compact size and at low cost.

[0026] In a further embodiment, the digital-to-analog converter module of the control unit has two output channels, each connected to two current-sharing resistors, thereby simultaneously driving the non-inverting inputs of four operational amplifiers. Correspondingly, the power dissipation unit includes four parallel dissipation branch structures to extend the load current and power capacity.

[0027] For example Figure 2 and Figure 3 The current shown is generated using four identical dissipation branches connected in parallel. Each branch includes a high-power MOSFET (such as an IRFP4368 field-effect transistor), a sampling resistor (such as R8), a current sensing amplifier (such as an INA180A2), and an operational amplifier (such as an OP07). During operation, the output channel of each digital-to-analog converter module simultaneously drives the non-inverting inputs of two operational amplifiers through a current-sharing resistor, for example... Figure 2 The DAC OUT1 output channel shown is connected to the non-inverting inputs of two operational amplifiers via current-sharing resistors. Figure 3 The DAC OUT2 output channel shown is also connected to the non-inverting inputs of the other two operational amplifiers.

[0028] When the microcontroller outputs a DAC voltage signal, this signal is sent to the non-inverting input of the operational amplifier in each branch through current-sharing resistors, and each branch independently forms a hardware negative feedback control loop. Since the structural parameters of each branch are identical, under ideal conditions, the current flowing through each branch is approximately equal, thus distributing the total load current across multiple branches. For example, if the total load current is set to 25A, and four parallel branches are used, each branch carries approximately 6.25A of current. The control unit adjusts the DAC output voltage so that the voltage across the sampling resistors in each branch, after being amplified by the current-sensing amplifier, is equal to the DAC output voltage, achieving current sharing and stable control.

[0029] The control unit acquires the voltage value of the power supply under test in real time through the measurement unit, and converts the set value of constant voltage, constant resistance, or constant power mode into the corresponding target current value based on received external commands, such as setting parameters sent via RS485 bus, and combined with internal algorithms. For example, in constant voltage mode, the set voltage value is Vset, and the control unit reads the current voltage Vactual of the power supply under test through the measurement unit. If Vactual is higher than Vset, the load current is increased to reduce Vactual to Vset. In constant resistance mode, the set resistance value is Rset, and the control unit calculates the target current Itarget = Vactual / Rset according to Ohm's law. In constant power mode, the set power value is Pset, and Itarget = Pset / Vactual is calculated. Subsequently, the control unit outputs the corresponding voltage value through the digital-to-analog converter module to drive the feedback current loop, making the actual current approach the target current, thereby realizing the load function of different modes.

[0030] In this way, the control unit achieves mode switching and parameter mapping through software calculations, rapidly establishes current using a hardware negative feedback loop, and then ensures that the actual load parameters are consistent with the set values ​​through periodic measurement and fine-tuning. This design allows a single electronic load device to flexibly support multiple operating modes and adapt to different testing needs.

[0031] To better illustrate this, a detailed description of an example will follow, along with accompanying diagrams:

[0032] Continue to refer to Figures 2-3 As shown, an exemplary structure of the feedback current loop is understood, and the four branches will be explained in turn.

[0033] Figure 2This is a partial circuit structure of the feedback current loop. This part of the circuit includes the first branch located in the upper half of the figure, the second branch located in the middle part, and the loop at the bottom of the figure.

[0034] Specifically, in the first branch, pin 7 of operational amplifier IC1 is connected to VCC 12V and capacitor C1, and then connected to GND through capacitor C1; pin 4 of IC1 is connected to VCC -5V and capacitor C38 to GND. Pin 3 of IC1 is connected to current sharing resistor R1 to DAC OUT1; pin 2 of IC1 is connected to the first bias resistor R9 to the output terminal (OUT pin) of current sensing amplifier U2; pin 6 of IC1 is connected to drive resistor R3 to the gate (G) of Q1, and the gate of Q1 is also connected to the second bias resistor R5 to GND; the source (S) of Q1 is connected to the IN+ of U2 and resistor R8 to I-test+; the VS pin of U2 is connected to VCC 3V3 and C3 to GND; the GND interface in U2 is directly grounded and connected to capacitor C3; the IN- of U2 is connected to I-test+.

[0035] In the second branch, pin 7 of operational amplifier IC3 is connected to VCC 12V and then to GND via capacitor C7; pin 4 is connected to VCC -5V and capacitor C40 to GND. Pin 3 of IC3 is connected to current sharing resistor R12 to DAC OUT 1; pin 2 of IC3 is connected to the first bias resistor R21 to the output terminal (OUT pin) of current sensing amplifier U4; pin 6 of IC3 is connected to drive resistor R13 and then to the gate (G) of Q3, which is also connected to the second bias resistor R16 to GND; the source (S) of Q3 is connected to the IN+ of U4 and then to I-test+ and the IN- of U4 via R19; the VS of U4 is connected to VCC 3V3 and then to GND via capacitor C9; the GND pin of U4 is directly grounded; the IN- of U4 is connected to I-test+.

[0036] Understandably, taking the first branch as an example, the non-inverting input (pin 3) of operational amplifier IC1 is connected to the DAC OUT1 signal source through a current-sharing resistor R1. Simultaneously, DAC OUT1 is grounded via a 0-ohm resistor R43 as a debugging monitoring point, ensuring the signal is smoothly input to pin 3 of IC1 through R1. The inverting input (pin 2 of IC1) is connected to the input of the current-sensing amplifier through a first bias resistor R9, and the output of IC1 (pin 6 of IC1) is connected to the gate (G) of Q1 through a drive resistor R3, thus forming a control path.

[0037] Therefore, in the high-power MOSFET Q1 section, its gate receives the output signal of IC1 through the drive resistor R3, and is grounded through the second bias resistor R5 to stabilize the gate voltage. The drain (D) is connected to the high-voltage side of the circuit or the load, while the source (S) is connected to pin 3 of U2 (INA180A2) (i.e., connected to IN+ of U2), and connected to the I-test+ terminal through a sampling resistor R8. In this way, current flows from the source of Q1 through R8 to I-test+, enabling U2 to detect the voltage across R8.

[0038] In other words, U2 is powered by VCC 3V3, and pin 5 of U2 (connected to the VS port) is grounded through capacitor C3. The IN+ pin of U2 receives the signal from the source of Q1, and the IN- pin (through pin 4 of U2) is connected to the I-test+ terminal. At the same time, its output terminal (OUT pin) is connected to the inverting input terminal of IC1 (i.e., pin 2 of IC1) through resistor R9, thus forming a voltage feedback loop. This ensures that the voltages at the non-inverting and inverting input terminals of IC1 are balanced, achieving precise control of the current of the high-power MOSFET Q1.

[0039] exist Figure 2 In the lower half of the circuit, pin 7 of operational amplifier IC7 is connected to VCC 12V and capacitor C44 to GND; pin 4 of IC7 is connected to VCC -5V and capacitor C46 to GND. Pin 3 is connected to resistor R45 and then to pin 2 of operational amplifier IC3 via the first bias resistor R21; pin 2 of IC7 is connected to resistor R48 to ADC-ch3, and pin 6 of IC7 is also connected to ADC-ch3. This allows the first and second branches to be connected back to pin PA3 of the microcontroller via the ADC-ch3 signal, enabling the microcontroller to read the current value of channel 2. In other words, ADC-ch3 is connected to an analog input pin on the microcontroller, allowing the microcontroller to directly receive the analog signal of the current in channel 2. Then, the microcontroller's internal ADC (analog-to-digital converter) converts the analog signal into a digital signal, which is then used by the subsequent measurement unit to acquire the current.

[0040] Figure 3 This is a partial circuit structure of the feedback current loop. Figure 2 Channel 1 and Channel 2 are connected to part A in the diagram, then refer to... Figure 3 Section A continues to connect to Figure 3 Channels 3 and 4 (part A in the diagram is only for illustrative purposes) Figure 2 and Figure 3 The connection relationship of the central channel has no special meaning. Figure 2 The circuit structures shown are basically the same. Figure 3 In the process, the signal input of DACOUT2 is split into two paths, namely the third branch and the fourth branch.

[0041] The third branch is connected to the non-inverting input (pin 3) of operational amplifier IC2 via current-sharing resistor R4. The positive power supply (pin 7) of IC2 is connected to VCC 12V and grounded via capacitor C2. The negative power supply (pin 4) of IC2 is connected to VCC -5V and grounded via capacitor C39. The output (pin 6) of IC2 is connected to drive resistor R6 and then back to the negative power supply of IC2 via capacitor C6. The other end of drive resistor R6 is connected to the second bias resistor R7, thus pulling it down to GND. This node is also connected to the gate (G) of MOSFET Q2.

[0042] The drain (D) of Q2 is connected from part A to channels 1 and 2 via channel 3. The source (S) is connected to one end of the sampling resistor R10 and the IN+ of the current sense amplifier U3. The power supply terminal VS of U3 is connected to VCC 3V3. The GND terminal of U3 is directly grounded and connected to VCC 3V3 through capacitor C5. The output terminal OUT is connected to the first bias resistor R11 and then back to pin 2 of IC2.

[0043] Let's continue with the explanation of the fourth branch.

[0044] The positive power supply terminal of IC4, pin 7, is connected to VCC 12V and grounded through capacitor C8. Pin 4 of IC4 is connected to VCC -5V and grounded through capacitor C41. The output terminal of IC4, pin 6, is connected to drive resistor R15. The other end of drive resistor R15 is connected to the second bias resistor R17 and pulled down to GND. This node is also connected to the gate (G) of MOSFET Q4. The drain (D) of Q4 is connected to the positive terminal of the power supply under test through channel 4, sharing the drain of Q2. The source (S) is connected to one end of sampling resistor R20 and the non-inverting input IN+ of current sense amplifier U5. The other end of R20 is connected to I-test+.

[0045] The power supply terminal VS of U5 is connected to VCC 3V3. The ground terminal GND of U5 is directly grounded and connected to VCC 3V3 through capacitor C11. Its negative input terminal IN- is connected to I-test+. At the same time, the output terminal OUT of U5 is connected to pin 2 of IC4 through the first bias resistor R22.

[0046] exist Figure 3 In the lower loop, resistor R47 is connected to the non-inverting input of operational amplifier IC8, i.e., pin 3 of IC8. The positive power supply of IC8, i.e., pin 7 of IC8, is connected to VCC 12V, and the negative power supply, i.e., pin 4 of IC8, is connected to VCC -5V. Its inverting input, i.e., pin 2 of IC8, is connected to ADC-ch6 through resistor R49, and the output pin 6 of IC8 is also connected to ADC-ch6, thereby sending the analog signal of the detected current to the ADC module of the microcontroller. The principle is the same as the first and second branches and will not be described in detail here.

[0047] After channels 3 and 4 merge, they are connected to the positive terminal of the power supply under test via a fuse, and are also connected to... Figure 3 Part B is shown, then refer to Figure 4 As shown, part B is connected to the measurement unit; part B here is only used to indicate the circuit connection and has no special meaning.

[0048] Next, the measurement unit will be described.

[0049] For example, the measurement unit uses the INA226 power monitoring chip as its core, and is equipped with external filtering circuits to complete the hardware circuit design. The measurement unit is as follows: Figure 4 As shown.

[0050] The pin connections of the INA226 chip are as follows: Pin 1 (A0 in the diagram) and pin 2 (A1 in the diagram) are both grounded (GND in the diagram). Pin 3 (Alert in the diagram) is pulled up to VCC_3V3 through resistor R27, and simultaneously outputs the Alert1 signal. Pin 4 (SDA in the diagram) is pulled up to VCC_3V3 through resistor R29, serving as the SDA line of the bus. Pin 5 (SCL in the diagram) is pulled up to VCC_3V3 through resistor R34, serving as the SCL line of the bus. The Alert1 signal line, SDA line, and SCL line can be connected to a microcontroller to transmit digital signals.

[0051] Pin 6 (Vs+) of U6 is directly connected to VCC_3V3 and grounded through capacitor C22. Pin 7 (GND) of U6 is directly grounded. Pin 8 (Vbus) of U6 is connected to one end of resistor R30, and also to ground through resistor R33; one end of R30 is connected to the power supply under test (in conjunction with...). Figure 3 and Figure 4 As shown in the diagram, it is connected to the power supply under test through point B, so that the Vbus terminal is connected to the power supply under test. Pin 9 (VIN-) of U6 is connected to one end of resistor R26, and pin 10 (VIN+) of U6 is connected to one end of resistor R23; the other end of resistor R23 is connected to the I-test+ signal point.

[0052] Resistor R24 ​​and capacitor C21 are connected between the VIN+ and VIN- pins. Resistors R30 and R33 form a voltage divider network connected to the Vbus pin of the chip, while capacitor C22 not only provides decoupling for the Vs+ pin, but also shares a connection point with one end of R33.

[0053] In this example, the INA226 is a voltage, current, and power measurement chip that monitors shunt voltage drop and bus power supply voltage. Its programmable calibration values, conversion time, and averaging functions, combined with an internal multiplier, enable direct reading of current, voltage, and power values. The chip has a built-in 16-bit ADC for acquisition. Shunt voltage drop acquisition uses one differential input with an input range of -81.92mV to 81.92mV. After sampling by the internal 16-bit ADC, the minimum resolution is 2.5uV. In the circuit design, the external sampling resistor (e.g., R24) is chosen to be 3mΩ. After filtering by the filter circuit, the signal is sent to the differential input pin. Theoretically, the maximum effective current measurement is 81.92mV / 3mΩ = 27.3A, meeting the module's technical requirement of a maximum load current of 25A. The maximum value of the chip bus voltage monitoring is about 40V. The voltage of the power supply being tested is divided by voltage divider resistors (such as R30 and R33) and then sent to the Vbus pin for bus voltage detection. It is calculated that the maximum voltage value that the power supply being tested can be connected to is about 120V, which meets the test requirement of the maximum load voltage of 75VDC required by the technical specifications of this module.

[0054] In this way, the INA226 simultaneously connects to the power supply under test and I-test+, thereby calculating the current and input voltage (such as the drain voltage of Q3) across the sampling resistor. These results, such as current and voltage, are then stored in the INA226's internal registers, allowing the microcontroller to directly obtain accurate digital values ​​through the interface. This enables real-time, accurate, and rapid detection of current and voltage using the obtained digital signal and the aforementioned analog signal.

[0055] It is understandable that in this embodiment, in order to ensure that the loop meets the technical specifications of load voltage, current and power requirements, four identical MOSFETs are used in parallel to form the power dissipation loop of the module. Since the circuit structure of the four load loops is exactly the same, the following only uses channel 1 as an example to explain the implementation process of the load current control loop.

[0056] The positive terminal of the power supply under test is input through the terminal block, and after current limiting protection by a one-time fuse, it is sent to the drain (D) of MOSFET Q1. The negative terminal of the power supply under test is input through the terminal block and finally connected to the source (S) of MOSFET Q1 through a sampling resistor. The voltage signal from DAC OUT1, generated by the microcontroller, is sent to the non-inverting input of OP07 operational amplifier IC1 after passing through the current sharing resistor R1. After amplification, the signal is sent to the gate (G) of MOSFET Q1 for driving after passing through the drive resistor R3. When MOSFET Q1 is turned on, the voltage of the power supply under test enters from the drain (D) of MOSFET Q1, flows out from the source (S), and returns to the negative terminal of the power supply under test after passing through the 10mΩ sampling resistor R8. This constructs a closed-loop current circuit from the positive terminal of the power supply under test to the drain (D) of MOSFET Q1, then to the source (S), and finally back to the negative terminal of the power supply under test. At this point, the current in the loop will form a voltage across the sampling resistor R8 that is linearly related to the loop current. This voltage is amplified 50 times by the INA180A2 current sensing amplifier U2 and then fed into the inverting input of the OP07 operational amplifier IC1. This ultimately forms a negative feedback loop from the operational amplifier output to the gate (G) of the MOSFET, and from the source (S) of the MOSFET to the sampling resistor R8. This negative feedback loop is then amplified by the current sensing amplifier and fed back to the operational amplifier. According to the characteristics of the operational amplifier, when the negative feedback loop is balanced, the voltage at the non-inverting input of the operational amplifier is equal to the voltage at the inverting input. Therefore, the functional relationship between the DACOUT1 voltage and the measured power supply current can be constructed: VDACOUT1 = I (measured power supply loop current) × R (sampling resistor value) × 50. The sampling resistor value can be selected based on the load current range and control accuracy. Thus, controlling the DACOUT1 voltage achieves the purpose of controlling the load current of the measured power supply. For example, if the load current of the power supply under test needs to be controlled to 4A, since there are 4 identical load loops, the current value allocated to load loop channel 1 is 1A. The sampling resistor in the circuit has a resistance of 10mΩ. The voltage formed across the sampling resistor by the 1A current is 0.01V. This voltage is amplified 50 times by the current sensing amplifier to 0.5V. That is, as long as the microcontroller controls the output voltage of DAC OUT1 and DAC OUT2 to be 0.5V, the current of the 4 load loop channels can be controlled to be 1A, thus achieving the effect of the load current of the power supply under test being 4A.

[0057] In the four load loops, two DAC output voltages are used for control, with each DAC voltage controlling two load loops through current-sharing resistors (such as R1 and R12). The DAC voltage source can be the two 12-bit DAC output channels built into the STM32F103RCT6 microcontroller. The microcontroller uses a 3.3VDC power supply, and its DAC output voltage resolution is approximately 3.3V / 2^12 = 3.3 / 4096 ≈ 0.0008V. Since the current sensing amplifier amplifies the voltage across the sampling resistor by 50 times, and the single-channel sampling resistor is 10mΩ, the resolution of the single-channel load current control can be calculated to be approximately 1.6mA, which meets the parameter setting accuracy requirement of 0.1% in the technical specifications.

[0058] Furthermore, when the DAC output voltage of the microcontroller reaches its maximum value, the output is approximately 3.3VDC. At this time, the single-channel load current can be calculated to be approximately 3.3VDC / 50 times / 10mR≈6.6A, that is, the maximum current of a single-channel load circuit is 6.6A, and the maximum load current of the 4-channel load circuit is approximately 26.4A, which meets the requirement of a maximum load current of 25A in the technical specifications.

[0059] In this example, the current-sharing resistor, the first bias resistor, the drive resistor, and the second bias resistor together constitute a bias circuit to maintain the stable operation and fast response of the feedback current loop. In the above example, current-sharing resistors such as R1 and R12 are suitable for connecting DAC OUT1 to the non-inverting input of the operational amplifier. The first bias resistors such as R9 and R21 are suitable for connecting to the output of the current-sensing amplifier. The drive resistors such as R3 and R13 are suitable for connecting the output of the operational amplifier to the gate of the MOSFET. The second bias resistors such as R5 and R16 are suitable for connecting the gate of the MOSFET to ground. It should be understood that these all belong to the same higher-level concept, and each specific resistor unit is suitable for the lower-level concept; this does not constitute a conflict of meaning.

[0060] It is understandable that the power consumption of the power supply under test is actually dissipated within the MOSFET. The preferred type is the IRFP4368 field-effect transistor, whose main technical specifications are: continuous drain current up to 195A; operating temperature range -55℃ to +175℃; drain-source voltage 75VDC; and power dissipation of 520W.

[0061] Therefore, by using a 4-way parallel connection as the load dissipation device, the theoretical maximum load voltage can reach 75VDC, the load current can reach 780A, and the load power can reach 2600W, which can fully meet the corresponding technical requirements of this module.

[0062] In some embodiments, the theoretical power that a high-power MOSFET can carry is generally very large. For example, the IRFP4368 field-effect transistor selected in this load module has a maximum single-transistor power dissipation of 520W. However, the maximum power dissipation is the limit parameter that can be achieved under ideal heat dissipation conditions, which cannot be achieved in practical applications. Therefore, it is particularly important to enhance the heat dissipation conditions of the electronic load and design over-temperature protection.

[0063] In some embodiments, reference is made to Figure 5 Understandably, the device in this application embodiment has an internal power supply module for converting externally input DC voltage into the voltage required for internal power supply, and provides real-time feedback on the working status of each power path through LED indicators, including two parts: 12 VDC to ±5 VDC and +5 VDC to +3.3 VDC.

[0064] In this example, refer to Figure 5 The upper part of the circuit, specifically the +5V to +3.3V conversion section, is primarily powered by the AMS1117-3V3 chip, denoted as V1. The input of this +5V to +3.3V section is connected to a filter capacitor C30 to remove high-frequency noise. The external 5V power supply is connected to the Vin pin via the VCC 5V node. The output voltage is further stabilized by capacitor C31, and the 3.3V power supply is led out from the VCC_3V3 node to power subsequent circuits. LED3 serves as the main power indicator, connected between VCC_3V3 and GND via a drive resistor R52, and illuminates when the 3.3V power supply is normal.

[0065] refer to Figure 5 The lower half of the circuit is the 12V to ±5V dual-output section. In the power input module, the JP6 terminal shown in the diagram serves as the module's power input interface, used to connect to an external 12V DC power supply. F3 is a 2A self-resetting fuse, which will cut off the circuit when the input current exceeds 2A and will automatically reset after the fault is cleared. LED5 is the power input indicator, which lights up when the input voltage is normal. Its current is controlled by the R46 drive resistor, with a typical current of approximately 0.34mA. C34 and C35 are input-side filter capacitors, used to filter out high-frequency ripple from the input power supply and stabilize the input voltage. Based on M1 (model WRA1205S-3WR2 DC-DC converter) shown in the diagram, the output is dual ±5V. In the M1 pins, pin 2 (Vin) is connected to the 12V input, pin 1 is grounded, pin 6 (Vo+) outputs +5V, and pins 7 (Vo-) and 8 (NC) output -5V.

[0066] In the +5V output section, the Vo+ output terminal is defined as the VCC_5V bus. C32 and C33 serve as output filter capacitors, used to filter out low-frequency and high-frequency ripple, respectively. LED2 is the +5V output indicator, and its drive resistor R53 controls the current to approximately 0.16mA. F4 is the -5V output resettable fuse, providing 0.3A overcurrent protection.

[0067] In the -5V output module, the Vo- output terminal is defined as the VCC_-5V bus. C36 and C37 are connected between Vo- and GND to filter out output ripple. LED4 indicates the -5V output status, and its connection is as follows: the positive terminal is grounded through R54, and the negative terminal is connected to Vo-, forming a positive bias, with a current limiting of approximately 0.16mA. The entire circuit's grounding system uses a common ground design, with all GND symbols representing the same potential reference point to ensure consistent system potential.

[0068] Therefore, this power supply circuit can convert 12VDC to ±5VDC. The +12VDC module power supply input voltage, after being protected by a one-time fuse current limiting, provides positive power to the operational amplifier in the module. On the other hand, after being filtered by capacitors (C34 and C35), it is sent to the WRA1205S-3WR2 dual-channel switching power supply module (M1) to be converted to ±5VDC voltage. After being filtered by the output and protected by a one-time fuse current limiting, the +5VDC is sent to the +5VDC to +3.3VDC circuit, and the -5VDC provides negative power to the operational amplifier in the module.

[0069] Furthermore, it can convert +5VDC to +3.3VDC. Specifically, the +5VDC is filtered by the input filter capacitor (C30) and then fed into the AMS1117-3V3 linear power chip (V1) to be converted into +3.3VDC voltage. After output filtering and current limiting protection by a one-time fuse, it powers the microcontroller and other functional circuits in the module.

[0070] In this embodiment, the device is designed such that the housings of four MOSFETs are directly mounted on an aluminum profile heat sink, and thermal grease is evenly applied to the mounting surface. An NTC thermistor is installed on the heat sink near the center of the four MOSFETs for temperature sensing and sampling. A 12V PWM high-power cooling fan is installed on the other side of the heat sink to directly cool the heat sink, thereby indirectly achieving the purpose of cooling the MOSFETs.

[0071] The temperature control circuit consists of two parts: a temperature sampling circuit and a cooling fan drive circuit. (Temperature sampling circuit reference) Figure 6 Understanding, reference for cooling fan drive circuit Figure 7 understand.

[0072] like Figure 6The temperature sampling thermistor interface JP2 shown is designed for external thermistors. Its interface 2 connects to the VCC 3V3 power supply, providing the operating voltage for the thermistor; interface 1 serves as the signal output terminal. Interface 1 is connected to both capacitor C20 and resistor R28, with the other ends of both grounded. C20 acts as a filter, while R28, together with the thermistor connected to interface 1, forms a voltage divider circuit, converting the thermistor's resistance change with temperature into a corresponding voltage change.

[0073] The voltage divider signal is then fed to the non-inverting input of operational amplifier IC5, pin 3, through an isolation and buffering resistor R25. Here, operational amplifier IC5 is configured as a voltage follower; its inverting input, pin 2, is directly connected to the output, pin 6, via a feedback resistor R31. The positive power supply of operational amplifier IC5, pin 7, is connected to VCC 12V, and pin 4 is connected to VCC -5V. Capacitors C42 and C13 are also connected to ground, respectively, at pins 7 and 4.

[0074] Finally, the voltage signal, buffered and processed by the operational amplifier, is directly output from the output terminal (pin 6) of IC5 to channel ADC-ch7 of the microcontroller's built-in analog-to-digital converter module. The system can calculate the current temperature information by reading the voltage value of this channel.

[0075] Therefore, the temperature sampling circuit uses an external thermistor in conjunction with the microcontroller's built-in ADC sampling function. The 3.3V voltage is divided by the thermistor and the voltage divider resistor (R28) and then sent to the non-inverting input of the operational amplifier (IC5). After being followed by the voltage of the operational amplifier, it is sent to the microcontroller's built-in ADC acquisition channel ADC-ch7 for voltage sampling. The microcontroller then calculates the current heat sink temperature value through an algorithm and performs automatic temperature control.

[0076] like Figure 7 The fan drive circuit shown includes components such as operational amplifiers, resistors, capacitors, and interfaces.

[0077] The core of the operational amplifier section is operational amplifier IC6, model OP07, with the following pin connections: Pin 3 (non-inverting input) is connected to one end of resistor R37; Pin 2 of IC6, the inverting input, is connected to one end of resistor R38 and also to one end of resistor R40; Pin 7 of IC6, the positive power supply, is connected to VCC 12V and also to one end of capacitor C26, which is grounded; Pin 4 of IC6, the negative power supply, is connected to VCC -5V and also to one end of capacitor C43, which is grounded; Pins 4 and 7 are connected to the other ends of capacitors C43 and C26, respectively.

[0078] One end of R37 is connected to the "Fan PWM" input signal, and the other end is connected to the non-inverting input terminal of IC6, i.e., pin 3 of IC6. Pin 6 of IC6, i.e. the output terminal, is connected to the "Fan PWM" signal terminal and then to the other end of resistor R40.

[0079] In the fan interface circuit, C28 is connected between VCC_12V and ground, and VCC_12V is connected to interface 1 in the fan interface JP4. Interface 4 of the fan interface is grounded.

[0080] Understandably, the fan drive circuit primarily amplifies the PWM signal output from the microcontroller via an operational amplifier (IC6) and outputs it to the cooling fan through a connector (JP4) to drive the fan for heat dissipation. When the microcontroller detects that the heatsink temperature is higher than 35°C, it starts outputting a PWM signal to drive the fan. When cooling is on, the microcontroller automatically adjusts the PWM pulse width based on real-time temperature data to control the fan speed. When cooling is on, if the detected heatsink temperature is lower than 25°C, the PWM signal is turned off, and cooling stops. When the detected heatsink temperature is higher than 85°C, the over-temperature protection function is triggered, automatically shutting off the load and issuing an alarm signal.

[0081] In some embodiments, the device includes a communication circuit connected to a serial communication interface of the control unit for receiving external control commands and uploading measurement data.

[0082] Specifically, the communication circuit uses an RS485 communication circuit with the SN65HVD1781DR chip (U7) as the core, and is equipped with external protection circuits to complete the circuit design. The specific circuit structure will not be described in detail here.

[0083] The device in this embodiment combines the advantages of both methods, employing hardware closed-loop control and software closed-loop fine-tuning to control the load current. This further reduces the gap between the load current and the set parameters, improving the load current control accuracy while ensuring stable and timely loop response. The control flow is as follows: when the load current parameter is set, this module calculates and outputs a preset DAC voltage output value based on the current measured load voltage. After the load is connected, a stable control loop is quickly established through the hardware closed-loop control circuit. The microcontroller periodically reads the current load current value and compares it with the preset current, gradually adjusting the output DAC voltage value until the measured load current is substantially consistent with the preset current.

[0084] Furthermore, to improve the accuracy of load parameter detection, this embodiment uses the INA226 power monitoring chip, which has a built-in 16-bit ADC converter. It can simultaneously measure the real-time voltage and current of the power supply under test, and can also directly measure the output power of the power supply under test through its internal multiplier, enabling simple and efficient acquisition of load parameter data. The programmable electronic load device has dimensions of approximately 150mm*90mm*100mm, and can be easily and quickly embedded into other devices as a sub-functional module via mounting holes. This programmable electronic load device uses an RS485 bus for networking with external functional components. It can be used in parallel with similar programmable electronic loads to expand the load power or current range; it can be used with an LCD screen with RS485 bus communication capabilities to develop a host computer interface, forming an independent, simple programmable DC electronic load device; it can also be paired with other main control modules with RS485 bus communication capabilities as a sub-functional module to complete functional tests, etc.

[0085] In some embodiments, as can be seen from the load current control circuit principle of the foregoing embodiments, the load voltage of this electronic load device is actually limited by the continuous drain-source power supply of the power MOSFET. By changing the MOSFET model, for example, replacing the IRFP4368 field-effect transistor with the IRFP4768 field-effect transistor with the same package, and adjusting the resistance value of the bus voltage measurement voltage divider resistor in the measurement unit, the circuit can be extended to a maximum load voltage of 250V without any other modifications.

[0086] Furthermore, this electronic module can handle a maximum current of 25A and a maximum power of 500W. It can be connected in parallel via an RS485 bus to expand the range of current and power to suit different applications.

[0087] This electronic load device can be controlled via RS485 bus to perform battery capacity testing. Specifically, the battery is connected to the power supply port of the module, constant current mode is selected, the load current value is set to 0.1C, and the real-time current consumption of the battery is read every 5ms. This current value is accumulated. When the real-time voltage of the tested battery drops to the discharge cutoff voltage, the accumulated current value is divided by the discharge time (in hours) to obtain the battery capacity of the tested battery pack.

[0088] A control method for a programmable DC electronic load device according to an embodiment of this application, used in the programmable DC electronic load device of the aforementioned embodiment, includes the following steps: S1: receiving target load parameters, the target load parameters including a target current value in constant current mode, a target voltage value in constant voltage mode, a target resistance value in constant resistance mode, or a target power value in constant power mode; S2: calculating the target current value according to the target load parameters and the voltage of the current power supply under test, using Ohm's law or a power formula; S3: outputting the initial digital-to-analog converter output voltage value to the feedback current loop to quickly establish the load current; S4: periodically acquiring the actual current value through the measurement unit; S5: comparing the actual current value with the target current value; S6: fine-tuning the output voltage value of the digital-to-analog converter module according to the comparison result, so that the actual current value approaches the target current value.

[0089] In some embodiments, a temperature control step is further included: acquiring the temperature value of the heat sink in real time through a temperature sensor; when the temperature value is greater than a first preset threshold, starting the cooling fan and outputting a PWM signal; dynamically adjusting the duty cycle of the PWM signal according to the temperature value to control the speed of the cooling fan; when the temperature value is lower than a second preset threshold, turning off the cooling fan; when the temperature value exceeds a third preset threshold, triggering over-temperature protection, shutting off the load and generating an alarm signal.

[0090] The above description is merely a preferred embodiment of the present invention. It should be understood that the present invention is not limited to the forms disclosed herein and should not be construed as excluding other embodiments. It can be used in various other combinations, modifications, and environments, and can be altered within the scope of the concept described herein through the above teachings or related technologies or knowledge. Modifications and variations made by those skilled in the art that do not depart from the spirit and scope of the present invention should be within the protection scope of the appended claims.

Claims

1. A programmable DC electronic load device, comprising a control unit, wherein the control unit is equipped with a digital-to-analog conversion module, characterized in that, It also includes a feedback current loop, which comprises a power dissipation unit and a measurement unit; The power dissipation unit includes at least one dissipation branch, and each of the dissipation branches includes a high-power MOSFET, a sampling resistor, a current sensing amplifier, and an operational amplifier. The high-power MOSFET is used to connect to the power supply under test to consume power; The sampling resistor is connected in series with the source of the high-power MOS transistor; The current sensing amplifier is connected to both ends of the sampling resistor; The non-inverting input of the operational amplifier is connected to the output of the digital-to-analog converter module, the inverting input is connected to the output of the current sensing amplifier, and the output of the operational amplifier is connected to the gate of the high-power MOSFET. The measuring unit is connected between the power supply under test and the power dissipation unit to measure the actual current value flowing through the power dissipation unit, and the measuring unit is connected to the input terminal of the control unit; The control unit controls the output voltage of the digital-to-analog converter module to establish a load current in the feedback current loop, periodically obtains the actual current value through the measurement unit, compares the actual current value with the target current value, and adjusts the output voltage of the digital-to-analog converter module to make the actual current value approach the target current value. The digital-to-analog converter module includes at least one output channel, each output channel is connected to multiple current-sharing resistors, each current-sharing resistor drives one of the dissipation branches, and the current-sharing resistors are connected to the non-inverting input terminal of the operational amplifier; In the feedback current loop: The non-inverting input of the operational amplifier is connected to the output of the digital-to-analog converter module through a current-sharing resistor; the inverting input of the operational amplifier is connected to the output of the current sensing amplifier after being connected in series with a first bias resistor. The output of the operational amplifier is connected to the gate of the high-power MOSFET via a drive resistor; The gate of the high-power MOSFET is grounded through a second bias resistor. The current sharing resistor, the first bias resistor, the driving resistor, and the second bias resistor constitute the bias circuit.

2. The programmable DC electronic load device according to claim 1, characterized in that, The control unit acquires the voltage of the power supply under test through the measurement unit, and can convert the set value of constant voltage mode, constant resistance mode or constant power mode into the corresponding target current value through calculation according to the received instructions, and maintain constant voltage by controlling the output voltage of the digital-to-analog conversion module.

3. The programmable DC electronic load device according to claim 1, characterized in that, It also includes a heat dissipation system, which includes a heat sink mounted on the high-power MOSFET, a temperature sensor mounted on the heat sink, and a cooling fan driven by the control unit via a PWM signal; the control unit is configured to adjust the duty cycle of the PWM signal according to the temperature value collected by the temperature sensor.

4. The programmable DC electronic load device according to claim 1, characterized in that, It also includes a communication circuit, which is connected to the serial communication interface of the control unit for receiving external control commands and uploading measurement data.

5. The programmable DC electronic load device according to claim 1, characterized in that, The measurement unit includes an integrated power monitoring chip, which simultaneously measures the bus voltage of the power supply under test and the current flowing through the sampling resistor.

6. A control method for a programmable DC electronic load device, characterized in that, A method for controlling the programmable DC electronic load device according to any one of claims 1-5 includes the following steps: S1. Receive target load parameters, the target load parameters including target current value in constant current mode, target voltage value in constant voltage mode, target resistance value in constant resistance mode or target power value in constant power mode. S2. Calculate the target current value based on the target load parameters and the voltage of the current power supply being measured, using Ohm's law or the power formula. S3. Output the initial digital-to-analog converter module output voltage value to the feedback current loop to quickly establish the load current; S4. The actual current value is periodically obtained through the measurement unit; S5. Compare the actual current value with the target current value; S6. Adjust the output voltage value of the digital-to-analog converter module according to the comparison result so that the actual current value approaches the target current value.

7. The control method for the programmable DC electronic load device according to claim 6, characterized in that, It also includes temperature control steps: The temperature value of the radiator is collected in real time by a temperature sensor; When the temperature value exceeds the first preset threshold, the cooling fan is activated and a PWM signal is output. The duty cycle of the PWM signal is dynamically adjusted according to the temperature value to control the speed of the cooling fan. When the temperature value is lower than the second preset threshold, the cooling fan is turned off; When the temperature value exceeds the third preset threshold, over-temperature protection is triggered, the load is shut down, and an alarm signal is generated.

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