High-precision time synchronization method based on microwave bidirectional measurement

Abstract

The invention discloses a high-precision time synchronization method based on microwave bidirectional measurement in the technical field of wireless equipment synchronization. The method comprises the following steps: S1, establishing a synchronization system, wherein the synchronization system is composed of a reference device and at least one synchronization device; s2, bidirectional timestamp measurement: the reference device sends a timestamp message to the synchronization device and marks the transmission time T1; the synchronization device receives the timestamp message and marks the receiving time T2; the synchronization device sends a timestamp message to the reference device and marks the transmission time T3; the reference device receives the timestamp message and marks the receiving time T4; s3, clock difference calculation: obtaining a clock difference between the reference equipment and the synchronization equipment according to a formula; and S4, clock correction: the synchronization device adjusts a local clock according to the clock difference to realize time synchronization. According to the method, asynchronous polling roll call type measurement is adopted, time slot occupation is reduced, the interference probability is reduced, and system communication time slots are saved; and link asymmetric errors are eliminated through bidirectional measurement, and the precision is improved.

Description

TECHNICAL FIELD

[0001] The present application relates to the field of wireless device synchronization technology, in particular to a high-precision time synchronization method based on microwave bidirectional measurement. BACKGROUND

[0002] The principle of time synchronization technology in different places is basically realized by bidirectional transmission ranging, and the transmission mode of signals is mainly wired and wireless. Common wired transmission comparison methods, such as network time service widely used in mobile networks, etc. The wired method can only be applied in a fixed network range, the area is fixed, and wiring (network cable or optical fiber) is required, which has limitations. Satellite common view and microwave common view are mainly wireless methods. In recent years, satellite common view, microwave common view and other time transfer synchronization technologies have been widely used. Satellite signals are easily disturbed and lose service capability in wartime or some complex environments due to the characteristics of satellite signals, or the satellite system is directly disturbed, and the working system loses the ability to obtain time synchronization.

[0003] Microwave common view is an effective time synchronization method in a certain distance range because its communication frequency can be customized and uses line-of-sight propagation. In a cooperative working system, precise time synchronization can be achieved, and it is not easy to be disturbed by human interference and industrial electromagnetic interference to cause system failure. Microwave common view adopts a wideband spread spectrum system to establish a wireless time transfer link for covert communication, which can work independently without relying on satellite systems, thereby improving the working ability of the system.

[0004] However, the traditional microwave common view uses a continuous comparison method, which occupies many communication time slots and is not suitable for radar and other bandwidth-constrained scenarios. In addition, the traditional BPSK modulation autocorrelation peak width used in the traditional microwave common view has a large tracking error, which leads to insufficient synchronization accuracy. SUMMARY

[0005] The present application provides a high-precision time synchronization method based on microwave bidirectional measurement, which solves the problem of "microwave common view occupying many communication time slots and insufficient precision" in the prior art, reduces the interference probability and communication time slot, and obtains accurate time difference measurement, thereby realizing high-precision time synchronization of the system.

[0006] The present application provides a high-precision time synchronization method based on microwave bidirectional measurement, which includes the following steps: S1: Assembling a synchronization system: composed of 1 reference device and at least 1 synchronization device, the reference device establishes a wireless link with each synchronization device in turn, and measures the time difference by using a polling method; S2: Bidirectional timestamp measurement: S2.1: The reference device sends a timestamp message to the synchronization device and marks the transmission time T1; the synchronization device receives the timestamp message and marks the reception time T2; S2.2: The synchronization device sends a time stamp message to the reference device and marks the transmission time T3; the reference device receives the time stamp message and marks the reception time T4; S3: Clock difference calculation: according to the formula ; obtain the clock difference Δt between the reference device and the synchronization device; S4: Clock correction: the synchronization device adjusts the local clock according to the clock difference Δt to achieve time synchronization.

[0007] The beneficial effects of the above embodiments are that in the microwave wireless mode, the continuous comparison mode and the network working mode of the prior art are improved, the asynchronous "polling roll call" type measurement is adopted, the time slot occupation is reduced, the interference probability is reduced, and the system communication time slot is saved; the link asymmetry error is eliminated through bidirectional measurement, the precision is improved, and it is especially suitable for network application scenarios with tight communication bandwidth and time slot of radar system.

[0008] On the basis of the above embodiments, the present application can be further improved, specifically as follows: In one of the embodiments of the present application, the reference device or the synchronization device includes a radio frequency processing unit, a baseband processing unit, and a communication antenna, the radio frequency transceiver unit is used for frequency conversion processing and accurate sampling of wireless microwave signals, the baseband signal processing unit is used for signal processing, time difference measurement, calculation and compensation, and the communication antenna is used for receiving and transmitting radio frequency microwave signals. Technical effect: the hardware / software of the reference device and the synchronization device are consistent, so that the roles of the reference device and the synchronization device can be dynamically switched, the hardware / software is multiplexed, the networking cost is reduced, and the system flexibility is improved.

[0009] In one of the embodiments of the present application, the radio frequency processing unit includes a radio frequency conditioning unit and a radio frequency agile frequency chip, the radio frequency conditioning unit is composed of a transmission channel and a receiving channel, the radio frequency processing unit performs radio frequency signal transceiver conversion processing, radio frequency direct sampling and mixing, and realizes the communication between the communication antenna and the baseband signal processing unit. Supplementary technical effect: direct frequency conversion reduces link loss, AD sampling rate 1.6GHz, 14bit orthogonal I / Q output ensures signal quality, and improves synchronization measurement precision.

[0010] In one of the embodiments of the present application, the baseband signal processing unit is a programmable SOC chip, the PL end is responsible for signal capture / tracking / synchronization / modulation / spreading, and the PS end is responsible for algorithm calculation / process control. The baseband signal processing unit includes a receiving signal processing module, a transmitting signal processing module and an information processing module, wherein the receiving signal processing module is used for processing the reception time of the measurement signal and analyzing the time information contained in the time stamp message, including: A pseudo code generator sub-module for generating a pseudo code sequence; An acquisition sub-module for acquiring a tracking start time according to the pseudo code sequence; a tracking sub-module, configured to realize carrier synchronization and pseudo-code synchronization based on the capturing sub-module; a decision sub-module, configured to make a decision on the synchronization result of the tracking sub-module; a synchronization sub-module, configured to detect a synchronization header information before data extraction, judge whether phase inversion occurs, and give a data validity identification; a data extraction sub-module, configured to perform data extraction control to form a data frame after detecting the data validity identification of the synchronization sub-module. Technical effects: single SOC integration simplifies hardware architecture, PL+PS division improves processing efficiency, supports complex synchronization algorithm, and reduces code phase tracking error.

[0011] In one of the embodiments of the present application, the baseband signal processing unit adopts a BOC signal modulation and demodulation mode, and the BOC modulated signal can be represented by the following formula: ; wherein, is a data source code, represents a pseudo-code for spread spectrum modulation, and the frequency thereof is , the subcarrier frequency is , is the intermediate frequency carrier signal frequency, and the BOC modulated signal can be recorded as , and a BPSK modulated signal with the same pseudo-code rate is recorded as ; After Fourier transform is performed on the above signals, the frequency spectrum of the BPSK and BOC signals is represented by the following formula: , ; ; The frequency spectrum of the BOC signal is obtained by shifting the frequency spectrum of the BPSK signal left and right by the subcarrier frequency , and then superimposing. Therefore, by selecting appropriate frequency parameters, the BOC signal can not only share the frequency spectrum resource with the BPSK signal of the same pseudo-code rate, but also can avoid mutual interference of BOC signals with different modulation parameters.

[0012] For the BPSK signal, the normalized autocorrelation function is: ; wherein, is the pseudo-code period. It can be known that the BPSK signal has strong autocorrelation, and only when the phase deviation of the pseudo-code is within 1 chip, the autocorrelation function can obtain a non-zero value.

[0013] For the BOC signal, the normalized autocorrelation function is: ; wherein, .

[0014] Technical effects: the autocorrelation function of the BOC signal has a multi-peak characteristic. As the modulation order increases, the main peak spacing of the autocorrelation function is narrower, the accuracy of the theoretical tracking code phase is higher. The anti-interference ability is enhanced, and it is suitable for complex electromagnetic environment.

[0015] In one of the embodiments of the present application, the baseband signal processing unit further comprises a bidirectional time comparison subunit, configured to receive the local measurement observation input by the modulation and demodulation subunit and the measurement observation of the device to be compared, and calculate the measurement values such as relative distance, speed and clock difference.

[0016] In one of the embodiments of the present application, the baseband signal acquisition adopts the ASPeCT algorithm, and the BOC signal autocorrelation side peak is eliminated by two-way correlation square subtraction. Technical effects: solve the BOC multi-peak ambiguity problem, improve the acquisition accuracy, and reduce the main peak width.

[0017] In one of the embodiments of the present application, the baseband signal tracking adopts the DLE double loop algorithm, and the carrier loop, code loop and subcarrier loop are independently tracked. Technical effects: reduce the code phase tracking error and improve the timestamp measurement accuracy.

[0018] In one of the embodiments of the present application, the data frame adopts the structure of "preamble + Barker code synchronization header + data". Technical effects: the Barker code has sharp autocorrelation, fast acquisition, high tracking accuracy; short frame period supports multiple measurements, and reduces random error.

[0019] In one of the embodiments of the present application, after obtaining the time difference of the two devices, the relative frequency deviation is calculated: ; In the above formula, is the sampling time, x (0) is the initial time difference of the two devices, is the time difference of the two devices after the sampling time, f A and f B are the average frequencies of the reference device and the synchronous device in the sampling time, respectively. Therefore, it has the ability to simultaneously complete the accurate measurement of the time difference and the frequency difference between the dynamic carriers. BRIEF DESCRIPTION OF DRAWINGS

[0020] In order to more clearly illustrate the technical solutions in the specific embodiments of the present application or the prior art, the drawings needed to be used in the description of the specific embodiments or the prior art will be briefly introduced below. In all the drawings, similar elements or parts are generally identified by similar reference numerals. In the drawings, the elements or parts are not necessarily drawn according to the actual proportions.

[0021] Figure 1 A flow chart of a high-precision time synchronization method based on microwave bidirectional measurement in an embodiment of the present application is shown; Figure 2 An application scenario schematic diagram of system time synchronization implementation in an embodiment of the present application is shown; Figure 3 A flow chart of a time correction parameter calculation method in an embodiment of the present application is shown.

[0022] Figure 4 A schematic diagram of a reference device / synchronization device in an embodiment of the present application is shown; Figure 5 A principle schematic diagram of a radio frequency processing unit in an embodiment of the present application is shown; Figure 6 A principle schematic diagram of a baseband signal processing unit in an embodiment of the present application is shown; Figure 7 A software function schematic diagram of a baseband signal processing unit in an embodiment of the present application is shown; Figure 8 A design block diagram of a BOC signal in an embodiment of the present application is shown; Figure 9 A schematic diagram of a capture circuit in an embodiment of the present application is shown; Figure 10 A simulation-self-correlation function comparison schematic diagram in an embodiment of the present application is shown; Figure 11 A tracking loop principle schematic diagram in an embodiment of the present application is shown; Figure 12 A simulation-tracing-carrier loop phase error diagram in an embodiment of the present application is shown; Figure 13 A simulation-tracing-carrier loop phase result diagram in an embodiment of the present application is shown; Figure 14 A simulation-tracing-code loop phase error diagram in an embodiment of the present application is shown; Figure 15 A simulation-tracing-code loop phase result diagram in an embodiment of the present application is shown; Figure 16 A data frame structure schematic diagram in an embodiment of the present application is shown. DETAILED DESCRIPTION

[0023] The present application will be further clarified by the following examples, which should not be construed as limiting the scope of the present application. After reading the present application, those skilled in the art will be able to affect various modifications to the application without departing from the scope of the application as defined by the accompanying claims.

[0024] Embodiments: The implementation method of the high-precision time synchronization system provided in the embodiments of the present application introduces a BOC modulation and demodulation mode in a signal processing system, realizes microwave bidirectional high-precision time difference measurement, and realizes high-precision time synchronization of the system in a flexible and practical manner, so as to solve the problems of low synchronization precision and poor anti-interference performance of the existing time synchronization implementation manner.

[0025] As shown in Figure 1 , a high-precision time synchronization method based on microwave bidirectional measurement, comprising the following steps: S1: Assembling a synchronization system: composed of one reference device and at least one synchronization device, the reference device establishes a wireless link with each synchronization device in time, and the subsequent time difference is measured by using a polling manner; The specific system networking work schematic diagram is shown in Figure 2 . The system is formed by one reference device and at least one synchronization device. The reference device establishes a wireless link with all synchronization devices in time, and each synchronization device establishes a wireless link with the reference device, and the synchronization devices do not build a link. The reference device and the synchronization device realize communication, relative distance measurement and other functions through the wireless microwave link of the unified channel. Taking the reference device as the time reference, multiple synchronization devices are compared with the reference device to realize the time synchronization between multiple devices.

[0026] S2: The flow of bidirectional timestamp measurement is shown in Figure 3 . S2.1: The reference device polls the synchronization devices with its own clock as the reference to send timestamp messages, and marks the transmission time T1; the synchronization device receives the timestamp message and marks the reception time T2; S2.2: The synchronization device sends a timestamp message to the reference device and marks the transmission time T3; the reference device receives the timestamp message and marks the reception time T4; S3: Clock difference calculation: according to the formula , the clock difference Δt between the reference device and the synchronization device is obtained; The specific clock difference calculation is as follows: It is known that T1, T2, T3, and T4 need to obtain Δt to adjust the clock of the slave device: (1); Since it is assumed that the two-way ranging and time synchronization system is highly symmetrical, the transmission delays along the path are considered to be equal, i.e. Then we can solve it as follows: (2); It can be seen that Δt and d are only related to the differences between T2 and T1, and the differences between T3 and T4. Therefore, the time difference Δt can be calculated from T1, T2, T3, and T4. By measuring in both directions, the clock difference Δt and distance d between the two devices can be obtained.

[0027] After obtaining the time difference between the two devices, the relative frequency deviation between them can also be easily calculated: (3); In the above formula, For sampling time, x (0) represents the initial time difference between the two devices. The time difference between the two devices after the sampling time. f A and f B These are the average frequencies of device A and device B during the sampling period, respectively. Therefore, this system has the capability to simultaneously and accurately measure the time difference and frequency difference between dynamic carriers.

[0028] S4: Clock Correction: The synchronization device adjusts the local clock according to the clock difference Δt to achieve time synchronization.

[0029] In this embodiment, the roles of the base preparation device and the synchronization device in the time synchronization system are interchangeable. For example... Figure 4 As shown, the reference / synchronization equipment mainly consists of a radio frequency (RF) processing unit, a baseband signal processing unit, and a communication antenna. The RF transceiver unit performs frequency conversion processing and precise sampling of the wireless microwave signal. The baseband signal processing unit performs signal processing and time difference measurement calculation and compensation. The communication antenna enables the reception and transmission of RF microwave signals. The baseband signal processing unit includes receiving, information processing, and transmitting modules. The receiving end performs signal acquisition and tracking, data frame synchronization, and information decoding and parsing, calculating local receiving timestamps, local time, and other information, performing measurement data parsing and local time correction and synchronization. The transmitting end performs measurement information framing, spread spectrum, and modulation. The receiving signal processing module processes the reception time of the measurement signal and parses the time information contained in the timestamp message, including: The pseudocode generator submodule is used to generate pseudocode sequences; A capture submodule is used to obtain the tracking start time based on the pseudocode sequence; The tracking submodule is used to achieve carrier synchronization and pseudocode synchronization based on the capture submodule; The decision submodule is used to make a decision on the synchronization results of the tracking submodule; The synchronization submodule is used to detect synchronization header information before data extraction, determine whether the phase has flipped, and provide a valid data identifier. The data extraction submodule is used to perform data extraction control and form a data frame after detecting the valid data identifier of the synchronization submodule.

[0030] Furthermore, the radio frequency (RF) processing unit provides uplink and downlink communication channels for the reference or synchronization equipment, and provides the required clock signal and the digital intermediate frequency (IF) signal obtained from the RF direct sampling to the baseband signal processing unit. For example... Figure 5 As shown, the radio frequency processing unit mainly includes a transmit channel, a receive channel, a clock management system, and a radio frequency agile RFSoC.

[0031] The RF processing unit operates in the S-band at a frequency of 3.3 GHz, providing uplink and downlink communication channels for reference and synchronization equipment. It manages the 100 MHz reference clock signal and provides the operating clock for the RFSoC chip and baseband processing unit. The AD sampling rate is 1.6 GHz, with 14-bit quadrature I / Q output. After direct frequency conversion by the RFSoC chip, a 10 MHz digital intermediate frequency signal is output to the baseband processing unit via the JESD204C interface; the transmitter directly converts the signal to a 3.3 GHz RF signal for the transmission channel.

[0032] Furthermore, such as Figure 6 As shown, the baseband signal processing unit mainly consists of a baseband signal processing chip, interface circuits, and power supply circuits, forming a hardware implementation platform. It uses a single-chip SOC to implement the core functions of baseband signal processing and measurement. The core processing chip of the baseband signal processing unit is the SOC chip, implemented using a single-chip FPGA with an integrated ARM hard core, such as Xilinx's FMQL7045. The SOC chip is divided into a PL (logic) side and a PS (ARM) side. The PS section contains four ARM9 CPUs. The PS side interacts with the PL side via an internal bus.

[0033] All software functions of the baseband signal processing unit are completed on the SOC chip. The overall software design architecture is as follows: Figure 7 As shown. The PL section receives the digital intermediate frequency signal and completes the acquisition, tracking, decision-making, synchronization detection, and data extraction of the carrier and pseudo-code signals. The PL section also implements the data encapsulation, spreading, and modulation functions of the baseband signal of the transmitting section, generates the local spreading code and digital intermediate frequency, and finally outputs the digital modulated signal. The PS section completes the control and configuration of the baseband PL section, and implements functions such as flow control, function scheduling, information parsing, and framing. At the same time, the PS section also implements functions such as bidirectional ranging algorithm and interface data protocol packaging.

[0034] The baseband signal processing unit introduces a BOC (Browser-Oriented) modulation and demodulation method. The BOC modulated signal can be represented by the following formula: (4); in, For data source code, The pseudocode representing spread spectrum modulation has a frequency of . The subcarrier frequency is , It is the intermediate frequency carrier signal frequency, therefore the BOC modulated signal can be denoted as: .

[0035] Performing a Fourier transform on the above signal, the spectrum of the BOC signal can be expressed by the following formula: (5); (6); The spectrum of the BOC signal is obtained by shifting the subcarrier frequency left and right of the spectrum of the BPSK signal. The signal is then superimposed. Therefore, by selecting appropriate frequency parameters, BOC signals can not only share spectrum resources with BPSK signals of the same pseudocode rate, but also avoid mutual interference between BOC signals with different modulation parameters.

[0036] For the BOC signal, its normalized autocorrelation function is: (7); in, (8).

[0037] The autocorrelation function of the BOC signal exhibits multi-peak characteristics. As the modulation order increases, the spacing between the main peaks of the autocorrelation function narrows, and the accuracy of the theoretical tracking code phase increases.

[0038] The baseband signal processing unit borrows the modulation method and dual-branch architecture of next-generation GNSS signals, adopting an orthogonal multiplexing dual-branch structure and a data:pilot power ratio of 1:1. It also uses BOC modulation as the signal pattern for data exchange between the master and slave units. The design block diagram of the BOC signal is shown below. Figure 8 As shown.

[0039] in, For data rate, It is a data source signal; , The signal is a dual-branch pseudocode signal generated by a pseudo-random code generator. For pseudocode rate; It is a sinusoidally modulated binary non-return-to-zero square wave subcarrier, i.e. , For subcarrier frequency; and It is a dual-branch carrier modulated at the transmitting end. It is the carrier frequency; The modulated transmitted signal is expressed as shown in equation (9), where C is the signal power of each branch.

[0040] (9); Traditional BPSK acquisition methods cannot address the code phase ambiguity and false locking issues caused by the multiple correlation peaks in BOC signals. This embodiment employs the ASPeCT algorithm, based on autocorrelation side-peak elimination technology, to eliminate the side peaks of the autocorrelation function, resulting in a single-peak characteristic in the reconstructed correlation function. Its mathematical principle is shown in the following equation: (10); in, For the relevant functions of reconstruction, This represents the squared value of the cross-correlation between the local BOC code and the received baseband signal. (Parameter) Used to adjust the structure to fit the derived BOC signal, for a BOC(n, n) signal, , This represents the square of the cross-correlation between the local spreading code (unmodulated square wave subcarrier) and the received signal. The BOC signal construction in this embodiment is shown in Table 1.

[0041]

[0042] The acquisition circuit of the ASPeCT algorithm combined with the parallel code phase algorithm is as follows: Figure 9 As shown, the local pseudocode signal is split into two paths. One path is directly correlated with the baseband signal, and the other path is modulated by a square wave subcarrier and then correlated with the baseband signal. The squares of the two paths are subtracted to obtain the detection value, which is then used for acquisition decision.

[0043] The above capture structure was simulated, and the results are as follows: Figure 10 As shown in the figure, the peak height of the reconstructed autocorrelation function is completely consistent with that of the BOC(10,10) signal, without any power loss; moreover, the peak width is narrower than that of the BOC(10,10) autocorrelation function, decreasing from 2 / 3 of a chip to 1 / 2 of a chip, theoretically improving the code phase tracking accuracy; and eliminating most of the ambiguity of the BOC(10,10) signal autocorrelation function. The acquisition algorithm in this embodiment significantly reduces the ambiguity of the autocorrelation function while slightly improving the phase resolution.

[0044] After the received signal is acquired, higher precision tracking of the Doppler frequency offset and pseudocode phase is required. For BOC signal tracking, since its modulation method only modulates a square wave subcarrier on the pseudocode, the basic structure of carrier tracking is the same as that of traditional BPSK signals. The problem to be solved is the side peak problem of the BOC signal code tracking loop, that is, finding a method to eliminate or weaken the side peak of the autocorrelation function.

[0045] This embodiment improves upon the traditional tracking loop architecture by modifying the basic structure of the code tracking loop to make it suitable for tracking signals with multi-peak autocorrelation functions. It employs a double-loop estimation algorithm (DLE). The square wave subcarrier is treated as another form of pseudo-code, and a subcarrier code loop (SLL) is added to track the subcarrier code phase independently. The carrier loop, code loop, and subcarrier loop are independent of each other. The basic structure of its tracking loop is as follows: Figure 11 As shown.

[0046] Depend on Figure 11 It can be seen that the carrier loop, code loop, and subcarrier loop operate independently, with three branches (E, P, and L) performing correlation and integration operations, followed by phase detection, filtering, and phase adjustment. This achieves tracking and locking of the carrier phase and code phase of the BOC signal.

[0047] In practice, the intermediate frequency (IF) signal first passes through a mixer to obtain the baseband signal. Then, each branch signal is correlated with the local CA code and BOC code, and after integration and accumulation, the result is sent to the discriminator. The three branches of EPL (lead, instant, and lag) in the loop are illustrated as one. After carrier phase detection, carrier frequency detection, and code loop phase detection, the discrimination results of the data and pilot branches are linearly superimposed and sent to the code loop filter and carrier loop filter for filtering. Finally, the filtered output value is used to control the local carrier and chip generation, forming a tracking closed loop.

[0048] For the tracking loop, the relevant parameters are shown in Table 2. Both the phase-locked loop and the code loop use second-order loops, and their damping coefficients are... The value is 0.707. On the other hand, code loops and phase-locked loops have smaller noise bandwidths, which is beneficial for achieving higher tracking accuracy; while frequency-locked loops have larger noise bandwidths and use a carrier loop structure with a first-order frequency-locked loop assisting a second-order phase-locked loop, thus improving the dynamic performance of the carrier loop.

[0049]

[0050] For BOC(10,10) signals, the key to tracking is also handling the ambiguity of the autocorrelation function. Therefore, a code ring phase detector based on the ASPeCT algorithm is proposed to identify the fine code phase. Its normalized incoherent lead-lag phase detection formula is as follows: (11); in, , These represent the received signals of the in-phase and quadrature branches, respectively. The subscripts E, P, and L represent the leading, immediate, and lagging branches, respectively. This indicates that the received BOC signal is correlated with the local BOC code. This indicates that the received signal is correlated with the local CA code.

[0051] The simulation conditions for the data source were set as follows: signal-to-noise ratio of -15 dB, code phase of 157.1 chips, Doppler frequency offset of 4350 Hz, and the data source code was set to 200 bits with a synchronization frame header of 1111100110101. The signal was tracked, and the tracking results of the phase-locked loop and code loop are as follows.

[0052] Depend on Figure 12 It can be seen that, since the initial Doppler error is only 150 Hz, the phase-locked loop begins to converge and enters the locked state after tracking for about 3.5 ms. Figure 13 In the loop-locked system, the peak-to-peak value of the carrier phase tracking error is 4.5° (in degrees), which is 1 / 80 of a carrier cycle. The calculated standard deviation is 1.8°. At this point, there is a 180-degree phase difference. The decoded synchronization code is 0000011001010, which is a flip. Therefore, the 180-degree phase difference can be compensated for and eliminated.

[0053] Figure 14 To delay the phase detection result of the locked loop, the loop begins to converge after 5 ms. Figure 15 It can be seen that the peak-to-peak value of the code phase tracking error is 0.0046 chip, that is, the error is within 1 / 200 chip, and its standard deviation is 0.0016 chip. Therefore, the tracking of the BOC(10,10) signal was successfully achieved, the tracking code phase accuracy is higher than 1 / 200 chip, and the final timestamp measurement error reaches the sub-nanosecond level accuracy.

[0054] This baseband signal processing unit introduces a synchronization header with better correlation to establish measurement information data frames, facilitating rapid acquisition and precise tracking. The time-slot data frame uses a preamble + synchronization header + data approach. The synchronization header employs Barker codes, which have sharp autocorrelation, high synchronization efficiency, and low inter-symbol interference. The design scheme of the time-slot data frame structure is as follows: Figure 16 As shown.

[0055] The data segment mainly includes time information, bidirectional comparison data, status information, etc.; other information is reserved fields that can be used for communication transmission of useful information of the system via data link and can be selectively loaded, such as time information such as year count, week count within the year, location information of master and slave devices, system status information, etc.

[0056] The frame period is designed to be 42.5ms, with each time slot containing one frame. The guard time slot interval between each time slot is 0.5ms. The frame time slot consisting of the preamble, header, and data is 20.1ms. Assuming a transmission time of no more than 0.35ms over 100km, we use 0.5ms, resulting in a total one-way transmission time of 21.1ms. Therefore, the time to complete one time comparison measurement is 42.2ms. To allow for a margin, we use 42.5ms. Simulation analysis shows that if 100 preambles are designed, the time to complete one comparison can be reduced to approximately 4ms. This reduction in time for a single measurement allows for multiple measurements. Data processing from multiple measurements can further reduce measurement errors.

[0057] The technical solutions described in the embodiments of this application have at least the following technical effects or advantages: 1. This invention establishes a measurement link between a reference device and a synchronization device based on a wireless microwave link. It employs a "point-to-point" networking mode and measurement method, enabling rapid calculation of measurement results with minimal system time slot occupation and high system resource utilization. Establishing the measurement link via a wireless microwave link offers flexibility and strong applicability, with low requirements for hardware resources and the operating environment.

[0058] 2. The hardware implementation is simple. The RF channel adopts an integrated frequency agile processing method, and the baseband processing adopts an integrated SOC method, realizing integrated design, improving system reliability and economy.

[0059] 3. The introduction of BOC modulation and demodulation in wireless microwave communication measurement improves the measurement accuracy of signal synchronization tracking and the anti-interference performance in the link, which is superior to the traditional BPSK spread spectrum system.

[0060] Although embodiments of the present invention have been shown and described above, it is understood that the above embodiments are exemplary and should not be construed as limiting the present invention. Those skilled in the art can make changes, modifications, substitutions and variations to the above embodiments within the scope of the present invention.

Claims

1. A high-precision time synchronization method based on microwave two-way measurement, characterized in that, The method comprises the following steps: S1: forming a synchronization system: composed of one reference device and at least one synchronization device, the reference device establishes a wireless link with each synchronization device in turn, and measures the time difference by polling; S2: bidirectional time stamp measurement: S2.1: the reference device sends a time stamp message to the synchronization device and marks the transmission time T1; the synchronization device receives the time stamp message and marks the reception time T2; S2.2: the synchronization device sends a time stamp message to the reference device and marks the transmission time T3; the reference device receives the time stamp message and marks the reception time T4; S3: Clock error calculation: According to the formula ; the clock difference Δt between the reference device and the synchronization device is obtained; S4: clock correction: the synchronization device adjusts the local clock according to the clock difference Δt to achieve time synchronization.

2. The high precision time synchronization method of claim 1, wherein: The reference device or the synchronization device comprises a radio frequency processing unit, a baseband processing unit and a communication antenna, the radio frequency transceiver unit is used for frequency conversion processing and accurate sampling of the wireless microwave signal, the baseband signal processing unit is used for signal processing, time difference measurement, solution and compensation, and the communication antenna is used for receiving and transmitting the radio frequency microwave signal.

3. The high precision time synchronization method of claim 2, wherein: The radio frequency processing unit comprises a radio frequency conditioning unit and a radio frequency agile frequency chip, the radio frequency conditioning unit is composed of a transmission channel and a receiving channel, the radio frequency processing unit performs radio frequency signal transceiver conversion processing, radio frequency direct sampling and mixing, and realizes the communication between the communication antenna and the baseband signal processing unit.

4. The high precision time synchronization method of claim 2, wherein: The baseband signal processing unit is a programmable SOC chip, the PL end is responsible for signal capture / tracking / synchronization / modulation / spreading, and the PS end is responsible for algorithm solution / process control; the baseband signal processing unit comprises a receiving signal processing module, a transmitting signal processing module and an information processing module, wherein the receiving signal processing module is used for processing the reception time of the measurement signal and analyzing the time information contained in the time stamp message, comprising: a pseudo code generator submodule for generating a pseudo code sequence; an acquisition submodule for acquiring a tracking start time according to the pseudo code sequence; a tracking submodule for realizing carrier synchronization and pseudo code synchronization based on the acquisition submodule; a decision submodule for judging the synchronization result of the tracking submodule; a synchronization submodule for detecting synchronization header information before data extraction, judging whether phase inversion occurs, and giving a data validity identifier; a data extraction submodule for performing data extraction control after detecting the data validity identifier of the synchronization submodule, and forming a data frame.

5. The high precision time synchronization method of claim 4, wherein: The baseband signal processing unit adopts a BOC signal modulation and demodulation mode, and the BOC modulated signal can be represented by the following formula: ； wherein, is a data source code, represents a pseudo code for spread spectrum modulation, whose frequency is , the subcarrier frequency is , is the intermediate frequency carrier signal frequency, then the BOC modulated signal can be recorded as , and the BPSK modulated signal with the same pseudo code rate is recorded as ; After Fourier transform is performed on the above signal, the frequency spectrum of the BPSK and BOC signals is represented by the following formula: ， ； ； For the BPSK signal, the normalized autocorrelation function is: ； wherein is a pseudo code period; For the BOC signal, the normalized autocorrelation function is: ； wherein, 。 6. The high precision time synchronization method of claim 4, wherein: The baseband signal acquisition adopts the ASPeCT algorithm, and the BOC signal autocorrelation side peak is eliminated by two-way correlation square subtraction.

7. The high precision time synchronization method of claim 4, wherein: The baseband signal tracking adopts the DLE double loop algorithm, and the carrier loop, the code loop and the subcarrier loop are independently tracked.

8. The high precision time synchronization method of claim 4, wherein: The data frame adopts the structure of preamble+ Barker code synchronization header+data.

9. The high precision time synchronization method of claim 1, wherein: After the time difference is obtained, the relative frequency deviation is calculated: ； In the above equation, is the sampling time, x (0) is the initial time difference between the two devices, is the time difference between the two devices after the sampling time, f A and f B are the average frequencies of the reference device and the synchronized device, respectively, over the sampling time.

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