Broadband magnetoelectric dipole antenna based on bowknot-shaped polarization conversion metasurface

Abstract

The invention discloses a broadband magnetoelectric dipole antenna based on a bowknot-shaped polarization conversion metasurface, and belongs to the technical field of magnetoelectric dipole antennas. In order to solve the problems that an existing antenna is high in profile, radiation and scattering performance is difficult to consider, and broadband RCS reduction is difficult to achieve in a Ka band, the antenna adopts three metallization layers and two dielectric substrates which are arranged in a staggered mode, and a PCM checkerboard array composed of bowknot-shaped PCM units is integrated above a magnetoelectric dipole radiation unit; the magnetoelectric dipole radiation unit adopts a coplanar design, comprises four radiation patches serving as electric dipoles and two perforated annular microstrip feeder lines serving as magnetic dipoles, and feeds through a 50-ohm coaxial feed port arranged at the bottom of the magnetoelectric dipole radiation unit. The overall thickness of the antenna is only 0.762 mm, the low profile is achieved, and meanwhile the antenna has the comprehensive excellent performance of wide impedance matching (23.26%, 30.4-38.4 GHz), high gain (the peak value is 11.9 dBi) and ultra-wideband RCS reduction (gt, 10 dB, and the relative bandwidth is 63.83%, 28.9-56 GHz).

Description

Technical Field

[0001] This invention relates to the field of magnetoelectric dipole antenna technology, and more particularly to a broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface. Background Technology

[0002] With the increasing complexity of the modern electromagnetic environment, the requirements for stealth performance of various combat platforms are constantly rising. As a key component of wireless communication and radar systems, antennas have strong electromagnetic scattering characteristics, which significantly increase the radar cross section (RCS) of stealth platforms. Therefore, designing antennas that combine good radiation performance with low RCS has become an important research direction.

[0003] Polarization conversion metasurfaces (PCMs) can suppress backscattering by changing the polarization state of incident waves, and are therefore widely used for antenna RCS reduction. However, existing PCM-based antennas still have the following limitations: some structures introduce air layers or multiple dielectric layers to broaden bandwidth, resulting in increased profile and complex manufacturing processes; some layouts generate grating lobes due to unreasonable element spacing, affecting radiation characteristics; and some designs, while achieving RCS reduction within a certain frequency band, have limited bandwidth, unstable low-frequency performance, or large overall thickness, making it difficult to meet the requirements of modern stealth platforms for low profile, wide bandwidth, and high integration. These problems are particularly pronounced in high-frequency bands such as Ka: the contradiction between structural complexity and electrical dimensions intensifies, and the difficulty of co-designing broadband RCS reduction and radiation performance increases significantly, making it difficult for existing solutions to achieve stable broadband high-gain radiation and ultra-wideband RCS suppression while maintaining a low profile and structural simplicity.

[0004] Therefore, how to achieve broadband operation, high-gain radiation, and ultra-wideband RCS reduction while maintaining a low antenna profile and simple structure in high-frequency applications such as the Ka band remains a pressing technical problem that needs to be solved. Summary of the Invention

[0005] To address the aforementioned problems, this invention aims to provide a broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface. Through a cross-stacked architecture of three metallization layers and two dielectric substrates, a PCM checkerboard array composed of bow-shaped PCM units is integrated above the magnetoelectric dipole radiating unit. Furthermore, the magnetoelectric dipole radiating unit employs a coplanar layout of radiating patches and perforated ring microstrip feed lines. This achieves high gain and wide impedance matching while possessing ultra-wideband RCS reduction and low profile characteristics, providing an effective solution for Ka-band satellite communication and radar stealth applications.

[0006] To achieve the above objectives, the technical solution adopted by the present invention is as follows: A broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface includes: a first dielectric substrate and a second dielectric substrate, wherein the first dielectric substrate is located above the second dielectric substrate; a PCM checkerboard array is provided on the top of the first dielectric substrate, wherein the PCM checkerboard array includes a plurality of bow-shaped PCM units arranged in a mirror-symmetric checkerboard array. The second dielectric substrate has a magnetoelectric dipole radiating unit on its top, and the magnetoelectric dipole radiating unit is located below the first dielectric substrate. The bottom of the second dielectric substrate is provided with a ground plane for grounding the PCM checkerboard array and the magnetoelectric dipole radiating unit.

[0007] Furthermore, the PCM checkerboard array includes 16×16 bow-shaped PCM units, and the PCM checkerboard array removes the 2×4 bow-shaped PCM units corresponding to those disposed above the magnetoelectric dipole radiation unit.

[0008] Furthermore, each 4×4 bow-shaped PCM unit forms a 4×4 PCM checkerboard subarray with the same rotation direction, and the rotation direction of each 4×4 PCM checkerboard subarray is opposite to the rotation direction of all directly adjacent 4×4 PCM checkerboard subarrays.

[0009] Furthermore, the magnetoelectric dipole radiating unit includes two magnetoelectric dipole antenna units arranged side by side. Each magnetoelectric dipole antenna unit includes an open-loop microstrip feed line and two radiating patches symmetrically arranged on the upper and lower sides of the open-loop microstrip feed line. The four radiating patches and the two open-loop microstrip feed lines are arranged in the same plane. The four radiating patches serve as electric dipoles, and the two open-loop microstrip feed lines serve as magnetic dipoles.

[0010] Furthermore, the magnetoelectric dipole radiating unit also includes: A 50-ohm coaxial feed port is located at the bottom of the magnetoelectric dipole radiating unit and extends through the second dielectric substrate and the ground plane.

[0011] Furthermore, the magnetoelectric dipole antenna also includes: Multiple first vias are disposed around the top of the magnetoelectric dipole antenna to connect the first dielectric substrate and the second dielectric substrate. Two second vias are provided through the top center of the magnetoelectric dipole antenna for mounting the feed coaxial connector.

[0012] Furthermore, the coaxial connector for power supply is equipped with a 50-ohm coaxial transmission line, which is connected to the 50-ohm coaxial power supply port for powering the magnetoelectric dipole radiating unit.

[0013] Furthermore, the dimensions of the magnetoelectric dipole antenna are 73.8 mm × 73.8 mm × 0.762 mm.

[0014] The beneficial effects of this invention are: (1) The scattering part of the antenna of the present invention is composed of a PCM checkerboard array and a first dielectric substrate. The PCM checkerboard array is composed of 16×16 bow-shaped PCM units arranged in a mirror-symmetric checkerboard array. The structure is simple, and by adjusting the geometric parameters of the bow-shaped PCM units, an RCS reduction of more than 10 dB (relative bandwidth of 63.83%) can be achieved in the ultra-wideband range of 28.9-56 GHz. At the same time, the bow-shaped PCM units of the present invention are stable under oblique incidence conditions and can still maintain a high polarization conversion rate (PCR>0.7) within an incidence angle of up to 40°, which has good angular robustness and significantly improves the wide-angle stealth capability of the stealth platform.

[0015] (2) The radiating part of the antenna of the present invention is composed of a magnetoelectric dipole radiating element and a second dielectric substrate. The magnetoelectric dipole radiating element adopts a coplanar design, including four radiating patches as electric dipoles and two perforated ring microstrip feed lines as magnetic dipoles. It is fed through a 50-ohm coaxial feed port located at the bottom of the magnetoelectric dipole radiating element. Based on the principle of complementary radiation of electric dipoles and magnetic dipoles, it not only achieves a broadband impedance matching of 23.26% (30.4-38.4 GHz), but also obtains a peak gain of up to 11.9 dBi. The perforated ring microstrip feed lines have both signal distribution and phase compensation functions, ensuring high radiation efficiency and good radiation pattern characteristics.

[0016] (3) The antenna of the present invention adopts a cross-stacked architecture of three metallization layers and two dielectric substrates, integrates the PCM checkerboard array on the top of the magnetoelectric dipole radiating unit, and the magnetoelectric dipole radiating unit adopts a layout of radiating patches and open-hole ring microstrip feed lines in a coplanar manner. The overall structure thickness is only 0.762 mm, which has a significant advantage of low profile.

[0017] (4) The antenna of the present invention has the excellent characteristics of high gain, wide bandwidth and low RCS, providing an efficient solution for Ka-band satellite communication and radar stealth applications, and has important theoretical and practical value. Attached Figure Description

[0018] Figure 1This is a schematic diagram of the overall structure of the broadband magnetoelectric dipole antenna based on the bow-shaped polarization conversion metasurface in this invention.

[0019] Figure 2 This is a side view of the broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface in this invention.

[0020] Figure 3 This is a top view of the first dielectric substrate in this invention.

[0021] Figure 4 This is a top view of the second dielectric substrate in this invention.

[0022] Figure 5 This is a three-dimensional structural schematic diagram of the magnetoelectric dipole radiation unit in this invention.

[0023] Figure 6 This is a schematic diagram showing the dimensional parameters of the magnetoelectric dipole radiation unit in this invention.

[0024] Figure 7 This is a schematic diagram showing the dimensional parameters of the bow-shaped PCM unit in this invention.

[0025] Figure 8 The simulation results show the reflection characteristics of the bow-shaped PCM unit in this invention under oblique incidence conditions.

[0026] Figure 9 The results show the simulation of the radiation performance of the broadband magnetoelectric dipole antenna based on the bow-shaped polarization conversion metasurface in this invention.

[0027] Figure 10 This shows the surface current and electric field distribution of the magnetoelectric dipole radiating unit in this invention at an operating frequency of 40 GHz.

[0028] Figure 11 This is the prototype of the magnetoelectric dipole antenna manufactured in this invention and the experimental test configuration.

[0029] Figure 12 This is a performance comparison of the reference antenna, the antenna of this invention, and the prototype of this invention.

[0030] Figure 13 The images show the normalized radiation patterns of co-polarization and cross-polarization of the antenna in this invention, as simulated at a working frequency of 38 GHz and as measured by the prototype of this invention.

[0031] Figure 14 The radiation and scattering characteristics of the antenna of the present invention at four operating frequencies of 35 GHz, 36 GHz, 37 GHz and 38 GHz within the operating frequency band are shown.

[0032] Figure 15This is a comparison of the monostatic RCS of the antenna, prototype, and equal-area metal planar reflector of the present invention under normal incidence conditions.

[0033] Among them, 1. PCM checkerboard array; 101. Bow-tie shaped PCM unit; 2. First dielectric substrate; 3. Magnetoelectric dipole radiating unit; 301. Magnetoelectric dipole antenna unit; 3011. 50-ohm coaxial feed port; 3012. Radiating patch; 3013. Open-hole ring microstrip feed line; 4. Second dielectric substrate; 5. Ground plane; 6. First via; 7. Second via. Detailed Implementation

[0034] To enable those skilled in the art to better understand the technical solutions of the present invention, the technical solutions of the present invention will be further described below in conjunction with the accompanying drawings and embodiments.

[0035] See attached document Figure 1 and attached Figure 2 As shown, a broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface employs a multi-layer architecture, comprising, from top to bottom, a PCM checkerboard array 1, a first dielectric substrate 2, magnetoelectric dipole radiating elements 3, a second dielectric substrate 4, and a ground plane 5. The PCM checkerboard array 1, magnetoelectric dipole radiating elements 3, and ground plane 5 are all metallized layers. The PCM checkerboard array 1 is located on top of the first dielectric substrate 2, forming the scattering portion of the magnetoelectric dipole antenna together with the first dielectric substrate 2. The magnetoelectric dipole radiating elements 3 are located on top of the second dielectric substrate 4, forming the radiating portion of the magnetoelectric dipole antenna together with the second dielectric substrate 4. The ground plane 5 is located at the bottom of the second dielectric substrate 4, serving to ground the PCM checkerboard array 1 and the magnetoelectric dipole radiating elements 3. Both the first dielectric substrate 2 and the second dielectric substrate 4 are fabricated using Rogers RO5880 material. The top perimeter of the magnetoelectric dipole antenna is provided with multiple first vias 6 for connecting the first dielectric substrate 2 and the second dielectric substrate 4 into a whole by bolts; the top center of the magnetoelectric dipole antenna is provided with two second vias 7 for installing a feed coaxial connector, and the feed coaxial connector is provided with a 50-ohm coaxial transmission line inside.

[0036] Specifically, the PCM checkerboard array 1 includes 16×16 bowtie-shaped PCM units 101, as shown in the attached diagram. Figure 3As shown; the magnetoelectric dipole radiating element 3 includes two magnetoelectric dipole antenna elements 301 arranged side by side. Each magnetoelectric dipole antenna element 301 includes an open-loop microstrip feed line 3013 and two radiating patches 3012 symmetrically arranged on the upper and lower sides of the open-loop microstrip feed line 3013. The two open-loop microstrip feed lines 3013 are arranged adjacent to each other in the horizontal direction and are connected to a 50-ohm coaxial feed port 3011. The 50-ohm coaxial feed port 3011 and The 50-ohm coaxial transmission line inside the power-feeding coaxial connector is connected; the two perforated annular microstrip feed lines 3013 and the four radiating patches 3012 are coplanarly arranged, the four radiating patches 3012 serve as electric dipoles, and the two perforated annular microstrip feed lines 3013 serve as magnetic dipoles. The 50-ohm coaxial power supply port 3011 is located at the bottom of the magnetoelectric dipole radiating unit 3 and penetrates the second dielectric substrate 4 and the ground plane 5; the structure of the magnetoelectric dipole radiating unit 3 is shown in the attached figure. Figure 4 and attached Figure 5 As shown in the attached diagram. Because the spacing between the magnetoelectric dipole radiating unit 3 and the PCM checkerboard array 1 is very small, coupling effects are easily generated. Therefore, to reduce the impact of the coupling effect, the 2×4 bow-shaped PCM units 101 correspondingly located above the magnetoelectric dipole radiating unit 3 are removed, as shown in the attached diagram. Figure 3 As shown.

[0037] In this invention, each bow-shaped PCM unit 101 is a bow-shaped structure formed by an integral copper-clad molding process, and the 16×16 bow-shaped PCM units 101 are arranged in a mirror-symmetrical checkerboard array. Each 4×4 bow-shaped PCM units 101 form a 4×4 PCM checkerboard sub-array with the same rotation direction, and the rotation direction of each 4×4 PCM checkerboard sub-array is opposite to the rotation direction of all directly adjacent (top, bottom, left, right) 4×4 PCM checkerboard sub-arrays, as shown in the attached figure. Figure 3 As shown.

[0038] The working principle of the magnetoelectric dipole antenna described in this invention is as follows: The four radiating patches 3012 in the magnetoelectric dipole radiating element 3 act as electric dipoles, generating vertically polarized electric field radiation with radiation characteristics similar to a short electric dipole. The two perforated ring microstrip feed lines 3013 act as magnetic dipoles, equivalent to a horizontal magnetic current source, generating radiation field components orthogonal to the electric dipoles. When the electric and magnetic dipoles are simultaneously excited and satisfy appropriate amplitude and phase relationships, their radiation fields in the forward direction of the magnetoelectric dipole antenna are superimposed and enhanced, while their radiation fields in the backward direction cancel each other out, thus forming a unidirectional radiation pattern and effectively suppressing the back lobe level. Simultaneously, since the electric and magnetic dipoles have complementary input impedance characteristics, their parallel connection can achieve good impedance matching over a wide frequency band. This invention uses a 50-ohm coaxial transmission line to feed the magnetoelectric dipole radiation unit 3. By rationally designing the position of the 50-ohm coaxial feed port 3011 and the geometric dimensions of the magnetoelectric dipole radiation unit 3, the electric dipole and the magnetic dipole maintain a stable amplitude ratio and a 90-degree phase difference within the operating frequency band, thereby achieving the technical effects of wide bandwidth, high gain, low cross-polarization, and stable radiation pattern.

[0039] The energy transmission path of the magnetoelectric dipole antenna in this invention is as follows: the electromagnetic energy generated by the radio frequency signal source is first fed into the magnetoelectric dipole antenna through a 50-ohm coaxial transmission line. The inner conductor of the 50-ohm coaxial transmission line is connected to the 50-ohm coaxial feed port 3011, and the outer conductor of the 50-ohm coaxial transmission line forms a good electrical connection with the ground plane 5. After the feed energy enters the magnetoelectric dipole antenna through the 50-ohm coaxial feed port 3011, it is distributed into two parallel transmission paths. In the first path, energy is coupled to an electric dipole unit composed of four radiating patches 3012, exciting a time-varying current distribution on the surface of the radiating patches 3012. This current flows along the longitudinal direction of the radiating patches 3012, generating an electric field radiation along the perpendicular polarization direction, radiating electromagnetic energy into free space in electric dipole mode. In the second path, energy is simultaneously coupled to a magnetic dipole unit composed of two perforated annular microstrip feed lines 3013, forming an equivalent magnetic current distribution at the perforation structure of the two perforated annular microstrip feed lines 3013, generating a magnetic field radiation component orthogonal to the electric dipole, radiating electromagnetic energy into free space in magnetic dipole mode. The two transmission paths of the fed energy converge in space. The electromagnetic fields radiated by the electric dipole and the magnetic dipole are superimposed in phase in front of the aperture of the magnetoelectric dipole antenna and canceled out of phase behind it, thereby achieving directional and efficient transmission of energy to the front half-space, completing the mode conversion from guided wave to radiated wave and spatial power synthesis.

[0040] Simulation Verification 1: To verify the performance of the magnetoelectric dipole antenna described in this embodiment, numerical simulation was performed using CST simulation software. After optimization via CST simulation, the specific dimensional parameters of the magnetoelectric dipole antenna in this embodiment are shown in Table 1 below (unit: mm). Correspondingly, the dimensional parameters of the magnetoelectric dipole radiating element 3 are shown in the attached figure. Figure 6 As shown, the dimensional parameters of the bow-shaped PCM unit 101 are attached. Figure 7 As shown.

[0041] Table 1 shows the size parameters of the magnetoelectric dipole antenna obtained through simulation optimization. Simulation Verification 2: To verify the reflection performance of the bow-shaped PCM unit 101 described in this embodiment under oblique incidence conditions, its characteristics within the incidence angle range of 0° to 40° were simulated and analyzed. The results are shown in the attached figure. Figure 8 As shown.

[0042] Appendix Figure 8 (a) illustrates the variation of the co-polarization reflection amplitude of the bowtie-shaped PCM unit 101 with the incident angle. At perpendicular incidence, the bowtie-shaped PCM unit 101 exhibits resonant points at 31.1 GHz and 38.6 GHz. As the incident angle increases, the two resonant points undergo frequency shifts: the low-frequency resonant point shifts towards lower frequencies, and the high-frequency resonant point shifts towards higher frequencies. Furthermore, the reflection amplitude at both resonant points increases with increasing incident angle.

[0043] Polarization conversion efficiency (PCR) is defined as: In the formula, Indicates the same polarization reflection coefficient. This represents the cross-polarization reflection coefficient.

[0044] Appendix Figure 8 (b) The PCR curve of the bow-shaped PCM unit 101 as a function of incident angle is shown. Simulation results show that when the incident angle is no greater than 30°, the PCR of the bow-shaped PCM unit 101 in the 28.9-43.3 GHz band is ≥0.8, and the polarization conversion bandwidth is 39.89%; even when the incident angle is increased to 40°, its PCR in the 28.6-41.8 GHz band remains consistently higher than 0.7. These results indicate that the bow-shaped PCM unit 101 exhibits excellent angular stability and broadband polarization conversion characteristics under oblique incidence conditions.

[0045] Appendix Figure 8(c) shows the surface current distribution of the bowtie-shaped PCM unit 101 at two resonant points of 31.1 GHz and 38.6 GHz. Simulation results show that at these two resonant points, the upper and lower surface currents of the bowtie-shaped PCM unit 101 are in opposite directions, reflecting the excitation of the magnetic resonance mode. Therefore, the polarization conversion function of this structure is mainly realized through the magnetic resonance mechanism.

[0046] Simulation Verification 3: To verify the impedance matching characteristics and radiation performance of the magnetoelectric dipole antenna described in this embodiment, numerical simulations were performed using electromagnetic simulation software. The simulation results of the radiation performance of the magnetoelectric dipole antenna are attached. Figure 9 As shown.

[0047] Appendix Figure 9 (a) illustrates the return loss of the magnetoelectric dipole antenna. Simulation curves of gain and amplitude are shown. Simulation results indicate that... With a standard of <-10 dB, the impedance bandwidth of the magnetoelectric dipole antenna is 23.74% (32.3-41 GHz). The magnetoelectric dipole antenna achieves a peak gain of 12.72 dBi at 40 GHz.

[0048] Appendix Figure 9 (b) shows the normalized far-field radiation pattern of the magnetoelectric dipole antenna in the E-plane (xz plane) at an operating frequency of 40 GHz, with both co-polarized and cross-polarized components also shown. The pattern indicates that its cross-polarization level is below -28 dB and its front-to-back ratio exceeds 24 dB, demonstrating that the magnetoelectric dipole antenna exhibits excellent polarization purity and strong directivity in the E-plane, effectively suppressing backscattering.

[0049] Appendix Figure 9 (c) shows the normalized far-field radiation pattern of the magnetoelectric dipole antenna in the H-plane (yz plane) at an operating frequency of 40 GHz, with both co-polarized and cross-polarized components also shown. The pattern exhibits good symmetry and a clear main lobe shape, indicating that the magnetoelectric dipole antenna has stable beam directivity and radiation symmetry in the H-plane.

[0050] The radiation characteristics of the E-plane and H-plane described above together demonstrate that the magnetoelectric dipole antenna achieves high gain and stable radiation performance at the target frequency.

[0051] The perforated ring microstrip feed 3013 of the magnetoelectric dipole radiating unit 3 serves both signal transmission and phase compensation functions in its design. Figure 10 The surface current and electric field distribution of the magnetoelectric dipole radiating element 3 at an operating frequency of 40 GHz are shown. The working principle of the magnetoelectric dipole antenna described in this invention can be understood through the attached diagram. Figure 10(a) provides a visual explanation of the surface current distribution. When the magnetoelectric dipole radiating unit 3 is excited, a characteristic current distribution pattern forms on its surface, which reveals the cooperative radiation mechanism of the electric dipole and the magnetic dipole.

[0052] Specifically, the four radiating patches 3012 exhibit a strong and consistent current distribution, as shown by the black arrows in the figure. The main current is concentrated along the positive x-direction, with a peak current intensity of up to 79.1 A / m, establishing the x-axis as the main radiation axis. This in-phase current distribution is equivalent to four parallel short electric dipoles. The in-phase superposition of the currents in the x-direction forms a wide-side radiation pattern, generating an electric field radiation component polarized along the y-direction in the far field. Although a reverse current can be observed along the y-axis, this reverse current is not part of the main radiation mode; its main function is to suppress mutual coupling between adjacent radiating patches 3012. Simultaneously, the two perforated annular microstrip feed lines 3013 exhibit a current distribution flowing along the y-direction. This current forms a closed loop at the edge of the perforation, which, according to the equivalence principle, can be equivalent to a magnetic current source distributed along the perforation direction, constituting a magnetic dipole radiation unit, generating a field component orthogonal to the electric dipole radiation in the far field. Among them, the leftmost and rightmost branches of the perforated annular microstrip feed line 3013 play a key role, acting as both radiation units and phase compensators, ensuring that the four radiating patches 3012 achieve coherent and in-phase excitation through the two middle perforated annular microstrip feed lines 3013.

[0053] Due to the rational design of the feeding structure of the magnetoelectric dipole radiating element 3, the currents excited by the electric dipole and the magnetic dipole maintain a phase difference of approximately 90 degrees in time. Their radiated fields are superimposed in phase in the normal direction of the magnetoelectric dipole antenna and cancel each other out in phase in the opposite direction, thus forming a stable unidirectional broadband radiation characteristic. (Appendix) Figure 9 (b) and appendix Figure 9 The simulation pattern in (c) verifies the above radiation mechanism: the main lobe points to the normal (z-axis direction), and no beam splitting phenomenon occurs, indicating that the magnetoelectric dipole antenna described in this invention operates in a normal resonance state rather than an anti-resonance state, achieving the technical effects of high gain, low back lobe, and wide bandwidth.

[0054] Appendix Figure 10 The electric field distribution in (b) further confirms the radiation mechanism described above. The open ends of the four radiating patches 3012 generate electromagnetic field discontinuities, which are crucial for generating an efficient radiation field. The x-direction branch of the apertured ring microstrip feed line 3013 also exhibits a strong electric field intensity, providing an additional contribution to the improvement of the radiation gain of the magnetoelectric dipole antenna.

[0055] The dual-function design of the aperture ring microstrip feeder 3013 combines signal distribution and phase compensation functions, enabling the magnetoelectric dipole antenna to achieve maximum radiation gain through multi-element coordinated in-phase radiation.

[0056] Simulation Verification 4: To verify the actual radiation performance and radar characteristic performance of the magnetoelectric dipole antenna described in this embodiment, simulation, fabrication, and testing were conducted. The prototype magnetoelectric dipole antenna fabricated according to the magnetoelectric dipole antenna described in this embodiment and the experimental test configuration are attached. Figure 11 As shown.

[0057] Appendix Figure 11 (a) A top view of the first dielectric substrate 2 is shown, clearly showing the PCM checkerboard array 1 on top of the first dielectric substrate 2. (See attached image) Figure 11 (b) A top view of the second dielectric substrate 4 is shown, with attached... Figure 11 (c) is a radiation performance testing device, with attachments Figure 11 (d) is the scattering performance testing device.

[0058] For a direct performance comparison, the following definitions are provided: the magnetoelectric dipole antenna without the first dielectric substrate 2 and the PCM checkerboard array 1 is referred to as the reference antenna; the magnetoelectric dipole antenna containing the first dielectric substrate 2 and the PCM checkerboard array 1 is referred to as the antenna of this invention; and the prototype of the magnetoelectric dipole antenna fabricated according to the antenna of this invention is referred to as the prototype of this invention.

[0059] The performance comparison results of the reference antenna, the antenna of this invention, and the prototype of this invention are attached. Figure 12 As shown, Sim.Ref. represents the reference antenna simulation, Sim.Pro. represents the antenna simulation of the present invention, and Mea.Pro. represents the actual measurement of the prototype of the present invention.

[0060] Appendix Figure 12 (a) Shows the S-parameter comparison curves of the reference antenna, the antenna of the present invention, and the prototype of the present invention. For the antenna of the present invention, due to the addition of the PCM checkerboard array 1 at the top, mutual coupling inevitably exists between the PCM checkerboard array 1 and the magnetoelectric dipole radiating element 3, which prolongs the effective current path. Therefore, the return loss of the antenna of the present invention differs significantly from that of the reference antenna, and the simulated operating frequency of the antenna of the present invention and the measured operating frequency of the prototype of the present invention are both shifted towards lower frequencies. (See attached diagram) Figure 12 (a) It can be seen that the relative bandwidth of the reference antenna simulation is 23.26% (32.3-40.8 GHz), the relative bandwidth of the antenna simulation of the present invention is 23.26% (30.4-38.4 GHz), and the relative bandwidth of the prototype of the present invention measured is 22.35% (30.2-37.8 GHz). The deviation from the simulation results is less than 1%, which is within an acceptable range. This high consistency strongly verifies the broadband characteristics of the antenna of the present invention.

[0061] Appendix Figure 12(b) Demonstrates the realized gain characteristics of the reference antenna, the antenna of the present invention, and the prototype of the present invention. The results show that the reference antenna achieves a maximum gain of 12.7 dBi at 40 GHz, while the antenna of the present invention achieves a maximum gain of 11.9 dBi at 38 GHz; below 34.7 GHz, the realized gain of the antenna of the present invention is greater than that of the reference antenna. This indicates that the addition of the PCM checkerboard array 1 generates new resonant modes, increasing the effective antenna radiating aperture and thus improving the realized gain; however, the opposite is true above 34.7 GHz. This suggests that the PCM checkerboard array 1 and the magnetoelectric dipole radiating element 3 interact, leading to a decrease in the realized gain of the antenna of the present invention.

[0062] Performance comparisons of the reference antenna, the antenna of this invention, and the prototype of this invention show that the prototype achieves a maximum gain of 12.2 dBi at 38 GHz, and the gain error between the simulated antenna and the measured prototype is less than 0.5 dB. This indicates that the antenna of this invention has good consistency and verifies the effectiveness of the antenna in maintaining high gain characteristics.

[0063] Appendix Figure 13 The simulated far-field radiation patterns of the antenna of this invention at a working frequency of 38 GHz and the measured far-field radiation patterns of the prototype are shown, exhibiting both co-polarized and cross-polarized normalized radiation patterns. (Attached is a note.) Figure 13 (a) is the far-field radiation pattern of the E-plane (xz plane), with appendix. Figure 13 (b) shows the far-field radiation pattern in the H-plane (yz-plane). The results show that the simulated cross-polarization level of the antenna in the E-plane and the measured level of the prototype in the E-plane are almost both below -31 dB. This indicates that the antenna of this invention possesses high signal quality, strong anti-interference capability, and excellent radiation pattern characteristics. This performance can effectively reduce inter-channel interference in polarization multiplexing systems and meet the stringent polarization purity requirements of high-demand applications such as satellite communication and 6G.

[0064] Appendix Figure 14 The radiation and scattering characteristics of the antenna of this invention at four operating frequencies of 35 GHz, 36 GHz, 37 GHz, and 38 GHz are demonstrated. (Attached...) Figure 14 (a) is the far-field radiation pattern of the antenna of the present invention, with appended diagrams. Figure 14 (b) is the bistatic RCS pattern of the antenna of the present invention, with appended diagrams. Figure 14 (c) is the bistatic RCS pattern of an equal-area metal plane reflector.

[0065] From the appendix Figure 14 As can be seen in (a), the antenna radiation of this invention maintains a stable high gain characteristic within the operating frequency band. From the scattering characteristics, the attached... Figure 14(b) shows that when a normally incident wave irradiates the antenna of this invention, electromagnetic energy is mainly scattered along the four diagonal directions, which effectively reduces the normal reflection energy, thereby achieving effective RCS reduction; Appendix Figure 14 (c) shows that for a metal planar reflector, the incident electromagnetic energy is mainly reflected along the normal direction, without producing a significant scattering effect. Therefore, the PCM checkerboard array 1 composed of bowtie-shaped PCM elements 101 can effectively reduce the antenna RCS.

[0066] To verify the scattering characteristics of the antenna of this invention, simulations were performed, and actual measurements were conducted on the prototype. A comparative analysis was also conducted with an equal-area metal planar reflector. During RCS measurements, a 50-ohm coaxial feed port 3011 of the antenna of this invention was connected to a 50-ohm matched load. The selection of the 50-ohm matched load was based on the following considerations: (1) 50 ohms is the characteristic impedance of a standard radio frequency system, and this condition best represents the actual working state of the antenna. (2) Under the condition of matching load, there is no reflection at the 50-ohm coaxial feed port 3011, which can suppress the antenna mode scattering to the greatest extent and make the measurement results more reflective of the RCS reduction effect of the PCM checkerboard array 1. (3) In practical applications, the antenna is usually connected to a matched transceiver system, so the RCS data under matched load conditions has higher engineering reference value.

[0067] The total RCS of an antenna can be decomposed into two parts: the structural mode RCS and the antenna mode RCS. The structural mode RCS is determined by the antenna's physical structure and is independent of the load terminated at the 50-ohm coaxial feed port 3011. The antenna mode RCS is related to the reflection coefficient of the load terminated at the 50-ohm coaxial feed port 3011. Under matched load conditions, the reflection coefficient of the 50-ohm coaxial feed port 3011 is zero, resulting in minimal antenna mode scattering, and the measured RCS primarily reflects structural mode scattering. Under open-circuit or short-circuit termination conditions, the reflection coefficient of the 50-ohm coaxial feed port 3011 is 1, resulting in maximum antenna mode scattering and a significantly increased total RCS. Under reactive load conditions, the 50-ohm coaxial feed port 3011 exhibits partial reflection, and the antenna mode scattering falls between the two aforementioned cases.

[0068] The antenna design of this invention employs a PCM checkerboard array 1 to achieve RCS reduction, a method primarily aimed at suppressing structured mode scattering. Therefore, verifying the RCS reduction effect of the PCM checkerboard array 1 under matched load conditions is the most reasonable approach. Due to the limitation of the test equipment's maximum test frequency of 40 GHz, the test frequency range is 25-40 GHz, and the simulation frequency range is 25-56 GHz. Figure 15The monostatic RCS comparison results of the antenna of this invention, the prototype of this invention, and the equal-area metal planar reflector under normal incidence conditions are presented. (Attached...) Figure 15 (a) shows the monostatic RCS comparison curves of the antenna of the present invention, the prototype of the present invention, and the equal-area metal planar reflector, with appended figures. Figure 15 (b) shows the RCS reduction curve of the antenna of the present invention. (See attached image.) Figure 15 In this context, "Proposed antenna (Sim.)" represents the actual measured antenna of the present invention, "Metal plane (Sim.)" represents the simulation of the equal-area metal plane reflector, and "Proposed antenna (Mea.)" represents the actual measured prototype of the present invention.

[0069] Measurement results show that, compared with a planar metal reflector of equal area, the antenna of the present invention achieves an RCS reduction of more than 10 dB in the entire 28.9-56 GHz frequency band, with an RCS reduction bandwidth of 63.83%, a maximum reduction of 19 dB, and an average reduction of 13 dB, verifying the effectiveness of the antenna design method of the present invention.

[0070] It should be noted that, attached Figure 15 (a) The simulation results of the antenna of the present invention differ significantly from the measured data of the prototype of the present invention. The main reasons include the following aspects: (1) The process requires welding a feed coaxial connector on the top of the second dielectric substrate 4, which introduces an air gap layer between the first dielectric substrate 2 and the second dielectric substrate 4, thereby affecting the scattering performance of the antenna of the present invention.

[0071] (2) There are differences between the simulation environment and the actual measurement environment.

[0072] (3) The influence of machining tolerances and manufacturing errors.

[0073] To demonstrate that the antenna of this invention has high gain, wide bandwidth and low RCS characteristics, Table 2 compares the antenna of this invention with existing antenna models.

[0074] Specifically, the PCM unit in reference [1] adopts a single-layer structure design, which achieves an RCS reduction of about 10 dB in three frequency bands, but the gain is relatively low; the antennas in references [2] and [3] have excellent RCS reduction bandwidth, but only achieve out-of-band RCS reduction, and the in-band RCS reduction effect is not obvious; compared with reference [4], the antenna of the present invention has the advantage of low profile. The above comparative analysis shows that the antenna of the present invention not only has excellent radiation performance, but also exhibits excellent scattering characteristics.

[0075] Table 2 Performance comparison between the antenna of this invention and existing antenna models Among them, Document [1] - Document [4] are respectively as follows: Document [1]: T. Hong, S. Wang, Z. Liu and S. Gong, "RCS Reduction and GainEnhancement for the Circularly Polarized Array by Polarization ConversionMetasurface Coating," in IEEE Antennas and Wireless Propagation Letters, vol.18, no. 1, pp. 167-171, Jan. 2019. Document [2]: J. Liu, J. -Y. Li and Z. N. Chen, "Broadband PolarizationConversion Metasurface for Antenna RCS Reduction," in IEEE Transactions onAntennas and Propagation, vol. 70, no. 5, pp. 3834-3839, May 2022. Document [3]: Q. Zheng, W. Liu, Q. Zhao, L. Kong, Y. -H. Ren and X. -X.Yang, "Broadband RCS Reduction, Antenna Miniaturization, and BandwidthEnhancement by Combining Reactive Impedance Surface and PolarizationConversion Metasurface," in IEEE Transactions on Antennas and Propagation,vol. 72, no. 9, pp. 7395-7400, Sept. 2024. Literature [4]: ​​Y. Hei, M. Wang, Z. Shen and W. Wu, "Radial Line Slot Arrayof Low Radar Cross Section for Target Detection and Radar Stealth in Missile-Borne Sensing Systems," IEEE Sensors Journal, vol. 23, no. 22, pp. 28084-28094, 15 Nov.15, 2023. In summary, this invention proposes a PCM checkerboard array composed of bowtie-shaped PCM elements, integrated above a magnetoelectric dipole radiating element, effectively reducing the antenna's RCS characteristics. The prototype magnetoelectric dipole antenna fabricated according to this invention has physical dimensions of 73.8 mm × 73.8 mm × 0.762 mm. Under a -10 dB impedance bandwidth condition, the antenna of this invention exhibits a relative bandwidth of approximately 23.26% and a peak gain of 11.9 dBi. Simultaneously, it achieves an RCS reduction of over 10 dB in the 28.9-56 GHz frequency band, with an RCS reduction bandwidth of 63.83%. Test results demonstrate that the antenna of this invention possesses excellent characteristics of high gain, wide bandwidth, and low RCS, making it valuable for applications in high-frequency wireless communication systems and radar stealth technology.

[0076] The foregoing has shown and described the basic principles, main features, and advantages of the present invention. Those skilled in the art should understand that the present invention is not limited to the above embodiments. The embodiments and descriptions in the specification are merely illustrative of the principles of the invention. Various changes and modifications can be made to the invention without departing from its spirit and scope, and all such changes and modifications fall within the scope of the present invention as claimed. The scope of protection of this invention is defined by the appended claims and their equivalents.

Claims

1. A broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface, comprising: A first dielectric substrate (2) and a second dielectric substrate (4), wherein the first dielectric substrate (2) is located above the second dielectric substrate (4); characterized in that: The top of the first dielectric substrate (2) is provided with a PCM checkerboard array (1), which includes a plurality of bow-shaped PCM units (101) arranged in a mirror-symmetric checkerboard array. The second dielectric substrate (4) is provided with a magnetoelectric dipole radiation unit (3) on its top, and the magnetoelectric dipole radiation unit (3) is located below the first dielectric substrate (2); The bottom of the second dielectric substrate (4) is provided with a ground plane (5) for grounding the PCM checkerboard array (1) and the magnetoelectric dipole radiation unit (3).

2. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 1, characterized in that: The PCM checkerboard array (1) includes 16×16 bow-shaped PCM units (101), and the PCM checkerboard array (1) removes 2×4 bow-shaped PCM units (101) that are disposed above the magnetoelectric dipole radiation unit (3).

3. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 2, characterized in that: Each 4×4 bow-shaped PCM unit (101) forms a 4×4 PCM checkerboard subarray with the same rotation direction, and the rotation direction of each 4×4 PCM checkerboard subarray is opposite to the rotation direction of all directly adjacent 4×4 PCM checkerboard subarrays.

4. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 2, characterized in that: The magnetoelectric dipole radiating unit (3) includes two magnetoelectric dipole antenna units (301) arranged side by side. Each magnetoelectric dipole antenna unit (301) includes an open-aperture ring microstrip feed line (3013) and two radiating patches (3012) symmetrically arranged on the upper and lower sides of the open-aperture ring microstrip feed line (3013). The four radiating patches (3012) and the two open-aperture ring microstrip feed lines (3013) are arranged in the same plane. The four radiating patches (3012) serve as electric dipoles, and the two open-aperture ring microstrip feed lines (3013) serve as magnetic dipoles.

5. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 4, characterized in that, The magnetoelectric dipole radiation unit (3) also includes: A 50-ohm coaxial feed port (3011) is located at the bottom of the magnetoelectric dipole radiation unit (3) and extends through the second dielectric substrate (4) and the ground plane (5).

6. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 5, characterized in that, The magnetoelectric dipole antenna also includes: Multiple first vias (6) are disposed around the top of the magnetoelectric dipole antenna to connect the first dielectric substrate (2) and the second dielectric substrate (4). Two second vias (7) are provided through the top center of the magnetoelectric dipole antenna for mounting the feed coaxial connector.

7. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 6, characterized in that: The power supply coaxial connector is equipped with a 50-ohm coaxial transmission line, which is connected to the 50-ohm coaxial power supply port (3011) for powering the magnetoelectric dipole radiation unit (3).

8. The broadband magnetoelectric dipole antenna based on a bow-shaped polarization conversion metasurface according to claim 7, characterized in that: The dimensions of the magnetoelectric dipole antenna are 73.8 mm × 73.8 mm × 0.762 mm.

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