W-band low-sidelobe slot waveguide array antenna

Abstract

The invention relates to a W-band low-sidelobe slot waveguide array antenna, and the antenna comprises a metal waveguide housing which is internally provided with a waveguide channel; the radiation slot array comprises a plurality of radiation slots formed in the wide wall surface of the metal waveguide shell, the plurality of radiation slots are arranged in a two-dimensional array, and the excitation amplitude of each radiation slot is determined by a two-dimensional Taylor distribution weighting coefficient; the hierarchical power division feed network comprises multi-stage power dividers which are branched stage by stage, the power division ratio of each stage of power divider is determined by a two-dimensional Taylor distribution weighting coefficient, each stage of power divider generates an inherent phase difference under the condition of unequal power division ratios, and the power division ratios, the geometric parameters of a bifurcation region and the length of a phase compensation section meet a constraint relation. The sum of the inherent phase difference and the compensation phase difference compensates for the path phase difference between the two output branches. Amplitude weighting and phase compensation are jointly achieved in the same power divider structure, the number of independent compensation sections is reduced, the total path of a feed network is shortened, and W-waveband transmission loss is reduced.

Description

Technical Field

[0001] This application relates to the field of radar antennas, and more particularly to a W-band low sidelobe slot waveguide array antenna. Background Technology

[0002] As one of the core sensors in autonomous driving perception systems, automotive radar faces ever-increasing demands for both range and angular resolution in target detection. Currently, mainstream automotive radar operates in the 77GHz band. With the rapid increase in the number of vehicles equipped with automotive radar, the device density in this band is constantly rising, leading to increasingly prominent co-channel interference issues between multiple radars. This results in a decrease in the radar's signal-to-noise ratio and an increase in the false alarm rate, affecting the reliability of the perception system. At the same time, the available bandwidth provided by the 77GHz band is limited, making it difficult to support the ultra-high range and angular resolution requirements of next-generation autonomous driving systems.

[0003] The W-band (75GHz to 110GHz) has become a candidate frequency band for the evolution of automotive radar towards higher performance due to its abundant spectrum resources, clean interference background, and higher angular resolution resulting from a shorter operating wavelength. However, the higher operating frequency of the W-band places more stringent requirements on antenna design and manufacturing.

[0004] Regarding antenna design, most automotive radars currently employ microstrip patch antennas. Microstrip patch antennas have already exhibited observable dielectric loss and reduced radiation efficiency in the 77GHz band. When the operating frequency is further increased to the W-band, the dielectric loss in the microstrip line approximately increases with the square of the frequency, leading to a significant deterioration in antenna gain and radiation efficiency. Waveguide structures, due to the propagation of electromagnetic waves within an air cavity surrounded by metal walls, suffer from much lower transmission loss than microstrip structures, offering significant advantages in low loss and high power capacity in the W-band. Slotted waveguide antennas, which use slots as radiating elements on the wide walls of the waveguide, combine the low-loss transmission characteristics of the waveguide with the high-gain characteristics of an array antenna, making them a competitive implementation for W-band automotive radar antennas.

[0005] A core challenge in practical applications of slotted waveguide array antennas is suppressing sidelobe levels. Sidelobes in the antenna's radiation pattern can cause radar to receive echo signals from non-target directions, generating false targets or reducing the ability to detect real targets. To suppress sidelobes, it is typically necessary to weight the excitation amplitude distribution of the array aperture, ensuring that the excitation amplitude of each radiating element exhibits a gradual change from high at the center to low at the edges. The Taylor distribution is a commonly used aperture weighting function that can achieve a near-narrow main lobe width under specified sidelobe level constraints. In slotted waveguide array antennas, implementing Taylor weighting requires precise control of the power distribution ratio of the feed network to each slot element. This necessitates that the power divider in the feed network can achieve unequal power distribution according to the Taylor distribution coefficients.

[0006] However, achieving accurate amplitude weighting in the W-band presents a significant technical challenge: controlling phase consistency. The operating wavelength in the W-band is approximately 3 millimeters. Phase deviations introduced by differences in physical path lengths, waveguide bends, and discontinuities in the feed network are significantly amplified when measured on a wavelength scale. To compensate for these phase deviations, existing solutions typically include separate phase compensation sections in each output branch of the power divider, adjusting the additional length of the waveguide in a specific branch to offset path differences. While this approach is straightforward, it has two drawbacks: firstly, each independent phase compensation section increases the total physical path length of the feed network. In the W-band, a longer path means greater cumulative transmission loss, weakening the original low-loss advantage of the waveguide structure; secondly, large-scale arrays have numerous power divider stages, requiring independent compensation sections for each branch of each stage, causing the structural complexity and fabrication difficulty of the feed network to increase rapidly with array size. Summary of the Invention

[0007] In order to achieve accurate Taylor weighting and complete phase compensation in a more compact structure in a W-band slot waveguide array antenna, thereby reducing the total path length and structural complexity of the feed network, this application provides a W-band low sidelobe slot waveguide array antenna.

[0008] This application provides a W-band low sidelobe slot waveguide array antenna, which adopts the following technical solution: A W-band low sidelobe slot waveguide array antenna, characterized in that it comprises: A metal waveguide housing, wherein a waveguide channel is formed inside the metal waveguide housing; A radiating slot array, comprising multiple radiating slots formed on a wide wall surface of the metal waveguide housing, the multiple radiating slots being arranged in a two-dimensional array along both row and column directions, the excitation amplitude of each radiating slot being determined by a two-dimensional Taylor distribution weighting coefficient; and A hierarchical power divider network is disposed within the waveguide channel and connected to the radiating slot array. The hierarchical power divider network includes multi-stage power dividers branching sequentially along the signal transmission direction. Each stage of the power divider splits the input signal into two output signals. The power division ratio of each stage is determined by the weighting coefficients of the two-dimensional Taylor distribution. Each stage of the power divider generates an inherent phase difference under unequal power division ratio conditions. At least some of the output branches of each stage of the power divider are provided with a phase compensation section. The power division ratio of each stage of the power divider, the geometric parameters of the bifurcation region of each stage of the power divider, and the length of the phase compensation section satisfy the following constraint: the sum of the inherent phase difference and the compensated phase difference generated by the phase compensation section compensates for the path phase difference between the two output branches of each stage of the power divider.

[0009] By adopting the above technical solution, the power division ratio of each stage of the power divider is determined by the weighting coefficients of the two-dimensional Taylor distribution, so that the power divider naturally generates an inherent phase difference related to the power division ratio while completing power distribution. This inherent phase difference, together with the compensated phase difference generated by the phase compensation section, is used to compensate for the path phase difference between the two output branches, so that the same power divider structure can simultaneously perform both amplitude weighting and phase compensation functions. Compared with the scheme where amplitude weighting and phase compensation are implemented by separate structures, the inherent phase difference of the power divider shares part of the compensation, thereby shortening the length of the phase compensation section and reducing the total physical path of the feed network. In the W-band, waveguide transmission loss accumulates linearly with path length. The shortening of the total path of the feed network directly reduces the insertion loss from the input port to each radiation slot, allowing the antenna to maintain high radiation efficiency and gain in the W-band. At the same time, the constraint relationship between the power division ratio, the geometric parameters of the bifurcation region, and the length of the compensation section provides multiple adjustable degrees of freedom for the design, allowing the feed network to simultaneously meet the requirements of amplitude accuracy and phase accuracy within a compact physical space.

[0010] Optionally, the hierarchical power distribution network is arranged in a biaxial mirror symmetry about the center of the radiating slot array. The biaxial mirror symmetry causes the hierarchical power distribution network to form four quadrants, one of which is a reference quadrant, and the other three quadrants are mirror copies of the reference quadrant.

[0011] By adopting the above technical solution, dual-axis mirror symmetry ensures that the two feed branches at symmetrical positions have the same physical path length, the same number of bends, and the same power divider topology. This naturally ensures that the radiating gaps at symmetrical positions have a consistent excitation phase, eliminating the need for additional compensation measures. Furthermore, only one reference quadrant requires parameter design for the feed network; the other three quadrants are obtained through mirror replication, reducing the number of independently designed power dividers to one-quarter of the total, thus lowering design complexity. In addition, the symmetrical layout ensures that phase deviations caused by path length differences in the four quadrants exist only between asymmetrical branches within each quadrant, without introducing additional phase mismatches between quadrants. This results in a more uniform and controllable phase error distribution across the entire array.

[0012] Optionally, the geometric parameters of the bifurcation region of each power divider include the bifurcation angle, the gradual length of the output branch cross section, and the fillet radius at the bifurcation start point; the geometric parameters of the bifurcation regions of different power dividers in the multi-stage power divider are different from each other, and the bifurcation angle and fillet radius of the bifurcation regions of different power dividers are set according to the power division ratio and target compensation phase difference of the corresponding stage.

[0013] By employing the above technical solution, the size of the bifurcation angle affects the coupling strength and field distribution symmetry between the two output branches, thus influencing the magnitude and sign of the inherent phase difference of the power divider. The size of the fillet radius affects the degree of field discontinuity at the bifurcation initiation; a greater degree of discontinuity results in greater reflections and additional phase shifts in the bifurcation region. By differentiating the bifurcation angle and fillet radius according to the power division ratio and target phase difference compensation of the corresponding stage, the geometry of the bifurcation region of each stage power divider is matched to the power division ratio and phase compensation amount required for that stage. For stages with a large power division ratio deviation from 1:1 (such as branches near the array edge), a larger bifurcation angle is used to achieve stronger asymmetric coupling; for stages with a large target phase difference compensation, the fillet radius is adjusted to generate a larger additional phase shift in the bifurcation region. This differentiated setting makes the bifurcation region itself one of the sources of phase compensation contribution, further reducing the dependence on independent phase compensation segments.

[0014] Optionally, the power divider located at the last stage in the multi-stage power divider is the final stage power divider. The inherent phase difference of the final stage power divider compensates for the path phase difference between the two output branches of the final stage power divider. The output branches of the final stage power divider do not have the phase compensation section. The two output branches of the final stage power divider are connected to the corresponding radiation slots.

[0015] By adopting the above technical solution, the final-stage power divider is the most numerous stage in the multi-stage power divider, occupying the largest number of branches in the entire feed network. The absence of a phase compensation section in the output branch of the final-stage power divider means that the waveguide length occupied by the compensation section is eliminated for the most numerous stage power dividers, maximizing the reduction in the total path length of the feed network. The spacing between adjacent radiating slots connected to the final-stage power divider is typically on the order of half a waveguide wavelength, corresponding to a small path phase difference. Since the power division ratio of the final-stage power divider is close to 1:1 (because the Taylor distribution changes gradually in the central region of the array), the resulting inherent phase difference is also small and matches the order of magnitude of the path phase difference, allowing the inherent phase difference to independently compensate for the path phase difference. Direct connection of the output branch to the radiating slot also eliminates transmission loss and discontinuity reflections in the compensation section waveguide, minimizing the signal transmission path and loss between the final-stage power divider and the radiating slot.

[0016] Optionally, the waveguide channel has a narrow wall adjacent to the wide wall; at least some of the power dividers in each stage have an inductive or capacitive diaphragm disposed on the inner wall of the bifurcation region; the inductive diaphragm protrudes along the narrow wall toward the interior of the waveguide channel, and the capacitive diaphragm protrudes along the wide wall toward the interior of the waveguide channel; the insertion depth and placement position of the inductive or capacitive diaphragm are jointly determined with the power division ratio of each stage of the power divider, the geometric parameters of the bifurcation region, and the length of the phase compensation section.

[0017] By employing the above technical solution, inductive and capacitive diaphragms introduce local reactive disturbances in the bifurcation region, altering the phase characteristics of the electromagnetic field within the region. The inductive diaphragm bulges along the narrow wall, equivalent to a series inductor, causing the electromagnetic wave passing through this region to have a phase lead; the capacitive diaphragm bulges along the wide wall, equivalent to a parallel capacitor, causing the electromagnetic wave to have a phase lag. The insertion depth of the diaphragm determines the magnitude of the introduced reactance, and its placement determines which segment of the field distribution in the bifurcation region the reactive disturbance affects. By jointly determining the diaphragm's insertion depth and placement along with the power division ratio, the bifurcation region's geometric parameters, and the compensation segment length, the diaphragm participates as an additional phase tuning degree of freedom in the overall constraint relationship optimization. Given a fixed power division ratio and bifurcation region geometric parameters, relying solely on the compensation segment length is insufficient to meet the target phase difference compensation. The diaphragm provides the ability to independently fine-tune the inherent phase difference without altering the power division ratio and the bifurcation region's shape, expanding the feasible solution space for joint optimization.

[0018] Optionally, the length of the phase compensation segment is an integer multiple of a preset fractional value of the waveguide wavelength, where the preset fractional value is one-eighth of the waveguide wavelength; the phase compensation segment of each stage of the multi-stage power divider selects the corresponding length value from the discrete value set of the preset fractional value.

[0019] By adopting the above technical solution, the length of the phase compensation section is limited to an integer multiple of one-eighth of the waveguide wavelength, ensuring that the length of the compensation section for each stage of the power divider is selected from only a finite number of discrete values. One-eighth of the waveguide wavelength corresponds to a phase step of 45 degrees, which translates to a length step of approximately 0.5 to 0.6 millimeters in the W-band (waveguide wavelength approximately 4 to 5 millimeters). This step size matches the typical tolerance level of W-band waveguide fabrication, ensuring that the phase quantization error introduced by discretization is within the range covered by the fabrication tolerance. Discrete values ​​allow compensation sections of different stages and branches to share the same waveguide length specification, reducing the number of types of compensation sections that need to be independently customized during fabrication, lowering the complexity of molds and tooling, and facilitating mass production.

[0020] Optionally, each of the plurality of radiation slots has a lateral offset relative to the centerline of the waveguide channel. The power division ratio of each stage of the power divider realizes the first-stage amplitude distribution between the two output branches. The lateral offset of each radiation slot realizes the second-stage amplitude calibration of the excitation amplitude of each radiation slot. The first-stage amplitude distribution and the second-stage amplitude calibration together realize the excitation amplitude required by the two-dimensional Taylor distribution weighting coefficient.

[0021] By adopting the above technical solution, the excitation amplitude required by the two-dimensional Taylor distribution weighted coefficients is decomposed into a two-stage implementation mechanism. The power divider's power ratio completes the amplitude distribution among branches at the feed network level, allocating power to each output branch according to the proportional relationship of the Taylor distribution coefficients, establishing an amplitude ratio framework among the branches. The lateral offset of the radiating slots completes the excitation amplitude calibration of each slot at the radiating element level. By adjusting the offset of the slot relative to the waveguide centerline, the coupling strength between the slot and the electromagnetic field within the waveguide is controlled, allowing for fine-tuning of the amplitude after power divider allocation on a slot-by-slot basis. This two-stage mechanism means that the power divider's power ratio does not need to bear all the amplitude control accuracy. The design of the power divider can better accommodate the inherent phase difference and phase compensation requirements while meeting the amplitude ratio framework, providing a larger feasible solution space for the joint constraint relationship between the power divider, geometric parameters, and compensation section length.

[0022] Optionally, at least some of the radial slits have microstructures on their edges. The microstructures are notches or protrusions, and the size of the microstructures is five to fifteen percent of the width of the corresponding radial slit. The microstructures cause the resonant frequency and coupling of the corresponding radial slit to shift without changing the length of the corresponding radial slit and the lateral offset.

[0023] By employing the above technical solution, the microstructure alters the current distribution path at the edge of the radiating slot. The notch locally extends the current path, effectively increasing the electrical length of the slot and shifting the resonant frequency towards lower frequencies; the protrusion locally shortens the current path, effectively reducing the electrical length of the slot and shifting the resonant frequency towards higher frequencies. The size of the microstructure is controlled within 5% to 15% of the slot width, ensuring that the resonant frequency shift and coupling variation introduced by the microstructure are within a controllable fine-tuning range. Without changing the slot length and lateral offset, the microstructure provides a third degree of tuning freedom independent of the first-stage amplitude distribution and the second-stage amplitude calibration. This compensates for deviations between the actual coupling and design values ​​of each slot caused by factors such as manufacturing tolerances, inter-slot coupling, and frequency shift, improving the fit between the array aperture excitation amplitude distribution and the target Taylor distribution.

[0024] Optionally, a matching pin or a matching blind hole is provided in the region between adjacent radiation slots on the wide wall surface of the metal waveguide housing. The matching pin or the matching blind hole extends from the outer surface of the wide wall surface toward the interior of the waveguide channel. The insertion depth and diameter of the matching pin or the depth and diameter of the matching blind hole are determined based on the disturbance to the input impedance of each radiation slot caused by the mutual coupling effect between adjacent radiation slots.

[0025] By employing the above technical solution, electromagnetic mutual coupling exists between adjacent radiating slots in a two-dimensional slot array. This mutual coupling causes the actual input impedance of each slot to deviate from its design value when it exists alone, resulting in an increase in the slot's reflection coefficient and a deterioration in the port VSWR of the feed network. Matching pins or matching blind holes extend from the outer surface of the wide wall towards the inside of the waveguide channel, forming a localized reactive disturbance inside the waveguide. The insertion depth and diameter of the pins or the depth and diameter of the blind holes determine the magnitude and nature (inductive or capacitive) of the introduced reactive value. By determining the size parameters of the pins or blind holes based on the disturbance to the input impedance of each slot caused by the mutual coupling effect, the introduced reactive disturbance is made to be opposite in direction and matched in magnitude to the impedance shift caused by the mutual coupling, thereby pulling the actual input impedance of each slot back to a level close to the design value. Matching pins or matching blind holes are placed in the region between adjacent slots. This region has no slots on the wide wall and is an unused structural space. Using this region to set up matching elements does not increase the area of ​​the antenna aperture, nor does it affect the radiation characteristics of the radiating slots.

[0026] Optionally, each of the plurality of radiation slots is a composite slot with multiple resonant modes. The composite slot includes a main slot and an additional arm connected to the end or middle of the main slot. The shape of the additional arm makes the composite slot H-shaped, cross-shaped, or dumbbell-shaped. The length of the additional arm of the radiation slot at different positions in the plurality of radiation slots is set differently according to the weighting coefficient of the two-dimensional Taylor distribution at the corresponding position.

[0027] By adopting the above technical solution, the additional arm introduces an extra resonant mode based on the main slot. The resonant frequency of the main slot and the resonant frequency of the additional arm correspond to different frequency points, and the impedance bandwidth of the slot is broadened within the frequency range between the two resonant frequency points. The broadening of the bandwidth allows the coupling of the radiating slots to remain stable over a wider frequency range. When the operating frequency deviates from the center frequency, the deviation between the actual coupling of each slot and the Taylor distribution design value is reduced, thereby maintaining the sidelobe suppression effect throughout the entire operating bandwidth. The length of the additional arm for the radiating slots at different positions is set differently according to the Taylor weighting coefficients at the corresponding positions, so that the additional resonant frequency of each slot is adapted to the coupling required by that slot, thus broadening the bandwidth without destroying the amplitude ratio between the slots required by the Taylor distribution.

[0028] In summary, this application includes at least one of the following beneficial technical effects: 1. By ensuring that the power division ratio of each stage of the power divider, the geometric parameters of the bifurcation region, and the length of the phase compensation section satisfy the joint constraint relationship, the inherent phase difference caused by the unequal power division ratio is transformed into a source of contribution for phase compensation. This allows the power divider to participate in phase compensation while completing Taylor weighted amplitude allocation, reducing the number and length of independent phase compensation sections and shortening the total physical path of the feed network. This reduces the cumulative transmission loss of the feed network in the W-band, enabling the antenna to maintain high radiation efficiency and gain in the W-band.

[0029] 2. A two-stage amplitude control mechanism is implemented by using the power divider to achieve the first-stage amplitude distribution and the lateral offset of the radiation gap to achieve the second-stage amplitude calibration. This allows the power divider design to better meet the phase compensation requirements while satisfying the amplitude ratio framework, expands the feasible solution space of the joint constraint relationship, and improves the ability of the power supply network to simultaneously meet the amplitude accuracy and phase accuracy requirements in a compact space.

[0030] 3. By designing the feed network as a dual-axis mirror symmetric layout, designing the final stage power divider as a direct connection structure with zero compensation section, and limiting the length of the compensation section to discrete values, the design difficulty and manufacturing cost of large-scale slot waveguide array antennas in the W-band are reduced in three aspects: simplification of symmetry, reduction of path length, and reduction of processing complexity. Attached Figure Description

[0031] Figure 1 This is a schematic diagram of the overall structure of a W-band low sidelobe slot waveguide array antenna in one embodiment of the present invention.

[0032] Figure 2 This is a top view of the layout of a radiating slot array in one embodiment of the present invention.

[0033] Figure 3 This is a schematic diagram of the topology of a hierarchical power distribution network in one embodiment of the present invention.

[0034] Figure 4 This is a schematic diagram of the direct connection between the final stage power divider and the radiation gap in one embodiment of the present invention.

[0035] Figure 5 This is a schematic diagram showing the arrangement of the sensitive diaphragm and the capacitive diaphragm in the bifurcation region according to an embodiment of the present invention. Figure 1 .

[0036] Figure 6 This is a schematic diagram showing the arrangement of the sensitive diaphragm and the capacitive diaphragm in the bifurcation region according to an embodiment of the present invention. Figure 2 .

[0037] Explanation of reference numerals in the attached figures: 100. Metal waveguide housing; 110. Waveguide channel; 120. Wide wall surface; 130. Narrow wall surface; 200. Radial slot array; 210. Radial slot; 211. Main slot; 213. Microstructure; 214. Lateral offset; 300. Hierarchical power divider network; 310. First-stage power divider; 320. Second-stage power divider; 330. Final-stage power divider; 340. Bifurcation area; 350. Output branch; 360. Phase compensation section; 400. Sensitive membrane; 410. Capacitive membrane. Detailed Implementation

[0038] The present application will be further described in detail below with reference to the accompanying drawings. It should be understood that the specific embodiments described herein are merely illustrative of the application and are not intended to limit the scope of the application.

[0039] This application discloses a W-band low sidelobe slot waveguide array antenna. Before describing the embodiments of this application in detail, some terms will be explained first.

[0040] The W band refers to the frequency range of approximately 75 GHz to 110 GHz in the electromagnetic spectrum. In this band, the free-space wavelength of electromagnetic waves is approximately 2.7 mm to 4 mm, and the corresponding waveguide wavelength (i.e., the equivalent wavelength of electromagnetic waves propagating in a rectangular waveguide) is approximately 4 mm to 5 mm. The specific value depends on the waveguide cross-sectional dimensions and the operating frequency.

[0041] The Taylor distribution is a mathematical function used for amplitude weighting at the aperture of an antenna array. By controlling the zero-point position of the amplitude distribution function at the array aperture, the Taylor distribution achieves a near-narrow main lobe width under specified sidelobe level constraints. The two-dimensional Taylor distribution is a two-dimensional weighting function obtained by multiplying the one-dimensional Taylor distribution along both row and column directions, resulting in a two-dimensional gradient amplitude distribution at the array aperture with a high center and low edges. The weighting coefficients of the two-dimensional Taylor distribution refer to the function values ​​of the two-dimensional Taylor distribution function at each radiating element position in the array. Each radiating element corresponds to a weighting coefficient, which determines the required excitation amplitude for that element.

[0042] Side lobes are radiation lobes in an antenna's radiation pattern other than the main lobe. Side lobe levels are typically expressed in decibels relative to the main lobe peak value. Lower side lobe levels indicate a stronger ability of the antenna to suppress signals from non-target directions.

[0043] A power divider is a passive microwave device that distributes the signal power from the input port to multiple output ports according to a preset ratio. The power ratio refers to the power distribution ratio among the output ports of the power divider. When the power ratio is not 1:1, it is called an unequal power ratio.

[0044] The inherent phase difference refers to the phase deviation that naturally occurs between the two output ports of a power divider under unequal power division ratios due to the asymmetry of the electromagnetic field distribution within the bifurcation region 340. The magnitude and sign of the inherent phase difference are related to the power division ratio and the geometry of the bifurcation region 340.

[0045] Figure 1 The overall structure of a W-band low sidelobe slotted waveguide array antenna according to some embodiments of this application is shown. Figure 1 As shown, the antenna comprises three components: a metal waveguide housing 100, a radiating slot array 200, and a hierarchical power divider feed network 300.

[0046] The metal waveguide housing 100 serves as the mechanical support structure for the antenna and the transmission carrier for electromagnetic waves. A waveguide channel 110 is formed within the metal waveguide housing 100, within which electromagnetic waves propagate with low loss. The waveguide channel 110 has a rectangular cross-section, with a wide wall 120 and a narrow wall 130 adjacent to the wide wall 120. As an example, the metal waveguide housing 100 can be made of aluminum alloy or copper alloy. When using aluminum alloy as the housing material, a layer of highly conductive metal (e.g., gold or silver) can be plated onto the inner wall of the waveguide channel 110 to further reduce transmission loss in the W-band.

[0047] The radiating slot array 200 is the radiating part of the antenna, responsible for coupling electromagnetic energy within the waveguide channel 110 into free space. The radiating slot array 200 includes multiple radiating slots 210 formed on the wide wall 120 of the metal waveguide housing 100. The multiple radiating slots 210 are arranged in a two-dimensional array along the row and column directions. As an example, the radiating slot array 200 can be an 8x8 array or a 16x16 array; no limitation is made here. The excitation amplitude of each radiating slot 210 is determined by a two-dimensional Taylor distribution weighting coefficient; that is, the radiating slots 210 located in the center region of the array are assigned larger excitation amplitudes, while the radiating slots 210 located in the edge regions of the array are assigned smaller excitation amplitudes, and the amplitude gradient follows a two-dimensional Taylor distribution function.

[0048] The hierarchical power divider feed network 300 is the feed section of the antenna, responsible for distributing the power of the input signal to each radiating slot 210 according to the weighting coefficients of the two-dimensional Taylor distribution, and performing phase compensation between branches while distributing the power. The hierarchical power divider feed network 300 is disposed in the waveguide channel 110 and connected to the radiating slot array 200, that is, the waveguide channel 110 and the radiating slots 210 of the feed network are integrated together in the same metal waveguide housing 100.

[0049] Figure 3 The topology of a hierarchical power distribution network 300 according to some embodiments of this application is shown. For example... Figure 3 As shown, the hierarchical power divider network 300 includes multi-stage power dividers that branch sequentially along the signal transmission direction. Each stage of the multi-stage power divider splits the input signal into two output signals, i.e., each power divider is a 1-to-2 structure. Taking an 8x8 array as an example, the hierarchical power divider network 300 includes three stages of power dividers along the column direction: the first-stage power divider 310 splits the input signal into two paths, the second-stage power divider 320 further splits each path into two paths, and the final-stage power divider 330 further splits each path into two paths, thus branching one input signal into eight output signals, which are fed into eight columns of radiating slots 210. The power divider structure along the row direction is similar.

[0050] The power division ratio of each stage of the power divider is determined by the two-dimensional Taylor distribution weighted coefficients. Taking the second-stage power divider 320 as an example, the two output branches 350 of the second-stage power divider 320 are respectively connected to the radiating slots 210 of different column groups in the array. The ratio of the sum of the Taylor weighted coefficients of the radiating slots 210 in the two column groups is the power division ratio of the second-stage power divider 320. For example, in an 8-column array, the second-stage power divider 320 divides four columns into two groups (two columns each). The sum of the two-dimensional Taylor distribution weighted coefficients of the two columns closer to the center of the array is 0.92, and the sum of the two-dimensional Taylor distribution weighted coefficients of the two columns closer to the edge of the array is 0.58. Then the power division ratio of the second-stage power divider 320 is 0.92:0.58, which is approximately 1.59:1.

[0051] Power dividers at different stages generate inherent phase differences under unequal power division ratios. Taking the second-stage power divider 320 with a power division ratio of 1.59:1 as an example, the phase carried by the electromagnetic wave in the higher-power output branch 350 (the branch with a two-dimensional Taylor distribution weighting coefficient sum of 0.92) when leaving the bifurcation region 340 differs from the phase carried by the electromagnetic wave in the lower-power output branch 350 (the branch with a two-dimensional Taylor distribution weighting coefficient sum of 0.58). This difference is the inherent phase difference. In this embodiment, the inherent phase difference of the power divider with a power division ratio of 1.59:1 is approximately 8 degrees.

[0052] At least some of the output branches 350 in each power divider stage are provided with a phase compensation section 360. The phase compensation section 360 is an additional waveguide in the output branch 350, and a specific amount of compensated phase difference is generated by adjusting the length of the additional waveguide. In conventional schemes where no inherent phase difference is involved in compensation, the phase compensation section 360 needs to compensate for the entire path phase difference between the two output branches 350 on its own. However, in the scheme of this application, the inherent phase difference undertakes a portion of the compensation for the path phase difference, and the phase compensation section 360 only needs to compensate for the difference between the path phase difference and the inherent phase difference, thus shortening the length of the compensation section.

[0053] Specifically, the power division ratio of each stage of the power divider, the geometric parameters of the bifurcation region 340 of each stage of the power divider, and the length of the phase compensation section 360 satisfy the following constraint: the sum of the inherent phase difference and the compensated phase difference generated by the phase compensation section 360 compensates for the path phase difference between the two output branches 350 of each stage of the power divider. Taking the second-stage power divider 320 as an example, the path phase difference between the two output branches 350 is 23 degrees (i.e., the phase difference caused by the different physical lengths of the two branches), and the inherent phase difference is 8 degrees. Therefore, the compensated phase difference that the phase compensation section 360 needs to generate is 23 degrees minus 8 degrees, which equals 15 degrees. In the traditional scheme, the compensation section needs to generate a 23-degree compensated phase difference on its own, while in this scheme, it only needs to generate 15 degrees, corresponding to a reduction of approximately 35% in the waveguide length of the compensation section.

[0054] It should be understood that the above constraints are not satisfied solely by adjusting the compensation section length. The power division ratio determines the magnitude of the inherent phase difference, the geometric parameters of the bifurcation region 340 affect the precise value of the inherent phase difference and the additional phase shift within the bifurcation region 340, and the compensation section length determines the magnitude of the compensated phase difference. All three factors work together to satisfy the constraints; that is, the power division ratio, geometric parameters, and compensation section length are jointly adjusted during the design phase to ensure that the constraints are satisfied at each stage of the power divider.

[0055] In some embodiments, the hierarchical power divider feed network 300 is arranged in a biaxial mirror symmetry about the center of the radiating slot array 200. Biaxial mirror symmetry means that the feed network is mirror symmetric about both the horizontal and vertical axes of the array center. This biaxial mirror symmetry results in the hierarchical power divider feed network 300 forming four quadrants. One quadrant is the reference quadrant, and the other three quadrants are mirror copies of the reference quadrant. Taking an 8x8 array as an example, the reference quadrant contains 16 feed branches corresponding to 4x4 radiating slots 210. The power division ratio, bifurcation region 340 geometric parameters, and compensation section length of each power divider in the reference quadrant are independently designed and optimized. The power dividers at corresponding positions in the other three quadrants directly adopt the mirror structure of the power dividers at the same position in the reference quadrant, eliminating the need for redesign.

[0056] The two-dimensional Taylor distribution function possesses the mathematical property of biaxial symmetry, meaning that the weighting coefficients of the two-dimensional Taylor distribution are symmetrically distributed about both the horizontal and vertical axes of the array center. The biaxial symmetric layout of the feed network is consistent with the biaxial symmetry property of the Taylor distribution, ensuring that the two feed branches in symmetrical positions have the same physical path length and the same power division ratio, thus naturally achieving consistent excitation phase and excitation amplitude.

[0057] In other embodiments, the power supply network may also be mirror-symmetric about only one axis (single-axis symmetry). In this case, the power supply network forms two half-regions, one half-region being the reference half-region and the other half-region being a mirror copy of the reference half-region. Single-axis symmetry can also utilize the symmetry of the Taylor distribution to reduce the number of independently designed power dividers, but the reduction is not as significant as that of dual-axis symmetry.

[0058] Figure 3The diagram also illustrates the geometric parameters of the bifurcation regions 340 of each power divider stage. In some embodiments, the geometric parameters of the bifurcation regions 340 of each power divider stage include the bifurcation angle, the gradient length of the output branch 350 cross-section, and the fillet radius at the bifurcation initiation. The bifurcation angle refers to the angle between the two output branches 350 of the power divider at the bifurcation initiation. The gradient length of the output branch 350 cross-section refers to the waveguide length traversed by the output branch 350 as it transitions from the bifurcation region 340 cross-section to the standard waveguide cross-section. The fillet radius at the bifurcation initiation refers to the fillet radius of the waveguide wall at the bifurcation point.

[0059] In a multi-stage power divider, the geometric parameters of the bifurcation regions 340 differ between different stages. The bifurcation angle and fillet radius of the bifurcation regions 340 of different stages are set according to the power division ratio of the corresponding stage and the target compensation phase difference. The size of the bifurcation angle affects the coupling strength between the two output branches 350: the larger the bifurcation angle, the weaker the electromagnetic coupling between the two output branches 350 in the bifurcation region 340, the more obvious the asymmetry of the field distribution in the bifurcation region 340, and the greater the inherent phase difference. The size of the fillet radius affects the degree of field discontinuity at the bifurcation initiation: the smaller the fillet radius, the stronger the discontinuity, and the greater the reflection and additional phase shift generated in the bifurcation region 340.

[0060] Taking a three-stage power divider structure as an example, the power division ratio of the first-stage power divider 310 is close to 1:1 (because the symmetry of the Taylor distribution makes the total power of the left and right groups nearly equal), and the target compensation phase difference is small. Therefore, the first-stage power divider 310 adopts a small bifurcation angle and a large fillet radius to make the field distribution in the bifurcation region 340 nearly symmetrical, with small inherent phase difference and additional phase shift. The power division ratio of the second-stage power divider 320 deviates from 1:1 (e.g., 1.59:1), and the target compensation phase difference is large. Therefore, the second-stage power divider 320 adopts a large bifurcation angle and a small fillet radius to generate a large inherent phase difference and additional phase shift in the bifurcation region 340, thus sharing more compensation.

[0061] Figure 4 A direct connection structure between the final-stage power divider 330 and the radiating slot 210 according to some embodiments of this application is shown. In some embodiments, the final-stage power divider in a multi-stage power divider is the final-stage power divider 330. The inherent phase difference of the final-stage power divider 330 compensates for the path phase difference between the two output branches 350 of the final-stage power divider 330. No phase compensation section 360 is provided in the output branches 350 of the final-stage power divider 330, and the two output branches 350 of the final-stage power divider 330 are directly connected to the corresponding radiating slot 210.

[0062] The final-stage power divider 330 is located at the very end of the power divider tree structure. Each final-stage power divider 330 has two output branches 350 connected to two adjacent radiating slots 210. The spacing between adjacent radiating slots 210 is typically on the order of half a waveguide wavelength, for example, approximately 2 mm to 2.5 mm in the W-band. Therefore, the path length difference between the two output branches 350 of the final-stage power divider 330 is very small, and the corresponding path phase difference is typically no more than 10 degrees. Simultaneously, the two-dimensional Taylor distribution weighting coefficients of the two adjacent slots connected to the final-stage power divider 330 are close (because the Taylor distribution is a continuous smooth function, the function values ​​at adjacent positions differ little), resulting in a power division ratio close to 1:1 for the final-stage power divider 330. The resulting inherent phase difference is also small, matching the order of magnitude of the path phase difference. For example, when the power division ratio of the final stage power divider 330 is 1.05:1, the inherent phase difference is about 2 degrees and the path phase difference is about 3 degrees. The inherent phase difference can compensate for most of the path phase difference, and the residual phase error is about 1 degree, which is within the acceptable range of W-band antennas.

[0063] like Figure 4 As shown, the two output branches 350 of the final-stage power divider 330 connect directly to the two radiating slots 210 on the wide wall 120 without any additional waveguide sections after leaving the bifurcation region 340. The final-stage power divider 330 is the stage with the most power dividers in the multi-stage power divider. Taking an 8-column array as an example, there are 4 final-stage power dividers 330 (4 in each half of the array), while the first-stage power divider 310 has only 1 and the second-stage power divider 320 has only 2. The final-stage power divider 330 completely omits the compensation section, so that none of the most numerous stage power dividers occupy the waveguide length of the compensation section, which contributes the most to the reduction of the total path length of the feed network.

[0064] Figure 5 and Figure 6 The positions of the sensitive diaphragm 400 and the capacitive diaphragm 410 within the bifurcation region 340 are shown in some embodiments of this application. Figure 5 This is a top view of the sensitive diaphragm in the bifurcation region in one embodiment of the present invention, i.e., a schematic diagram of its placement position in the wide wall direction. Figure 6 This is a side view of the capacitive diaphragm in the bifurcation region in the direction of the narrow wall surface, which is a schematic diagram of the setting position in the direction of the narrow wall surface, according to an embodiment of the present invention.

[0065] In some embodiments, at least a portion of the bifurcation regions 340 of the power dividers in each stage are provided with an inductive diaphragm 400 or a capacitive diaphragm 410 on their inner wall surfaces. The inductive diaphragm 400 protrudes along the narrow wall surface 130 toward the interior of the waveguide channel 110, and in the circuit equivalent model, it acts as a series inductor, causing the electromagnetic waves passing through this region to have a phase lead. The capacitive diaphragm 410 protrudes along the wide wall surface 120 toward the interior of the waveguide channel 110, and in the circuit equivalent model, it acts as a parallel capacitor, causing the electromagnetic waves passing through this region to have a phase lag.

[0066] The insertion depth and placement of the inductive diaphragm 400 or capacitive diaphragm 410 are jointly determined by the power division ratio of each stage of the power divider, the geometric parameters of the bifurcation region 340, and the length of the phase compensation section 360. The insertion depth determines the magnitude of the introduced reactance, which in turn determines the amount of fine-tuning of the diaphragm to the inherent phase difference. The placement determines which section of the field distribution within the bifurcation region 340 the reactive disturbance acts upon. When the diaphragm is placed near the bifurcation start, its phase influence on the two output branches 350 is relatively balanced; when the diaphragm is placed near the entrance of a particular output branch 350, it mainly affects the phase of that output branch 350.

[0067] Taking the second-stage power divider 320 as an example, after the power division ratio and bifurcation angle are determined, if the constraint relationship cannot be accurately satisfied at this stage by simply adjusting the length of the compensation section (e.g., the residual phase error is 4 degrees), a sensitive diaphragm 400 with an insertion depth of 0.3 mm can be set on the inner wall surface of the bifurcation region 340 to increase the inherent phase difference by about 4 degrees, thereby eliminating the residual error.

[0068] In some embodiments, an inductive diaphragm 400 and a capacitive diaphragm 410 may be simultaneously disposed within the same bifurcation region 340. The combination of the inductive diaphragm 400 and the capacitive diaphragm 410 provides phase tuning capability in both positive and negative directions, enabling the diaphragm to both increase and decrease the inherent phase difference, thereby further expanding the feasible solution space for joint optimization.

[0069] In some embodiments, the length of the phase compensation segment 360 is an integer multiple of a preset fractional value of the waveguide wavelength. The preset fractional value is one-eighth of the waveguide wavelength. The phase compensation segment 360 of each stage of the power divider in the multi-stage power divider selects the corresponding length value from a discrete set of preset fractional values.

[0070] Taking the second-stage power divider as an example, the path phase difference between the two output branches is 23 degrees, and the inherent phase difference is 8 degrees. Therefore, the compensation stage needs to generate a compensation phase difference of 15 degrees. The closest value selected from the discrete value set is 0.56 mm (corresponding to 45 degrees). The quantization error between this value and the target value of 15 degrees is 30 degrees. This quantization error is absorbed by adjusting the geometric parameters of the bifurcation region (bifurcation angle and fillet radius), increasing or decreasing the additional phase shift of the bifurcation region by 30 degrees, thereby satisfying the constraint relationship under discretization conditions.

[0071] It should be understood that the preset fraction value is not limited to one-eighth of the waveguide wavelength. In scenarios with higher phase accuracy requirements, one-sixteenth of the waveguide wavelength can be used as the preset fraction value, corresponding to a phase step of approximately 22.5 degrees. This reduces the quantization error accordingly, but increases the number of selectable compensation segments. In scenarios with lower phase accuracy requirements or limited manufacturing conditions, one-quarter of the waveguide wavelength can be used as the preset fraction value, corresponding to a phase step of approximately 90 degrees. This reduces the number of selectable compensation segments to one-quarter.

[0072] Figure 2 A top view layout of a radial slot array 200 according to some embodiments of this application is shown. For example... Figure 2 As shown, each of the multiple radiating slots 210 has a lateral offset 214 relative to the centerline of the waveguide channel 110. The lateral offset 214 refers to the deviation of the center of the radiating slot 210 from the longitudinal centerline of the waveguide channel 110 in the wide-wall direction. The larger the lateral offset 214, the stronger the coupling between the radiating slot 210 and the dominant mode of the electromagnetic field propagating in the waveguide, and the larger the excitation amplitude of the slot; the smaller the lateral offset 214, the weaker the coupling and the smaller the excitation amplitude.

[0073] In some embodiments, the power division ratio of each power divider achieves the first-stage amplitude distribution between the two output branches 350, and the lateral offset 214 of each radiation slot 210 achieves the second-stage amplitude calibration of the excitation amplitude of each radiation slot 210. The first-stage amplitude distribution and the second-stage amplitude calibration together achieve the excitation amplitude required by the two-dimensional Taylor distribution weighting coefficient.

[0074] Taking the radiating slot 210 in the 4th column (from left) of the 8-column array as an example, the two-dimensional Taylor distribution weighting coefficient at this location is 0.95 (close to the center, with a relatively large amplitude). The first-stage amplitude distribution allocates the total power step by step according to the proportional relationship of the two-dimensional Taylor distribution weighting coefficient through the power division ratio of each power divider, so that the power reaching the branch where the radiating slot 210 in the 4th column is located accounts for approximately 13.2% of the total input power (i.e., 0.95 divided by the sum of the two-dimensional Taylor distribution weighting coefficients of all 8 columns, 7.2). The second-stage amplitude calibration adjusts the lateral offset 214 of the radiating slot 210 in the 4th column to precisely match the actual power coupled from the waveguide by this slot with the 13.2% allocation value.

[0075] The two-stage amplitude control mechanism means that the power divider's power ratio does not need to bear the entire amplitude control accuracy. During the design process, the power divider can be offset within a certain range, provided that the approximate proportional relationship of the first-stage amplitude distribution is met, so that the inherent phase difference can better match the target compensation phase difference. The slight deviation in the power divider caused by the offset results in a small deviation between the first-stage amplitude distribution and the two-dimensional Taylor distribution weighting coefficients, which is compensated by the corresponding adjustment of the lateral offset 214 of the radiation slot 210 in the second-stage amplitude calibration.

[0076] In some embodiments, at least a portion of the edges of the plurality of radial slits 210 are provided with microstructures 213. The microstructures 213 are notches or protrusions. A notch refers to a small area cut inward from the long or short edge of the radial slit 210, and a protrusion refers to a small protrusion extending outward from the long or short edge of the radial slit 210. The size of the microstructure 213 is five percent to fifteen percent of the slit width of the corresponding radial slit 210. Taking a radial slit 210 with a slit width of 0.4 mm as an example, the size of the microstructure 213 ranges from 0.02 mm to 0.06 mm.

[0077] Microstructure 213 causes the resonant frequency and coupling of the corresponding radiating slot 210 to shift without changing the length and lateral offset 214 of the corresponding radiating slot 210. The notch locally lengthens the current path at the edge of the slot, effectively increasing the electrical length of the slot and shifting the resonant frequency towards lower frequencies; the boss locally shortens the current path, effectively reducing the electrical length of the slot and shifting the resonant frequency towards higher frequencies. Microstructure 213 provides a third degree of tuning freedom independent of the power ratio and lateral offset 214, which can compensate for deviations between the actual coupling and the design value caused by factors such as manufacturing tolerances and inter-slot coupling.

[0078] In other embodiments, the excitation amplitude of the radiating slot 210 can also be controlled solely by the power division ratio of the power divider, without differentiated adjustment of the lateral offset 214. In this embodiment, the lateral offset 214 of all radiating slots 210 relative to the waveguide centerline is the same, and all amplitude control accuracy is borne by the power division ratio of each stage of the power divider. This method simplifies the fabrication of the radiating slots 210 (the offset of all slots is consistent), but requires higher accuracy of the power division ratio of the power divider, imposing stricter requirements on the fabrication tolerance of the power divider. Both implementation methods (two-stage control of power division ratio plus lateral offset 214, and single-stage control of power division ratio only) are specific implementations of the overarching scheme that "the excitation amplitude of each radiating slot 210 is determined by a two-dimensional Taylor distribution weighted coefficient."

[0079] In some embodiments, a mating pin or a mating blind hole is provided in the region between adjacent radiation slots 210 on the wide wall 120 of the metal waveguide housing 100. The mating pin or the mating blind hole extends from the outer surface of the wide wall 120 toward the interior of the waveguide channel 110. The mating pin is a metal post inserted into the waveguide channel 110 from the outer surface, and the mating blind hole is a blind hole drilled from the outer surface toward the waveguide channel 110 but not penetrating the wide wall 120.

[0080] In a two-dimensional slot array, electromagnetic mutual coupling exists between adjacent radiating slots 210. This mutual coupling causes the actual input impedance of each slot to deviate from its design value when it exists alone, manifesting as an additional inductive or capacitive component in the imaginary part of the impedance. After the matching pin is inserted into the waveguide channel 110 from the outer surface, it creates a local capacitive disturbance inside the waveguide; the greater the insertion depth, the stronger the introduced capacitive disturbance. The matching blind via creates a local inductive disturbance on the wide wall 120; the greater its depth, the stronger the introduced inductive disturbance. By selecting the type and size of the matching pin or matching blind via based on the disturbance to the input impedance of each slot caused by the mutual coupling effect, the introduced reactive disturbance is offset against the impedance shift caused by the mutual coupling.

[0081] Taking the adjacent gaps in the central region of an 8x8 array as an example, the mutual coupling effect causes an inductive shift of approximately 5 ohms in the input impedance of the gap in the central region. Placing a matching pin with a diameter of 0.3 mm and an insertion depth of 0.5 mm in the region between these two gaps introduces a capacitive disturbance of approximately 5 ohms, which cancels out the inductive shift, pulling the actual input impedance of the gap back to near the design value.

[0082] In some embodiments, each of the plurality of radiating slots 210 is a composite slot with multiple resonant modes. The composite slot includes a main slot 211 and additional arms connected to the ends or middle of the main slot 211. The shape of the additional arms causes the composite slot to be H-shaped, cross-shaped, or dumbbell-shaped. The additional arms of the H-shaped slot are connected to the two ends of the main slot 211 and extend perpendicular to the length of the main slot 211. The additional arms of the cross-shaped slot are connected to the middle of the main slot 211 and extend perpendicular to the length of the main slot 211. The additional arms of the dumbbell-shaped slot are connected to the two ends of the main slot 211 and extend in a circular or elliptical shape.

[0083] The additional arm introduces an additional resonant mode based on the main slot 211. The resonant frequency of the main slot 211 is determined by its length, and the resonant frequency of the additional arm is determined by its length. The impedance bandwidth of the slot is broadened within the frequency range between the two resonant frequencies. This bandwidth broadening allows the coupling of the radiating slot 210 to remain stable over a wider frequency range.

[0084] The lengths of the additional arms of the multiple radiating slots 210 at different locations are set differently based on the weighting coefficients of the two-dimensional Taylor distribution at the corresponding locations. The radiating slot 210 located at the center of the array has a larger two-dimensional Taylor distribution weighting coefficient, requiring a larger coupling amount. The length of the additional arm is set to ensure a larger interval between the additional resonant frequency and the main resonant frequency, thereby obtaining a wider impedance bandwidth. The radiating slots 210 located at the edge of the array have a smaller two-dimensional Taylor distribution weighting coefficient, requiring a smaller coupling amount. The length of the additional arm is set to ensure a smaller interval between the additional resonant frequency and the main resonant frequency, thereby obtaining more precise coupling control within a narrower bandwidth.

[0085] In other embodiments, the radiation slot 210 can also be a single rectangular slot without additional arms. The single rectangular slot has a simpler structure and is easier to manufacture, but its impedance bandwidth is narrower. When the operating frequency deviates from the center frequency, the actual coupling of each slot may deviate from the Taylor distribution design value, and the sidelobe level may increase. Both single rectangular slots and composite slots are specific implementations of the radiation slot 210.

[0086] The technical solutions of various embodiments of this application have been described above. It should be understood that the features in the above embodiments can be combined according to actual needs. For example, features such as dual-axis mirror symmetry layout, zero-compensation section design of the final stage power divider 330, fine-tuning of inductive diaphragm 400 / capacitive diaphragm 410, discretization of compensation section length, two-stage amplitude control, microstructure 213 tuning, matching pin / blind hole impedance compensation, and composite slot bandwidth broadening can all be applied to the same antenna to obtain optimal overall performance, or some features can be selected and combined according to specific performance requirements and processing constraints.

[0087] The above embodiments are only used to illustrate the technical solutions of the present invention, and are not intended to limit it. Although the present invention has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that modifications can still be made to the technical solutions described in the foregoing embodiments, or equivalent substitutions can be made to some of the technical features. Such modifications or substitutions do not cause the essence of the corresponding technical solutions to deviate from the spirit and scope of the technical solutions of the embodiments of the present invention, and should all be included within the protection scope of the present invention.

Claims

1. A W-band low sidelobe slot waveguide array antenna, characterized in that, include: A metal waveguide housing (100) has a waveguide channel (110) formed inside it. A radiating slot array (200) includes a plurality of radiating slots (210) formed on a wide wall (120) of the metal waveguide housing (100). The plurality of radiating slots (210) are arranged in a two-dimensional array along the row and column directions. The excitation amplitude of each radiating slot (210) is determined by a two-dimensional Taylor distribution weighting coefficient. as well as A hierarchical power divider network (300) is disposed within the waveguide channel (110) and connected to the radiation slot array (200). The hierarchical power divider network (300) includes a multi-stage power divider that branches sequentially along the signal transmission direction. Each stage of the multi-stage power divider divides the input signal into two output signals. The power division ratio of each stage of the power divider is determined by the weighting coefficient of the two-dimensional Taylor distribution. Each stage of the power divider generates an inherent phase difference under unequal power division ratio conditions. At least some of the output branches (350) of each stage of the power divider are provided with a phase compensation section (360). The power division ratio of each stage of the power divider, the geometric parameters of the bifurcation region (340) of each stage of the power divider, and the length of the phase compensation section (360) satisfy the following constraint relationship: the sum of the inherent phase difference and the compensated phase difference generated by the phase compensation section (360) compensates for the path phase difference between the two output branches (350) of each stage of the power divider.

2. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, The hierarchical power distribution network (300) is arranged in a biaxial mirror symmetry about the center of the radiating slot array (200). The biaxial mirror symmetry makes the hierarchical power distribution network (300) form four quadrants. One of the four quadrants is the reference quadrant, and the other three quadrants are mirror copies of the reference quadrant.

3. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, The geometric parameters of the bifurcation region (340) of each power divider include the bifurcation angle, the gradual length of the output branch (350) cross section, and the radius of the rounded corner at the bifurcation start point; the geometric parameters of the bifurcation region (340) of different power dividers in the multi-stage power divider are different from each other, and the bifurcation angle and the radius of the bifurcation region (340) of different power dividers are set according to the power division ratio and the target compensation phase difference of the corresponding stage.

4. The W-band low sidelobe slot waveguide array antenna according to claim 3, characterized in that, The power divider located at the last stage in the multi-stage power divider is the final stage power divider (330). The inherent phase difference of the final stage power divider (330) compensates for the path phase difference between the two output branches (350) of the final stage power divider (330). The output branch (350) of the final stage power divider (330) does not have the phase compensation section (360). The two output branches (350) of the final stage power divider (330) are connected to the corresponding radiation slot (210).

5. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, The waveguide channel (110) has a narrow wall (130) adjacent to the wide wall (120); at least some of the power dividers in each stage have an inductive diaphragm (400) or a capacitive diaphragm (410) disposed on the inner wall of the bifurcation region (340) of the power divider; the inductive diaphragm (400) protrudes along the narrow wall (130) toward the inside of the waveguide channel (110), and the capacitive diaphragm (410) protrudes along the wide wall (120) toward the inside of the waveguide channel (110); the insertion depth and placement position of the inductive diaphragm (400) or the capacitive diaphragm (410) are jointly determined with the power division ratio of the power dividers in each stage, the geometric parameters of the bifurcation region (340), and the length of the phase compensation section (360).

6. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, The length of the phase compensation segment (360) is an integer multiple of a preset fractional value of the waveguide wavelength, where the preset fractional value is one-eighth of the waveguide wavelength; the phase compensation segment (360) of each stage of the multi-stage power divider selects the corresponding length value from the discrete value set of the preset fractional value.

7. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, Each of the plurality of radiation slots (210) has a lateral offset (214) relative to the centerline of the waveguide channel (110). The power division ratio of each stage of the power divider realizes the first-stage amplitude distribution between the two output branches (350). The lateral offset (214) of each radiation slot (210) realizes the second-stage amplitude calibration of the excitation amplitude of each radiation slot (210). The first-stage amplitude distribution and the second-stage amplitude calibration together realize the excitation amplitude required by the two-dimensional Taylor distribution weighting coefficient.

8. The W-band low sidelobe slot waveguide array antenna according to claim 7, characterized in that, At least some of the radial slits (210) have microstructures (213) on their edges. The microstructures (213) are notches or bosses. The size of the microstructures (213) is five percent to fifteen percent of the slit width of the corresponding radial slit (210). The microstructures (213) cause the resonant frequency and coupling of the corresponding radial slit (210) to shift without changing the length of the corresponding radial slit (210) and the lateral offset (214).

9. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, Matching pins or matching blind holes are provided in the region between adjacent radiation slots (210) on the wide wall (120) of the metal waveguide housing (100), the matching pins or the matching blind holes extending from the outer surface of the wide wall (120) toward the interior of the waveguide channel (110); the insertion depth and diameter of the matching pins or the depth and diameter of the matching blind holes are determined according to the amount of disturbance to the input impedance of each radiation slot (210) by the mutual coupling effect between adjacent radiation slots (210).

10. The W-band low sidelobe slot waveguide array antenna according to claim 1, characterized in that, Each of the plurality of radiation slots (210) is a composite slot with multiple resonant modes. The composite slot includes a main slot (211) and an additional arm connected to the end or middle of the main slot (211). The shape of the additional arm makes the composite slot H-shaped, cross-shaped or dumbbell-shaped. The length of the additional arm of the radiation slot (210) at different positions in the plurality of radiation slots (210) is set differently according to the weighting coefficient of the two-dimensional Taylor distribution at the corresponding position.

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